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Research Report 2012-39 Open-Winding Power Conversion Systems Fed by Half-Controlled Converters Y. Wang, D. Panda, T.A. Lipo, D. Pan Dept. of Elect. & Comp. Engr. University of Wisconsin-Madison 1415 Engineering Drive Madison, WI 53706 University of Wisconsin-Madison College of Engineering Wisconsin Power Electronics Research Center 2559D Engineering Hall 1415 Engineering Drive Madison WI 53706-1691 © Confidential IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 28, NO. 5, MAY 2013 2427 Open-Winding Power Conversion Systems Fed by Half-Controlled Converters Yang Wang, Student Member, IEEE, Debiprasad Panda, Member, IEEE, Thomas A. Lipo, Life Fellow, IEEE, and Di Pan, Student Member, IEEE Abstract—An open-winding power conversion system with two conventional six-switch voltage-source converters (VSC) affords operation with a lower volt–ampere (VA) rating of each device for a given power rating, as well as a degree of fault tolerance. The disadvantages of such a configuration include a higher total device VA rating and increased conduction loss as compared to a single Y/Δ-connected VSC. In certain ac–dc applications, such as telecommunications, wind, and aerospace generator drives, it is desired to have regulated input currents and output voltage, but regenerative operation is either not required or prohibited. For such applications, an alternative open-winding power converter is proposed in this paper where half-controlled converter (HCC) is employed at each end of an open-winding structure. The resultant total switch count and total VA rating are reduced by half, compared to using full bridges. Besides, the basic advantages of an open-winding configuration, use of HCCs also guarantees immunity to dc-bus shootthrough and simplifies the gate drive circuit. The total VA rating of the proposed topology is found to be 42% less than a six-switch VSC. The operating principle, control method, and analysis with simulation and experimental results of the proposed topology in both a grid-tied rectifier application and a PM generator application are illustrated in this paper. Index Terms—Common mode (CM), half-controlled converter, open winding. I. INTRODUCTION HE conceptual structure of a typical open-winding power converter is shown in Fig. 1. A variety of realizations, control schemes, and modulation techniques related to this system have been discussed in the literature [1]–[16]. In particular, different types of converters can be adopted [1]–[7], and there are different combinations of dc voltage levels if two isolated dc-links are adopted [8], [9]. Several unique applications of open-winding systems have been presented in [10]–[12]. In related papers, it was determined that if a common dc-link is used in an open-winding system, then a common-mode (CM) path is present in the circuit, and means must be taken to suppress T Manuscript received June 3, 2012; revised July 30, 2012; accepted September 5, 2012. Date of current version November 22, 2012. Recommended for publication by Associate Editor A. Muetze. Y. Wang is with United Technologies Research Center, University of Wisconsin-Madison, East Hartford, CT 06108 USA (e-mail: wangyangnb@ gmail.com). D. Panda was with the Trans-Coil International LLC, Milwaukee, WI 53224 USA. He is now with GE Aviation Systems, Vandalia, OH 45431 USA (e-mail: [email protected]). T. A. Lipo and D. Pan are with University of Wisconsin-Madison, Madison, WI 53706 USA (e-mail: [email protected]; [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TPEL.2012.2218259 Fig. 1. Conceptual structure of a typical open-winding system. the CM current [5], [7], [16]. The main advantage of an openwinding topology is the lower VA rating of each individual device for a given overall power rating. Another advantage of the open-winding configuration is its fault tolerance [17]–[20]. The disadvantages of an open-winding full-bridge configuration include a higher total device VA rating and increased conduction loss as compared to the conventional closed-winding full bridge. Three-phase current-source and voltage-source halfcontrolled-converters (HCCs) were introduced in [21] and [22], [23] respectively, and their potential as a lower VA rating faulttolerant controlled rectifier was explored in [24]–[28]. The HCC also has found application as open-winding permanent magnet synchronous generator (PMSG) drives [29]. A system with a HCC-fed open-winding configuration combines some features of both HCC and open-winding structure, and it eliminates some disadvantages of existing open-winding systems. The apparent drawback of any HCC-based system is that it does not support regenerative operation [24]–[28]. However, in certain ac–dc applications, such as wind turbines [29], telecommunication equipment [30]–[33], and aerospace applications [34] regenerative operation is either not required [29]–[33] or prohibited [34], in which case circuits with reduced VA rating become viable [29]–[34]. For such applications, a new openwinding system with HCC is proposed in this paper to replace the existing full-bridge rectifier. This new solution will provide acceptable input harmonic distortion, regulated output power, better fault-tolerance, and the system cost will be less since the converter VA rating will be significantly less than a full-bridge. Other benefits include being free of shootthrough and possessing a relatively simple gate drive circuitry. This paper explores the implementation of the proposed topology for two typical ac–dc systems, i.e., 1) an open-winding transformer rectifier; and 2) an open-winding PMSG drive. After serving TCI, LLC for almost six years, he has very recently joined with GE Aviation Vandalia. He has authored more than 20 peer reviewed articles in various IEEE and equivalent international journals and conference proceedings. His interests include electronic power conversion, variable-speed drives, 0885-8993/$31.00 © 2012 IEEE 2428 Fig. 2. IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 28, NO. 5, MAY 2013 Generalized HCC-fed open-winding power-transfer system. Fig. 4. Examples of current distortion. Fig. 3. Per-phase equivalent circuit of the HCC-fed open-winding powertransfer system with a common dc-link. real-time hardware in loop modeling, energy efficient systems, and system integration. II. HCC-FED OPEN-WINDING POWER TRANSFER SYSTEM A generalized HCC-fed open-winding power converter is shown in Fig. 2, representing several typical realizations. While not necessary, a common dc-link is adopted in this paper, leading to benefits such as minimized dc-link voltage ripple and reduced system size and component count. The drawback is the existence of a CM path, which requires that the CM current be minimized. The operating principles of the HCC-fed open-winding system are explained using the per-phase equivalent circuit shown in Fig. 3. Here, positive current i flowing through D4 is controlled by S3 . If i is reversed, then switch S4 regulates the current through D3 . The voltage modulation signal and the phase current must have the same polarity in order to guarantee fully controlled phase current, and a failure to meet this requirement results in current distortion. Two examples are illustrated in Fig. 4, where es is the phase back EMF, vcm d1 is the desired phase-voltage modulation, vcm d2 is the achievable phase-voltage modulation for, icm d is the desired phase current, and is is the achievable phase current with minimum distortion. Back EMF and voltage waveforms are scaled by 50% for a clearer view. Fig. 4(a) shows the case where icm d lags vcm d1 . Immediately after vcm d1 crosses zero, icm d and vcm d1 have opposite polarities, so that vcm d1 cannot be synthesized, causing distortion in is . The best that the HCCs can do at that moment is to maintain 0V across the ac terminals until icm d crosses zero, as highlighted by vcm d2 . The situation of icm d leading vcm d1 can be derived similarly and the leading displacement may result in more distortion than with a lagging condition. The above discussion reveals that Fig. 5. All possible voltage vectors of an HCC-fed open-winding system. Highlighted is the available voltage vector space for current vectors located in 1. the modulated voltage and current must have same polarity to guarantee full controllability. Due to the need for unidirectional power flow, all the existing CM suppressing techniques reported in [5], [7], [16] do not apply, and a new means for low-frequency CM compensation is required. III. VOLTAGE VECTORS AND VOLT–SECONDS CONSTRAINTS All possible voltage vectors of the proposed HCC-fed openwinding system are shown in Fig. 5 as black dots, which resemble the voltage vector plane of a three-level VSC. Note that not every voltage vector is available at each instant. For example, if the current vector falls in 1 (ia > 0, ib < 0, ic < 0), the available voltage vector is limited to the shaded area. State (1 0 0–0 0 0) designates that HCC1 resides at voltage vector (1 0 0) and HCC2 resides at (0 0 0). To minimize the CM current, CM voltage must be eliminated over each PWM cycle. For this purpose, define the two available switching states for WANG et al.: OPEN-WINDING POWER CONVERSION SYSTEMS FED BY HALF-CONTROLLED CONVERTERS Fig. 6. 2429 Example of switching sequence with minimum switching loss. HCC1 as S1 = (1 0 0) and S2 = (0 0 0) with dwell times T1 and T2 , respectively, and define the four available switching states of HCC2 as S3 = (0 1 1), S4 = (0 1 0), S5 = (0 0 1), and S6 = (0 0 0) with dwell times T3−T6 , respectively. The modulation for a given voltage vector V α β (in the stationary reference frame) is shown in (1), where Ts is the PWM switching period. The CM component is eliminated over each PWM cycle ⎧ 2 2 1 1 ⎪ T1 + T3 + T4 + T5 Vdc = Vα ⎪ ⎪ ⎪ 3 3 3 3 ⎪ ⎪ ⎪ √ √ ⎪ ⎪ ⎪ 3 3 ⎪ ⎨ T5 − T4 Vdc = Vβ 3 3 (1) ⎪ ⎪ ⎪ ⎪ T1 − (T3 + T4 ) − (T3 + T5 ) = 0 (zero CM voltage) ⎪ ⎪ ⎪ ⎪ ⎪ T1 + T2 = Ts ⎪ ⎪ ⎩ T3 + T4 + T5 + T6 = Ts . The set of equations in (1) can be solved as (2)–(3) yields Vα 3Vα − 2T1 = 2T3 + T4 + T5 = T1 −−→ T1 = Vdc Vdc T2 = Ts − T1 = Ts − Vα . Vdc (2) (3) While there is not a unique solution to T3−T6 , they can be written as functions of one variable, e.g., T6 , as shown in (4) ⎧ Vα ⎪ ⎪ T3 (T6 ) = − Ts + T6 ⎪ ⎪ V dc ⎪ ⎪ √ ⎪ ⎨ 3 Vβ 1 Vα (4) T4 (T6 ) = Ts − T6 − − ⎪ 2 V 2 Vdc dc ⎪ ⎪ √ ⎪ ⎪ ⎪ ⎪ ⎩ T5 (T6 ) = Ts − T6 − 1 Vα + 3 Vβ . 2 Vdc 2 Vdc The assignment of T6 can now be used to reduce the switching losses. For example, one switching sequence and the corresponding conduction paths at each state are given in Fig. 6, which requires minimum switching action. Note the above results only apply to situations where the back EMF does not have a CM component. Otherwise, it will be desirable to compensate actively using converters in order to minimize the net zero-sequence voltage. Fig. 7. D–Q–0 equivalent circuit model of an open-winding power transfer system, with generator convention. Fig. 8. Proposed PWM current regulator. IV. SYSTEM MODELING AND CONTROLLER DESIGN The generalized D–Q–0 equivalent circuit model of the openwinding power transfer system is shown in Fig. 7 where a synchronous reference frame and generating convention are used, and the corresponding state equations are shown in (5) ⎧ d ⎪ ⎪ ⎪ ⎪ v dq 1 − v dq 2 = eq − Rs idq − dt λdq − jωλdq ⎨ λdq = Lq iq − jLd id (5) ⎪ ⎪ ⎪ ⎪ ⎩ Pe = 3 [eq iq − ω(Ld − Lq )iq id ]. 2 At steady state, the polarity constraint is essentially equivalent to Id /Iq = Vd /Vq . Neglecting resistance and assuming a sinusoidal back EMF, this expression can be further reduced to (6). However, again, (6) may not hold true if a CM component is present E − E 2 − 4Xd Xq Iq2 . 0 = Xd Id2 − EId + Xq Iq2 ⇒ Id = 2Xd (6) The proposed PWM current regulator is shown in Fig. 8. The current controller consists of PI regulators in D–Q coordinates and D–Q decoupling terms. The corresponding low-frequency CM compensator is shown in Fig. 9. In general, there exist several means of generating current command, e.g., power control 2430 Fig. 9. IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 28, NO. 5, MAY 2013 Active CM compensator. and dc-link voltage control. The final duty cycle commands are applied to HCCs depending on the current polarity with regard to the reference direction defined in Fig. 7, as shown in (7): m1a,1b,1c = ma,b,c , m2a,2b,2c = 0 (if ia,b,c > 0) (7) m1a,1b,1c = 0, m2a,2b,2c = −ma,b,c (if ia,b,c < 0) The active CM compensator shown in Fig. 9 consists of two major parts, the third harmonic phase-locked loop (PLL), and the CM current regulator. The output of the compensator v0∗ is included in the three-phase voltage modulation commands. The third harmonic PLL is used to detect and lock onto the third harmonic current that is caused by not only the third harmonic back EMF, but also the operating quadrant limitation. The basic principle of the PLL is explained by (8), where i0 is the measured CM current signal, I3 and θ3 are the amplitude and phase angle of the third harmonic current, and θ3∗ is the estimated phase angle of the third harmonic current Fig. 10. Simulated phase-A current with sinusoidal back EMF: (a) Almost unity displacement angle without CM compensation. (b) At theoretical MPP (i.e., unaligned voltage and current vectors) without CM compensation. (c) At theoretical MPP with CM compensation. i0 cos(θ3∗ ) ≈ I3 sin(θ3 ) cos(θ3∗ ) I3 [sin(θ3 + θ3∗ ) + sin(θ3 − θ3∗ )]. (8) 2 Since the third harmonic current dominates in the CM component, it is sufficient to make the approximation that i0 ≈ I3 sin(θ3 ). The PI controller in the PLL minimizes the last term of (8), i.e., sin(θ3 − θ3∗ ), so that θ3∗ will successfully track θ3 . The output of (8) oscillates at six times the fundamental frequency; therefore, the bandwidth of the PLL should be sufficiently low, or a low-pass filter could be added before the PI controller. The other portion of the active CM compensator is the CM current regulator, which further consists of two parts: the PI controller and the P controller. The principle of the PI compensator is explained in equation (9): = i0 sin(θ3∗ ) ≈ I3 sin(θ3 ) sin(θ3∗ ) = −I3 [cos(θ3 + θ3∗ ) − cos(θ3 − θ3∗ )]. 2 (9) Fig. 11. Power penalty and phase current THD due to unity displacement, for various MPP-suggested operating displacement angles, sinusoidal back EMF assumed. WANG et al.: OPEN-WINDING POWER CONVERSION SYSTEMS FED BY HALF-CONTROLLED CONVERTERS 2431 The PI controller is used to minimize the last term of (9), i.e., I3 cos(θ3 − θ3∗ ), and, assuming cos(θ3 − θ3∗ ) = 1, the last term becomes I3 . The PI output is multiplied by a cosine function and then injected into the PWM duty cycles. Again the sixth harmonic oscillation in the resultant signal of (9) calls for careful attention to the gain selection for the PI compensator. The third harmonic voltage injection from the PI controller only compensates for the inductive voltage drop, but not for the third harmonic voltage drop across a resistance. Therefore, the P controller is necessary, since it compensates for the resistive voltage drop in the CM path. Theoretically, the same CM compensation technique is also suitable for other measurable odd harmonics. V. SIMULATIONS Fig. 12. Simulation results with 0.09 pu third harmonic in back EMF without CM compensation: (a) phase current, and (b) CM current. The simulation results are organized and presented in two subsections for comparing and contrasting the impact of sinusoidal and nonsinusoidal back EMF, and highlighting the effect of saliency. A. Open-Winding Power Transfer System With Sinusoidal Back EMF With a sinusoidal back EMF, the polarity restriction is equivalent to forcing a unity displacement factor between the converter voltage modulation signal and the phase current. Fig. 10 shows the simulated phase current during three different scenarios, i.e., (a) at almost unity displacement angle without CM compensation, (b) at the theoretical maximum-power-point MPP without CM compensation, and (c) at the theoretical MPP with CM compensation. The D–Q–0 inductances are 0.31, 0.5, and 0.078 pu, respectively. The results presented in Fig. 10 demonstrate the fact that a misalignment of the voltage and current vectors results in severe current distortion. It is also shown that the CM compensation is very effective for suppressing the current distortion caused by nonunity displacement factor. Simulations have been carried out for both nonsalient and salient systems with different parameters running at two operating conditions: (1) at unity displacement factor point, and (2) at the theoretical MPP. The summarized results are plotted in Fig. 11. Fig. 11(a) shows the reduction in power transfer at unity displacement factor, which is a penalty for achieving minimum current distortion. It is plotted as a function of the displacement angle between phase current and voltage modulation signal as suggested by the MPP. The power penalty increases as the MPP-suggested displacement angle becomes more leading. The relationship is almost independent of the saliency. Fig. 11(b) shows that the current THD at the MPP increases as the MPPsuggested displacement angle becomes more leading, regardless of the saliency. The above results imply that for the HCC-fed open-winding systems with sinusoidal excitation, it is desirable to design the system MPP residing close to a unity displacement factor point. Fig. 13. Simulation results with 0.09 pu third harmonic in back EMF with CM compensation: (a) phase current, and (b) CM current. B. Open-Winding Power Transfer System with Nonsinusoidal Back EMF This section focuses on the impact of the third harmonics in the back EMF, which is a common occurrence in PM machines. Due to the third harmonic, the polarity restriction is no longer equivalent to unity displacement factor, and seeking the operating point of minimum current distortion is no longer a straightforward problem. Assuming that 0.09 pu third harmonic exists in the back EMF and no CM compensation, significant CM current is induced as shown in Fig. 12. However, if the compensator shown in Fig. 9 is incorporated and the remainder unchanged, there is immediate improvement as shown in Fig. 13. These results prove the effectiveness of the CM compensator. Fig. 14(a) compares the simulated power output with the ideal value as the commanded current vector varies, and (b) shows the trend of the corresponding current THD and the peak-topeak power ripple. As the current angle changes, the simulated average power throughput follows quite closely the ideal value (i.e., the power throughput assuming a four-quadrant converter) except at the MPP, which is a penalty due to limited operating 2432 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 28, NO. 5, MAY 2013 Fig. 15. Configuration of open-winding transformer rectifier. Fig. 16. Configuration of HCC-fed open-winding PMSG system. Fig. 14. Simulated variations of system performances as functions of commanded current vector angles (i.e., internal power factor angle): (a) Ideal power throughput (P_est) and simulated power throughput (P_sim). (b) Current THD and peak-to-peak power ripple. quadrants of the HCCs. Although the average power throughput is regulated, the current waveform may actually have severe distortion as shown in Fig. 14(b). A small current angle results in a large leading displacement angle between the desired current and voltage vectors, causing severe distortion, which agrees with the results illustrated in Fig. 4. It should be apparent that the best current quality point does not necessarily reside at the MPP. Although one can always reduce the zero-sequence current by increasing the CM inductance, a compromise between the power throughput and current quality usually has to be made. Fortunately, the distortion in the current does not add much to the power throughput fluctuation since the back EMF does not have the corresponding harmonic. VI. EXPERIMENTS Experiments have been carried out for (1) an HCC-fed openwinding transformer rectifier as shown in Fig. 15, and (2) an HCC-fed open-winding salient PMSG as shown in Fig. 16. The experiments of the open-winding transformer rectifier are done with grid excitation. The main purpose of this part of the experiment was to first demonstrate the functionality of the HCC-fed open-winding system under pure sinusoidal excitation, and second to show the effects of misalignment of the voltage and current vectors. The experiments of the open-winding salient PMSG were done at fixed speed operation. The main focus is to first investigate the effects of the harmonics in the back EMF, and second to test the effectiveness of active CM compensator. Fig. 17. Measured converter phase-A current ia and source current ia s of the open-winding transformer rectifier: (a) Converter voltage and current in phase. (b) Current leading converter voltage. A. Open-Winding Transformer Rectifier A 1 kW open-winding transformer rectifier was tested in this part of the experiment. The D–Q current commands were generated via an outer dc-voltage control loop. The parameters of the hardware setup can be found in the appendix and the results are presented in Fig. 17. The source voltage contains nontriplen WANG et al.: OPEN-WINDING POWER CONVERSION SYSTEMS FED BY HALF-CONTROLLED CONVERTERS Fig. 18. Measured PMSG current, without active third harmonic compensation: (a) Phase currents. (b) CM current. (c) Spectrum of CM current. odd order harmonics, resulting in the corresponding current harmonics. A three-phase filter capacitor bank is installed at the grid side of the transformer. As the displacement angle between the converter voltage and current increases, the current distortion increases. The results agree very well with the analysis and simulation and confirm the functionality of the HCC-fed open-winding system with sinusoidal excitation. B. Open-Winding PMSG System A 1.7-kW open-winding PMSG was tested in this part of the experiment. Parameters of the open-winding PMSG can be found in the appendix. The back EMF mainly consists of 3.03 V/Hz fundamental, 12.3% third harmonic, and 0.6% fifth harmonic. First, the system was tested without the active CM compensator. A 5 mH (∼ 0.09 pu) CM choke was installed in series with the machine winding for suppressing CM current. The resultant phase current is subject to zero-clamping effect. Note that due to the slightly unbalanced common-choke there is small CM current at fundamental frequency as shown in Fig. 18(c). Next, the active CM compensator was activated (with all other conditions unchanged) and the results are plotted in Fig. 19. Although the CM compensator cannot eliminate the zeroclamping, comparison of the spectrums presented in Figs. 18(c) and 19(c) shows that the third harmonic has been eliminated. The experiments were carried out for a series of operating points and the results are presented in Fig. 20. Fig. 20(a) shows the estimated and measured power outputs. The symbol “Pdes” designates the theoretical air-gap power calculated with the commanded current vector and estimated machine parameters. The 2433 Fig. 19. Measured PMSG current, with active third harmonic compensation: (a) Phase currents. (b) CM current. (c) Spectrum of CM current. symbol “P_DSP” is the air-gap power calculated with the measured average D–Q current and estimated machine parameters. The symbol “P_SG” is the measured average terminal power output of the PMSG. It is seen that “P_DSP” and “Pdes” match very well indicating that the average D–Q currents are well regulated. However, the “P_SG” is lower than the estimated values, and the measured MPP is somewhat different than the estimation. This is mainly caused by resistive loss, parameter estimation errors (including saturation), and possibly negative torque. The experimental results imply that the real MPP may differ from the theoretical value. The THD of the machine phase current and the output power ripple are plotted in Fig. 20(b). The trends of the measured THD and the power ripple agree with that shown in Fig. 14(b). The dynamic response to 25% step changes in the D–Q current command was also demonstrated in the experiments. The results are shown in Fig. 21 and the traces are power output and three-phase currents. The system is completely stable through abrupt transients. C. Observations and Discussions The basic operating principles of the HCC-fed open-winding systems with sinusoidal input voltage have been demonstrated in the experiments of the open-winding transformer rectifier, which confirm the theoretical analysis. A practical openwinding PMSG possessing both a salient rotor and back EMF harmonics was also tested with HCCs. Although there is slight discrepancy between the estimated and measured results due to real world limitations, the effectiveness and stability of the proposed controller, as well as the active CM compensator, have been well demonstrated. 2434 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 28, NO. 5, MAY 2013 tion (50% reduction compared to open-winding structure with conventional full-bridges) in device VA rating compared to a single full-bridge VSC with Y/Δ connection, simplified gate drive circuitry, and immunity to dc-link shootthrough. On the other hand, the open-winding system may have a small current distortion near its zero-crossing and the unidirectional power flow may not always closely follow the MPP operation. The operating principles of the proposed topology have been investigated in detail. The effects of system saliency and harmonics in the back EMF were explored in the case of the open-winding PMSG system. A PWM current regulator was proposed. Since a common dc-link for both HCCs reduces component cost but introduces a zero-sequence path, an active CM compensator was implemented to minimize the low-frequency CM current. The switching-frequency CM behaviors require future work. This new concept has been proven effective in both simulations and experiment work. The proposed system was shown to be stable through any transient changes in the current commands. Given the robust and satisfactory performance of HCC-fed open-winding systems as well as the explained advantages, the proposed concepts are expected to be suitable for nonregenerative power conversion applications that place emphasis on both performance and reliability such as in wind generators, telecommunication equipment and aerospace applications. Fig. 20. Measured results: (a) steady-state power outputs, and (b) current THD and power output ripple. Fig. 21. Dynamic response to 25% step changes in current commands. The major benefits of the proposed systems are (1) a reduced power device VA rating, and (2) an improved reliability due to shootthrough immunity and much simpler gate drive configuration. The limitations, on the other hand, are mainly the slight reduction in the power output, an increase in the current harmonics under certain circumstances, and unidirectional power flow. However, as mentioned previously, unidirectional power flow is not a drawback in certain applications where reverse power flow is not needed and blocked using additional diodes. VII. CONCLUSION This paper has proposed a HCC-fed open-winding power converter and demonstrated operation with two experimental setups where regenerative operation is not required. Apart from all the reported benefits of open-winding structures, the major benefits of the proposed solution includes a 42% reduc- APPENDIX Open-winding transformer rectifier parameters Source: voltage amplitude 80 V, frequency 60 Hz, input current amplitude 12 A. Converter-side inductor: three single-phase 3 mH. Load: DC-bus voltage 90 V, resistor 8 Ω, load power 1 kW. PWM frequency: 10 kHz. Open-winding PMSG and drive parameters PMSG: fundamental frequency 40 Hz, back-emf amplitude 120 V, phase current amplitude 8 A. Inductance: D-axis 19 mH, Q-axis 30 mH, zero-sequence 4 mH. Load: DC-bus voltage 150 V, load power 1.7 kW. PWM frequency: 10 kHz. REFERENCES [1] H. Stemmler and P. Guggenbach, “Configurations of high-power voltage source inverter drives,” in Proc. IEEE Conf. Power Electron. Appl., vol. 5, Sep. 1993, pp. 7–14. [2] T. Kawabata, E. C. Ejiogu, Y. Kawabata, and K. Nishiyama, “New openwinding configurations for high-power inverters,” in Proc.IEEE Symp. Ind. Electron., vol. 2, Jul. 1997, pp. 457–462. [3] Y. Kawabata, M. Nasu, T. Nomoto, E. C. Ejiogu, and T. Kawabata, “Highefficiency and low acoustic noise drive system using open-winding AC motor and two space-vector-modulated inverters,” IEEE Trans. Ind. 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Yang Wang (S’08) was born in Ningbo, Zhejiang, China. He received the B.S. degree from Zhejiang University, Zhejiang, China, in 2007, and the M.S. and Ph.D. degrees from Wisconsin Electric Machines and Power Electronics Consortia, University of Wisconsin–Madison, Madison, both in electrical engineering, in 2009 and 2011, respectively. He is currently with United Technologies Research Center, Hartford, CT. His research interests include power electronics and machine drives. Debiprasad Panda (M’99) was born in Doldere, India. He received the B.E. and M.E. degrees from Bengal Engineering College (Presently Bengal Engineering and Science University), Kolkata, India, and the Ph.D. degree from the Indian Institute of Science, Bangalore, India. During 2000–2003, he was a Postdoctoral Fellow with Wisconsin Electric Machines and Power Electronics Consortia, University of Wisconsin-Madison, Madison. From 2003 to 2006, he held a Senior Principal Engineer position with the Rockwell Automation Advanced Technology Laboratory, Richmond Heights, OH. Thereafter, he joined Trans-Coil International (TCI), LLC., Milwaukee, WI, as the Senior Principal Engineer for the Electronic Product Division. He was instrumental in the design and development of active filter product for TCI, LLC. After serving TCI, LLC for almost six years he has very recently joined with GE Aviation Systems at Dayton, OH. He has authored more than 20 peer reviewed articles in various IEEE and equivalent international journals and conference proceedings. His interests include electronic power conversion, variable-speed drives, real-time hardware in loop modeling, energy efficient systems, and system integration. 2436 Thomas A. Lipo (M’64–SM’71–F’87–LF’05) was born in Milwaukee, WI. He has spent his career in the technical field of ac motor drives. From 1969 to 1979, he was an electrical engineer in the Power Electronics Laboratory, Corporate Research and Development, General Electric Company, Schenectady, NY, where he participated in some of the earliest work in this field. In 1981, he joined the University of Wisconsin-Madison, Madison, where he co-founded the industry consortium Wisconsin Electric Machines and Power Electronics Consortia and served for 28 years as its Co-Director and as the W. W. Grainger Professor for power electronics and electrical machines. He is currently an Emeritus Professor at University of Wisconsin-Madison and WCU Chair Professor at Hanyang University, Ansan, Korea. Dr. Lipo has received the Outstanding Achievement Award from the IEEE Industry Applications Society in 1986, the William E. Newell Award of the IEEE Power Electronics Society in 1990, and the Nicola Tesla IEEE Field Award from the IEEE Power Engineering Society in 1995 for his work. In 2002, he was elected as a Fellow of the Royal Academy of Engineering (U.K.) and in 2008 a Fellow of the U.S. National Academy of Engineering. In 2004, he was the recipient of the Hilldale Award in Physical Sciences from the University of Wisconsin, the most prestigious award given by the university for scientific research. IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 28, NO. 5, MAY 2013 Di Pan (S’08) received the B.S. degree from Zhejiang University, Zhejiang, China, in 2007, and the M.S. degree from University of WisconsinMadison, Madison, in 2009, both in electrical engineering. He is currently working toward the Ph.D. degree at the University of Wisconsin-Madison. His research interests include modeling, design, and control of adjustable ac drives, and power electronics converters.