Download Open-Winding Power Conversion Systems Fed by Half

Research Report
2012-39
Open-Winding Power Conversion Systems Fed
by Half-Controlled Converters
Y. Wang, D. Panda, T.A. Lipo, D. Pan
Dept. of Elect. & Comp. Engr.
University of Wisconsin-Madison
1415 Engineering Drive
Madison, WI 53706
University of Wisconsin-Madison
College of Engineering
Wisconsin Power Electronics Research Center
2559D Engineering Hall
1415 Engineering Drive
Madison WI 53706-1691
© Confidential
IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 28, NO. 5, MAY 2013
2427
Open-Winding Power Conversion Systems Fed
by Half-Controlled Converters
Yang Wang, Student Member, IEEE, Debiprasad Panda, Member, IEEE, Thomas A. Lipo, Life Fellow, IEEE,
and Di Pan, Student Member, IEEE
Abstract—An open-winding power conversion system with two
conventional six-switch voltage-source converters (VSC) affords
operation with a lower volt–ampere (VA) rating of each device for
a given power rating, as well as a degree of fault tolerance. The
disadvantages of such a configuration include a higher total device VA rating and increased conduction loss as compared to a
single Y/Δ-connected VSC. In certain ac–dc applications, such as
telecommunications, wind, and aerospace generator drives, it is
desired to have regulated input currents and output voltage, but
regenerative operation is either not required or prohibited. For
such applications, an alternative open-winding power converter is
proposed in this paper where half-controlled converter (HCC) is
employed at each end of an open-winding structure. The resultant
total switch count and total VA rating are reduced by half, compared to using full bridges. Besides, the basic advantages of an
open-winding configuration, use of HCCs also guarantees immunity to dc-bus shootthrough and simplifies the gate drive circuit.
The total VA rating of the proposed topology is found to be 42% less
than a six-switch VSC. The operating principle, control method,
and analysis with simulation and experimental results of the proposed topology in both a grid-tied rectifier application and a PM
generator application are illustrated in this paper.
Index Terms—Common mode (CM), half-controlled converter,
open winding.
I. INTRODUCTION
HE conceptual structure of a typical open-winding power
converter is shown in Fig. 1. A variety of realizations,
control schemes, and modulation techniques related to this system have been discussed in the literature [1]–[16]. In particular,
different types of converters can be adopted [1]–[7], and there
are different combinations of dc voltage levels if two isolated
dc-links are adopted [8], [9]. Several unique applications of
open-winding systems have been presented in [10]–[12]. In related papers, it was determined that if a common dc-link is used
in an open-winding system, then a common-mode (CM) path
is present in the circuit, and means must be taken to suppress
T
Manuscript received June 3, 2012; revised July 30, 2012; accepted September 5, 2012. Date of current version November 22, 2012. Recommended for
publication by Associate Editor A. Muetze.
Y. Wang is with United Technologies Research Center, University of
Wisconsin-Madison, East Hartford, CT 06108 USA (e-mail: wangyangnb@
gmail.com).
D. Panda was with the Trans-Coil International LLC, Milwaukee, WI 53224
USA. He is now with GE Aviation Systems, Vandalia, OH 45431 USA (e-mail:
[email protected]).
T. A. Lipo and D. Pan are with University of Wisconsin-Madison, Madison,
WI 53706 USA (e-mail: [email protected]; [email protected]).
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TPEL.2012.2218259
Fig. 1.
Conceptual structure of a typical open-winding system.
the CM current [5], [7], [16]. The main advantage of an openwinding topology is the lower VA rating of each individual
device for a given overall power rating. Another advantage of
the open-winding configuration is its fault tolerance [17]–[20].
The disadvantages of an open-winding full-bridge configuration
include a higher total device VA rating and increased conduction loss as compared to the conventional closed-winding full
bridge.
Three-phase current-source and voltage-source halfcontrolled-converters (HCCs) were introduced in [21] and [22],
[23] respectively, and their potential as a lower VA rating faulttolerant controlled rectifier was explored in [24]–[28]. The HCC
also has found application as open-winding permanent magnet
synchronous generator (PMSG) drives [29]. A system with a
HCC-fed open-winding configuration combines some features
of both HCC and open-winding structure, and it eliminates some
disadvantages of existing open-winding systems. The apparent drawback of any HCC-based system is that it does not
support regenerative operation [24]–[28]. However, in certain
ac–dc applications, such as wind turbines [29], telecommunication equipment [30]–[33], and aerospace applications [34]
regenerative operation is either not required [29]–[33] or prohibited [34], in which case circuits with reduced VA rating
become viable [29]–[34]. For such applications, a new openwinding system with HCC is proposed in this paper to replace
the existing full-bridge rectifier. This new solution will provide
acceptable input harmonic distortion, regulated output power,
better fault-tolerance, and the system cost will be less since the
converter VA rating will be significantly less than a full-bridge.
Other benefits include being free of shootthrough and possessing a relatively simple gate drive circuitry.
This paper explores the implementation of the proposed topology for two typical ac–dc systems, i.e., 1) an open-winding
transformer rectifier; and 2) an open-winding PMSG drive. After serving TCI, LLC for almost six years, he has very recently joined with GE Aviation Vandalia. He has authored more
than 20 peer reviewed articles in various IEEE and equivalent
international journals and conference proceedings. His interests include electronic power conversion, variable-speed drives,
0885-8993/$31.00 © 2012 IEEE
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Fig. 2.
IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 28, NO. 5, MAY 2013
Generalized HCC-fed open-winding power-transfer system.
Fig. 4.
Examples of current distortion.
Fig. 3. Per-phase equivalent circuit of the HCC-fed open-winding powertransfer system with a common dc-link.
real-time hardware in loop modeling, energy efficient systems,
and system integration.
II. HCC-FED OPEN-WINDING POWER TRANSFER SYSTEM
A generalized HCC-fed open-winding power converter is
shown in Fig. 2, representing several typical realizations. While
not necessary, a common dc-link is adopted in this paper, leading to benefits such as minimized dc-link voltage ripple and
reduced system size and component count. The drawback is the
existence of a CM path, which requires that the CM current be
minimized.
The operating principles of the HCC-fed open-winding system are explained using the per-phase equivalent circuit shown
in Fig. 3.
Here, positive current i flowing through D4 is controlled by
S3 . If i is reversed, then switch S4 regulates the current through
D3 . The voltage modulation signal and the phase current must
have the same polarity in order to guarantee fully controlled
phase current, and a failure to meet this requirement results in
current distortion. Two examples are illustrated in Fig. 4, where
es is the phase back EMF, vcm d1 is the desired phase-voltage
modulation, vcm d2 is the achievable phase-voltage modulation
for, icm d is the desired phase current, and is is the achievable
phase current with minimum distortion. Back EMF and voltage
waveforms are scaled by 50% for a clearer view. Fig. 4(a) shows
the case where icm d lags vcm d1 . Immediately after vcm d1 crosses
zero, icm d and vcm d1 have opposite polarities, so that vcm d1
cannot be synthesized, causing distortion in is . The best that
the HCCs can do at that moment is to maintain 0V across the
ac terminals until icm d crosses zero, as highlighted by vcm d2 .
The situation of icm d leading vcm d1 can be derived similarly
and the leading displacement may result in more distortion than
with a lagging condition. The above discussion reveals that
Fig. 5. All possible voltage vectors of an HCC-fed open-winding system.
Highlighted is the available voltage vector space for current vectors located
in 1.
the modulated voltage and current must have same polarity to
guarantee full controllability. Due to the need for unidirectional
power flow, all the existing CM suppressing techniques reported
in [5], [7], [16] do not apply, and a new means for low-frequency
CM compensation is required.
III. VOLTAGE VECTORS AND VOLT–SECONDS CONSTRAINTS
All possible voltage vectors of the proposed HCC-fed openwinding system are shown in Fig. 5 as black dots, which resemble the voltage vector plane of a three-level VSC. Note
that not every voltage vector is available at each instant. For
example, if the current vector falls in 1 (ia > 0, ib < 0, ic < 0),
the available voltage vector is limited to the shaded area. State
(1 0 0–0 0 0) designates that HCC1 resides at voltage vector
(1 0 0) and HCC2 resides at (0 0 0). To minimize the CM current, CM voltage must be eliminated over each PWM cycle.
For this purpose, define the two available switching states for
WANG et al.: OPEN-WINDING POWER CONVERSION SYSTEMS FED BY HALF-CONTROLLED CONVERTERS
Fig. 6.
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Example of switching sequence with minimum switching loss.
HCC1 as S1 = (1 0 0) and S2 = (0 0 0) with dwell times T1
and T2 , respectively, and define the four available switching
states of HCC2 as S3 = (0 1 1), S4 = (0 1 0), S5 = (0 0 1),
and S6 = (0 0 0) with dwell times T3−T6 , respectively. The
modulation for a given voltage vector V α β (in the stationary
reference frame) is shown in (1), where Ts is the PWM switching period. The CM component is eliminated over each PWM
cycle
⎧
2
2
1
1
⎪
T1 + T3 + T4 + T5 Vdc = Vα
⎪
⎪
⎪
3
3
3
3
⎪
⎪
⎪
√
√
⎪
⎪
⎪
3
3
⎪
⎨
T5 −
T4 Vdc = Vβ
3
3
(1)
⎪
⎪
⎪
⎪
T1 − (T3 + T4 ) − (T3 + T5 ) = 0 (zero CM voltage)
⎪
⎪
⎪
⎪
⎪
T1 + T2 = Ts
⎪
⎪
⎩
T3 + T4 + T5 + T6 = Ts .
The set of equations in (1) can be solved as (2)–(3)
yields
Vα
3Vα
− 2T1 = 2T3 + T4 + T5 = T1 −−→ T1 =
Vdc
Vdc
T2 = Ts − T1 = Ts −
Vα
.
Vdc
(2)
(3)
While there is not a unique solution to T3−T6 , they can
be written as functions of one variable, e.g., T6 , as shown
in (4)
⎧
Vα
⎪
⎪
T3 (T6 ) =
− Ts + T6
⎪
⎪
V
dc
⎪
⎪
√
⎪
⎨
3 Vβ
1 Vα
(4)
T4 (T6 ) = Ts − T6 −
−
⎪
2
V
2
Vdc
dc
⎪
⎪
√
⎪
⎪
⎪
⎪
⎩ T5 (T6 ) = Ts − T6 − 1 Vα + 3 Vβ .
2 Vdc
2 Vdc
The assignment of T6 can now be used to reduce the switching losses. For example, one switching sequence and the corresponding conduction paths at each state are given in Fig. 6,
which requires minimum switching action. Note the above results only apply to situations where the back EMF does not
have a CM component. Otherwise, it will be desirable to compensate actively using converters in order to minimize the net
zero-sequence voltage.
Fig. 7. D–Q–0 equivalent circuit model of an open-winding power transfer
system, with generator convention.
Fig. 8.
Proposed PWM current regulator.
IV. SYSTEM MODELING AND CONTROLLER DESIGN
The generalized D–Q–0 equivalent circuit model of the openwinding power transfer system is shown in Fig. 7 where a synchronous reference frame and generating convention are used,
and the corresponding state equations are shown in (5)
⎧
d
⎪
⎪
⎪
⎪ v dq 1 − v dq 2 = eq − Rs idq − dt λdq − jωλdq
⎨
λdq = Lq iq − jLd id
(5)
⎪
⎪
⎪
⎪
⎩ Pe = 3 [eq iq − ω(Ld − Lq )iq id ].
2
At steady state, the polarity constraint is essentially equivalent to Id /Iq = Vd /Vq . Neglecting resistance and assuming a
sinusoidal back EMF, this expression can be further reduced to
(6). However, again, (6) may not hold true if a CM component
is present
E − E 2 − 4Xd Xq Iq2
.
0 = Xd Id2 − EId + Xq Iq2 ⇒ Id =
2Xd
(6)
The proposed PWM current regulator is shown in Fig. 8. The
current controller consists of PI regulators in D–Q coordinates
and D–Q decoupling terms. The corresponding low-frequency
CM compensator is shown in Fig. 9. In general, there exist several means of generating current command, e.g., power control
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Fig. 9.
IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 28, NO. 5, MAY 2013
Active CM compensator.
and dc-link voltage control. The final duty cycle commands are
applied to HCCs depending on the current polarity with regard
to the reference direction defined in Fig. 7, as shown in (7):
m1a,1b,1c = ma,b,c , m2a,2b,2c = 0 (if ia,b,c > 0)
(7)
m1a,1b,1c = 0, m2a,2b,2c = −ma,b,c (if ia,b,c < 0)
The active CM compensator shown in Fig. 9 consists of two
major parts, the third harmonic phase-locked loop (PLL), and
the CM current regulator. The output of the compensator v0∗ is
included in the three-phase voltage modulation commands. The
third harmonic PLL is used to detect and lock onto the third
harmonic current that is caused by not only the third harmonic
back EMF, but also the operating quadrant limitation. The basic
principle of the PLL is explained by (8), where i0 is the measured
CM current signal, I3 and θ3 are the amplitude and phase angle
of the third harmonic current, and θ3∗ is the estimated phase
angle of the third harmonic current
Fig. 10. Simulated phase-A current with sinusoidal back EMF: (a) Almost
unity displacement angle without CM compensation. (b) At theoretical MPP
(i.e., unaligned voltage and current vectors) without CM compensation. (c) At
theoretical MPP with CM compensation.
i0 cos(θ3∗ ) ≈ I3 sin(θ3 ) cos(θ3∗ )
I3
[sin(θ3 + θ3∗ ) + sin(θ3 − θ3∗ )].
(8)
2
Since the third harmonic current dominates in the CM
component, it is sufficient to make the approximation that
i0 ≈ I3 sin(θ3 ). The PI controller in the PLL minimizes the
last term of (8), i.e., sin(θ3 − θ3∗ ), so that θ3∗ will successfully
track θ3 . The output of (8) oscillates at six times the fundamental frequency; therefore, the bandwidth of the PLL should be
sufficiently low, or a low-pass filter could be added before the
PI controller.
The other portion of the active CM compensator is the CM
current regulator, which further consists of two parts: the PI controller and the P controller. The principle of the PI compensator
is explained in equation (9):
=
i0 sin(θ3∗ ) ≈ I3 sin(θ3 ) sin(θ3∗ )
=
−I3
[cos(θ3 + θ3∗ ) − cos(θ3 − θ3∗ )].
2
(9)
Fig. 11. Power penalty and phase current THD due to unity displacement, for
various MPP-suggested operating displacement angles, sinusoidal back EMF
assumed.
WANG et al.: OPEN-WINDING POWER CONVERSION SYSTEMS FED BY HALF-CONTROLLED CONVERTERS
2431
The PI controller is used to minimize the last term of (9),
i.e., I3 cos(θ3 − θ3∗ ), and, assuming cos(θ3 − θ3∗ ) = 1, the last
term becomes I3 . The PI output is multiplied by a cosine function and then injected into the PWM duty cycles. Again the
sixth harmonic oscillation in the resultant signal of (9) calls for
careful attention to the gain selection for the PI compensator.
The third harmonic voltage injection from the PI controller
only compensates for the inductive voltage drop, but not for
the third harmonic voltage drop across a resistance. Therefore,
the P controller is necessary, since it compensates for the resistive voltage drop in the CM path. Theoretically, the same CM
compensation technique is also suitable for other measurable
odd harmonics.
V. SIMULATIONS
Fig. 12. Simulation results with 0.09 pu third harmonic in back EMF without
CM compensation: (a) phase current, and (b) CM current.
The simulation results are organized and presented in two
subsections for comparing and contrasting the impact of sinusoidal and nonsinusoidal back EMF, and highlighting the effect
of saliency.
A. Open-Winding Power Transfer System With Sinusoidal
Back EMF
With a sinusoidal back EMF, the polarity restriction is equivalent to forcing a unity displacement factor between the converter
voltage modulation signal and the phase current. Fig. 10 shows
the simulated phase current during three different scenarios,
i.e., (a) at almost unity displacement angle without CM compensation, (b) at the theoretical maximum-power-point MPP
without CM compensation, and (c) at the theoretical MPP with
CM compensation. The D–Q–0 inductances are 0.31, 0.5, and
0.078 pu, respectively. The results presented in Fig. 10 demonstrate the fact that a misalignment of the voltage and current vectors results in severe current distortion. It is also
shown that the CM compensation is very effective for suppressing the current distortion caused by nonunity displacement
factor.
Simulations have been carried out for both nonsalient and
salient systems with different parameters running at two operating conditions: (1) at unity displacement factor point, and
(2) at the theoretical MPP. The summarized results are plotted
in Fig. 11. Fig. 11(a) shows the reduction in power transfer
at unity displacement factor, which is a penalty for achieving
minimum current distortion. It is plotted as a function of the displacement angle between phase current and voltage modulation
signal as suggested by the MPP. The power penalty increases as
the MPP-suggested displacement angle becomes more leading.
The relationship is almost independent of the saliency. Fig. 11(b)
shows that the current THD at the MPP increases as the MPPsuggested displacement angle becomes more leading, regardless
of the saliency. The above results imply that for the HCC-fed
open-winding systems with sinusoidal excitation, it is desirable
to design the system MPP residing close to a unity displacement
factor point.
Fig. 13. Simulation results with 0.09 pu third harmonic in back EMF with
CM compensation: (a) phase current, and (b) CM current.
B. Open-Winding Power Transfer System with Nonsinusoidal
Back EMF
This section focuses on the impact of the third harmonics
in the back EMF, which is a common occurrence in PM machines. Due to the third harmonic, the polarity restriction is
no longer equivalent to unity displacement factor, and seeking the operating point of minimum current distortion is no
longer a straightforward problem. Assuming that 0.09 pu third
harmonic exists in the back EMF and no CM compensation,
significant CM current is induced as shown in Fig. 12. However, if the compensator shown in Fig. 9 is incorporated and
the remainder unchanged, there is immediate improvement as
shown in Fig. 13. These results prove the effectiveness of the CM
compensator.
Fig. 14(a) compares the simulated power output with the ideal
value as the commanded current vector varies, and (b) shows
the trend of the corresponding current THD and the peak-topeak power ripple. As the current angle changes, the simulated
average power throughput follows quite closely the ideal value
(i.e., the power throughput assuming a four-quadrant converter)
except at the MPP, which is a penalty due to limited operating
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 28, NO. 5, MAY 2013
Fig. 15.
Configuration of open-winding transformer rectifier.
Fig. 16.
Configuration of HCC-fed open-winding PMSG system.
Fig. 14. Simulated variations of system performances as functions of commanded current vector angles (i.e., internal power factor angle): (a) Ideal power
throughput (P_est) and simulated power throughput (P_sim). (b) Current THD
and peak-to-peak power ripple.
quadrants of the HCCs. Although the average power throughput
is regulated, the current waveform may actually have severe
distortion as shown in Fig. 14(b). A small current angle results in
a large leading displacement angle between the desired current
and voltage vectors, causing severe distortion, which agrees with
the results illustrated in Fig. 4.
It should be apparent that the best current quality point does
not necessarily reside at the MPP. Although one can always
reduce the zero-sequence current by increasing the CM inductance, a compromise between the power throughput and current
quality usually has to be made. Fortunately, the distortion in the
current does not add much to the power throughput fluctuation
since the back EMF does not have the corresponding harmonic.
VI. EXPERIMENTS
Experiments have been carried out for (1) an HCC-fed openwinding transformer rectifier as shown in Fig. 15, and (2) an
HCC-fed open-winding salient PMSG as shown in Fig. 16. The
experiments of the open-winding transformer rectifier are done
with grid excitation. The main purpose of this part of the experiment was to first demonstrate the functionality of the HCC-fed
open-winding system under pure sinusoidal excitation, and second to show the effects of misalignment of the voltage and
current vectors.
The experiments of the open-winding salient PMSG were
done at fixed speed operation. The main focus is to first investigate the effects of the harmonics in the back EMF, and second
to test the effectiveness of active CM compensator.
Fig. 17. Measured converter phase-A current ia and source current ia s of the
open-winding transformer rectifier: (a) Converter voltage and current in phase.
(b) Current leading converter voltage.
A. Open-Winding Transformer Rectifier
A 1 kW open-winding transformer rectifier was tested in this
part of the experiment. The D–Q current commands were generated via an outer dc-voltage control loop. The parameters of
the hardware setup can be found in the appendix and the results
are presented in Fig. 17. The source voltage contains nontriplen
WANG et al.: OPEN-WINDING POWER CONVERSION SYSTEMS FED BY HALF-CONTROLLED CONVERTERS
Fig. 18. Measured PMSG current, without active third harmonic compensation: (a) Phase currents. (b) CM current. (c) Spectrum of CM current.
odd order harmonics, resulting in the corresponding current
harmonics. A three-phase filter capacitor bank is installed at the
grid side of the transformer. As the displacement angle between
the converter voltage and current increases, the current distortion increases. The results agree very well with the analysis
and simulation and confirm the functionality of the HCC-fed
open-winding system with sinusoidal excitation.
B. Open-Winding PMSG System
A 1.7-kW open-winding PMSG was tested in this part of
the experiment. Parameters of the open-winding PMSG can
be found in the appendix. The back EMF mainly consists of
3.03 V/Hz fundamental, 12.3% third harmonic, and 0.6% fifth
harmonic.
First, the system was tested without the active CM compensator. A 5 mH (∼ 0.09 pu) CM choke was installed in series with
the machine winding for suppressing CM current. The resultant
phase current is subject to zero-clamping effect. Note that due
to the slightly unbalanced common-choke there is small CM
current at fundamental frequency as shown in Fig. 18(c).
Next, the active CM compensator was activated (with all other
conditions unchanged) and the results are plotted in Fig. 19.
Although the CM compensator cannot eliminate the zeroclamping, comparison of the spectrums presented in Figs. 18(c)
and 19(c) shows that the third harmonic has been eliminated.
The experiments were carried out for a series of operating
points and the results are presented in Fig. 20. Fig. 20(a) shows
the estimated and measured power outputs. The symbol “Pdes”
designates the theoretical air-gap power calculated with the commanded current vector and estimated machine parameters. The
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Fig. 19. Measured PMSG current, with active third harmonic compensation:
(a) Phase currents. (b) CM current. (c) Spectrum of CM current.
symbol “P_DSP” is the air-gap power calculated with the measured average D–Q current and estimated machine parameters.
The symbol “P_SG” is the measured average terminal power
output of the PMSG. It is seen that “P_DSP” and “Pdes” match
very well indicating that the average D–Q currents are well
regulated. However, the “P_SG” is lower than the estimated
values, and the measured MPP is somewhat different than the
estimation. This is mainly caused by resistive loss, parameter
estimation errors (including saturation), and possibly negative
torque. The experimental results imply that the real MPP may
differ from the theoretical value. The THD of the machine phase
current and the output power ripple are plotted in Fig. 20(b). The
trends of the measured THD and the power ripple agree with
that shown in Fig. 14(b).
The dynamic response to 25% step changes in the D–Q current command was also demonstrated in the experiments. The
results are shown in Fig. 21 and the traces are power output and
three-phase currents. The system is completely stable through
abrupt transients.
C. Observations and Discussions
The basic operating principles of the HCC-fed open-winding
systems with sinusoidal input voltage have been demonstrated
in the experiments of the open-winding transformer rectifier, which confirm the theoretical analysis. A practical openwinding PMSG possessing both a salient rotor and back EMF
harmonics was also tested with HCCs. Although there is slight
discrepancy between the estimated and measured results due
to real world limitations, the effectiveness and stability of the
proposed controller, as well as the active CM compensator, have
been well demonstrated.
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 28, NO. 5, MAY 2013
tion (50% reduction compared to open-winding structure with
conventional full-bridges) in device VA rating compared to a
single full-bridge VSC with Y/Δ connection, simplified gate
drive circuitry, and immunity to dc-link shootthrough. On the
other hand, the open-winding system may have a small current
distortion near its zero-crossing and the unidirectional power
flow may not always closely follow the MPP operation. The
operating principles of the proposed topology have been investigated in detail. The effects of system saliency and harmonics
in the back EMF were explored in the case of the open-winding
PMSG system. A PWM current regulator was proposed. Since
a common dc-link for both HCCs reduces component cost but
introduces a zero-sequence path, an active CM compensator was
implemented to minimize the low-frequency CM current. The
switching-frequency CM behaviors require future work. This
new concept has been proven effective in both simulations and
experiment work. The proposed system was shown to be stable
through any transient changes in the current commands.
Given the robust and satisfactory performance of HCC-fed
open-winding systems as well as the explained advantages,
the proposed concepts are expected to be suitable for nonregenerative power conversion applications that place emphasis
on both performance and reliability such as in wind generators,
telecommunication equipment and aerospace applications.
Fig. 20. Measured results: (a) steady-state power outputs, and (b) current THD
and power output ripple.
Fig. 21.
Dynamic response to 25% step changes in current commands.
The major benefits of the proposed systems are (1) a reduced
power device VA rating, and (2) an improved reliability due
to shootthrough immunity and much simpler gate drive configuration. The limitations, on the other hand, are mainly the
slight reduction in the power output, an increase in the current harmonics under certain circumstances, and unidirectional
power flow. However, as mentioned previously, unidirectional
power flow is not a drawback in certain applications where reverse power flow is not needed and blocked using additional
diodes.
VII. CONCLUSION
This paper has proposed a HCC-fed open-winding power
converter and demonstrated operation with two experimental
setups where regenerative operation is not required. Apart from
all the reported benefits of open-winding structures, the major benefits of the proposed solution includes a 42% reduc-
APPENDIX
Open-winding transformer rectifier parameters
Source: voltage amplitude 80 V, frequency 60 Hz, input
current amplitude 12 A.
Converter-side inductor: three single-phase 3 mH.
Load: DC-bus voltage 90 V, resistor 8 Ω, load power 1 kW.
PWM frequency: 10 kHz.
Open-winding PMSG and drive parameters
PMSG: fundamental frequency 40 Hz, back-emf
amplitude 120 V, phase current amplitude 8 A.
Inductance: D-axis 19 mH, Q-axis 30 mH, zero-sequence
4 mH.
Load: DC-bus voltage 150 V, load power 1.7 kW.
PWM frequency: 10 kHz.
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Yang Wang (S’08) was born in Ningbo, Zhejiang,
China. He received the B.S. degree from Zhejiang
University, Zhejiang, China, in 2007, and the M.S.
and Ph.D. degrees from Wisconsin Electric Machines and Power Electronics Consortia, University
of Wisconsin–Madison, Madison, both in electrical
engineering, in 2009 and 2011, respectively.
He is currently with United Technologies Research
Center, Hartford, CT. His research interests include
power electronics and machine drives.
Debiprasad Panda (M’99) was born in Doldere,
India. He received the B.E. and M.E. degrees from
Bengal Engineering College (Presently Bengal Engineering and Science University), Kolkata, India, and
the Ph.D. degree from the Indian Institute of Science,
Bangalore, India.
During 2000–2003, he was a Postdoctoral Fellow
with Wisconsin Electric Machines and Power Electronics Consortia, University of Wisconsin-Madison,
Madison. From 2003 to 2006, he held a Senior Principal Engineer position with the Rockwell Automation Advanced Technology Laboratory, Richmond Heights, OH. Thereafter, he
joined Trans-Coil International (TCI), LLC., Milwaukee, WI, as the Senior Principal Engineer for the Electronic Product Division. He was instrumental in the
design and development of active filter product for TCI, LLC. After serving TCI,
LLC for almost six years he has very recently joined with GE Aviation Systems
at Dayton, OH. He has authored more than 20 peer reviewed articles in various
IEEE and equivalent international journals and conference proceedings. His
interests include electronic power conversion, variable-speed drives, real-time
hardware in loop modeling, energy efficient systems, and system integration.
2436
Thomas A. Lipo (M’64–SM’71–F’87–LF’05) was
born in Milwaukee, WI.
He has spent his career in the technical field of ac
motor drives. From 1969 to 1979, he was an electrical
engineer in the Power Electronics Laboratory, Corporate Research and Development, General Electric
Company, Schenectady, NY, where he participated in
some of the earliest work in this field. In 1981, he
joined the University of Wisconsin-Madison, Madison, where he co-founded the industry consortium
Wisconsin Electric Machines and Power Electronics
Consortia and served for 28 years as its Co-Director and as the W. W. Grainger
Professor for power electronics and electrical machines. He is currently an
Emeritus Professor at University of Wisconsin-Madison and WCU Chair Professor at Hanyang University, Ansan, Korea.
Dr. Lipo has received the Outstanding Achievement Award from the IEEE Industry Applications Society in 1986, the William E. Newell Award of the IEEE
Power Electronics Society in 1990, and the Nicola Tesla IEEE Field Award
from the IEEE Power Engineering Society in 1995 for his work. In 2002, he
was elected as a Fellow of the Royal Academy of Engineering (U.K.) and in
2008 a Fellow of the U.S. National Academy of Engineering. In 2004, he was
the recipient of the Hilldale Award in Physical Sciences from the University
of Wisconsin, the most prestigious award given by the university for scientific
research.
IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 28, NO. 5, MAY 2013
Di Pan (S’08) received the B.S. degree from
Zhejiang University, Zhejiang, China, in 2007, and
the M.S. degree from University of WisconsinMadison, Madison, in 2009, both in electrical engineering. He is currently working toward the Ph.D.
degree at the University of Wisconsin-Madison.
His research interests include modeling, design,
and control of adjustable ac drives, and power electronics converters.