Download Wireless Power and Data Transfer

Wireless Power and Data Transfer
for
Sonar Array Applications
Final Report for the Period Jan.-Dec. 2003
By:
Ricardo M. Silva
Advised by:
Dr. Rajeev Bansal (Univ. of Connecticut)
Mr. Michael Sullivan (Electric Boat)
Sponsored by:
Electric Boat
Lockheed Martin
In cooperation with:
EDO
NUWC
Table of Contents
Topic
Page Number
Executive Summary
3
Introduction
4
Waveguide Layout
6
Waveguide Terminations
11
Power Accounting
22
Future research
35
Conclusion
35
References
36
Acknowledgements
37
2
Executive Summary
This project will identify an efficient method to power multiple hydrophones wirelessly.
In this research, the inside of an aluminum waveguide is energized with continuous wave, CW
(unmodulated), radio frequency (RF) signals. Power will be extracted from the waveguide with
stub monopole antennas located along the length of the waveguide. The CW RF will be
converted to direct current voltage (VDC) via a rectifying antenna (rectenna) which is composed
of an impedance matching circuit, a shunt diode, and a low pass filter. The VDC will be used to
power the hydrophone and data telemetry circuitry. The sound acquired from each hydrophone
will be processed and sent wirelessly to another location in the waveguide for further processing.
It is shown that the corporate/parallel layout is a low-cost method to distribute power to
various staves in the array. Also, several schemes which will increase the number of elements on
each stave have been discussed.
To prevent standing waves, it is necessary to terminate a waveguide with a
multilayered/tapered material. A sample of material from ARC-Tech has been able to lower the
SWR to about 1.17. It is a wedge fabricated from ML-79 material.
By using high-speed diodes it is possible to increase the efficiency of a rectenna. These
diodes should have a low junction capacitance, Cjo, and a low series resistance, Rs. To handle
large power fluctuations, a diode with a high reverse breakdown voltage, Vbr, should be used.
The Skyworks SMS3924 diode with a Vbr of 70 Volts, Rs of 11 ohms, and Cjo of 1.5pF has
been chosen as a good candidate for this project.
3
Introduction
This annual report will update the research on wireless power and data transfer, being
conducted at the University of Connecticut. The report will provide possible schemes for
extending the current research into a large scale design with the possibility of hundreds of
hydrophone elements.
The specific issues addressed in this report are:
1)
Waveguide Layout. This portion of the report will give possible solutions on
how to distribute power to multiple elements.
2)
Waveguide Terminations. These may be necessary in order to prevent
standing waves in the waveguide
3)
Power Accounting. This is necessary in order to properly and safely power
each hydrophone-element.
As a quick refresher, this project will identify an efficient way to power multiple
hydrophone elements wirelessly. The proposed research consists of illuminating the inside of an
aluminum waveguide with continuous wave, CW (unmodulated), radio frequency (RF) signals.
The TE10 mode will be excited in the waveguide. This mode was selected due to its simpler
power density distribution along the x-axis of the waveguide (Fig. 1). Power from the
waveguide will be extracted by stub monopole antennas located in the center of the waveguide.
The CW RF will be converted to direct current voltage (VDC) via a rectifying antenna (rectenna)
to power the audio pre-amplifier and the data telemetry circuitry. The acoustical data acquired
from each hydrophone will be digitized and it will be sent wirelessly to a receive antenna via the
waveguide.
4
A simple graphical representation of the setup is shown below:
HYDROPHONE/TELEMETRY
6
6
x
.
RAM
z
48
z=L
z=0
POLYETHYLENE
DIELECTRIC
3
6
900 MHz Stub
ANTENNA
(DATA XMIT.)
SIDE VIEW
1 GHz ¼ wave
1 GHz Stub MONOPOLE
ANTENNA ANTENNA
(POWER
(POWER XMIT.)
RCV.)
y
x
Fig. 1 Simplified Stave Layout
5
1)
Waveguide Layout
The current design consists of a small waveguide with 4 elements, where each element is
composed of a rectenna (power supply), hydrophone, audio pre-amplifier, and wireless telemetry
circuitry. With this setup it is easy to assign a different RF frequency to each element in order to
have multiple channels operating simultaneously. The current prototype uses the TI TRF6901
demo board which has 16 channels preconfigured to be tuned from 902-917 MHz, with a 1 MHz
channel separation. The TRF 6901is capable of transmitting at a maximum baud rate of 96 kbps
and its RF spectrum is displayed below:
Fig. 2 TRF6901 RF Spectrum (Ref. 13)
The total bandwidth of the channel, BWC, required to prevent cross-modulation is
approximately 200 kHz. To increase the signal to noise ratio, the bandwidth has been increased
6
to 1 MHz. The increase to 1 MHz is necessary to avoid interference from spurious frequencies
which occur at the center frequency plus 350 kHz and at the center frequency minus 350 kHz.
Given the channel separation of 1 MHz for the TRF6901 and a need for 400 elements, a
bandwidth of 400 MHz will be required to accommodate the data telemetry. One may argue that
there are more “economical” modulation schemes which may provide more channels, such as
Single-Side-Band (SSB), but they also require more elaborate circuitry and circuitry demands
more power consumption. Therefore, frequency management becomes important as the number
of channels increases.
Also, the CW power density inside the waveguide decreases along the z-axis as each
element extracts a small amount of power from the waveguide. This implies that the elements
near the CW transmitting antenna in the waveguide, z=0, will have to cope with a very large
electric field while elements near the end of the waveguide, z=L, will see a much smaller electric
field.
These two problems, frequency and power management, have led to the explorations of
different methods that reuse RF channels and allow the electric field to be much smaller at z=0 .
The proposed concept is similar to what the cellular telephone companies use by dividing a very
big array into smaller cell arrays. By having a smaller number of elements per cell, say 50, it
becomes easier to manage both the power and the frequency assignments. Such an
implementation is envisioned below:
7
Hull Penetrator
Hull
Cell
Array
Staves
Fig. 3 Hull penetrator and cell arrays
Each cell array would then be made with one of the following architectures:
A) Serpentine/Series Feed (Ref. 2, page 478)
TX
Ant.
Fig. 4 Serpentine (series) architecture
With the serpentine configuration, a very long waveguide is folded at various lengths.
The CW power is injected at one end of the waveguide (fig.4). It then travels down the first
stave where it smoothly couples to the beginning of the next stave as long as the angle of the
bend is not too steep. If the angle of the bend is too steep, it will cause some of the incoming
CW RF to reflect back to the source antenna. The radius (r) of the bend must be large,
approximately greater than 1.5 * wavelength in W/G (Ref. 3 page 39).
8
CW RF
r
Fig. 5 Serpentine bend radius
B) Manifold/Space Feed (Ref. 4, page 166)
Fig. 6 Manifold /Space Feed
This design consists of a small piece of waveguide which is then coupled to multiple n
waveguides by allowing the CW RF to split between n waveguides. The transition between the
first short waveguide and the manifold should be gradual to prevent the first waveguide from
behaving like an open ended waveguide which in turn would lead to high standing waves in the
first waveguide.
9
C) Corporate/Parallel Feed (Ref. 4, page 166)
Fig. 7 Corporate/Parallel Feed
With this configuration, the power is split in a simple power divider with the output of
the power divider being fed to each stave. The power divider will provide impedance matching
between the 1 to n waveguides and it may be built with microstrip technology or regular off-theshelf coaxial power dividers.
This design is the easiest to build and it is also one of the easiest to analyze, and unlike
the serpentine design, this design allow the power density at z=0 to be substantially lower than
with z=0 for the serpentine design with the same number of elements. Additionally, due to the
small size of the power distribution device, the staves can be accommodated side by side without
any decrease in performance.
Overall, this design seems to be the most efficient and the most practical of the three. As
a result, this design looks like a very feasible solution for future multi-stave arrays.
10
2)
Waveguide Terminations
When an electromagnetic wave travels in a waveguide, it has an e-jβz behavior with β
being the phase constant and z being distance along the axis of propagation. Therefore, the
power density along the z axis remains constant. If the end of the waveguide is shorted (no
termination) the traveling wave will reflect off the end and it will create standing waves with a
sin(βz) property, leading to localized voltage maxima and voltage minima.
¼ wave Exciting Antenna
z=0
Voltage Minima
Voltage Maxima
z=L
Waveguide Side View
Fig. 8 Waveguide with standing waves
Ideally, the power density would decrease as a function of z because each element would
extract a small amount of power from the waveguide and eventually the last element would
extract the remaining power in the waveguide. This would also imply that the elements working
at the very end of the waveguide would have to be designed for much smaller power levels in
order to operate properly.
To simplify this project and to maintain symmetry amongst the rectennas, it is important
to minimize/cancel any standing waves in the waveguide. Again, the lack of standing waves in
the waveguide will represent a constant power density along the z-axis and it will facilitate the
positioning of the rectenna devices.
11
By inserting a termination at the end of each waveguide, reflected waves will be greatly
attenuated with a reduction of standing waves. Terminations are either lossy or resonant. In
lossy type terminations, energy is given off in the form of heat. When properly matched
(impedance) the reflected waves from the lossy material will be greatly attenuated. With resonant
type terminations, the thickness of the material is made to be a suitable fraction of the incoming
wavelength. A portion of the signal is reflected at the face of the material and another portion is
transmitted into the material. At the right thickness, the resonant material will transmit a wave
which is 180 degrees out of phase with the face reflected wave. The two waves cancel each other
and it appears as if the material prevents EM waves from reflecting off the end of the waveguide.
A) Conical Rod
l2
l1
Fig. 9, Conical Rod (Ref. 5 page 316)

The conical rod is made of a lossy material.

If l1 is greater than a couple of wavelengths, the SWR could be less than 1.04

The length l2 is adjusted to provide a total absorption greater than 20 dB

Low power applications

Broadband, bandwidth of about 40%
12
B) High Power Load
l1
l2
Top View
Side View
Fig. 10, High Power Load (Ref. 5 page 316)

The termination is made of a lossy material.

l1 is greater than a couple of wavelengths

The length l2 is adjusted to provide a total absorption greater than 20 dB

High power applications

Broadband, bandwidth of about 40%
C) Water Load
l1
Glass Rod with
Circulating Water
Top View
Side View
Fig. 11, Water Load (Ref. 5 page 316)
13

The termination is made of a glass rod with circulating water.

l1 is greater than a couple of wavelengths

By measuring the water temperature rise, power in the waveguide can be calculated.

Incident power is dissipated by the circulating water
D) Step Load
Lightly Doped Lossy
Material
l1
Heavily Doped
Lossy Material
l2
Top View
b1
Side View
Fig. 12, Step Load (Ref. 5 page 316)

The termination is made of a lossy material.

l1 acts as a ¼ wave transformer

l1 is determined experimentally

The length l2 is adjusted to provide a total absorption greater than 20 dB

Bandwidth of about 10% with a SWR < 1.1.

Compact
14
E) Resonant Tile
d
Incident
Wave
Reflected
Waves
R1
R2
Top View
¼ Wave
Layer
Side View
Metal
Fig. 13, Resonant Tile (Ref. 6)

The ¼ wave layer produces an emergent wave (R2) which cancels out the face
reflected wave (R1).

Depending on the dielectric constant of the ¼ wave layer, d may be very thin.

Reflections are 20 dB lower than normal incident waves

Reflections increase as the angle from normal incidence increases.

Very compact
F) Multilayer
Lightly Doped Side
l1
Heavily Doped Side
Top View
Side View
Fig. 14, Multilayer (Ref. 7)
15

The termination is made of a lossy material with a very low concentration at the
wave/material interface and a gradual increase in concentration towards the back of
the material.

l1 is a function of how much absorption is required

l1 is determined experimentally/look up tables

Compact
G) Physical Tapering
l1
Evenly Doped
Top View
Side View
Fig. 15, Physical Tappering (Ref. 7)

The termination is made of an evenly lossy material.

l1 should be greater than the wavelength to provide a smooth impedance change for
the incoming wave.

l1 may be increased to reduce the SWR
16
H) ¼ Wave Antenna w/Load
¼ Wave monopole antenna with
resistor
¼Wave
Top View
Side View
Fig. 16, ¼ Wave Antenna w/Load

Energy is coupled out of the W/G via the ¼ monopole and it is dissipated in the
resistor

Inexpensive

Simple to build
Experimental data has been obtained for different waveguide terminations. Localized
power measurements were obtained by using a waveguide with a perforated top. The SWR was
then calculated with the following test setup:
17
Function
Generator
Pout = 10dBm
@ 1004 MHz
Power
Meter
Pr
Sample
Directional
Pf
Coupler
Sample
Waveguide
Spectrum
Analyzer
Fig. 17 Test Setup for measuring forward power (Pf) and reflected power (Pr).
Pick-up
probe
consists of
a 1.5cm
antenna
RAM
under test
Power
Meter
Holes along the zaxis to allow local
power measurements
Waveguide –
Top View
Function
Generator
Pout = 10dBm
@ 1004 MHz
Directional
Coupler
Pf
Sample
Spectrum
Analyzer
Fig. 18 Test setup for measuring power along the W/G z-axis.
18
The data collected, suggested that standing waves were present inside of the waveguide.
A quick check was made to determine if the power fluctuations along the W/G z-axis separation
occurred at a distance predicted by theory. The following was performed:
g 
2
z
 m   n 
    
 

 a   b 
2
 z TE
2
2
TE10
m=1
n=0
ω= 2πf = 2π1GHz
μ=4*π*10-7 H/m (air)
ε=8.854*10-12 F/m (air)
a= 8” = 20.32 cm
b= 4”= 10.16 cm
So,
 1 
)
  14.15rads / m
 0.2032 
2
 z TE  (2 *  *1GHz ) (4 *  *10 )(8.854 *10
2
g 
7
12
2
= 0.444m
14.15
Therefore, there should be a repetition of maxima and minima every ½ λg = 0.222m. By
analyzing the Excel data, one can see that the maximum power values appear every 22 cm to
every 23cm.
Since the experimental data coincides with theoretical data, it is now safe to say that there
is a standing wave present in the waveguide. Please see chart below:
19
Power Distribution in a Terminated W/G
40.00
Noise Floor is
at 3.6 uW
None/Short
ARC Wedge RAM
35.00
Dense Urethane
ARC Flat
microWatts
30.00
25.00
20.00
RAM
Material
on this side
TX antenna
here
122.3
15.00
10.00
5.00
0.00
0.00
Noise Floor
20.00
40.00
60.00 cm
80.00
100.00
120.00
Fig. 19, Power Fluctuations in a Waveguide
The SWR based on the z-axis power measurements was determined from Fig. 19. One of
the power measurements for the shorted waveguide sits at the noise floor (lowest possible
reading by the power meter), Pmin =0. This means that there is no measurable power at this
location and if we compute the SWR as a ratio of Emax/Emin = (sqrt(Pmax)*k) / (sqrt(Pmin)*k),
where k is a constant (let k=1), then we can see that the SWR is equal to infinity for when the
waveguide is shorted.
20
The following SWR values were then computed:
Material
SWR
Short
ARC Wedge
Dense Urethane
ARC Perpendicular
31622
1.93
2.75
16788
(no termination)
(resonant type material in a triangular shape similar to fig. 14)
(lossy material in a triangular shape similar to fig. 14)
(setup similar to fig. 13)
Based on the experimental data obtained, the ARC wedge has provided decent results so
far. ARC technologies (Ref. 7) has since recommended a combination of Fig. 14 and Fig. 15
(Multilayer and Physical tapering) to further reduce the SWR to values close to 1.1-1.3. A
vertical-wedge shaped termination was fabricated from ARC ML-79 multi-layer foam and the
following results were obtained:
Power Distribution in a Terminated W/G
30.00
25.00
None/Short
ARC ML-79 Wedge
microWatts
20.00
15.00
RAM Material
on this side
TX antenna
here
122.3
10.00
5.00
Noise Floor is at
3.6 uW
0.00
0.00
20.00
40.00
60.00 cm
80.00
100.00
120.00
Fig. 20, Power Fluctuations with ARC ML-79 RAM
The corresponding SWR was then computed to a very good value of 1.17. Based on
these results, future terminations will be manufactured with this material.
21
3)
Power Accounting.
Power accounting is necessary in order to properly and safely power each hydrophone-
element. It should be done in an efficient manner to maximize the number of sensors per
available power.
A) Sensor power requirements
The sensor will be composed of a hydrophone which produces an audio output in the
order of a couple of millivolts. This output will be amplified by an OP-AMP, LMV751, which
consumes 20 mW. The output of the LMV751 will be fed to the TI-Demo Board. From the
Texas Instruments documentation, (Ref. 1), the board should consume approximately 79 mW
when transmitting at a 0 dBm level. The TI board consists of an RF module and an analog to
digital converter (ADC). The total power required will be approx. 100 mW, and to allow a
margin of safety, assume that the total power required is 150mW. This power will be supplied
through a switching regulator, which based on past work, will operate at 50% efficiency while
regulating a voltage. Higher efficiencies maybe achieved if there is a small difference between
the input and the output voltages. Based on these values, the required input power to the voltage
regulator will have to be around 300 mW. With proper selection, the rectenna is expected to
have an RF to DC conversion efficiency greater than 30%. Again, to be safe, assume that the
rectenna has an efficiency of 30%. Based on these assumptions, the input power to the sensor
will have to be in the order of 1 Watt. Please see Fig. 21 below:
22
Rectenna
Antenna
1W
Impedance
Matching
+
Filter
Network
300 mW
Diode
Desired
effic. >30%
150 mW
Switch.
Regulator
50 % effic.
w/large I/O
Voltage
Difference
Load
3V
TI-Board
LMV751
Fig. 21, Power Requirements for Sensor
B) Rectenna efficiency
One of the key parts of this project is the development of an efficient rectenna. The
rectenna receives RF energy (antenna) and converts it to DC via a shunt diode. The diode
configuration is similar to a diode clamper circuit, where it takes the input waveform and it shifts
it down to a level where the diode just barely turns on. If the rectenna didn’t have a load, the
input waveform would shift down to a point where the diode would just barely turn on (Fig.22A). But with the introduction of a load, the input waveform turns the diode on for longer periods
of time. Please see the pictures below:
Fig. 22, A) No Load
B)Rectenna voltage and input wave w/load (Ref. 8)
23
V = input wave
Vf= diode turn-on voltage
Vd= voltage across the diode
 = Phase when diode is turned on, function of load value
In Fig. 22-B, an input waveform is represented by a full sinusoid with a DC reference
shifted down to a level represented by a dashed line. This signal would be present only if the
diode didn’t exist in the circuit. A second waveform is then superimposed to represent the diode
switching action in the circuit and it is represented by the sinusoid with a chopped upper half.
The missing upper half, θ, is due to the amount of charge being discharged by the load and with
an increase in the load there will be an increase in θ. Ideally, with no load, the input waveform
would be completely shifted to the dashed line and θ would be very small. As can be seen from
Fig. 22-B, the diode almost sees a peak-peak input waveform while it is turned off (reverse-bias).
This leads to one dilemma; it isn’t possible to simply raise the input voltage in order to increase
the output power. There is a point when the input voltage will exceed the reverse breakdown
voltage (Vbr) and the diode will begin to conduct while reverse biased. If the input voltage
continues to increase, the diode will eventually suffer a catastrophic breakdown in its junction
layer. As a general rule of thumb developed by McSpadden (Ref. 9), the output DC voltage
should be less than Vbr / (2.2) to avoid operating the diode at its breakdown points. McSpadden
has also developed a formula which takes into account the max. output power of a rectenna as a
function of Vbr and as a function of the load resistance (RL) , PDC 
Vbr 2
, from the fact that, P
4 RL
= V2 / R, and that the output DC voltage should be ½ Vbr. Without a load, the diode will see
approx. 2*Vpeak which should be less than the Vbr since without a load, the charge held in the
capacitor will not discharge rapidly and the “Phase-on” (Fig. 22-B) will be very small, shifting
the Vd signal on Fig. 22-B further down.
24
Assuming 100% efficiency, PDC_Load =Pin, but to be conservative, it is assumed that the
conversion efficiency of the rectenna is only 30%, so Pin=3.3* PDC_Load. From Fig. 19, RL for the
rectenna is the switching regulator and it is very roughly estimated to be
RL 
5.992V ( from _ previous _ measurements)
 120 (the resistance of the switching regulator
300mW
is very dynamic). With this value for RL, Vbr is then calculated by
Vbr  4*120*300mW  12V (minimum) for a 100% efficiency, which means that the input
voltage would be ½ of Vbr = 6V(Max). But since the rectenna is only 30% efficient, the input
(6V * k ) 2 Solving _ for _ k
1W *120

 k  1.83 , therefore,
voltage will have to be scaled by 1W 
120
6V
the input voltage should be a 1.83*6V=11V and the Vbr of the diode has to be at least
2*11V=22V. Again, for safe operation of the diode, the voltage across the load, VL, will have to
½ of the diode’s reverse breakdown voltage, Vbr. Assuming that the antenna is properly
matched to the load via an impedance matching network, it is possible to estimate the required
antenna impedance for maximum power transfer by RA 
Voc 2
(2*11) 2

 60 based on the
8* Pin 8*1Watt
following circuit:
ZA=ZL*
Vant
22 V
ZL
Fig. 23, Circuit for Antenna Impedance
25
This presents a new problem; the ideal antenna resistance would have to be about 60
ohms which would indicate that the antenna length would be approx. ½ wavelength. But since it
is desired to minimize the antenna length to prevent mutual coupling between sensors, the
antenna length is going to be approximately 1/10 of the wavelength or ¾”. At this length, the
antenna will behave highly capacitive and the real part of the impedance will be determined by
2
Rr _ monopole
 2* lmonopole 
1
1
2  2*1 
 20 2 
  20 
  3.95 (Ref. 14 p.51).
2

2
 10 


2
One may argue that one should operate the diode at very low input power levels in order
to keep input voltages much lower than Vbr, but at low power levels, the input voltage is low and
the diode turn-on voltage (Vf) is very large compared with the input. The ratio of Vf/Vinput is
large and losses across the diode will be large compared with the amount of available rectified
power. Therefore, the rectenna will operate very inefficiently at low power levels.
The figure below indicates optimal input power levels to optimize efficiency:
Fig. 24, Optimal Power Levels (Ref. 8)
26
Additional items should be considered in the future to make the rectennas more efficient.
In the past, experimental work was done in the lab with a commercially available 1N5711 Si
Schottky diode. The parameters for this diode are:
Fig. 25, 1N5711 Diode Specifications (Ref. 10)
Based on the specifications for this diode, and from McSpadden’s work (Ref. 9), the
maximum operating frequency for this diode is
f c _ diode

1
2 *  * Rs * Cjo

1
2 *  * 25 *1.6 *10
12
 3.96GHz . According to McSpadden, the fc_diode for a
rectenna diode should be at least 10*frequency of operation, fop, to allow the diode to rectify
properly and to have good conversion efficiencies. The 10* fop factor is due to the fact that it
takes an RC network 5 time constants to charge and 5 time constants to discharge. So the
operating frequency should be at least 10 times slower than the fc_diode. This is one of the reasons
why the recorded conversion efficiencies for this diode haven’t exceeded 35% since the fc_diode1N5711=3.96
GHz and the frequency of operation was 1 GHz.
In order to have high efficiencies, the Rs*Cjo should be a small number, and with a
frequency of 1 GHz, Rs*Cjo should be less then 159*10-12Ω-F.
27
GaAs diodes should also be used instead of Si diodes due to the higher electron mobility
in GaAs. Essentially, GaAs diodes are inherently much faster than Si diodes, but unfortunately,
most commercially available GaAs diodes have a very low Vbr in the order of 10-25 Volts due
to the lower E field breakdown voltage of GaAs.
For good thermal stability of the rectifying diodes, it should be essential that they operate
at non-scorching temperatures in order to preserve proper rectification of the input signal.
Diodes with ceramic bodies would be ideal instead of the glass bodied commercially available
1N5711 due to their ability to dissipate higher levels of heat.
W. C. Brown who worked for
Raytheon in the 70’s was able to achieve RF-DC conversion efficiencies in the order of 90-92%
because he had access to custom designed diodes (Ref. 15).
Recent Development:
With the continuous effort of searching for a more suitable diode than the 1N5711, a new
candidate, although untested, has been found. It is the SMS3924-011 (Ref.16) manufactured by
Skyworks. Based on its specs, the Vbr is the same as the 1N5711 but Rs is only 11Ω and Cjo is
only 1.5pF. Using the previous equation,
fc _ diode 
1
1

 9.64GHz . This is a very good fc_diode since it
2*  * Rs * Cjo 2*  *11*1.5*1012
is roughly ten times greater then the frequency of operation. These parameters were obtained
from the figures below:
28
Fig. 26, SMS3924 Diode Specifications (Ref. 16)
29
With these new values, a new goal of 50% for the rectenna conversion efficiency has
been set based on the fact that this diode should be able to switch a lot faster than the 1N5711.
Again, it is a goal and there is no experimental data yet to back this up.
A couple of new values can be calculated with this new conversion efficiency.
Figure 21 becomes:
0.6 W
Impedance
Matching
+
Filter
Network
300 mW
Diode
Desired
effic. >50%
150 mW
Switch.
Regulator
50 % effic.
w/large I/O
Voltage
Difference
Load
3V
TI-Board
LMV751
Fig. 27, SMS3924 Diode Power Distribution
By re-using some of the previous values, and allowing the rectenna to be 50% efficient,
the input voltage will have to be scaled by
0.6W 
(6V * k ) 2 Solving _ for _ k
0.6W *120

 k  1.41 , therefore, the load voltage should be a
120
6V
minimum of 1.41*6V=8.46V and the Vbr of the diode has to be at least 2*8.46V=17V.
C) Maximum number of 1N5711 diode powered sensors in an Array
As discussed previously, in order to rectify 1 watt of power at 30% efficiency it is
required that the diode have a Vbr greater than 22Volts (Vbr1N5711=70V). To calculate the
electric field required to deliver Vin=11V to a ¾” (1.9cm) antenna we solve for Eo.
From fig. 21, and plugging in values for Rantenna=10 ohms, you can calculate the new voltage at
the antenna, Vant:
30
Step 1
RA= 4 Ω
VRa=4 Ω *.09A=.3V
Vant
Step 3 11.3V
RL
Step 2
RL=121 (Load resistance (11V)2/1W
Vin=11V (voltage present at the load)
Pin=1W
IL=0.09A
Vant=VRa+VL
Fig. 28, Calculating Vant (antenna voltage)
Eo 
Vant
11.3V

 11.8V / cm  1180V / m . This would be smallest electric
Leff 0.5*(1.905cm)
field required to power the last sensor on an array.
The very first sensor on the array would be able to tolerate a maximum Eo determined
by the Vbr of the first diode,Vbr1N5711=70V.
Vant
35.1V
RA=4
VRa=4*28.5mA
=.113V
RL
RL=1225
Vin=35V=Vbr/2=70/2
Pin=1W (idealistic)
IL=28.5mA
Fig. 29, Calculating Vant
Eo 
Vant
35.1V

 3685V / m
Leff 0.5*(1.905cm)
The very first sensor would be able to tolerate a maximum Eo of 3685 V/m with the
1N5711 diodes.
Given the waveguide dimensions 6”x3” (15.24cm x 7.62cm) and the breakdown voltage
of HDPE 480V/mil (480kV/1”= 18.8MV/m), the diodes will break down before the dielectric
will.
31
To find what the average power is at the first sensor, we use:
Eo 2 ab 36852 (.1524)(.0762)

 120Watts
(Ref. 11 page 624)
4
4*330
  TE _ waveguide _ impedance
Pave 
and at he very last sensor the average power is computed to 12 Watts (assuming no losses due to
the waveguide walls and no attenuation by the HDPE filler). Theoretically, it would be possible
to power (120W – 12W)/1W= 108 of these elements in series with 1 watt per element and a Vbr
of 70V (1N5711) (assuming that each element would extract exactly-only 1watt from the
waveguide due to the control by the voltage regulator).
D) Maximum number of SMS3924-011 diode powered sensors in an Array
Since Vbr is the same for the 1N5711 and the SMS3924 diodes, the max. Efield possible
in the WG remains at 3685 V/m or an average power of 120 Watts.
Vant
35V
RA=4
VRa=4*17.5mA
=0.07V
RL
RL=2000
Vin=35V=Vbr/2=70/2
Pin=0.6W (idealistic)
IL=17.5mA
Fig. 30, Calculating Vant High Side for the SMS3924
But the smallest Efield required to power the last sensor in the waveguide drops to
Eo 
Vant
8.7V

 9.1V / cm  910V / m , or, 7.28Watts required for the last sensor.
Leff 0.5*(1.905cm)
32
RA==4
VRa=4*.07=.3V
Vant
8.7V
RL
RL=119 (Rectenna resistance (11V)2/1W
Vin=8.46V (voltage present at the diode)
Pin=0.6W
IL=0.07A
Fig. 31, Calculating Voc Low Side for the SMS3924
The total number of sensors possible with the SMS3924 configuration becomes
120W  7.28W
 186 sensors.
0.6W / sensor
F) Extending the maximum number of elements in a stave
It is possible to extend the max. number of elements in an array by:
I)
Offsetting the power antenna.
The E field is max. at the center of the waveguide and drops off from the center as a
function of sin(x). Therefore, it is possible to increase the input power / E field without
damaging the rectifying diodes.
y
New
Antenna
Location
x
Fig. 32, Efield as a sine function
33
II)
Different sensors.
In this case, you partition the stave into three or more sections: high power, mid power,
and low power. The sensors in the high power section use state of the art, high Vbr diodes, while
the diodes in the low power region use regular 1N5711/SMS3924 diodes. This allows the
custom diodes to operate efficiently when they are fully turned on with the high Eo values and it
also prevents these high power diodes from suffering due to low power levels which decrease the
efficiency as discussed earlier.
It is also possible to vary the antenna size on each sensor in order to control the amount
of power that each element extracts; a shorter antenna symbolizes a smaller voltage induced on
the antenna.
III)
Adaptive sensors
Previously, it was discussed that each rectenna would have an impedance/filter network
before the shunt diode to maximize efficiency. It is possible to build such a network with a
varactor diode that will sense very high input power levels and it will purposely detune the
network by causing an impedance mismatch. The mismatch would result in lower power levels
being transferred to the shunt diode, and preventing high voltages from destroying the diodes
junction.
IV)
Multiple diodes
There are researchers who have experimented with multiple diodes in a possible series
configuration to increase the power handling capability of their rectennas. With two diodes in
series, it will increase the Vbr by two, but it will also increase the turn-on voltage (Vf) by two.
34
Future Research
In the next few months, research emphasis will be on design layout, design simulation
with Ansoft HFSS, and experimentation of the prototype rectenna. Towards the second half of
the year, all of the pieces of the project will come together and additional electrical/acoustical
testing will be conducted in EDO testing facilities.
Conclusion
It is possible to increase the efficiency of a rectenna by using high-speed diodes, possibly
custom GaAs diodes. A high Vbr, low capacitance diode is required in order to be able to have
multiple elements in a stave. The Skyworks diode is good candidate for this project. There are
multiple schemes which will increase the number of elements in each stave. It was found that it
is necessary to terminate a waveguide to prevent standing waves and this was accomplished with
a multilayered/tapered material. And finally, the corporate/parallel layout seems like a
good/inexpensive method to distribute power to the staves.
35
References
1) TI-TRF6901 Demo Board – Application notes, www.ti.com
2) Elliot, Robert S. “Antenna Theory an Design”, Prentice-Hall, New Jersey, 1981
3) Cook, Nigel P “Microwave Principles and Systems”, Prentice-Hall, NJ, 1986
4) Stutzman, Warren L. and Thiele, Gary A. “Antenna Theory and Design”, John Wiley and
Sons, 1981.
5) Rizzi, Peter A. “Microwave Engineering, Passive Circuits”, Prentice-Hall, NJ 1988
6) RF Products Website, http://www.randf.com/ramapriaas.html
7) ARC, http://www.arc.com
8) Tae-Whan Yoo and Kai Chang “ Theoretical and experimental development of 10 and 35
GHz Rectennas”, IEEE MTTS, vol 40, no6 June 1992, page 1259
9) McSpadden, James O “Design and experiments of a high conversion efficiency 5.8GHz
rectenna”, IEEE MTTS, vol 46, no12, December 1998
10) Agilent, http://www.agilent.com
11) Sadiku, Mathew “Elements of Electromagnetics”, Sauders Co. Pub., Or. Fl. 1994
12) Rodger Ziemer and William Tranter “Principles of Communications”, John Wiley &
Sons, NY, 2002
13) Texas Instruments, http://focus.ti.com/lit/an/swra035/swra035.pdf
14) Stutzman & Thiele “Antenna Theory an Design”, Wiley & Sons, USA, 1981
15) Brown, W.C. “Free-Space Microwave Power Transmission Study, Phase 3” US Dep. Of
Commerce, Doc. # N7616619, 10 Sep. 1975.
16) http://www.skyworksinc.com
36
Acknowledgements
1) Dr. Rajeev Bansal, University of Connecticut, for his technical help and patience.
2) Mr. Michael Sullivan, Electric Boat, again, for his technical help and patience.
3) Robin Padden, EDO, for his technical expertise in telemetry.
4) Skyworks, for semiconductor samples.
5) ARC, for RAM samples.
6) And numerous others which will remain untold.
37