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Application Note 84
April 2000
Linear Technology Magazine Circuit Collection, Volume IV
Power Products
Richard Markell, Editor
INTRODUCTION
Application Note 84 is the fourth in a series that excerpts
useful circuits from Linear Technology magazine to preserve them for posterity. This application note highlights
“power” circuits from issue VI:1 (February 1996) through
issue VIII:4 (November 1998). Another application note
will feature data conversion, interface and signal processing circuits from the same era. Like its predecessor, AN 66,
this Application Note includes circuits that can power
most any system you can imagine, from “server” power
supplies that generate in excess of 50 amps to micropower systems for portable and handheld equipment.
Also included are power converters that can be voltage
programmed using Intel’s VID code. Charge pump converters, linear regulators and battery charger circuits are
included here, with Li-Ion batteries receiving extra attention. There are, of course, circuits that cannot be so simply
categorized. Come browse. I’ll get out of the way and let
the authors describe their creations.
Note: Article Titles appear in this application note exactly
as they originally appeared in Linear Technology magazine. This may result in some inconsistency in the usage
of terminology.
TABLE OF CONTENTS
Introduction ......................................................................................................................................................... 1
REGULATORS—SWITCHING (BUCK)
New LTC®1435–LTC1439 DC/DC Controllers Feature Value and Performance ..................................................... 4
The LTC1266 Operates From ≥12V and Provides 3.3V Out at 12A ....................................................................... 7
The New LTC1435 Makes a Great Microprocessor Core Voltage Regulator ......................................................... 8
LTC1433/LTC1434: High Efficiency, Constant-Frequency Monolithic Buck Converter........................................ 10
24 Volt to 14 Volt Converter Provides 15 Amps ................................................................................................. 12
LTC1553 Synchronous Regulator Controller Powers Pentium® Pro and Other Big Processors ......................... 13
Synchronizing LTC1430s for Reduced Ripple .................................................................................................... 16
Combine a Switching Regulator and an UltraFast™ Linear Regulator for a High Performance 3.3V Supply ....... 18
The LTC1624: a Versatile, High Efficiency, SO-8 N-Channel Switching Regulator Controller ............................. 19
Low Cost 3.3V to 1.xV 6 Amp Power Supply ..................................................................................................... 21
The LT®1374: New 500kHz, 4.5A Monolithic Buck Converter............................................................................. 23
LTC1504: Flexible, Efficient Synchronous Switching Regulator Can Source or Sink 500mA .............................. 24
High Efficiency Distributed Power Converter Features Synchronous Rectification ............................................. 26
Fixed Frequency, 500kHz, 4.5A Step-Down Converter in an SO-8 Operates from a 5V Input ............................. 29
VID Voltage Programmer for Intel Mobile Processors........................................................................................ 32
New DC/DC Controller Enables High Step-Down Ratios ..................................................................................... 34
LTC1627 Monolithic Synchronous Step-Down Regulator Maximizes Single or Dual Li-Ion Battery Life ............ 36
The LTC1625 Current Mode DC/DC Controller Eliminates the Sense Resistor ................................................... 38
PolyPhase™ Switching Regulators Offer High Efficiency in Low Voltage, High Current Applications ................. 39
LTC1622: Low Input Voltage, Current Mode PWM Buck Converter.................................................................... 43
www.BDTIC.com/Linear
AN84-1
Application Note 84
Wide Input Range, High Efficiency Step-Down Switching Regulators ................................................................ 46
REGULATORS—SWITCHING (BOOST)
±12 Volt Output from the LT1377 ...................................................................................................................... 51
The LT1370: New 500kHz, 6A Monolithic Boost Converter ................................................................................ 53
Bootstrapped Synchronous Boost Converter Operates at 1.8V Input ................................................................. 55
REGULATORS (SWITCHING)—BUCK-BOOST
500kHz Buck-Boost Converter Needs No Heat Sink ........................................................................................... 56
Battery-Powered Buck-Boost Converter Requires No Magnetics ....................................................................... 57
REGULATORS—SWITCHING (INVERTING)
Making –5V 14-Bit Quiet .................................................................................................................................... 57
Negative-to-Positive Telecommunication Supply ............................................................................................... 60
Positive-to-Negative Converter Powers –48V Telecom Circuits ......................................................................... 61
Low Noise LT1614 DC/DC Converter Delivers –5V at 200mA from 5V Input ..................................................... 62
–48V to 5V DC/DC Converter Operates from the Telephone Line ....................................................................... 63
REGULATORS—SWITCHING (FLYBACK)
The LT1425 Isolated Flyback Controller ............................................................................................................. 65
High Isolation Converter Uses Off-the-Shelf Magnetics ..................................................................................... 68
Wide-Input-Range, Low Voltage Flyback Regulator ........................................................................................... 69
REGULATORS—SWITCHING (LOW NOISE)
The LT1533 Heralds a New Class of Low Noise Switching Regulators ............................................................... 70
LT1533 Ultralow Noise Switching Regulator for High Voltage or High Current Applications .............................. 74
REGULATORS—SWITCHING (MULTIOUTPUT)
LTC1538-AUX: a New Addition to LTC’s Adaptive Power™ Controller Family .................................................... 77
High Efficiency, Low Power, 3-Output DC/DC Converter .................................................................................... 77
Dual-Output Voltage Regulator ........................................................................................................................... 78
Switcher Generates Two Bias Voltages without Transformer ............................................................................. 80
New IC Features Reduce EMI from Switching Regulator Circuits ....................................................................... 81
REGULATORS—SWITCHING (MICROPOWER)
Power Management and High Efficiency Switcher Maximize Nine-Volt Battery Life ........................................... 85
LT1307 Micropower DC/DC Converter Eliminates Electrolytic Capacitors .......................................................... 86
An Ultralow Quiescent Current, 5V Boost Regulator........................................................................................... 89
Capacitive Charge Pump Powers 12V VPP from 5V Source ............................................................................... 90
LTC1474 and LTC1475 High Efficiency Switching Regulators Draw Only 10µA Supply Current ........................ 91
Free Digital Panel Meters from the Oppressive Yoke of Batteries ....................................................................... 94
The LTC1514/LTC1515 Provide Low Power Step-Up/Step-Down DC/DC Conversion without Inductors ........... 95
LTC1626 Low Voltage Monolithic Step-Down Converter Operates from a Single Li-Ion Cell ............................. 96
12V Wall Cube to 5V/400mA DC/DC Converter is 85% Efficient......................................................................... 99
Micropower 600kHz Fixed-Frequency DC/DC Converters Step Up from a 1-Cell or 2-Cell Battery ................... 100
LT1610 Micropower Step-Up DC/DC Converter Runs at 1.7MHz ..................................................................... 103
Low Noise 33V Varactor Bias Supply ............................................................................................................... 105
The LTC1516 Converts Two Cells to 5V with High Efficiency at Extremely Light Loads ................................... 106
REGULATORS—LINEAR
Low Dropout Regulator Driver Handles Fast Load Transients and Operates on A Single 3V–10V Input ........... 107
AN84-2
www.BDTIC.com/Linear
Application Note 84
The LT1575/LT1577 UltraFast Linear Regulator Controllers Eliminate Bulk Tantalum/Electrolytic Output
Capacitors ........................................................................................................................................................ 108
LT1579 Battery-Backup Regulator Provides Uninterruptible Power ................................................................. 111
BATTERY CHARGERS
The LT1511 3A Battery Charger Charges All Battery Types, Including Lithium-Ion .......................................... 114
LT1512/LT1513 Battery Chargers Operate with Input Voltages Above or Below the Battery Voltage ............... 116
Li-Ion Battery Charger Does Not Require Precision Resistors .......................................................................... 118
LT1510 Charger with –∆V Termination ............................................................................................................ 119
Constant-Voltage Load Box for Battery Simulation........................................................................................... 121
High Efficiency, Low Dropout Lithium-Ion Battery Charger Charges Up to Five Cells at 4 Amps or More ........ 122
Battery Charger IC Can Also Serve as Main Step-Down Converter ................................................................... 127
LT1635 1A Shunt Charger ................................................................................................................................ 129
800mA Li-Ion Battery Charger Occupies Less Volume than Two Stacked Quarters ......................................... 130
Single-Cell Li-Ion Battery Supervisor ............................................................................................................... 132
POWER MANAGEMENT
LTC1479 PowerPath™ Controller Simplifies Portable Power Management Design .......................................... 134
The LTC1473 Dual PowerPath Switch Driver Simplifies Portable Power Management Design ........................ 137
Short-Circuit-Proof Isolated High-Side Switch ................................................................................................. 139
Tiny MSOP Dual Switch Driver is SMBus Controlled ........................................................................................ 140
LTC1710: Two 0.4Ω Switches with SMBus Control Fit into Tiny MSOP-8 Package ......................................... 141
MISCELLANEOUS
VID Voltage Programmer for Intel Mobile Processors...................................................................................... 141
Battery Charger IC Doubles as Current Sensor ................................................................................................. 145
100V, 2A, Constant-Voltage/ Constant-Current Bench Supply ......................................................................... 146
A Complete Battery Backup Solution Using a Rechargeable NiCd Cell ............................................................. 147
What Efficiency Curves Don’t Tell ..................................................................................................................... 149
APPENDIX A: COMPONENT VENDOR CONTACTS ......................................................................... 153
INDEX............................................................................................................................ 157
, LTC, and LT are registered trademarks of Linear Technology Corporation; Adaptive Power, Burst Mode, No RSENSE, PolyPhase, PowerPath and UltraFast are trademarks of Linear Technology
Corporation. Gelcell is a trademark of Johnson Controls, Inc.; Kool Mµ is a registered trademark of Magnetics, Inc.; Pentium is a registered trademark of Intel Corp.; VERSA-PAC is a trademark of
Coiltronics, Inc.
www.BDTIC.com/Linear
AN84-3
Application Note 84
Regulators—Switching (Buck)
NEW LTC1435–LTC1439 DC/DC CONTROLLERS
FEATURE VALUE AND PERFORMANCE
by Randy Flatness, Steve Hobrecht and Milton Wilcox
Introduction
The new LTC1435–LTC1439 multiple-output DC/DC controllers bring unprecedented levels of value to supplies for
notebook computers and other battery-powered equipment, while eliminating previous performance barriers.
For example, a new Adaptive Power™ output stage allows
two previously incompatible parameters, constant frequency operation and good low current efficiency, to
coexist in the same power supply. A second breakthrough
allows N-channel power MOSFETs to be used exclusively,
while maintaining low dropout operation previously available only with P-channel MOSFETs. Other innovations
include an auxiliary linear regulator loop, a phase-locked
loop (PLL) to synchronize the oscillator to an external
source, a self-contained power-on-reset (POR) timer and
programmable run delays useful for staging output
voltages.
VIN 28V (MAX)
+
2.2µF
INT VCC
VIN
*CMDSH-3
DR VCC
+
BOOST
VPROG
0.1µF
56pF
EXT.
CLOCK
10k
COSC
TGS
PLL IN
SW
PLL LPF
BG
IRLML2803
T1
0.01µF
MBRS1100
MBRS140
IRF7403
+
100
SENSE+
POR
10k
22µF
35V
×2
IRF7403
TGL
510pF
1000pF
LTC1437
0.033Ω
100
SENSE –
ITH
51pF
VOSENSE
0.1µF
RUN/SS
3.3µF
35V
0.1µF
EXT VCC
+
VOUT1
5V/3A
100µF
10V
×2
LBI
47k
LBO
AUX DR
SFB
AUX FB
SGND
ZETEX
FZT749
+
PGND AUX ON
1MEG
100k
26V
VOUT2
12V/200mA
4.7µF
25V
T1 = DALE LPE-8562-A092
(650) 665-9301
*CENTRAL SEMICONDUCTOR
(516) 435-1110
Figure 1. High Efficiency, Constant Frequency, Dual-Output Supply Delivers 3A at 5V and 250mA at 12V
AN84-4
www.BDTIC.com/Linear
Application Note 84
Cost Effective LTC1437 Switcher/Linear Combination
with 5V/3A and 12V/200mA Outputs
The main switcher loop, shown in the schematic in Figure␣ 1, is set to 5V by strapping the VPROG pin high. Other
output options include 3.3V (VPROG low) and adjustable
(VPROG open).
The 12V output in Figure 1’s circuit is provided by the
auxiliary linear regulator operating in conjunction with a
secondary winding feedback loop using the SFB pin. The
turns ratio for the transformer is 1:2.2, resulting in a
secondary output voltage of approximately 15V. The secondary resistive divider causes the SFB pin voltage to drop
below the internal 1.19V reference if the secondary output
is loaded and the 5V output has little or no load. This forces
continuous operation as necessary to guarantee sufficient
headroom for the linear regulator to maintain 12V regulation independent of the 5V load. The auxiliary output is
turned on and off with the AUX ON pin.
The auxiliary regulator can also be used in an adjustable
mode, determined by the voltage on the AUX DR pin.
When the AUX DR voltage is higher than 9.5V, as is the
VIN 5.2V-25V
10
+
0.1µF
2.2µF
*CMDSH-3
+
Si4412DY
22µF
35V
×2
SFB1 INT VCC
BOOST1
IRLML2803
VIN
*CMDSH-3
VPROG1 VPROG2
BOOST2
TGL1
TGL2
TGS1
TGS2
SW1
SW2
BG1
BG2
+
IRLML2803
0.1µF
10µH
0.1µF
10µH
MBRS140
MBRS140
Si4412DY
100
0.033Ω
SENSE1+
SENSE2+
SENSE1–
SENSE2–
Si4410DY
100
VOUT2
3.3V/3A
VOSENSE2
1000pF
100µF
10V
×2
LTC1439
0.02Ω
1000pF
100
1000pF
100
+
22µF
35V
×2
Si4410
1000pF
1000pF
10k
ITH1
220pF
0.1µF
10k
+
ITH2
0.1µF
0.05µF
RUN/SS1
100µF
10V
×2
220pF
RUN/SS2
20
56pF
EXT.
CLOCK
10k
EXT VCC
4.7nF
COSC
AUX DR
PLL IN
AUX ON
PLL LPF
AUX FB
0.01µF
47k
51pF
316k
LB1
LB0
SGND
PGND
POR2
MMBT2907ALT1
ZETEX
ZTX849
100
+
221k
VOUT3
2.9V/2.5A
330µF
6.3V
VOUT1
5V/3A
*CENTRAL SEMICONDUCTOR (516) 435-1110
Figure 2. High Efficiency, Constant-Frequency, Triple-Output Supply Features 200mV Dropout
www.BDTIC.com/Linear
AN84-5
Application Note 84
case in Figure 1, the regulator automatically configures
itself for fixed 12V operation using an internal AUX FB
resistive divider. When AUX DR is less than 8.5V, the
internal divider is removed and the user can adjust the
output voltage via an external divider referenced to 1.19V.
The external auxiliary regulator PNP pass transistor is
sized for the desired output current; in this case a SOT-223
device is used to deliver up to 200mA.
PLL␣ IN signal is present, the PLL LPF pin goes low,
causing the oscillator to run at its minimum frequency
(fMIN = 180kHz with COSC = 56pF). Applying a 3.3V or 5V
logic signal of any duty cycle to the PLL IN pin will cause
the oscillator frequency to lock to the external frequency
and to track it up to a maximum of fMAX = 2 • fMIN. A logic
signal may also be coupled to PLL LPF to effect a 2:1
frequency shift, provided that the initial frequency has
been set to less than 200kHz.
Synchronizable, Triple-Output, Low Dropout Supply
Figure 3 is a photograph showing the 3.3V output staged
to start 10ms before the 5V output when power is first
applied to Figure 2’s circuit.
The LTC1439-based supply shown in Figure 2 is an
example of how three logic supply voltages, 5V, 3.3V and
2.9V, can be easily derived using only two simple inductors. The two main DC/DC controller loops are used to
supply 5V/3A and 3.3V/5.5A. Up to 2.5A of the 3.3V output
current is then used to supply a 2.9V output using the
adjustable capability of the auxiliary linear regulator.
The 2.9V output also illustrates the use of an external NPN
pass transistor with the auxiliary regulator. Because only
0.4V is dropped across the NPN transistor, 2.9V efficiency
remains in the 85% range. And thanks to the 99% duty
cycle capability of the switcher loops, Figure 2’s supply
can maintain all three output voltages in regulation down
to VIN = 5.2V with a 2A load on the 5V output.
The phase-locked loops built into the LTC1437/LTC1436PLL and LTC1439 offer a convenient means of synchronization for the applications in Figures 1 and 2. The internal
oscillator is actually a voltage-controlled oscillator (VCO)
controlled by the voltage on the PLL LPF pin. When no
An internal regulation monitor is continually monitoring
the main controller output in the LTC1436/LTC1437, and
the controller 2 output (3.3V in Figure 2) in the LTC1438/
LTC1439. When out of regulation or in shutdown mode,
the POR open drain output pulls low. At start-up, once the
output voltage has reached 5% of its final value, an internal
timer is started, after which the POR pin is released. The
timer is accomplished by counting 216 oscillator cycles,
yielding a delay-to-release reset of approximately 300ms
in a typical application.
The EXT VCC pin is normally connected to the 5V output to
allow INT VCC power to be derived from the regulator itself.
Quiescent current is then reduced because driver and
control currents are scaled by a factor approximately equal
to the 5V controller duty cycle. EXT VCC can also be
connected to other external high efficiency sources, up to
a maximum of 10V.
Figure 3. Start-Up of 3.3V and 5V Supplies is Easily Staged
Upon Initial Application of Input Power
AN84-6
www.BDTIC.com/Linear
Application Note 84
THE LTC1266 OPERATES FROM ≥12V AND PROVIDES
3.3V OUT AT 12A
by Craig Varga
Circuit Description and Operation
to nearly 24V above ground. When the LTC1266 takes
pin␣ 1 high, Q4 turns on, pulling charge from the gate
capacitance of Q1 through D3. This back biases the baseemitter junction of Q6, forcing the pull-up circuit, and
therefore Q1, off.
The design in Figure 4 relies on a floating high-side driver
that provides enough gate-drive capability to easily switch
a large power MOSFET. The LTC1266 is configured to
drive a P-channel MOSFET by tying pin 3 (PINV) to ground.
This is required because there will be a net inversion by the
floating driver. Q4 controls the driver stage and provides
gate-discharge capability through D3. When the low-side
switches are on, C16 charges to 12V through D1. When
the LTC1266 signals Q1 to turn on, Q4 is turned off. R11
provides base current for Q6, which, in conjunction with
Q5, acts like an SCR. Once fired, the regenerative behavior
of Q5 and Q6 rapidly charges the gate of Q1. Since C16 is
referenced to the source of Q1, the top of C16 rises above
the 12V supply rail as Q1 turns on, forcing the gate of Q1
Since the input voltage is high relative to the output, the
nominal duty factor of the high-side switch is small (in this
case approximately 31%). As a result, the RMS current
through Q1 is relatively low. By contrast, the low-side
switches are on nearly 70% of the time, and therefore see
a much higher RMS current. This explains why the lowside switch employs two MOSFETs, whereas the high-side
switch uses only one. Schottky diode D2 is used to help
keep the body diodes of Q2 and Q3 from turning on during
the short dead time before switching transitions. These
body diodes exhibit relatively long reverse recovery times,
contributing to commutation losses. The Schottky diode
improves overall efficiency several percent, but the circuit
will function correctly without it. Switching losses in the
12V
+
D1
MBR120T3
R11
4.3k
R10
220Ω
Q5
2N3906
C7 TO C12
100µF, 16V
×6
C16
0.1µF
R9
220Ω
Q6
MPS2222
R8
51Ω
1
2
3
4
5
6
7
R6
1k
C13
1µF
8
TDRV
BDRV
PWRVIN
PGND
PINV
BINH
VIN
LBO
ITH
SENSE–
C14
C3
300pF 1000pF
D3
MBR0520LT3
16
C17
0.001µF
S/D
VFB
5
6
7
1
2
3
8
R4
0.015Ω
L1
4µH
R13
1.0Ω
15
R5
0.015Ω
VOUT
3.3V
12A
14
R1
100Ω
13
LBIN
U1
LTC1266
12
SGND
CT
R10
10k
1%
R2
100Ω
11
+
10
9
SENSE+
C1
1000pF
C2
3300pF
E1
S/D
Q4
VN2222LL
Q1
Si4410
4
R12
5.1Ω
Q3
Si4410
4
5
1
6
2
7
3
Q2
8 Si4410
4
5
6
7
8
D2
MBRS320T3
1
2
C4 TO C6, C15
330µF, 6.3V
×4
R3
6.04k, 1%
3
ALL POLARIZED CAPACITORS ARE AVX TYPE TPS (207) 282-5111 OR EQUIVALENT
Figure 4. 12V In, 3.3V/12A Out Supply
www.BDTIC.com/Linear
AN84-7
Application Note 84
two low-side switches are nearly zero, since these devices
are turned on and off into nearly zero volts (the forward
drop of the Schottky).
Figure 5 shows a 24V input design. As the input supply
voltage is increased, one thing to watch for is the potential
for overlap in the high- and low-side turn-on/turn-off
transitions. The LTC1266 is designed to prevent shootthrough by actually waiting until the gate voltage of one
switch is low before allowing the other switch to be turned
on. Using the floating driver defeats this capability, so this
condition must be checked for. The high-side drive turnon time may be reduced by lowering the value of R11.
Using a larger device for Q4 will speed up the turn-off
transition. The value of C16 may also need to be a bit larger
if R11 is reduced to limit drooping of the bootstrap supply
voltage.
There is no fundamental limitation on how high the maximum input voltage can be with this approach. The drive
level shift is limited by the breakdown rating of Q4.
Obviously, the power transistors and input capacitors
must be rated for the intended input voltage. A low power
12V supply is needed to provide power for the LTC1266
and voltage for the bootstrap supply.
24V IN
+ C12
R14
20k
D4
MBR0540LT3
R11
4.3k
Q5
2N3906
D3
1N759
12V
1
2
3
4
5
6
7
R6
1k
C11
1µF
C9
0.1µF
R9
220Ω
Q6
MPS2222
R8
51Ω
8
Q4
VN2222LL
TDRV
BDRV
PWRVIN
PGND
PINV
BINH
VIN
LBO
Q1
Si4410
4
R12
5.1Ω
D2
MBR0520LT3
16
C10
0.001µF
S/D
ITH
VFB
SENSE–
6
7
1
2
3
330µF
35V
SEE NOTE 3
8
R4
0.015Ω
L1
7µH
R5
0.015Ω
VOUT
3.3V
12A
14
R1
100Ω
R7
10k
1%
R2
100Ω
11
+
10
9
SENSE+
C7
C3
470pF 1000pF
5
R13
1.0Ω, 1/4W
15
13
LBIN
U1
LTC1266
12
SGND
CT
330µF
35V
SEE NOTE 3
R10
220Ω
Q7
MPS2222A
+ C13
Q3
Si4410
4
C1
1000pF
5
1
C2
3300pF
6
2
7
Q2
8 Si4410
4
3
C4 TO C6, C15
330µF 6.3V
×4
5
6
7
8
D1
MBRS340T3
1
2
R3
6.04k, 1%
3
E1
S/D
1. ALL POLARIZED CAPACITORS ARE AVX TYPE TPS
OR EQUIVALENT UNLESS NOTED OTHERWISE.
(207) 282-5111
2. L1 CONSISTS OF 15 TURNS OF #16 AWG ON
MAGNETICS, INC. 77848-A7 Kool Mµ CORE
(800) 245-3984
3. C12 AND C13 ARE PANASONIC TYPE HF
OR EQUIVALENT
(201) 348-7522
Figure 5. 24V In, 3.3V/12A Out Supply
THE NEW LTC1435 MAKES A GREAT
MICROPROCESSOR CORE VOLTAGE REGULATOR
by John Seago
Current microprocessor architectures require different
voltages for the core and the I/O ring. For portable computer applications, the microprocessor core voltage is
reduced for lower power consumption. Three high current
AN84-8
regulated voltages, 5V, 3.3V and 2.9V, are commonly
required. Several IC manufacturers offer two-output controllers, like the LTC1438, which are normally used for 5V
and 3.3V. Another controller is required to generate the
2.9V. Figure 6 shows a simple circuit using the LTC1435
to provide 2.9V at 2.65 amps for the Intel portable Pentium®
processor.
www.BDTIC.com/Linear
Application Note 84
The circuit’s 165kHz switching frequency was selected as
a compromise between transient response and circuit
efficiency. This frequency is determined by the value of C1.
Output voltage transient response is shown in Figure 7.
The transient response can be adjusted for other applications by changing the values of compensation components R1, C3 and C14. Efficiency curves for different input
voltages and load currents up to 3.2 amps are shown in
Figure 8.
Another feature of the LTC1435 is the option to maintain
constant switching frequency under all load conditions or
to select Burst Mode™ operation for the highest efficiency
at light loads. Pulling the SFB pin high enables Burst Mode
when load current drops to a low value. However, Burst
Mode can degrade transient response at low input voltages
and should not be used for pulsed load applications where
good transient response at low input voltage is required.
100
5.5V INPUT
EFFICIENCY (%)
90
100mVP-P
50mV/DIV
4A
2A/DIV
10V INPUT
15V INPUT
80
70
20V INPUT
0.0A
60
500µs/DIV
28V INPUT
50
0.01A
Figure 7. Output Voltage vs Transient Response
0.1A
1.0A
10A
Figure 8. LTC1435 Efficiency Curves for Different Input Voltages
5.5V-28V
+
C9
22µF
35V
C1
68pF
1
TG
COSC
+
C10
22µF
35V
16
Q1
SI4412
C2
0.1µF
2
R1
10k
RUN/SS
BOOST
15
C3
330pF
3
ITH
SW
SFB
VIN
14
C6
0.1µF
C14
47pF
4
C4
100pF
6
7
SGND
INT VCC
VOSENS
BG
SENSE–
PGND
SENSE+
EXT VCC
12
C5
0.001µF
8
C9, C10 =
C12, C13 =
D1 =
D2 =
C7
0.1µF
2.9V/
2.65A
C11
470pF
11
10
R2
0.033Ω
D1
MBRS0530
13
LTC1435
5
L1
10µH
R3
35.7k
+
C13
100µF
10V
Q2
SI4412
+
+
C8
4.7µF
D2
MBRS140T3
9
C12
100µF
10V
R4
24.9k
AVX, TPSE226M035
L1 = SUMIDA, CDRH125-10
AVX, TPSD107M010
Q1 = Q2 = SILICONIX, SI4412DY
MOTOROLA, MBRS0530
R2 = IRC, LR2010-01-R033-F
MOTOROLA, MBRS140T3
Figure 6. 2.9V Regulator for Portable Pentium Processor
www.BDTIC.com/Linear
AN84-9
Application Note 84
2.9V
10V
1V/DIV
0.0V
0.0V
1.25A
5V/DIV
4A
1A/DIV
0.0A
0.0A
2A/DIV
2µs/DIV
200µs/DIV
Figure 9. Inductor Input Voltage and Current Waveforms
Figure 10. Soft-Start Output Voltage and Inductor Current
The SFB pin in the circuit of Figure 6 is grounded, which
will defeat the Burst Mode and ensure constant frequency
operation.
age during this soft-start period depends on the load
impedance. If soft-start is not required, capacitor C2 is not
used and the current limit setting of the regulator determines the maximum load current during start-up.
It is sometimes necessary to shut down power to the load.
RUN/SS is a dual-function pin on the LTC1435 that provides both output voltage on/off control and output current soft-start capability. When RUN/SS (pin 2) is pulled
low by an open collector or open drain device, the output
voltage is turned off and the controller shuts down. The
soft-start feature takes over when the low is removed from
pin 2. Figure 9 shows the output voltage under no-load
conditions at turn-on, with the soft-start capacitor C2
equal to 0.1µF. This simulates the start up conditions of a
microprocessor held in standby until after the input voltage has stabilized. If the regulator is started under full-load
conditions, the output current ramp time will be approximately 0.5s/µF of soft-start capacitance. The output volt-
In order to properly enhance the top MOSFET (Q1), INT
VCC is level shifted by charge pumping capacitor C6 to INT
VCC minus one diode drop. C6 provides the power to turn
Q1 on and off. The INT VCC of the LTC1435 is regulated to
5V, but will increase with higher voltage applied to EXT
VCC, up to a maximum of 10V. For outputs between 5V and
10V, the output should be connected to EXT VCC. The
power loss of the INT VCC linear regulator will be replaced
by the more efficient switcher output and the gate-drive
voltage of both MOSFETs will be increased for lower “ON”
resistance. Figure 10 shows L1 input voltage and current
with a 10 volt input, 2.9 volt output, and 2.65 amp load
current.
LTC1433/LTC1434: HIGH EFFICIENCY, CONSTANTFREQUENCY MONOLITHIC BUCK CONVERTER
by San-Hwa Chee
Typical Application: Buck Converter
Supplies 3.3V at 600mA
Figure 11 shows a practical LTC1433 circuit that can be
used for cellular telephone applications. Efficiency curves
for this circuit at various input voltages are shown in
Figure 12. Note that the efficiency reaches 93% at a supply
voltage of 5V and a load current of about 150mA. This high
efficiency makes the LTC1433 and LTC1434 attractive for
power-sensitive applications. The circuit works all the way
down to 3.6V at a load current of 250mA before dropping
out and the oscillator frequency is a constant 210kHz
down to 20mA load current.
AN84-10
Typical Application: Positive-to-Negative Converter
Both the LTC1433 and LTC1434 can easily be set up for a
negative output voltage. Figure 13 shows the schematic
using the LTC1433. The efficiency curve is shown in Figure
14. This circuit is set up so that the output is taken from the
device ground. Components that are normally referenced
back to the device ground, such as the Run/SS capacitor,
oscillator frequency capacitor and the ITH compensation
network, are connected to the output instead of to the
circuit ground.
www.BDTIC.com/Linear
Application Note 84
0.1µF
* MBRS130LT3
** COILCRAFT DO3316-104
†
AVX TPSD107M010R0100
††
AVX TPSE686M020R0150
100
68µF††
20V
VIN = 5V
D1*
100µH
L1**
+
†
100µF
10V
INPUT VOLTAGE
3.6V TO 12V
10k
13
C
LTC1433 OSC
5 SGND
POR 12
6 RUN/SS
ITH 11
4 NC
7 LBO
8 LBI
0.1µF
80
VIN = 12V
VIN = 9V
70
60
POWER ON RESET
50
680pF
VOSENSE 10
5.1k
VPROG 9
47pF
40
0.001
6800pF
0.01
0.10
1.00
LOAD CURRENT (A)
Figure 12. Efficiency vs Load Current
for Figure 11’s Circuit
Figure 11. LTC1433 Typical Application: 3.3V Output at 600mA
* MOTOROLA MBRS130LT3
** COILCRAFT DO3316 SERIES
†
AVX TPSD107M010R0100
††
AVX TPSE107M016R0100
+
VOUT
–5.0V
VIN (V)
L1**
68µH
100µF
10V
†
D1*
0.01µF
IOUT MAX (mA)
3.0
180
4.0
240
5.0
290
6.0
340
7.0
410
7.5
420
1
SSW
2
NC
3
BSW
4
NC
5
SGND
6
RUN/SS
7
LBO
8
LBI
LTC1433
PWRVIN
16
PGND
15
SVIN
14
COSC
13
POR
12
ITH
11
VOSENSE
10
VPROG
INPUT VOLTAGE
3V TO 7.5V
100µF††
16V
100pF
+
0.1µF
6800pF
9
680pF
5.1k
Figure 13. Positive-to-Negative (–5.0V) Converter
100
90
EFFICIENCY (%)
VOUT
3.3V
PWRVIN 16
PGND 15
SVIN 14
1 SSW
2 NC
3 BSW
EFFICIENCY (%)
+
90
VIN = 7V
80
70
VIN = 3.5V
60
50
40
0.001
VOUT = –5.0V
COSC = 100pF
0.01
0.10
1.00
LOAD CURRENT (A)
Figure 14. Efficiency Curves for Figure 13’s Positive-to-Negative Converter
www.BDTIC.com/Linear
AN84-11
Application Note 84
24 VOLT TO 14 VOLT CONVERTER
PROVIDES 15 AMPS
by John Seago
efficiency is required, adding a second power MOSFET for
synchronous switching will improve efficiency by
about␣ 1%.
Combining the LTC1435 with a large geometry power
MOSFET and good PCB layout allows large currents to be
processed easily and efficiently. With the use of a current
sense transformer, output voltages greater than 10V can
be implemented. The circuit in Figure 15 shows an LTC1435
configured as a conventional buck regulator using a single
N-channel MOSFET to control an output voltage greater
than 10V with load current exceeding 15 amps. The
efficiency of the breadboard measured 94% with a 24V
input, 14V output and 15A of load current. If maximum
This circuit’s 100kHz switching frequency was selected to
reduce switching losses so that PCB mounted heat sinks
could be used without requiring additional air flow. The
switching frequency can be set from 50kHz to 400kHz by
selecting an appropriate value for C1. The current sense
transformer T1 uses a 1:100 turns ratio to scale down the
buck inductor input current and develop the voltage across
R9, used by the ±SENSE inputs for regulation. Shortcircuit protection is provided by Q4 and Q5. When the
current transformer secondary voltage developed across
D5
1N4148
INPUT
18V TO 28V
+
C10
1000µF
35V
+
C11
1000µF
35V
T1
R3
10Ω
D1
1T
100T
R8
R7 1.2Ω
1K
C15
100pF
D2 1N758
Q1
2N3904
C1
120pF
Q3
1
C2, 0.1µF
2
R1, 10k
3
C3, 330pF
C4, 47pF
4
5
C5, 100pF
6
R2
11.8k
COSC
TG
RUN/SS
ITH
SFB
BOOST
SW
U1
LTC1435
VIN
INT VCC
SGND
BG
VOSENS
7
SENSE –
8
SENSE +
PGND
EXT VCC
15
C7, 0.1µF
C12
470pF
D3
C8, 0.1µF
12
11
R10
100Ω
D6
C17
0.001µF
Q5
VN2222LL
R11
100k
L1
10µH
14
13
Q4
2N3906
R9
0.62Ω
D7
1N751
Q2
2N3906
16
C16
0.001µF
+
C9
4.7µF
D4
14V
AT 15A
R6
127k
C13
470µF
25V
+
10
R4, 100Ω
D8
1N4148
R5, 100Ω
R12, 100Ω
C18
1µF
R14
430Ω
R13
2.2k
R16
16k
Q6
2N3904
C19
0.01µF
R15
470Ω
C20
0.001µF
C10, C11 = NICHICON, UPL1V102MHH6
(847) 843-7500
C13, C14 = NICHICON, UPL1E471MHH6
D1, D3, D6 = MOTOROLA, MBRS0540
(800) 441-2447
D4 = MOTOROLA MBR2045 WITH
THERMALLOY #7020 HEAT SINK
Q3 = INTERNATIONAL RECTIFIER, IRL3803
(310) 322-3331
WITH THERMALLOY #6299 HEAT SINK
(972) 243-4321
L1: CORE = MAGNETICS, 55930-AZ
(800) 245-3984
WINDING = 8T #14 BIFILAR
T1: CORE = MAGNETICS W-41406-TC
WINDING = PRI = 1T #18 SEC = 100T #32
Figure 15. 14V, 15A Buck Regulator
AN84-12
C14
470µF
25V
GND
9
C6, 0.001µF
+
+
www.BDTIC.com/Linear
Application Note 84
A = Q3 SWITCH VOLTAGE
20V/DIV
B = L1 CURRENT
10A/DIV
0.0V
0.0A
C = T1 PRIMARY CURRENT
10A/DIV
0.0A
D = OUTPUT VOLTAGE RIPPLE
14VDC
0.2V/DIV
2µS/DIV
Figure 16. Buck Regulator Circuit Waveforms
R8 and R9 is enough to turn on Q4, Q5 temporarily pulls
the RUN/SS pin low, turning off the regulator. Output
current soft-starts when Q5 releases the RUN/SS pin. This
results in frequent attempts to establish output voltage if
a short exists, without high current continuously flowing
through the power elements. The power elements consist
of input capacitors C10 and C11, Current sense transformer T1, buck inductor L1, power MOSFET Q3, commutating diode D4 and output capacitors C13 and C14.
Although the wide 3.6V–36V input voltage range and 99%
duty cycle operation of the LTC1435 are ideal for battery/
wall adapter input applications, operating above 95% duty
cycle causes problems for the current sense transformer.
To avoid transformer saturation, the Q6 stage limits duty
cycle to approximately 90%. Current through R16 tries to
LTC1553 SYNCHRONOUS REGULATOR CONTROLLER
®
POWERS PENTIUM PRO AND
OTHER BIG PROCESSORS
by Y.L. Teo, S.H. Lim and Craig Varga
The LTC1553 provides current-limit and short-circuit protection without the use of an external sense resistor. It has
excellent (±1%) output regulation over temperature, line
voltage and load current variations. To compliment the
main voltage-feedback loop, the LTC1553 includes two
additional feedback loops to provide good large-signal
transient response. The LTC1553 adds additional internal
circuits to conform to the Intel Pentium Pro processor
charge C20 to the 3V base voltage of Q6. If the switch cycle
terminates at less than a 90% duty cycle, C20 is reset by
D8. If the duty cycle exceeds 90%, C20 charges until Q6
turns on, ending the switch cycle.
Switch voltage, inductor current, T1 primary current, and
output voltage ripple waveforms are shown in Figure 16.
These waveforms were measured with a 24V input, 14V
output, and 15A load current. When MOSFET Q3 turns on,
the switch voltage (Trace A) goes high, the inductor
current (Trace B) increases, as does the T1 primary
current (Trace C) and the output ripple voltage (Trace D).
When Q3 turns off, the switch voltage goes low, inductor
current decreases as its stored energy supplies load
current through D4, T1 primary current goes to zero and
the output voltage decreases slightly.
power converter requirements while minimizing the number of external components. An on-chip 5-bit digital-toanalog converter (DAC) provides output voltages conforming to Intel’s specifications. This allows the LTC1553
to read the code sent by the processor and provide it with
the requested voltage. The LTC1553 also provides a
power-good indication (PWRGD) to the system. There is
also an on-chip overvoltage protection circuit that latches
the regulator in an off state if the output voltage ever rises
15% or more above the DAC-requested voltage.
In applications with other processors, the four DAC inputs
can be routed to a jumper block, zero ohm resistors or a
www.BDTIC.com/Linear
AN84-13
Application Note 84
Typical Application
DIP switch, or hard wired, to set the desired output
voltage. This allows the output voltage to be programmed
easily in steps while eliminating the need to stock an
assortment of precision resistors. This flexibility in output
voltage setting is cheap insurance against last-minute
power supply voltage changes by microprocessor manufacturers.
A typical application for LTC1553 is converting 5V to
1.8V–3.5V in a Pentium Pro processor based personal
computer. The supply may be in the form of a voltage
regulator module (VRM) or may be implemented directly
on the motherboard. The output is used to power the
Pentium Pro processor and the input is taken from the
system’s 5V supply. The circuit shown in Figure 17 provides 1.80V–3.5V at 14A while maintaining output regulation within ±1%. The output voltage is determined by
connecting the five DAC inputs to the VID pins of the
processor. The power MOSFETs are sized to minimize
board space and allow operation without the need of a heat
sink. With proper airflow, ambient temperature conditions
of up to 50° Celsius are acceptable. Typical efficiency is
above 90% from 1A to 10A at 3.3V out. (see Figure 18).
Achieving higher output currents from LTC1553 based
designs is simply a matter of selecting appropriate MOSFETs and passive components.
LTC1553 Overview
The on-chip, 5-bit digital-to-analog converter (DAC) allows
the output voltage to be adjusted from 1.80V to 3.5V, as
shown in Table 1. Current limiting is maintained by sensing the voltage drop across the RDS(ON) of the high-side
MOSFET. The DAC accuracy, initial reference voltage
tolerance and internal feedback resistor tolerances result
in a maximum initial output voltage error of ±1% of the
selected output voltage. The line and load regulation plus
temperature drift over the 0°C to 70°C temperature range
will contribute another ±1% to the output error budget.
This gives a total static operating error of less than ±2%,
providing sufficient headroom (3%) for the dynamic
response to remain within a ±5% output voltage tolerance,
while still requiring a reasonable amount of output
capacitance.
It pays to look at the regulator design from two perspectives: electrical and thermal. Most processor applications
operate at average currents that are approximately 80% or
less of the specified peak current. As such, the thermal
VIN = 5V
10µF +
0.1µF
5.6k
5.6k
+
1N5817
RIMAX
5.6k
VCC
IMAX
PWRGD
PVCC
0.1µF
Q1A, Q1B
(2 IN PARALLEL)
G1
LO
2.0µH/18A
FAULT
Pentium Pro
Processor
SYSTEM
CIN
990µF
3 × 330µF
OT
VOUT
IFB
LTC1553
VID0–VID4
Q2
5V
C0
2310µF
7 × 330µF
G2
OUTEN
COMP
SS
GND
PGND
SENSE
+
1.8k
CONNECTING
VID0–VID4
TO DIP SWITCH
TO SET VOUT
DALE NTHS-1206N02
(605) 665-9301
C1
100pF
RC
20k
CC
0.01µF
CSS
0.01µF
0.1µF
Q1A, Q1B, Q2: MOTOROLA MTD20N03HDL
(800) 441-2447
1552_06.eps
Figure 17. Typical 5V to 1.8V–3.5V/14A LTC1553 Application
AN84-14
www.BDTIC.com/Linear
Application Note 84
100
Table 1. Output Voltage vs VIDx Code
VID3
VID2
VID1
VID0
1
1
1
1
1
1
1
0
1
1
0
1
1
1
0
0
1
0
1
1
1
0
1
0
1
0
0
1
1
0
0
0
0
1
1
1
0
1
1
0
0
1
0
1
0
1
0
0
0
0
1
1
0
0
1
0
0
0
0
1
0
0
0
0
1
1
1
1
1
1
1
0
1
1
0
1
1
1
0
0
1
0
1
1
1
0
1
0
1
0
0
1
1
0
0
0
0
1
1
1
0
1
1
0
0
1
0
1
0
1
0
0
0
0
1
1
0
0
1
0
0
0
0
1
0
0
0
0
*Reserved for future expansion
90
(VDC)
*
*
*
*
*
*
*
*
*
*
1.80
1.85
1.90
1.95
2.00
2.05
No CPU
2.1
2.2
2.3
2.4
2.5
2.6
2.7
2.8
2.9
3.0
3.1
3.2
3.3
3.4
3.5
design can be based on the lower current level. Higher
currents, while present, are typically not of sufficient
duration to significantly heat the power devices. The
design does, however, need to be capable of delivering the
peak current without entering current limit or resulting in
device failures. Keep in mind that the power dissipation in
a resistive element, such as a MOSFET, varies as the
square of load current. As such, raising the load current
from 80% to 100% translates to approximately 56% more
power dissipation (1/0.82). Designing for this higher ther-
80
EFFICIENCY (%)
VID4
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
70
60
50
40
30
20
10
0
0.1
1
10
LOAD CURRENT (A)
100
1552_07.eps
Figure 18. Efficiency Plot for Figure 17’s Circuit
mal load results in a huge, and most likely unnecessary,
design margin. A good understanding of your system
requirements can result in substantial savings in the size
and cost for the power supply.
RIMAX sets current limit to the desired level. Add one-half
of the inductor ripple current to the maximum load current
to determine the peak switch current. Multiply this current
by the maximum on-resistance of the selected MOSFET
switch to determine the minimum current limit threshold
voltage. It’s a good idea to add at least a 10% margin to this
limit. Also, be sure to use the hot on-resistance of the
MOSFET. A multiplier of about 1.4 times the room temperature RDS(ON) should be used to determine the hot
resistance. In the case of two parallel MTD20N03HDLs
(Q1A and Q1B), the cold resistance is approximately
0.035Ω each; therefore, assume the hot resistance to be
approximately 0.050Ω. Divide this by two because the
FETs are in parallel. The threshold voltage is programmed
by multiplying the IMAX pin’s sink current by the value of
RIMAX. Since we now can determine the required threshold, we need to calculate the value of RIMAX. Use the
specified minimum sink current, 150µA, to calculate the
resistor value.
The soft-start time is programmed by the 0.01µF cap
connected to the SS pin. The larger the value of this
capacitor, the slower the turn-on ramp.
Inductor LO is sized to handle the full load current, up to the
onset of current limit, without saturating. A value of
between 2µH and 3µH is adequate for most processor
supply designs. Be careful not to overspecify the inductor.
www.BDTIC.com/Linear
AN84-15
Application Note 84
The inductor need not retain its no-load inductance up to
the current-limit threshold. If the inductor still retains on
the order of 25% to 30% of its initial inductance under
worst-case short-circuit current conditions, the circuit
should prove reliable. However, you do want to ensure
that approximately 60% to 75% of the initial inductance is
retained at nominal full load. Excessive inductance roll-off
will result in higher than expected output ripple voltage at
high loads, along with increased dissipation in the power
FETs and the inductor itself.
Proper loop compensation is critical for obtaining optimum transient response while ensuring good stability
margins. The compensation network shown here gives
good response when used with the inductor and the
output capacitors values shown in Figure 17. Several low
ESR capacitors are placed in parallel to reduce the total
output ESR, resulting in lower output ripple and improved
transient performance. Generally speaking, low ESR, high
value output capacitors should be chosen to optimize the
use of board space. However, if the ESR value is too low
for a given capacitor value, loop stability problems can
occur. The feedback loop depends on the frequency of the
ESR “zero” being well below the loop crossover frequency. There is 45° of positive phase shift at the frequency where the capacitive reactance equals the ESR of
the capacitor. Without this phase shift, the loop would be
impossible to stabilize. Low ESR, AVX TPS-series tantalum capacitors are a very good compromise between
ESR, capacitance value and physical size.
Input capacitors are included to suppress the input switching noise and to keep the input 5V supply variation to a
minimum during the Q1 ON/OFF cycle. Excessive conSYNCHRONIZING LTC1430s FOR REDUCED RIPPLE
by Craig Varga
The recent move to split-plane microprocessors by several
CPU makers has led to the inclusion of multiple switching
regulators on many motherboard designs. These regulators typically provide 3.3V for system logic and a separate
supply for the processor core. Current requirements of
5A–10A or more per supply are not unusual. The LTC1430
synchronous buck regulator is commonly used to provide
these tightly regulated supplies. By nature, the input
current waveform in the buck topology is discontinuous,
AN84-16
ducted emissions are usually traced back to inadequate
input capacitance or poor layout of the power-path traces.
The crucial parameter for the input capacitors is ripple
current rating. A reasonable rule of thumb says that the
input capacitor ripple current is going to be approximately
50% of the load current. Therefore, in a typical Pentium
Pro processor application, the input capacitors should be
rated for close to 7ARMS. An excellent choice for the input
capacitors are Sanyo OS-CONs or the equivalent. They
have extremely high ripple current ratings for their size
and have demonstrated excellent reliability in this type of
application. Low ESR aluminium electrolytic capacitors
are a viable option from both input and output. Although
lower in cost than OS-CONs or tantalum capacitors, their
long-term reliability is not as good. Using 105°C capacitors and keeping operating temperatures low will help to
obtain reasonable capacitor life.
The combination of the Dale NTHS-1206N02 thermistor
and the 1.8k resistor are for overtemperature monitoring.
The OT flag trips if the ambient temperature at Q1 reaches
about 90°C; at 100°C the G1 and G2 drivers stop operating. If the system monitors the OT flag, there should be
ample time to take precautions, saving data and system
configuration information prior to an overtemperature
shutdown. Alternatively, CPU activity could be reduced,
lowering power supply current and allowing the supply to
cool down.
The PWRGD pin gives the CPU rail-voltage OK indication.
If, for any reason, the output regulation falls out of the ±5%
limit (including an overtemperature shutdown), PWRGD
will provide a logic low signal to the system monitor.
resulting in large input ripple current. By synchronizing a
pair of supplies out of phase, it is possible to achieve a
degree of ripple current cancellation. This results in less
stress on the input capacitors (the number of input capacitors could be reduced) and lower EMI. The ripple is easier
to filter since the frequency is effectively doubled and the
peak-to-peak current is reduced.
It is extremely simple to synchronize a pair of LTC1430s
in an appropriate phase relationship. Simply connect a
resistor divider from the low gate drive of a “master”
www.BDTIC.com/Linear
Application Note 84
12V
5V
U1
LTC1430
14
11
12
10
PVCC1
Vcc
G1
FSET
IFB
IMAX
SD
G2
PGND
COMP
9 SS
4 SGND
–SENS
1
MASTER
Q1
L1
13
OUT 1
16
Q2
C3
+
8
R1
130k
PVCC2
C2
2
+
15
3
5
SLAVE
+SENS 7
FB 6
R2
15k
R3
10k
12V
5V
Figure 20. Phase Relations Between the Switching Nodes
of the Two Regulators
U2
LTC1430
14
11
12
10
PVCC1
Vcc
G1
FSET
IFB
IMAX
G2
SD
COMP
9 SS
4 SGND
PGND
–SENS
1
Q3
L2
13
16
3
OUT 2
Q4
C4
+
8
PVCC2
C1
2
+
15
5
DI1430_01.eps
+SENS 7
FB 6
Figure 19. Simplified Schematic Diagram
of Synchronization Circuitry
regulator to the sync pin of a “slave” regulator. The
resistors should divide the gate-drive voltage down to
something slightly less than the VCC supply of the slave
regulator, typically from 12V down to approximately 4.5V.
Total divider resistance of 20k to 30k is adequate. Also, the
slave regulator must be set up to free run slower than the
master regulator. If, for example, the master is configured
to run at approximately 300kHz (a 130k resistor from
FSET to ground) the slave can be left to run at its natural
frequency of 200kHz. The slave frequency will be forced
up to that of the master.
The sync function on the LTC1430 works as follows: when
the shutdown pin is pulled low, the high-side switch turns
off; normal duty factor control determines when the highside switch will turn back on. As long as the shutdown pin
is held low for less than approximately 40µs, the chip will
not shut down.
The simplified schematic (Figure 19) shows the synchronization circuitry. For a detailed description of LTC1430based regulator designs, see the LTC1430 data sheet. The
scope photo (Figure 20) shows the voltage at the common
connection of the two FETs of each regulator.
www.BDTIC.com/Linear
AN84-17
Application Note 84
COMBINE A SWITCHING REGULATOR AND AN
ULTRAFAST LINEAR REGULATOR FOR A HIGH
PERFORMANCE 3.3V SUPPLY
by Craig Varga
power state to full load in several clock cycles. Generally,
switching regulators are used to supply such high power
devices, because of the unacceptable power losses associated with linear regulators. Unfortunately, switching
regulators exhibit much slower transient response than
linear regulators. This greatly increases the output capacitor requirements for switchers.
Introduction
It is becoming increasingly necessary to provide low
voltage power to microprocessor loads at very high
current levels. Many processors also exhibit high speed
load transients. The Pentium® Pro processor from Intel
exhibits both of these requirements. This processor
requires 3.3V ±5% at approximately 14A peak (9A average) and is capable of making the transition from a low
Circuit Operation
The circuit shown in Figure 21 takes advantage of a new,
ultrahigh speed linear regulator combined with a switching regulator to get the best of both worlds. An LTC1435
synchronous buck regulator is combined with an LT1575
12V
+
C11
150µF
16V
+
C12
150µF
16V
+
C13
150µF
16V
C17
1µF
+
C14, 150µF, 16V
C16
1µF
13
9
1
C8, 68pF
C7, 0.1µF
2
3
C10, 1000pF 4
C9
1500pF
R5
16.5k
5
6
C15
1µF
VIN
EXTVCC
TG
COSC
SW
RUN/SS
BOOST
ITH
INTVCC
SFB
U2
LTC1435
SGND
VOS
BG
S
PGND
10
16
Q2
L1
4µH
14
C3, 0.1µF
15
12
D1, CMDSH-3
11
R6
7.5mΩ
R3
100
Q3
C18
1000µF
10V
C20 +
1000µF
10V
+
C19
1000µF
10V
+ 8
S–
C2, 1000pF
7
+
C4, 4.7µF
D2
MBRS330T3
C5
0.1µF
R8
15K
2
IPOS
S/D
8
7
U1
INEG
LT1575
3
6
GND
GATE
12V
4
C6
0.1µF
VIN
FB
COMP
Q1
IRLZ44
5
C21, 10pF
R9
2k
C22
1000pF
R1
2.1k, 1%
C1, 470pF
R2
1.21k
1%
Figure 21. 12V to 3.3V/9A (14A Peak) Hybrid Regulator
AN84-18
C23
1µF
R7
35.7k
1
L1 = COILTRONICS CTX01-13199-X2
(561) 241-7876
Q2, Q3 = SILICONIX SUD50N03-10
(800) 544-5565
+
R4
100
www.BDTIC.com/Linear
DI1575_01.eps
3.3V
VCORE
40 × 1µF
X7R
CERAMIC
0805 CASE
Application Note 84
100
SWITCHER EFFICIENCY
EFFICIENCY (%)
90
80
50mV/DIV
TOTAL EFFICIENCY
70
60
50
0
2
4
6
8
10
LOAD CURRENT (A)
12
200µs/DIV
14
DI1575_02.eps
Figure 22. Efficiency of Figure 21’s Circuit
linear regulator to generate a 3.3V output from a 12V input
with an overall conversion efficiency of approximately
72%. The output is capable of current slew rates of
approximately 20A per microsecond.
The LT1575 uses an IRLZ44 MOSFET as the pass transistor, allowing the dropout voltage to be less than 550mV.
Setting the switching supply’s output to only 700mV
above the output of the linear regulator ensures output
regulation. The switcher is therefore set up to deliver 4.0V
at 14A from the 12V supply. Conversion efficiency of the
switcher is around 90% (depending on load), whereas the
LT1575’s efficiency is 82.5% (see Figure 22). The 12V
input current is only about 5.5A. At an average current of
9A, the power dissipation in the linear pass transistor is
only 6.3W. A small stamped aluminum heat sink is adequate.
THE LTC1624: A VERSATILE, HIGH EFFICIENCY, SO-8
N-CHANNEL SWITCHING REGULATOR CONTROLLER
by Randy G. Flatness
Introduction
The LTC1624 is a current mode switching regulator controller operating at an internally set frequency of 200kHz.
This versatile 8-pin controller uses the same constant
frequency current mode architecture and Burst Mode
operation as the LTC1435–LTC1439 controllers, but without the synchronous switch. The LTC1624, like the other
members of the family, drives a cost-effective, external Nchannel MOSFET for the topside switch and maintains low
dropout operation previously available only with P-channel MOSFETs.
Figure 23. Transient Response of Figure 21’s Circuit
to a 10A Load Step
Figure 23 shows the transient response to a 10A load step
with a rise time of approximately 50ns. The only output
capacitance is 40, 1µF ceramic capacitors. No additional
bulk capacitance is required at the processor. The circuit
eliminates approximately a dozen low ESR tantalum
capacitors at the load, which would be required without
the linear postregulator. The switching supply’s output is
decoupled with three aluminum electrolytic capacitors.
Because the transient response at this point is much less
critical than at the load, the long-term degradation of the
aluminum capacitors will not be as detrimental to the
circuit’s performance as it would be if they were used for
load decoupling.
The LTC1624 can be configured to operate in all standard
switching configurations, including boost, step-down,
inverting, SEPIC and flyback, without a limitation on the
output voltage. A wide input voltage range of 3.5V to 36V
allows operation from a variety of power sources, from as
few as four NiCd cells up though high voltage wall adapters. Tight load regulation, coupled with a reference voltage
trimmed to 1%, provides very accurate output voltage
control.
Application Circuits
The LTC1624 can be used in a wide variety of switching
regulator applications, the most common being the stepdown converter. Other switching regulator architectures
discussed here include step-up and SEPIC converters.
www.BDTIC.com/Linear
AN84-19
Application Note 84
100
VIN
1000pF 4.5V TO 25V
RC
5.1k
SENSE –
CC, 570pF
2 I /RUN
TH
3
100pF
4
VIN
BOOST
LTC1624
VFB
TG
SW
GND
8
VIN = 5V
90
+
RSENSE
0.05Ω
7
6
M1
Si4412DY
CB
0.1µF
5
CIN
22µF
35V
x2
EFFICIENCY (%)
1
L1
10µH
D1
MBRS340T3
VOUT
3.3V/2A
R2
35.7k
80
VIN = 10V
70
60
+
R1
20k
VIN = 20V
COUT
100µF
10V
x2
50
1mA
10mA
100mA
1A
LOAD CURRENT
10A
1624_07.eps
Figure 25. Efficiency Plot
of Figure 24’s Circuit
1624_06.eps
Figure 24. High Performance 3.3V/2A Step-Down DC/DC Converter
The basic step-down converter is shown in Figure 24. This
application shows a 3.3V/2A converter operating from an
input voltage range of 4.5V to 25V. The efficiency for this
circuit is shown in Figure 25.
Step-up and SEPIC applications require a low-side switch
pulling the inductor to ground (see Figures 26 and 28).
Since the source of the MOSFET must be grounded, the
switch pin (SW) on the LTC1624 is also grounded in order
for the driver to supply a gate-to-source signal to control
the MOSFET. In these applications, the voltage on the
boost pin is a constant 5V, resulting in a 0V–5V gate-drive
level. A capacitor from boost to switch is still required,
since this capacitor supplies the gate-charge currents.
The basic step-up converter is shown in Figure 26. The
LTC1624 is used to create 12V/1A from a 5V source with
the efficiency shown in Figure 27. Efficiency is above 90%
from 20mA up to close to full load, dropping only to 89%
at 1A.
In order to allow input voltages both above and below the
output voltage, a SEPIC converter can be used. An example
of the LTC1624 used as a 12V/0.5A SEPIC converter
operating from an input range of 5V to 20V is shown in
Figure 28.
CIN
22µF
35V x2
VIN
5V
CC, 330pF
2
3
100pF
4
SENSE –
ITH/RUN
VIN
GND
1000pF
RSENSE
0.04Ω
7
BOOST
LTC1624
VFB
8
L1
20µH
SW
5
VOUT
12V/1A
D1
MBRS130LT3
6
TG
+
1
RC
5k
CB
0.1µF
M1
Si4412DY
R2
35.7k
1%
1624_08.eps
+
R1
3.92k
1%
Figure 26. 12V/1A Step-Up Converter
AN84-20
www.BDTIC.com/Linear
COUT
100µF
16V
x2
Application Note 84
100
EFFICIENCY (%)
90
80
70
60
50
1mA
10mA
100mA
LOAD CURRENT
1A
1624_09.eps
Figure 27. Efficiency Plot for Figure
26’s Circuit
1000pF
2
3
100pF
4
SENSE –
ITH/RUN
VIN
GND
RSENSE
0.082Ω
7
BOOST
LTC1624
VFB
8
+
CC, 330pF
L1a
6
TG
SW
D1
MBRS130LT3
VOUT
12V/0.5A
+
1
RC
10k
CIN
22µF
35V x2
VIN
5V TO 15V
5
CB
0.1µF
M1
Si4412DY
22µF
35V
L1b
R2
35.7k
1%
L1a, L1b:CTX50-4
1624_10.eps
R1
3.92k
1%
+
COUT
100µF
16V
x2
Figure 28. 12V/0.5A DC/DC Converter Operates from 5V–15V Inputs
LOW COST 3.3V TO 1.XV 6 AMP POWER SUPPLY
by Sam Nork
As voltage requirements for microprocessors drop, the
need for high power DC/DC conversion from a 3.xV supply
to a lower voltage keeps growing. The LTC1430 is a very
attractive choice for such DC/DC applications, due to its
low cost, high efficiency and high output power capability.
However, there are two problems: first, 3.xV does not
provide enough gate drive to ensure low RDS(ON) using
external logic-level FETs; and second, the LTC1430 has a
4V minimum input requirement. These obstacles are both
overcome by using an LTC1517-5 regulated charge pump
to generate the input voltage for the LTC1430.
The circuit shown in Figure 29 uses the LTC1430 to
produce a synchronous 3.3V to 1.9V step-down DC/DC
converter. The circuit achieves 90.5% efficiency at 3 amps
of output current and has a 6 amp maximum output
capability. (Refer to the LTC1430 data sheet for detailed
description of LTC1430-based designs). Power for the
LTC1430 is derived from the output of the LTC1517-5.
The LTC1517-5 is a switched capacitor charge pump
available in a tiny, 5-pin SOT-23 package. The part uses
Burst Mode operation to generate a 5V output from a 2.7V
to 5V input.The regulated 5V supply powers the internal
circuitry of the LTC1430 and ensures that the LTC1430 can
www.BDTIC.com/Linear
AN84-21
Application Note 84
overall shutdown current below 10µA plus external FET
leakage. (For further reductions in shutdown current, an 8pin LTC1522 may be used in place of the LTC1517-5; the
LTC1522 is the same as an LTC1517-5 with shutdown.)
The additional LTC1517-5 circuitry will not take up much
board space. The entire circuit consumes only 0.045 in2.
100
90
Pulling the SHDN pin on the LTC1430 low will shut down
the power supply. Q1 and Q2 will be forced off and the
LTC1430 quiescent current will drop to 1µA. Although the
LTC1517-5 does not have a shutdown feature, the no-load
operating current is an extremely low 6µA. This keeps the
70
60
50
40
0.1
1
LOAD CURRENT (A)
10
1517_02.EPS
Figure 30. Efficiency of Figure 29’s Circuit
VIN
3.3V
C1
Y5V CERAMIC
3.3µF
2
+
5
1
LTC1517-5
3
4
C3
0.22µF
C6 TO C9*
330µF
6.3V
C10
×4
0.1µF
R2
24k
C2
Y5V CERAMIC
10µF
C11
1µF
L1
2.4µH, 8A
SUMIDA
CDRH127-2R4
Q1
Si4410
5V
D1
BAT54
C5
1µF
OFF
R1
100Ω
Q2
Si4410
R3
1k
C4
0.1µF
ON
VIN = 3.3V
VOUT = 1.9V
80
EFFICIENCY (%)
provide adequate gate drive to the external N-channel
FETs. With insufficient gate drive, output power and efficiency will be significantly reduced due to high RDS(ON) of
the FETs. In this circuit, typical supply current drawn by the
LTC1430 is between 25mA and 30mA, the vast majority of
which is needed to charge and discharge the external FETs.
Because the LTC1517-5 has a maximum effective output
impedance of 50Ω, this current can be comfortably supplied from a 3.3V input. If the input voltage drops to 3V or
lower, the LTC1517-5 output may also drop. However,
with the FETs shown in Figure 29, the LTC1517-5 will
provide a 4.5V minimum supply to the LTC1430 at input
voltages down to 3V. The circuit’s efficiency is shown in
Figure 30.
1
2
3
4
5
6
7
8
LTC1430CS
16
G1
G2
15
PVCC1
PVCC2
14
PGND
VCC
13
IFB
GND
12
IMAX
SENSE –
11
FB
FREQSET
10
SENSE +
COMP
9
SHDN
SS
C12
0.1µF
D2
MBRS120
C16
0.018µF
R5
4.99K
1%
VOUT
1.9V
6A
+
R6
10K
1%
C17 TO C21 *
330µF
6.3V
×5
R4
5.2k
C13
390pF
C14
0.012µF
+
C15
10µF
10V
1517 TA03
*AVX TPS TANTALUM
(207) 282-5111
Figure 29. 3.3V to 1.9V/6A Power Supply
AN84-22
www.BDTIC.com/Linear
Application Note 84
THE LT1374: NEW 500kHz, 4.5A MONOLITHIC
BUCK CONVERTER
by Karl Edwards
Introduction
The LT1374 is a 4.5A buck converter using an on-chip
80mΩ switch. With its 500kHz operating frequency and
integral switch, only a few external, surface mount components are required to produce a complete switching regulator. The LT1374’s features include current mode control,
external synchronization and a low current (typically 20µA)
shutdown mode. Improvements have been made to reduce
start-up headroom and switching noise. A novel power
device layout makes it possible to fit a high speed, bipolar,
80mΩ switch into a surface mount SO-8 package. The
LT1374 is also available in DD and TO-220 packages for
higher power applications.
Application: 5V/4.25A Buck Converter
With its 25V input and 4.5A minimum switch current, the
LT1374 will fit into a wide range of applications. Figure 31
shows a typical buck converter with a 6V to 25V input
range, a 5V output and 4.25A of output current capability.
Due to the low on-resistance of the switch, efficiency
remains high over a wide range of currents, as shown in
Figure 32. To reduce power dissipation, both the BIAS pin
and boost circuit are supplied from the 5V output.
Several factors, including maximum current, core and
copper losses, size and cost, affect the choice of inductor,
L1. A high value, high current inductor gives the highest
output current with the lowest ripple, at the expense of a
large physical size and cost. Lower inductance values tend
to be physically smaller, have higher current ratings and
are cheaper, but output ripple current, and hence ripple
voltage, increases.
The input capacitor, C3, experiences very high ripple
currents, up to IOUT/2, so low ESR tantalum capacitors are
needed. At 4.25A output current, two capacitors in parallel
are required to meet the ripple current requirement. The
ripple current in the output capacitor, C1, is lower, but its
ESR still needs to be low to limit output voltage ripple. The
voltage drop across the catch diode, D1, has a significant
effect on overall converter efficiency, especially at higher
input voltages when the switch duty cycle is low. Its ability
to survive short-circuit conditions may increase its power
rating. For good electrical performance, D1 must be placed
close to the LT1374. The power dissipated in D1 will raise
the PC board’s temperature around the LT1374. This must
be taken into account when modeling or taking bench
measurements of die temperature.
D2
1N914
100
C3*
10µF TO
50µF
BOOST
+
R1
56k
LT1374-5 BIAS
SHDN
GND
R2
33k
L1**
5µH
FB
VC
CC
3.3nF
D1
MBRS330T3
95
OUTPUT**
5V/4.25A
VSW
VIN
+
C1
100µF, 10V
SOLID
TANTALUM
EFFICIENCY (%)
INPUT
6V TO 25V
C2
0.27µF
90
85
80
75
* RIPPLE CURRENT RATING > IOUT/2
** L1 = COILTRONICS UP2-4R7; (561) 241-7876
INCREASE L1 TO 10µH FOR LOAD CURRENTS ABOVE 3.5A AND TO 20µH ABOVE 4A
1374_02.EPS
70
0
0.5
2.5
1.0 1.5 2.0
LOAD CURRENT (A)
3.0
3.5
1374_03
Figure 31. 5V Buck Converter
Figure 32. Efficiency of Figure 31’s Circuit:
10V In, 5V Out
www.BDTIC.com/Linear
AN84-23
Application Note 84
PCB Layout
The loop compensation capacitor, CC, produces a pole in
the frequency response at 240Hz. Unity-gain phase margin can be further improved with the addition of a resistor,
typically 2k, in series with CC, adding a zero to the
frequency response. This, however, can cause a largesignal subharmonic problem in the loop. The output ripple
voltage feeds back through the error amplifier to the VC
pin, changing the current trip point of the next cycle. This
changes the voltage ripple at the output, and the loop is
closed. Adding a second capacitor directly from the VC pin
to ground to form a pole at one-fifth the switching frequency solves the problem.
All high current, high speed circuits require careful layout
to obtain optimum performance. When laying out the
PCB, keep the trace length around the high frequency
switching components as short as possible. This minimizes the EMI and RFI radiation from the loop created by
this path. These traces have a parasitic inductance of
approximately 20nH/inch, which can cause an additional
problem at higher operating voltages. At switch-off, the
current flowing in the trace inductance causes a voltage
spike. This is in addition to the input voltage across the
switch transistor. At higher currents, the additional voltage can potentially cause the output switching transistor
to exceed its absolute maximum voltage rating.
LTC1504: FLEXIBLE, EFFICIENT
SYNCHRONOUS SWITCHING REGULATOR
CAN SOURCE OR SINK 500mA
by Dave Dwelley
is included. The diminutive SO-8 package minimizes the
amount of space the LTC1504 fills while allowing adequate
thermal dissipation for 500mA load current levels. The
LTC1504 allows previously impossible (or at least awkward) tasks to be completed with ease.
Introduction
The LTC1504 is an 8-pin step-down switching regulator.
It consists of a 200kHz fixed frequency, voltage-feedback,
buck-mode switching regulator controller and a pair of
1.5Ω power switches in an 8-pin SO package. The LTC1504
also includes a synchronous rectifier on-chip, maximizing
efficiency and minimizing external parts count while allowing the output to both sink and source current: it can
source or sink up to 500mA with input voltages from 3.3V
to 10V and output voltages as low as 1.26V. The LT1504
can achieve 100% duty cycle at the output switch, maximizing dropout performance with low input-to-output
voltage differentials. The LTC1504 includes an onboard
precision reference and user-programmable currentlimit and soft-start circuits, allowing implementation of
full-featured power conversion circuits with a minimum of
external components.
Minimum Component-Count Circuits
Figure 33 shows a fully functional LTC1504 5V to 3.3V
regulator, including current limit and soft-start, using the
fixed-output LTC1504-3.3 and only six external components. Efficiency is above 90% with load currents between
SHUTDOWN
RIMAX**
68k
IMAX
VIN
5V
+
VCC
CIN
22µF
AN84-24
VOUT
3.3V/500mA
SW
LTC1504-3.3
GND
SS
SENSE
+
COMP
CSS*
O.1µF
The LTC1504 architecture is optimized for maximum
efficiency at loads above 50mA and does not include a
light-load Burst Mode™ circuit. This penalizes efficiency
at very light loads but allows the device to seamlessly shift
between sourcing and sinking current, opening up a whole
new class of applications. A micropower shutdown mode
LEXT 47µH
SHDN
COUT
100µF
CC
1000pF
CIN = AVX TPSC226M016R0375
COUT = SANYO 16CV100GX
LEXT = SUMIDA CD54-470
* OPTIONAL: DELETE TO DISABLE SOFT START
** OPTIONAL: DELETE TO DISABLE CURRENT LIMIT
1504_01.EPS
Figure 33. Minimum Parts-Count 5V–3.3V Converter
www.BDTIC.com/Linear
Application Note 84
110Ω
SHUTDOWN
RIMAX
68k
110Ω
IMAX
I
IMAX
VIN
5V
+
VCC
CIN
22µF
SHDN
MAX SHDN
LEXT 47µH
SHDN
VOUT
3.3V/500mA
SW
10µF
CERAMIC
LTC1504-3.3
GND
SS
CSS
O.1µF
+
SENSE
COMP
RC
7.5k
VCC
VCC
TERMPWR
COUT
100µF
LEXT 47µH
LTC1504
LTC1504
SENSE
GND
GND
FB
SS
COMP
SS COMP
15k
COUT = AVX TPSC107M006R0150
LEXT = SUMIDA CD54-470
110Ω
+
COUT
100µF
12k
7.5k
CF
220pF
110Ω
SW
SW
18
TO
27
LINES
220pF
0.01µF
1540_03.EPS
CC
0.01µF
1504_02EPS
CIN = AVX TPSC226M016R0375
COUT = AVX TPSE107M016R0125
LEXT = SUMIDA CD54-470
Figure 34. Improved Transient Response
50mA and 200mA, peaking at 92% at 100mA and remaining above 82% all the way to the maximum 500mA load.
Current limit is set at 500mA in this example; it can be
reduced by lowering the value of RIMAX. CSS sets the startup time at approximately 25ms.
The circuit in Figure 33 relies on the ESR of the output
capacitor to maintain loop stability with just a single
capacitor at the COMP pin. Figure 33 uses a surface mount
electrolytic capacitor with about 400mΩ ESR. A low ESR
tantalum output capacitor can improve the transient
response at the output but requires a more complex
compensation network at the COMP pin (Figure 34). There
is a tradeoff to be made here: the minimum component
count solution is the simplest and uses the least expensive
components but pays a penalty in transient response. The
low ESR circuit in Figure 34 has improved transient
response and actually uses less board space: the tantalum
output capacitor is smaller than the electrolytic device
used in Figure 33 and the additional compensation components are tiny 0603 surface mount devices.
Note that the input bypass capacitor in both Figures 33 and
34 is an AVX TPS type, a relatively costly surge-tested
tantalum capacitor. This is a small, surface mount device
that has a surge current rating adequate to support the
500mA maximum load current of the LTC1504. Buck
regulators (like the LTC1504) inherently draw large RMS
currents from the input bypass capacitor, and the capacitor
type chosen must be capable of withstanding this current
Figure 35. SCSI-2 Active Terminator
without overheating. As with all switching regulator circuits, layout is critical to obtaining maximum performance; if in doubt, contact the LTC Applications Department for component selection and layout advice.
Sink/Source Capability Improves SCSI Terminators
and Supply Splitters
Figure 35 shows an adjustable-output LTC1504 connected as a 2.85V regulator for use as a SCSI terminator.
The ability of the LTC1504 circuit to sink current makes it
ideal for use in terminator applications, where the load is
just as likely to be putting current into the regulator as
taking it out. The synchronous-buck architecture of the
LTC1504 allows it to shift cleanly between sourcing and
sinking current, making it ideal for such applications. The
small number of tiny external components required
minimizes the space used by the terminator circuit. A low
ESR output capacitor is used along with an optimized
compensation network to improve output transient
response and maintain maximum data fidelity.
SHUTDOWN
IMAX
I
SHDN
MAX SHDN
VCC
VCC
5V
10µF
CERAMIC
LEXT
47µH (22µH)*
SPLIT SUPPLY
2.5V ±500mA
SW
SW
LTC1504
LTC1504
SENSE
GND
GND
FB
SS
COMP
SS COMP
7.5k
11.8k
+
COUT
47µF
12.1k
220pF
0.01µF
1540_04.EPS
COUT = TAJC476M016R
LEXT = SUMIDA CDRH73-470 (LOWER RIPPLE/HIGHER EFFICIENCY)
*CDRH73-220 (FASTER TRANSIENT RESPONSE)
Figure 36. 5V Supply Splitter
www.BDTIC.com/Linear
AN84-25
Application Note 84
Substituting a different set of feedback resistors (Figure
35) creates a 5V supply splitter, which creates a 2.5V
“ground” to allow analog circuitry to operate from split
supplies. Op amp circuits and data converters like to
operate from dual supplies, and the sink/source capability
of the LTC1504 allows load currents to be returned directly
to the 2.5V “ground” supply.
HIGH EFFICIENCY DISTRIBUTED
POWER CONVERTER FEATURES SYNCHRONOUS
RECTIFICATION
by Dale Eagar
For input voltages ranging from 12V to 48V and output
voltages ranging from 1.3V to 36V, the LT1339 is a simple,
robust solution to your power-conversion problems. The
LT1339 is ideal for power levels ranging from tens of watts
to tens of kilowatts. The LT1339 is straightforward and
remarkably easy to use. This is one power converter that’s
not afraid of 20A, 50A or even 150A of load current.
Introducing the LT1339
The LT1339 is the buck/boost converter that needs no
steroids. As a full-featured switching controller, the LT1339
incorporates the features needed for system-level solutions. The LT1339 has an innovative slope-compensation
function that allows the circuit designer freedom in controlling both the slope and offset of the slope-compensation ramp. Additionally, the LT1339 has an average current limit loop that yields a constant output current limit,
regardless of input and/or output voltage. The LT1339’s
RUN pin is actually the input to a precision comparator,
giving the designer freedom to select an undervoltage
lockout point and hysteresis appropriate for the design.
The SYNC and SS (soft-start) pins allow simple solutions
to system-level design considerations. Like all Linear
Technology controllers, the LT1339 has anti-shootthrough circuitry that ensures the robustness that is
demanded in real-world applications for medium and high
power conversion.
10V TO
18V
+
CINPUT
1000µF
16V ×2
OS-CON
R1
100k
13
2
C1
1µF
RT
15k
CT 1500pF
RCOMP
4.7k
CCOMP
2200pF
4
3
5
CAVG 2200pF
7
10
C2
0.1µF
D1 1N914
*
20
12VIN
BOOST
RUN
TG
5VREF
TS
SLOPE
BG
IAVG
SENSE –
12
SYNC SGND PGND
15
RSENSE
0.002Ω
Q2–Q5
IRL3103D2
×4
D3
1N5817
ILIMIT
0.01Ω
0.005Ω
0.002Ω
10A
20A
50A
9
RREF
1k
5V
COUTPUT 50A
2200µF
6.3V
OS-CON
RFB
3K
1.66K
1.25K
450Ω
40Ω
RFB
3k
1339_01.EPS
Figure 37. 10V–18V In, 5V/50A Out Buck Converter
AN84-26
RSENSE
+
16
VC
FB
Q1
IRL3803
L1
10µH
50A
18
11
8
The circuit shown in Figure 37 is limited to 20V because of
the maximum rating (Abs Max) of the LT1339 VIN pin. The
input voltage can be extended above 20V by inserting a
10V Zener diode where the asterisk (*) is shown in Figure
37. This will extend the input voltage of Figure 37’s circuit
up to 30V (the Abs Max rating of the MOSFETs).
19
SENSE +
1
Higher Input Voltages
D2
1N5817
LT1339
VREF
Figure 36 details a typical low voltage buck converter. This
circuit has a VIN range of 10V to 18V with configurable
output current and voltage. This simple circuit delivers
250W of load power into a 5V load while maintaining
efficiencies in the mid-nineties.
C3
1µF
17
CT
Distributed Power
www.BDTIC.com/Linear
VOUT
5V
3.3V
2.8V
1.8V
1.3V
*SEE TEXT
Application Note 84
12V
(SUPPLIED SEPARATELY)
D1
1N914
+
C5
47µF
R1
100k
C3
1µF
17
13
2
C1
1µF
RT
15k
48V
4
3
Q8
FMMT720
20
RUN
TG
5VREF
TS
SLOPE
BG
19
Q9
FMMT619
18
CAVG 2200pF
RCOMP
4.7k
7
CT
10
C3
0.1µF
C2
2200pF
IAVG
VREF
8
10A
20A
50A
5V
50A
COUT
2200µF
6.3V
OS-CON
×4
Q3–Q6
IRFZ44
×4
R1
10k
D3 1N914
RFB
18.2K
8.66K
3K
1.66k
1.25k
RFB
SYNC SGND PGND
1
ILIMIT
+
D2 3.3V
C4
1µF
Q10
FMMT720
11
SENSE +
12
SENSE –
9
FB
VC
12V
16
LT1339
5
+
BOOST
12VIN
CT
1500pF
RSENSE
CINPUT
0.01Ω
1500µF
0.005Ω
Q1–Q2
63V
0.002Ω
IRFZ44
×4
×2
L1
D4 TO D14
10µH 50A RSENSE
0.002Ω
3A ×10
Q7
FMMT619
3k
RREF
1k
15
1339_02.EPS
VOUT
24V
12V
5V
3.3V
2.8V
Figure 38. 48V In, 5V/50A Out, High Power Buck Converter
VIN
15V–25V
CIN
1000µF 35V
×2
+
R13
2k
Q7 FZT849
C10
R1
100k 0.1µF
C1
0.1µF
2
R2
1k
RT
47k
4
6
CAVG
2.2nF
5
9
14
10
C11
22µF
35V
12V
SLOPE
SENSE+
11
SENSE–
12
BOOST
TG
TS
C3
3.3µF
Q3 Si4450
18
R9
10Ω
C6
1µF
T2**
BG
16
FB
VC
SYNC
1
SGND
8
Q5
8
7
4
6
2
C5
1µF
9
R5
PGND
15
U3
CNY17-3
Q1, Q2 = SUD50N03
L1 = 15 TURNS AWG20 77130-A7
* T1 = POWER TRANSFORMER
** T2 = GATE-DRIVE TRANSFORMER
(SEE FIGURE 4 FOR DETAILS)
+
COUT
220µF 10V
OS-CON
R11
10Ω
R12 1k
10
Si4539DY
Q6
PHASE
VREF
R10
10Ω
VOUT
5V/6A
Q2
12V
R14 10Ω
SS
IAVG
Q4 Si4450
19
U1
LT1339
CT
D4
MURS
120
C3
1µF
20
L1 7µH
T1*
D2
IN4148
3.3Ω
R4
33k
D3
MURS120
Q1
5VREF
SECONDARY
GROUND
+
13
17
RUN 12VIN
R3
560Ω
3
CT
230pF
RSENSE
0.02Ω
1/2W
D1
12V
ISOLATION
BARRIER
PRIMARY
GROUND
D5
1N914
R8
1k
D6 1N914
C7 1µF
C9 0.1µF
R7 1k
3
C4
220pF
V+
R6 560Ω
1
COLL
RFB
2.49k
8
U2
REF
LT1431
GND-S
5
GND-F
6
RREF
2.49k
Figure 39. Galvanically Isolated Synchronous Forward Converter (see Figure 40 for Details of T1 and T2)
www.BDTIC.com/Linear
AN84-27
Application Note 84
localized gate voltages above VT, the threshold voltage of
the bottom MOSFET. To defeat the physicists, we add 3.3V
of negative offset to the bottom gate drive, effectively
making the threshold of the bottom MOSFETs 3.3V harder
to reach (see Figure 38). This offset is provided by the 3.3V
Zener, 1µF capacitor, 10k resistor and the 1N914 diode
preceding the gate of the bottom MOSFETs.
T2: COILTRONICS VP1-1400
(500V ISOLATION)
T1: PHILIPS EFD20-3F3 CORE
Lp = 93µH, Al = 1150nH/T2 (NO GAP)
2MIL
POLYESTER
FILM
SECONDARY, 9 TURNS
TRIFILAR 26AWG
PRIMARY, 9 TURNS
TRIFILAR 26AWG
4
10
1
7
5
11
2
8
1500VDC ISOLATION
TUCK TAPE ENDS
6
The Synchronous Forward Converter
3
12
Figure 39 details a Galvanically isolated LT1339 synchronous forward converter. Operating at its rated load of 6V
at 5A, this circuit achieves 87% efficiency with a 15V input
and 85% efficiency with a 24V input. Figure 40 shows
details of the transformers used in Figure 39’s circuit.
9
1339 04 .eps
Figure 40. Transformer Details of Figure 39’s Circuit
Blame it on the Physicists
The Synchronous Boost Converter
As the input voltage approaches 30V, the bottom MOSFETs will begin to exhibit “phantom turn-on.” This
phenomenon is driven by the instantaneous voltage step
on the drain, the ratio of CMILLER to CINPUT, and yields
The LT1339 becomes a synchronous boost controller
when the PHASE pin is grounded. Figure 41 details a 250W
boost converter that outputs 28V at 9A from a 5V supply.
Q1 TO Q2
IRF3205 ×2
Q7 FMMT720
VIN
5V/60A
RSENSE
0.002Ω
D1
1N914
Q8 FMMT619
+
L1
40µH
C3
1µF 20
BOOST
19
CINPUT +
220µF 6.3V
×4
18
VOUT
28V/8.5A
12V (SUPPLIED SEPARATELY)
+
+
17
12VIN
TG
RUN
R1
100k
13
C7
47µF
16V
TS
RFB
27k
12V
Q9
FMMT619
C4
1µF
U1
LT1339
16
Q3 TO Q6
IRF3205 ×4
BG
FB
5VREF
SLOPE
9
C5
10pF
4
R3 100Ω
RT
10k
12
SENSE+
CT
11
SENSE–
VC
10
14
C6
0.1µF
VREF
PHASE
IAVG
SS
SYNC PGND SGND
15
1
8
RREF
1.2k
2
Q10
FMMT720
R2 100Ω
CT
2200pF
3
7
RCOMP
5
7.5k
6
+
CSS
10µF
CAVG
2.2nF
CCOMP
1.5nF
L1 = 12T 4× AWG12 ON 77437-A7
Figure 41. This 5V to 28V Synchronous Boost Converter Limits Input Current at 60A (DC)
AN84-28
COUT
2200µF
35V ×6
www.BDTIC.com/Linear
C1
1µF
Application Note 84
FIXED FREQUENCY, 500kHz, 4.5A STEP-DOWN
CONVERTER IN AN SO-8 OPERATES FROM A 5V INPUT
by Karl Edwards
Introduction
The LT1506 is a 500kHz monolithic buck mode switching
regulator, functionally identical to the LT1374 but optimized for lower input voltage applications. Its high 4.5A
switch rating makes this device suitable for use as the
primary regulator in small to medium power systems. The
small SO-8 footprint and input operating range of 4V to
15V is ideal for local onboard regulators operating from 5V
or 12V system supplies. The 4.5A switch is included on the
die, along with the necessary oscillator, control and logic
circuitry to simplify design. The part’s high switching
frequency allows a considerable reduction in the size of
external components, providing a compact overall solution.
The LT1506 is available in standard 7-pin DD and fusedlead SO-8 packages. It maintains high efficiency over a
wide output current range by keeping quiescent supply
current to 4mA and by using a supply-boost capacitor to
saturate the power switch. The topology is current mode
for fast transient response and good loop stability. Full
cycle-by-cycle short-circuit protection and thermal shutdown are provided. Both fixed 3.3V and adjustable output
voltage parts are available.
INPUT
5V
C3
10µF TO
50µF
CERAMIC
BOOST
VIN
+
OPEN
OR
HIGH
= ON
The circuit in Figure 44 uses multiple LT1506s to produce
a 5V, 12A power supply. There are several advantages to
using a multiple switcher approach compared to a single
larger switcher. The inductor size is considerably reduced.
Inductor size is proportional to the energy that needs to be
stored in the core. Three 4A inductors store less energy
(1/2Li2) than a single 12A coil, so they are much smaller.
In addition, synchronizing three converters 120° out of
phase with each other reduces input and output ripple
currents. This reduces the ripple rating, size and cost of
the filter capacitors.
90
L1
5µH
85
OUTPUT
3.3V
4A
LT1506-3.3
SENSE
VC
+
CC
1.5nF
Current Sharing Multiphase Supply
D2
1N914
VSW
SHDN
GND
The circuit in Figure 42 is a step-down converter suitable
for use as a local regulator to supply 3.3V logic from a 5V
power bus. The high efficiency, shown in Figure 43, removes the need for bulky heat sinks or separate power
devices, allowing the circuit to be placed in confined
locations. Since the boost circuit only needs 3V to operate,
the boost diode can still be connected to the output,
improving efficiency. Figure 42’s circuit shows the shutdown pin option. If this pin is pulled to a logic low, the
output is disabled and the part goes into shutdown mode,
reducing supply current to 20µA. An internal pull-up ensures
correct operation when the pin is left open. The SYNC pin,
an option for the DD package, can be used to synchronize
the internal oscillator to a system clock. A logic-level clock
signal applied to the SYNC pin can synchronize the switching frequency in the range of 580kHz to 1MHz.
D1
MBRS330T3
C1
100µF, 10V
SOLID
TANTALUM
1506 TA01
EFFICIENCY (%)
C2
0.68µF
5V to 3.3V Buck Converter
80
75
70
0
Figure 42. 5V to 3.3V Step-Down Converter
1
2
3
LOAD CURRENT (A)
4
Figure 43. Efficiency vs Load Current for Figure 42’s Circuit
www.BDTIC.com/Linear
AN84-29
Application Note 84
C1, C3: MARCON THCS50E1E106Z
D1: ROHM RB051L-40
D2: 1N914
L1: DO3316P-682
3-BIT RING
COUNTER
1.8MHz
INPUT
6V TO 15V
LT1506-SYNC
LT1506-SYNC
LT1506-SYNC
VC SYNC SW GND VIN BOOST FB
VC SYNC SW GND VIN BOOST FB
VC SYNC SW GND VIN BOOST FB
R1
5.36k
1% +
+
+
C3A
10µF
25V
D1A
+
C3B
10µF
25V
+
D1B
D2A
L1B
6.8µH
C2B
330nF
10V
+
+
C2A
330nF
10V
C3C
10µF
25V
C1
10µF
25V
D1C
+
L1A
6.8µH
C4
68nF
25V
R2
4.99k
1%
5V
12A
D2B
L1C
6.8µH
C2C
330nF
10V
D2C
1506 F15
Figure 44. Current-Sharing 5V/12A Supply
Current Sharing/Split Input Supplies
Synchronized Ripple Currents
Current sharing is accomplished by connecting the VC
pins to a common compensation capacitor. The output of
the error amplifier is a gm stage, so any number of devices
can be connected together. The effective gm of the composite error amplifier is the product of the individual
devices. In Figure 44, the compensation capacitor, C4, has
been increased by 3×. Tolerances in the reference voltages
cause small offset currents to flow between the VC pins.
The overall effect is that the loop regulates the output at a
voltage somewhere between the minimum and maximum
references of the devices used. Switch-current matching
between devices will be typically better than 300mA over
the full current range. The negative temperature coefficient
of the VC-to-switch-current transconductance prevents
current hogging.
A ring counter generates three synchronization signals at
600kHz, 33% duty cycle, phased 120° apart. The sync
input will operate over a wide range of duty cycles, so no
further pulse conditioning is needed. At full load, each
device’s input ripple current is a 4A trapezoidal wave at
600kHz, as shown in Figure 45. Summing these waveforms gives the effective input ripple for the complete
system. The resultant waveform, shown at the bottom of
Figure 45, remains at 4A but its frequency has increased
to 1.8MHz. The higher frequency eases the requirements
on the value of input filter without the 3× increase in ripple
current rating that would normally occur. Although only a
single input capacitor is required, practical layout restrictions usually dictate an individual capacitor at each device.
Figure 46 shows the output ripple current waveforms. The
resultant 1.8MHz triangular waveform has a maximum
amplitude of 350mA at an input voltage of 10V. This is
significantly lower than would be expected for a 12A
output. Interestingly, at inputs of 7.6V and 15V, the
theoretical summed output ripple current cancels completely. To reduce board space and ripple voltage, C1 and
C3 are ceramic capacitors. Loop compensation capacitor
C4 must be adjusted when using ceramic output capacitors, due to the lack of effective series resistance (ESR).
A common VC voltage forces each LT1506 to operate at the
same switch current, not at the same duty cycle. Each
device operates at the duty cycle defined by its input
voltage. This is a useful feature in a distributed power
system. The input voltage to each device could vary due to
drops across the backplane, copper losses, connectors
and so on. The common VC signal ensures that loading is
still shared between the devices.
AN84-30
www.BDTIC.com/Linear
Application Note 84
Redundant Operation
The typical tantalum compensation value of 1.5nF is
increased to 22nF (×3) for the ceramic output capacitor. If
synchronization is not used and the internal oscillators
free run, the circuit will operate correctly, but ripple
cancellation will not occur. Input and output capacitors
must be ripple rated for the individual output currents.
The circuit shown in Figure 44 is fault tolerant when
operating at less than 8A of output current. If one power
stage fails open circuit, the output will remain in regulation. The feedback loop will compensate by raising the
voltage on the VC pin, increasing the switch current of the
two remaining devices.
PHASE 1
CURRENT
CURRENT
PHASE 1
TIME
TIME
PHASE 2
CURRENT
CURRENT
PHASE 2
TIME
TIME
PHASE 3
CURRENT
CURRENT
PHASE 3
TIME
TIME
TOTAL
CURRENT
CURRENT
TOTAL
TIME
Figure 45. Input Current
TIME
Figure 46. Output Current
www.BDTIC.com/Linear
AN84-31
Application Note 84
VID VOLTAGE PROGRAMMER
FOR INTEL MOBILE PROCESSORS
by Peter Guan
Each VID pin must be grounded or driven low to produce
a digital low input, whereas a digital high input can be
generated by either floating the VID pin or connecting it to
VCC. The LTC1706-19 is fully TTL compatible and operational over a VID input voltage range that is much higher
than VCC.
Figure 47 shows a VID-programmed DC/DC converter for
an Intel mobile processor that uses the LTC1435A and
LTC1706-19 to deliver 7A of output current with a programmable VOUT of 1.3V to 2.0V from a VIN of 4.5V to 22V.
Simply connecting the LTC1706-19’s FB and SENSE pins
to the LTC1435A’s VOSENSE and SENSE– pins, respectively, closes the loop between the output voltage sense
and the feedback inputs of the LTC1435A regulator with
the appropriate resistive divider network, which is controlled by the LTC1706-19’s four VID input pins.
VIN
4.5V TO 22V
LTC1435A
COSC
43pF
1
CSS
0.1µF
2
3
CC2
220pF
COSC
VIN
RUN/SS
TG
ITH
SW
CC
1000pF
INTVCC
RC
10k
BOOST
5
6
51pF
SGND
VOSENSE
BG
PGND
Table 2 shows the VID inputs and their corresponding
output voltages. VID3 is the most significant bit (MSB) and
VID0 is the least significant bit (LSB). When all four inputs
are low, the LTC1706-19 sets the regulator output voltage
to 2.00V. Each increasing binary count is equivalent to
decreasing the output voltage by 50mV. Therefore, to
RF
4.7Ω
13
CF
0.1µF
16
+
M1
Si4410DY
CIN
10µF, 30V
×2 R
SENSE
VOUT
1.30V TO
2.00V/7A
0.015Ω
14
L1 3.3µH
DB*
12
0.22µF
SENSE
VCC
15
+
+
11
10
SENSE– SENSE+
7
8
4.7µF
D1
MBRS
-140T3
M2
Si4410DY
LTC1706-19
FB
VID VID VID VID
0 1 2 3
GND
*DB = CMDSH-3
FROM µP
1000pF
L1: COILCRAFT D05022P-332HC
Figure 47. Intel Mobil Pentium II VID Power Converter
VIN
4.8V TO 20V
VCC
2.7V TO 5.5V
LTC1624 1000pF
LTC1706-19 3
0.1µF
VCC
7
6
VID0
SENSE
8
VID1
100pF
1
VID2
2
5
VID3
FB
470pF
GND
4
6.8k
1
2
3
4
SENSE–
VIN
ITH/RUN BOOST
VFB
GND
TG
SW
8
7 0.1µF
6
RSENSE
0.033Ω
+
Si4412DY
VOUT
1.3V–3.0V
5
L1 10µH
MBRS340T3
CIN
22µF
35V
×2
+
COUT
100µF
10V
×2
L1: SUMIDA CDRH125-10
Figure 48. High Efficiency SO-8, N-Channel 3A Switching Regulator with Programmable Output
AN84-32
www.BDTIC.com/Linear
COUT
820µF
4V
×2
Application Note 84
Table 2. VID Inputs and Coresponding Output Voltages
Code
VID3
VID2
VID1
VID0
Output
0000
GND
GND
GND
GND
2.00V
0001
GND
GND
GND
Float
1.95V
0010
GND
GND
Float
GND
1.90V
0011
GND
GN D
Float
Float
1.85V
0100
GND
Float
GND
GND
1.80V
0101
GND
Float
GND
Float
1.75V
0110
GND
Float
Float
GND
1.70V
0111
GND
Float
Float
Float
1.65V
1000
Float
GND
GND
GND
1.60V
1001
Float
GND
GND
Float
1.55V
1010
Float
GND
Float
GND
1.50V
1011
Float
GND
Float
Float
1.45V
1100
Float
Float
GND
GND
1.40V
1101
Float
Float
GND
Float
1.35V
1110
Float
Float
Float
GND
1.30V
Figure 48 shows a combination of the LTC1624 and the
LTC1706-19 configured as a high efficiency step-down
switching regulator with a programmable output of 1.3V
to 2.0V from an input of 4.8V to 20V. Using only one
N-channel power MOSFET, the two SO-8 packaged LTC
parts offer an extremely versatile, efficient, compact regulated power supply.
Figure 49 shows the LTC1436A-PLL and the LTC1706-19,
a combination that yields a high efficiency low noise
synchronous step-down switching regulator with programmable 1.3V to 2V outputs and external frequency
synchronization capability.
Besides the LTC family of 1.19V-referenced DC/DC converters, the LTC1706-19 can also be used to program the
output voltages of regulators with different onboard references. Figure 50 shows the LTC1706-19 programming
the output of the LT1575, an UltraFast™ transient response,
low dropout regulator that is ideal for today’s powerhungry desktop microprocessors. However, since the
LT1575 has a 1.21V reference instead of a 1.19V reference, the output will range from 1.27V to 2.03V in steps of
50.8mV.
obtain a 1.30V output, the three MSBs are left floating
while only VID0 is grounded. In cases where all four VID
inputs are tied high or left floating, such as when no
processor is present in the system, a regulated 1.25V
output is generated at VSENSE.
10k
EXTERNAL
FREQUENCY
SYNCHRONIZATION
0.1µF
COSC
39pF
2
CSS
0.1µF
3
4
COSC
VIN
RUN/SS
TGL
TGS
LTC1436A-PLL
ITH
CC
510pF
SW
INTVCC
RC
10k
BOOST
6
100pF
VIN
4.5V–22V
1
24
PLL LPF PLLIN
8
SGND
VOSENSE
BGL
PGND
18
+
M1
Si4412DY
21
19
M3
IRLML2803
20
L1
3.3µH
CIN
22µF, 35V
×2
RSENSE
0.02Ω
VOUT
1.30V–
2.00V/5A
DB*
17
0.22µF
+
+
16
15
SENSE– SENSE+
9
10
1000pF
SENSE
VCC
22
4.7µF
M2
Si4412DY
D1
MBRS
-140T3
FB
LTC1706-19
VID VID VID VID
0 1 2 3
COUT
100µF
10V
×2
GND
*DB = CMDSH-3
FROM µP
Figure 49. High Efficiency, Low Noise, Synchronous Step-Down Switching Regulator with Adjustable Output Voltage
www.BDTIC.com/Linear
AN84-33
Application Note 84
VIN
12V
VCC 3.3V
LT1575
LTC1706-19 3
VCC
7
6
VID0
SENSE
8
VID1
1
VID2
2
5
VID3
FBK
GND
4
1
2
1µF
3
4
SHDN
IPOS
VIN
INEG
GND
GATE
FB
COMP
8
3.3V
7
6
5.1Ω
IRFZ24
220µF
5
+
VOUT
1.27V TO 2.03
IN 50.8mV STEPS
7.5k
24µF
10pF
1000pF
Figure 50. UltraFast Transient Response, Low Dropout Regulator with Adjustable Output Voltage
NEW DC/DC CONTROLLER ENABLES
HIGH STEP-DOWN RATIOS
by Greg Dittmer
high. This occurs because tON = VOUT/(VIN • f); thus, at low
duty ratios, frequency must be decreased to keep tON >
tON(MIN). Lowering the operating frequency is usually not
desirable because it increases noise and componnent
size.
Capabilities of the LTC1435
The LTC1435 high efficiency synchronous DC/DC controller has been extremely popular for notebook computers
and other battery-powered equipment due to its low noise,
constant-frequency operation and its dual N-channel drive
for outstanding high current efficiency without sacrificing
low dropout operation. However, its 400ns to 500ns
minimum on-time requires lower operating frequencies
(<150kHz) to regulate output voltages below 2.0V if VIN is
What happens if minimum on-time is violated in the
LTC1435? If VIN is increased so that the on-time falls
below tON(MIN), the LTC1435 will begin to skip cycles to
remain in regulation. During this “cycle-skipping” mode,
the output remains in regulation but the operating frequency decreases, causing the inductor ripple current and
output ripple voltage to increase.
400
35
RECOMMENDED
REGION FOR
MIN ON-TIME
AND MAX
EFFICIENCY
30
MINIMUM ON-TIME (ns)
MAXIMUM VIN (V)
LTC1435A
25
MOSFET VDS LIMIT
20
LTC1435
15
10
f = 250kHz
ILOAD = 0A
L = 4.7µH
T = 25°C
5
0
1.25
300
250
IMAX =
0.1
RSENSE
200
1.5
1.75
2.0
2.25
OUTPUT VOLTAGE (V)
2.5
AN70 F52
Figure 51. LTC1435/LTC1435A Maximum VIN Comparison
AN84-34
350
0
10
20
30
40
50
60
70
INDUCTOR RIPPLE CURRENT (% OF IMAX)
Figure 52. LTC1435A Minimum On-Time
vs Inductor Ripple Current
www.BDTIC.com/Linear
AN70 F52
Application Note 84
VIN
4.5V TO 22V
LTC1435A
COSC
43pF
1
CSS
0.1µF
2
3
CC
330pF
COSC
VIN
RUN/SS
TG
ITH
SW
INTVCC
CC2
51pF
RC
10k
BOOST
5
100pF
6
SGND
VOSENSE
BG
PGND
13
+
M1
Si44412DY
16
CIN
10µF, 30V
×2
14
L1 4.7µH
DB*
12
RSENSE
0.033Ω
0.1µF
VOUT
1.60V/3A
35.7k
1% +
15
+
11
10
4.7µF
M2
Si44412DY
D1
MBRS
-140T3
102k
1%
SENSE– SENSE+
7
8
COUT
100µF, 6.3V
×2
*DB = CMDSH-3
CENTRAL
(516) 435-1110
1000pF
Figure 53. LTC1435A 22V to 1.6V/3A Converter (f = 250kHz)
Enter the LTC1435A
22V to 1.6V Converter at 250kHz
The operating envelope has been substantially expanded
with the introduction of the new LTC1435A DC/DC controller,
which has all the outstanding features of the LTC1435 with
a reduced minimum on-time of 300ns or less and improved
noise immunity at low output voltages. With these improvements, high performance at output voltages down to
1.3V can be achieved with operating frequencies in excess
of 250kHz from input supply voltages above 22V. Figure 51
shows the resulting improvement of maximum VIN vs
output voltage as a result of the reduced minimum on-time.
Figure 53 shows the LTC1435A configured in an all
N-channel synchronous buck topology as a 22V to 1.6V/
3A converter running at 250kHz. The 43pF COSC capacitor
sets the internal oscillator frequency at 250kHz and the
33mΩ sense resistor sets the maximum load current at
3A. For a 22V to 1.6V converter, the on-time required is:
The LTC1435A’s minimum on-time is dependent on the
speed of the internal current comparator, which in turn is
dependent on the amplitude of the signal the comparator
is monitoring: inductor ripple current. Thus, the higher the
ripple current, the lower the minimum on-time. Figure 52
shows how minimum on-time varies as a function of the
inductor ripple amplitude. At higher amplitudes, tON(MIN)
is less than 250ns; at low amplitudes it can be 350ns or
more. This means that for low duty cycle applications
where the on-time is approaching tON(MIN), there may be
a minimum ripple current amplitude, and hence, a maximum inductance necessary to prevent cycle skipping. Or,
expressed differently, the lower the inductance, the higher
the maximum VIN that can be achieved before the minimum on-time is violated and cycle skipping occurs. For
most applications, 40% ripple not only reduces the minimum on-time but also optimizes efficiency.
tON = 1.6/(22 × 250kHz) = 291ns
Can the LTC1435A do this? At maximum VIN the inductor
ripple is
∆IL =
=
VOUT • (1 – VOUT/VIN)
F•L
1.6 • (1 – 1.6/22)
250kHz • 4.7µH
= 1.3A
which is 43% of the 3A maximum load. From Figure 52,
43% ripple gives a minimum on-time of 235ns, which is
well below the 291ns required by this application, so no
cycle skipping will occur. If a 10µH inductor is used, the
ripple amplitude drops to 0.6A or 20% and the minimum
on-time increases to 280ns. This does not provide much
margin below the 291ns on-time required, and thus the
4.7µH inductor is a better choice.
www.BDTIC.com/Linear
AN84-35
Application Note 84
Intel Mobile Processor VID Power Converter
in 50mV increments at 250kHz and a 7A load current. The
selectable output voltage is implemented by replacing the
conventional feedback resistor network with the
LTC1706-19, which provides the appropriate feedback
resistor ratios internally. The proper ratio is selected with
the 4-bit digital input pins.
Figure 54 shows the LTC1435A used with an LTC1706-19
to implement an Intel Mobile Pentium® II Processor VID
power converter. This DC/DC converter provides digitally
selectable output voltages over the range of 1.3V to 2.0V
VIN
4.5V TO 22V
LTC1435A
COSC
43pF
1
CSS
0.1µF
2
3
CC2
220pF
COSC
RUN/SS
ITH
CC
1000pF
TG
SW
INTVCC
RC
10k
BOOST
5
51pF
VIN
6
SGND
VOSENSE
BG
PGND
RF
4.7Ω
13
CF
0.1µF
16
+
M1
Si4410DY
CIN
10µF, 30V
×2 R
SENSE
VOUT
1.30V TO
2.00V/7A
0.015Ω
14
L1 3.3µH
DB*
12
6
0.22µF
3
15
10
VCC
+
+
11
SENSE
4.7µF
M2
Si4410DY
D1
MBRS
-140T3
5
LTC1706-19
FB
VID VID VID VID
0 1 2 3
7 8 1 2
COUT
820µF
4V
×2
GND
4
*DB = CMDSH-3
CENTRAL
(516) 435-1110
SENSE– SENSE+
7
8
1000pF
FROM µP
FIgure 54. Intel Mobil Pentium II VID Power Converter
LTC1627 MONOLITHIC SYNCHRONOUS
STEP-DOWN REGULATOR MAXIMIZES SINGLE
OR DUAL LI-ION BATTERY LIFE
by Jaime Tseng
Introduction
The LTC1627 is a new addition to a growing family of
power management products optimized for Li-Ion batteries. Li-Ion batteries, with their high energy density, are
becoming the chemistry of choice for many handheld
products. As the demand for longer battery operating time
continues to increase and the operating voltages of submicron DSPs and microcontrollers decreases, more demands
are placed on DC/DC conversion. The LTC1627 monolithic, current mode synchronous buck regulator was
specifically designed to meet these demands.
The LTC1627, with its operating supply range of 2.65V to
8.5V, can operate from one or two Li-Ion batteries as well
as 3- to 6-cell NiCd and NiMH battery packs.
AN84-36
The LTC1627 incorporates power saving Burst Mode
operation and 100% duty cycle for low dropout to maximize the battery operating time. In Burst Mode operation,
both power MOSFETs are turned off for increasing intervals as the load current drops. Along with the gate-charge
savings, unused circuitry is shut down between burst
intervals, reducing the quiescent current to 200µA. This
extends operating efficiencies exceeding 90% to over two
decades of output load range.
Typical Applications
The LTC1627, with its synchronous switching and attendant circuitry, provides the means of easily constructing a
secondary flyback regulator, as shown in Figure 55. This
flyback regulator is regulated by the secondary feedback
resistive divider tied to the SYNC/FCB pin. This pin forces
continuous operation whenever it drops below its groundreferenced threshold of 0.8V. Power can then be drawn
from the secondary flyback regulator whether the main
output is loaded or not.
www.BDTIC.com/Linear
Application Note 84
R3
249k
CITH
47pF
1 I
TH
VIN ≤ 8.5V
R4
80.6k
2 RUN/SS
DR
LTC1627
6
VFB 3
VIN
CSS
0.1µF
CIN*
22µF
16V
1%
SYNC/ 8
FCB
7
V
+
4 GND
SW
+
D1
MBR0520LT1
1%
D1††
1.8V
22µF***
6.3V
5
+
25µH† R1
1:1 100k
1%
* AVX TPSC226M016R0375
** AVX TPSC107M006R0150
*** AVX TAJA226M006R
(207) 282-5111
VSEC†††
3.3V/100mA
VOUT
1.8V/0.3A
COUT**
100µF
6.3V
R2
80.6k
† COILTRONICS CTX25-1
(561) 241-7876
†† MMSZ4678T1
††† 10mA MIN LOAD CURRENT
RECOMMENDED
1%
Figure 55. Dual-Output 1.8V/0.3A and 3.3V/100mA Application
1 or 2 Li-Ion Step-Down Converter
Figure 56 is a schematic diagram showing the LTC1627
being powered by one or two Li-Ion batteries. All the
components shown in this schematic are surface mount
and have been selected to minimize the board space and
height. The output voltage is set at 3.3V, but is easily
programmed to other voltages.
CITH
47pF
1 I
TH
CSS
0.1µF
VIN
≤ 8.4V
CITH
47pF
C1
0.1µF
VIN
2.8V–4.5V
BAT54S**
D1
D2
SW
5
+
CIN††
22µF
16V
The circuit in Figure 57 is intended for input voltages below
4.5V, making it ideal for single Li-Ion battery applications.
Diodes D1 and D2 and capacitors C1 and C2 comprise the
bootstrapped charge pump to realize a negative supply at
the VDR pin, the return pin for the top P-channel MOSFET
driver. This allows Figure 57’s circuit to maintain low
switch RDS(ON) all the way down to the UVLO trip voltage.
+
2 RUN/SS
DR
LTC1627
3
6
VFB
VIN
4 GND
Single Li-Ion Step-Down Converter
CIN††
22µF
16V
SYNC/ 8
FCB
7
V
CSS
0.1µF
1 I
TH
C2
0.1µF
R1
249k
1%
VOUT
3.3V/0.5A
+
R2
80.6k
1%
COUT†
100µF
6.3V
Figure 56. Lithium-Ion to 3.3V/0.5A regulator
SYNC/ 8
FCB
7
V
2 RUN/SS
DR
LTC1627
6
VFB 3
VIN
4 GND
*SUMIDA CD54-250
(847) 956-0666
†AVX TPSC107M006R0150
††AVX TPSC226M016R0375
(207) 282-5111
25µH*
SW
5
* SUMIDA CD54-150
(847) 956-0666
** ZETEX BAT54S
(516) 543-7100
† AVX TPSC107M006R0150
††
AVX TPSC226M016R0375
(207) 282-5111
15µH*
R1
169k
1%
VOUT
2.5V/0.5A
+
R2
80.6k
1%
COUT✝
100µF
6.3V
Figure 57. Single Lithium-Ion to 2.5V/0.5A Regulator
www.BDTIC.com/Linear
AN84-37
Application Note 84
100
THE LTC1625 CURRENT MODE
DC/DC CONTROLLER ELIMINATES
THE SENSE RESISTOR
VIN = 20V
VOUT = 2.5V
EFFICIENCY (%)
95
by Christopher B. Umminger
Introduction
LTC1625
90
LTC1435
85
Power supply designers have a new tool in their quest for
ever higher efficiencies. In the past, when designing a
step-down DC/DC converter, one had to choose between
the high efficiency of voltage mode control and the many
benefits of current mode control. Although voltage mode
control offers high efficiency and a simple topology, it is
difficult to compensate, has poor rejection of inputvoltage transients and does not inherently limit output
current under fault conditions, such as an output short
circuit. Current mode control overcomes these problems
by adding a control loop to regulate the inductor current
in addition to the output voltage. Unfortunately, a sense
resistor is required to measure this current, which adds
cost and complexity while reducing converter efficiency.
However, with the new LTC1625 No RSENSE™ controller,
one can enjoy all of the benefits of current mode control
without the penalties of using a sense resistor.
80
0
1
2
3
LOAD CURRENT (A)
full input voltage. The controller provides synchronous
drive for N-channel power MOSFETs and retains the
advantage of low dropout operation typically associated
with P-channel MOSFETs. Burst Mode™ operation maintains efficiency at low load currents, but can be overridden
to assist secondary-winding regulation by forcing continuous operation. In addition to eliminating the sense
resistor, the LTC1625 further reduces the external parts
count by incorporating the oscillator timing capacitor. The
oscillator frequency can be set to 150kHz, 225kHz, or can
be injection locked to any frequency between these points.
Design Examples
Figure 58 shows the LTC1625 in an application supplying
a 2.5V output using an external feedback divider. Si4410DY
MOSFETs from Siliconix allow this converter to deliver up
RF 1Ω
2
CSS
0.1µF
3
4
CC1
820pF
RC1
5
10k
6
CC2 220pF 7
8
LTC1625
EXTVCC
VIN
SYNC
TK
RUN/SS
SW
FCB
TG
ITH
BOOST
SGND
INTVCC
VOSENSE
VPROG
BG
PGND
16
15
M1
Si4410DY
14
CB
O.22µF
13
11
VIN
5V TO 28V
CIN
10µF
30V
×2
L1 7µH
D1
MBRS140T3
*DB
12
10
+
CF
0.1µF
+
1
CVCC
4.7µF
R2
11k
1% +
R1
10k
1%
9
M2
Si4410DY
* DB = CMDSH-3
Figure 58. 2.5V/5A Adjustable-Output Supply
AN84-38
5
DI_1068_02a. EPS
Figure 59. Efficiency vs Load Current
The LTC1625 is a step-down DC/DC switching regulator
controller that is capable of a wide range of operation with
inputs from 3.7V to 36V. Fixed output voltages of 5V and
3.3V can be selected or an external resistive divider can be
used to obtain output voltages from 1.19V up to nearly the
5V
4
www.BDTIC.com/Linear
VOUT
2.5V/5A
COUT
100µF
10V
0.065Ω ESR
×3
Application Note 84
RF 4.7Ω
CSS
INTVCC
2
0.1µF
3
4
CC1
1nF
RC1
5
10k
6
CC2 220pF
7
8
LTC1625
16
EXTVCC
VIN
SYNC
TK
RUN/SS
SW
FCB
TG
ITH
BOOST
SGND
INTVCC
VOSENSE
VPROG
BG
PGND
15
M1
Si4412DY
14
13
12
**DB
11
10
+
1
+
CF
0.1µF
CB
O.1µF
VIN
12V TO 28V
CIN
22µF
35V
×2
L1* 39µH
D1
MBRS140T3
CVCC
4.7µF
R2
35.7k
1% +
R1
3.92k
1%
9
VOUT
12V/2.2A
COUT
100µF
16V
0.030Ω ESR
M2
Si4412DY
* L1 = SUMIDA CDRH127-390MC
** DB = CMDSH-3
Figure 60. 12V/2.2A Adjustable-Output Supply
to 5A of load current. Ripple current is 1.8A (36% of full
load) and current limit occurs around 6A. Note also that
the EXTVCC pin is connected to an external 5V supply. This
increases efficiency by drawing the roughly 7mA gate
charge current from a supply lower than VIN.
greater than 90% at high load current. The benefit of
reduced I2R loss is readily apparent at the highest loads.
The controller makes a transition to Burst Mode operation
below around 1.1 A which keeps the efficiency high at
moderate loads.
An efficiency plot of this circuit is shown in Figure 59. An
LTC1435 with identical components in the power path is
also plotted for comparison. At lower output voltages such
as this, the sense resistor is responsible for an increasing
share of the total power loss. By eliminating this source of
loss, the LTC1625 is easily able to deliver an efficiency
A circuit demonstrating the wide output range of the
LTC1625 is shown in Figure 60. This application uses
Si4412DY MOSFETs to deliver a 12V output at up to 2.2A.
Note that the SYNC pin is tied high for 225kHz operation
in order to reduce the inductor size and ripple current.
PolyPhase SWITCHING REGULATORS OFFER
HIGH EFFICIENCY IN LOW VOLTAGE,
HIGH CURRENT APPLICATIONS
by Craig Varga
transient response requirements also become much more
severe. The question that arises is: “is there a topology that
can solve all of these problems simultaneously? ” The
answer is “PolyPhase™.”
Introduction
What is PolyPhase, Anyway?
In recent years, there has been a tendency in the digital
world toward smaller device geometries and higher gate
counts. This has led to requirements for lower voltages
and higher currents for logic supplies. As this trend
continues, to levels under 2V and over 30A, the conventional buck regulator approach ceases to be viable. Switch
currents are too high for a single device to handle, inductor
energy storage exceeds what is available in surface mount
technology and ripple current requirements on input capacitors dictate the use of many capacitors in parallel.
Although all this may seem like enough of a challenge, the
Since it is apparent that multiple FETs need to be paralleled
to handle the current requirements, the question is whether
there is a way to drive them intelligently, rather than by
brute force. The solution is to stagger the turn-on times so
that the dead bands in the input current waveform are
“filled up,” so to speak. In the simplest implementation,
there are essentially two independent synchronous buck
regulators operating 180° out of phase. The net effect of
this is that the input and output ripple currents of the two
channels tend to cancel during steady-state operation.
This results in significant reductions in both input and
www.BDTIC.com/Linear
AN84-39
AN84-40
Figure 61. 2-Phase Synchronous Buck Regulator
www.BDTIC.com/Linear
ISENSE1
ISENSE2
R6
3.09k
1%
C28
0.1µF
C38
3300pF
R31
1k
13
11
10
R8
4.3k
R9
4.3k
C37
3300pF
12V R32
10Ω
C26 +
22µF
25V
R30
1k
Q
Q
OSC
CD4047
RX
RST
CX
AST
AST
–T
+T
RET
RCC
U4
(POWER FROM 5V)
9
5
4
6
8
12
3
1
C6, 100pF,
NPO, 5% 2
5V
CLOCK
3
2
+
–
C31
0.022µF
C29
1µF
16V
SYNC2
×
R4
1k
R3
1k
SYNC1
4
7
C3
22µF
25V
8
×
+
+
R7
51k
C16
180pF
C2
1µF
16V
2
PVCC1
1
G1
4
FB
3
GND
LTC1430ACS8
C15
1500pF
R15
10k
U3
7
PVCC2
8
G2
5
SHDN
6
COMP
C24
1µF
16V
R16
10Ω
12V
2
PVCC1
1
G1
4
FB
3
GND
LTC1430ACS8
C14
1500pF
R14
10k
7
8
5
6
U2
PVCC2
G2
SHDN
COMP
C25
1µF
16V
5V
Q8
R26 Si4410DY
1Ω
R27
1Ω
Q5
Si4410DY
L2
0.8µH
+
ISENSE1
L1
0.8µH
+
C20
6800pF
R29
1Ω
C9 +
470µF
6.3V
+
C12
470µF
6.3V
C7
470µF
6.3V
+
C8
470µF
6.3V
R18
10k
1%
R2
9.76k
1%
R12
10k
C17
1000pF
C34
1µF
10V
C36
1µF
10V
C35
1µF
10V
R24
39k
Q1
MMBT3906LT1
D1
BAW56LT1
C18
1000pF
SYNC2
SYNC1
R17
10k, 1%
C33
470µF
6.3V
R11
0.002Ω
TRACE
+
+
+VIN
C32
470µF
6.3V R5
9.76k
1%
R13
0.002Ω
TRACE
C11
470µF
6.3V
ISENSE2
C10 +
470µF
6.3V
ETQP1F0R8LB
C4
1µF
10V
C19
6800pF
R28
1Ω
ETQP1F0R8LB
C5
1µF
10V
(408) 986-0424
Q9
Si4410DY
Q7
Si4410DY
CHARGE PUMP
D3
OPTIONAL
BAT54
C27, 0.47µF
Q6
R19 Si4410DY
1Ω
R25
1Ω
R23
1Ω
Q4
R22 Si4410DY
1Ω
Q3
Si4410DY
CHARGE PUMP
OPTIONAL
C23
D2
BAT54 0.47µF
Q2
R20 Si4410DY
1Ω
R21
1Ω
D4
BAT54
5V
NOTES:
1. ALL RESISTORS = ±5% UNLESS NOTED OTHERWISE.
2. INPUT/OUTPUT CAPACITORS = KEMET T510 SERIES
3. TRACE RESISTORS R11, R13 = 0.1" WIDE x 0.675" LONG
R1
51Ω
C1
1µF
16V
C13
180pF
C21
47µF
10V
(4)
×
U1
LT1006
SHARE AMPLIFIER
6
1
C30
0.022µF
C22
1µF
10V
5V
R10
10Ω
12V
INPUT
RTN
+VIN
OUTPUT
RTN
+VOUT
2.5V/30A
Application Note 84
Application Note 84
100
VIN = 5V
95
EFFICIENCY (%)
VOUT = 3.3V
90
VOUT = 2V
85
∆V =
160mV
100mV/DIV
VOUT = 2.5V
80
75
70
0
5
10
15
20
CURRENT (A)
25
10µs/DIV
30
Figure 62. Efficiency of Figure 61’s Circuit, VIN = 5V
DC201 F01b
output capacitor requirements. There is also a fourfold
reduction in the total inductor energy storage requirement, which means much smaller inductors and vastly
improved transient dynamics. During a large load step, the
two channels operate at maximum duty factor in an
attempt to maintain the desired output voltage. Both
inductor currents slew rapidly and are now additive, since
they are going in the same direction. Hence, the slew rate
is double what a single channel could do for equal inductor
values. However, due to the ripple current cancellation
during steady-state conditions, the two inductors can be
reduced to approximately one-half the value that a single
channel design would require for equal ripple currents.
Since during slew they appear to be operating in parallel,
the actual slew rate is four times that of a single channel
design with equal steady-state output ripple current. Both
input and output ripple frequencies are double those of a
single-channel design, further simplifying filtering
requirements.
Figure 64. Transient Response with
10A Load Step (100ns Rise Time)
Why Stop at Two?
If two channels are good, aren’t more channels better? In
a word, yes. In principle, there is no limit to the number of
parallel channels that can be added. As the number of
channels, n, increases, the ripple frequency increases to n
times the single-channel frequency. Input and output
RMS ripple currents continue to decrease. Diminishing
returns are reached as n rises above three. At three stages,
the ripple reductions are very substantial and dynamic
performance is excellent. Adding more channels produces
slight improvements but the dramatic gains will have been
realized by n = 3. The only real penalty is added complexity.
The bottom line is that PolyPhase designs offer a considerable reduction in the cost and volume of the power
devices at the expense of a little added complexity in the
control circuitry.
CHANNEL A
5A/DIV
CHANNEL A + B =
TOTAL INPUT
RIPPLE CURRENT,
UNFILTERED
20mV/DIV
CHANNEL B
5A/DIV
IO = 15A
f = 306kHz
VO = 2.5V
VIN = 5V
2µs/DIV
Figure 63. Output Ripple with 30A Load
2µs/DIV
Figure 65. Ripple Cancellation—Input
www.BDTIC.com/Linear
AN84-41
Application Note 84
A+B
5A/DIV
CHANNEL A
CHANNEL B
IO = 25A
2µs/DIV
nize the two LTC1430s. Unfortunately, simply connecting
two regulators in parallel is a recipe for instant disaster.
The output voltages of the two regulators will be slightly
different due to normal component tolerances. Therefore,
the higher output voltage channel will attempt to supply
the full load current, while the lower voltage output will
sink current from the output in a desperate attempt to
reduce the output voltage to where it thinks it should be.
The result is like a dog chasing its tail, with large currents
running around in a circle and going nowhere.
Figure 66. Ripple Cancellation—Output
2-Phase Design Example
The circuit shown in Figure 61 is a 2-phase, voltage mode–
control, synchronous buck regulator designed for a 5V
input and output voltages below 3.3V. It is intended to
power large memory arrays, ASICs, FPGAs and the like in
server and workstation applications. The output is capable
of more than 30 amps continuous at outputs of 2.5V and
below, with peak current capability of greater than 40
amps. The design is entirely surface mount and the
maximum height above the board is 5.5mm. Overall board
area is only 4.24 in2. Efficiency is excellent, as can be seen
in the curve in Figure 62. Output ripple voltage is shown in
Figure 63. The circuit’s dynamic response to a 10 amp load
step is shown in Figure 64. The response is dominated by
the output capacitor’s ESR and shows the output voltage
recovered to the original level in under 10µs. Figures 65
and 66 show how the input and output ripple currents
cancel.
Circuit Operation
The basic design consists of two LTC1430CS8-based
synchronous buck regulators connected in parallel and
operated 180° out of phase. U4, the CD4047 oscillator, is
used to generate the required clock signals and synchro-
AN84-42
Op amp U1 solves this problem. Because the two channels
are identical, if the output currents are the same, the input
currents will be also. Low value sense resistors are included in the input power path to allow the circuit to
measure input current. U1 then forces the input current of
channel two to match the input current of channel one by
making small adjustments in channel two’s output voltage. It does this by adding or subtracting a small amount
of current from channel two’s feedback divider. The two
sense resistors are short lengths of PCB trace and only
need to be ratiometrically accurate. The absolute value of
these resistors is not important (see Linear Technology
Application Note 69, Appendix A, for a discussion on how
to design trace resistors).
The only remaining trick in the circuit is the role of Q1 and
its associated circuitry. At start-up, the LTC1430’s clock
frequency is slowed down to approximately 10kHz until
the output voltage rises to approximately 50% of the
desired level. If, during this start-up phase, an attempt is
made to synchronize the controller to a very high frequency, the oscillator ramp amplitude never rises to a level
sufficiently high to trip the PWM comparator and enable
the FET drivers. Therefore, the output gets stuck on
ground. Q1 fixes this by forcing the sync signals high
during the turn-on transient. Once the output voltage
nears its final level, the clock signals are allowed to
synchronize the two PWM controllers.
www.BDTIC.com/Linear
Application Note 84
100
LTC1622: LOW INPUT VOLTAGE,
CURRENT MODE PWM BUCK CONVERTER
by San-Hwa Chee
EFFICIENCY (%)
90
Introduction
The 8-pin LTC1622 step-down DC/DC controller is designed to help system designers harness all of the available energy from lithium-ion batteries in several ways. Its
wide operating input-voltage range (2.0V to an absolute
maximum of 10V) and 100% duty cycle allows low
dropout for maximum energy extraction from the battery.
The part’s low quiescent current, 400µA, with a shutdown
current of 15µA, extends battery life. Its user-selectable
Burst Mode operation enhances efficiency at low load
current.
VIN = 4.2V
VIN = 3.3V
80
VIN = 8.4V
70
VIN = 6V
60
VOUT = 2.5V
RSENSE = 0.03Ω
50
40
0.100
0.010
LOAD CURRENT (A)
0.001
1.000
Figure 68. Efficiency vs Load Current for Figure 67’s
Circuit (Burst Mode Operation Enabled)
supplying 1.5A at a low input voltage. In addition, a
sublogic threshold MOSFET is used, since the circuit
operates at input voltages as low as 2.7V. The circuit
operates at the internally set frequency of 550kHz. A 4.7µH
inductor is chosen so that the inductor’s current remains
continuous during burst periods at low load current. For
low output voltage ripple, a low ESR capacitor (100mΩ) is
used.
For portable applications where board space is a premium, the LTC1622 operates at a constant frequency of
550kHz and can be synchronized to frequencies of up to
750kHz. High frequency operation allows the use of small
inductors, making this part ideal for communications
products. The LTC1622 comes in a tiny 8-lead MSOP
package, providing a complete power solution while
occupying only a small area.
Efficiency Considerations
The efficiency curves for Figure 67’s circuit are shown in
Figures 68 and 69. Figure 68 shows the efficiency with
Burst Mode enabled, whereas Figure 69 has Burst Mode
defeated. (Burst Mode is defeated by connecting the
SYNC/Mode pin to ground.) Note that, at low load currents, the efficiency is higher with Burst Mode operation.
However, constant frequency operation is still achievable
2.5V/1.5A Step-Down Regulator
A typical application circuit using the LTC1622 is shown
in Figure 67. This circuit supplies a 1.5A load at 2.5V with
an input supply between 2.7V and 8.5V. The 0.03Ω sense
resistor is selected to ensure that the circuit is capable of
R2 0.03Ω
VIN
2.5V–8.5V
LTC1622
1
SENSE–
7
2
ITH
PDRV
6
5
SYNC/MODE
GND
3
4
RUN/SS
VFB
8
C1 +
10µF
16V
R1
10k
C3
220pF
VIN
Si3443DV
L1 4.7µH
D1
470pF
C1: MURATA CERAMIC GRM235Y5V106Z
(814) 236-1431
C2: SANYO POSCAP 6TPA47M
(619) 661-6835
L1: MURATA LQN6C-4R7M04
(814) 237-1431
R3
159k
VOUT
+
C2 2.5V/1.5A
47µF
6V
R4
75k
D1: IR10BQ015
(310) 322-3331
R2: DALE, 0.25W
(605) 665-9301
Figure 67. LTC1622 Typical Application: 2.5V/1.5A Converter
www.BDTIC.com/Linear
AN84-43
Application Note 84
100
VIN = 4.2V
VIN = 3.3V
EFFICIENCY (%)
90
80
70
60
VIN = 6V
VOUT = 2.5V
RSENSE = 0.03Ω
50
40
0.001
OUTPUT VOLTAGE
(AC COUPLED)
0.1V/DIV
VIN = 8.4V
0.010
0.100
1.000
LOAD CURRENT (A)
Figure 69. Efficiency vs Load Current for Figure 66’s Circuit
(Burst Mode Operation Disabled)
0.1ms/DIV
Figure 70. Transient Response with Burst Mode
Operation Enabled; Load Step = 50mA to 1.2A
at a lower load currents with Burst Mode operation defeated. The kinks in the efficiency curves indicate the
transition out of Burst Mode operation.
enhanced significantly. However, this comes at the expense
of slightly reduced efficiency at low load currents, as
indicated by the efficiency curves of Figures 75 and 76.
The components of Figure 67 have been carefully chosen
to provide the amount of output power using a minimum
of board space. Efficiency is also a prime consideration in
selecting the components, as illustrated in Figures 68 and
69. Figures 70 and 71 show the transient response of VOUT
with a load step from 50mA to 1.2A. Figure 70 has Burst
Mode enabled, while Figure 71 has it defeated. Note that
the output voltage ripple (in the middle portion of the
photographs) is higher for Burst Mode operation than with
Burst Mode disabled at 50mA load current.
OUTPUT
VOLTAGE
(AC COUPLED)
0.1V/DIV
Applications that require better transient response can use
the circuit in Figure 72, whose components are selected
specifically for this requirement. Figures 73 and 74 show
the response with and without Burst Mode operation,
respectively. Note that the transient response has been
0.1ms/DIV
Figure 71. Transient Response with Burst Mode
Operation Inhibited; Load Step = 50mA to 1.2A
R2 0.03Ω
VIN
2.5V TO
8.5V
8
C1 +
47µF
16V
R1
22k
C3
100pF
2
5
4
LTC1622
1
VIN
SENSE–
7
ITH
PDRV
6
SYNC/MODE
GND
3
RUN/SS
VFB
470pF
Si3443DV
L1 1.3µH
D1
C1: AVX TPSD476M016R0150
(803) 946-0362
C2: AVX TPSD476M016R0065
L1: MURATA LQN6C-1R5M04
(814) 237-1431
R3
159k
+
VOUT
C2 2.5V/1.5A
100µF
6V
R4
75k
Figure 72. 2.5V/1.5A Converter with Improved Transient Response
AN84-44
www.BDTIC.com/Linear
D1: IR10BQ015
(310) 322-3331
R2: DALE, 0.25W
(605) 665-9301
Application Note 84
OUTPUT
VOLTAGE
(AC COUPLED)
0.1V/DIV
OUTPUT
VOLTAGE
(AC COUPLED)
0.1V/DIV
0.1ms/DIV
0.1ms/DIV
Figure 73. Transient Response with Burst Mode
Operation Enabled; Load Step = 50mA to 1.2A
Figure 74. Transient Response with Burst Mode
Operation Inhibited; Load Step = 50mA to 1.2A
100
VIN = 4.2V
100
VIN = 3.3V
VIN = 3.3V
90
80
VIN = 8.4V
70
VIN = 6V
60
50
40
0.001
VOUT = 2.5V
RSENSE = 0.03Ω
0.010
0.100
LOAD CURRENT (A)
1.000
Figure 75. Efficiency vs Load Current for Figure 72’s
Circuit (Burst Mode Operation Enabled)
EFFICIENCY (%)
EFFICIENCY (%)
90
VIN = 4.2V
80
VIN = 8.4V
70
VIN = 6V
60
VOUT = 2.5V
RSENSE = 0.03Ω
50
40
0.001
0.010
0.100
LOAD CURRENT (A)
1.000
Figure 76. Efficiency vs Load Current for Figure 72’s
Circuit (Burst Mode Operation Disabled)
www.BDTIC.com/Linear
AN84-45
Application Note 84
WIDE INPUT RANGE, HIGH EFFICIENCY
STEP-DOWN SWITCHING REGULATORS
by Jeff Schenkel
Introduction
The LT1676/LT1776 are current mode switching regulator
ICs optimized for high efficiency operation in high input
voltage, low output voltage buck topologies. These two
parts are pin-for-pin compatible and virtually identical in
operation, the only difference being their internal oscillator frequencies—100kHz for the LT1676 vs 200kHz for
the LT1776. They operate in a fixed frequency mode (as
opposed to constant off-time or on-time, for instance) and
can be externally synchronized to a higher switching
frequency.
The internal output switch is rated at a nominal peak
current of 700mA, which typically accommodates DC
output currents of up to 500mA. The input voltage range
is 7.4V to 60V. Maintaining acceptable efficiency in the
upper half of this input voltage range requires very fast
output-switch edge rates. The LT1676/LT1776 contain
specialized output circuitry to deliver this performance.
Additionally, they contain circuitry to monitor output load
level and reduce leading-edge switch rate (turn-on) when
the output load is light. This arrangement helps avoid
pulse skipping at light load, with its consequent subharmonic behavior.
+
C1
39µF
63V
1
C5
100pF
6
Applications
Minimum Component-Count Application
Figure 77 shows a basic “minimum component count”
application using the LT1676. The circuit produces 5.0V at
up to 500mA IOUT with input voltages in the range of 12V
to 48V. The typical POUT/PIN efficiency is shown in Figure
78. No pulse skipping is observed down to zero external
load. (The several milliamperes drawn by the VCC pin acts
as a sufficient preload.) As shown, the SHDN and SYNC
pins are unused, however either (or both) can be optionally driven by external signals as desired.
5
VIN
SHDN
VCC
VSW
LT1676
FB
SYNC
VC
GND
L1
220µH
2
3
+
D1
MBRS1100
7
8
4
C1: PANASONIC HFQ
(201) 348-7522
C2: AVX D CASE TPSD107M010R0080
(803) 946-0362
C4, C5: X7R OR COG/NPO
D1: MOTOROLA 100V, 1A, SMD SCHOTTKY
(800) 441-2447
L1: COILCRAFT DO3316P-224
(847) 936-6400
C3
2200pF
X7R
R3
22k
5%
90
C2
100µF
10V
C4
100pF
FOR 3.3V VOUT VERSION:
R1: 24.3k, R2: 14.7k
L1: 150µH, DO3316P-154
IOUT: 0mA TO 500mA
R1
36.5k
1%
VOUT
5V
0mA to 500mA
R2
12.1k
1%
1676 F04a
80
70
EFFICIENCY (%)
VIN
12V TO
48V
True current mode operation is supported, with all its well
known advantages for switching regulator operation. The
shutdown pin implements a pair of functions. Pulling it
down to near ground turns off the part almost completely
and reduces the quiescent current to a few tens of microamperes. The second shutdown pin function acts at a
threshold of roughly 1.25V. Below this level, the part
operates normally, except that output switching action is
inhibited. This allows the implementation of an undervoltage lockout function set by, for instance, an external
resistor divider. The LT1676/LT1776 are available in both
8-pin SO and PDIP packages.
60
50
40
VIN = 12V
VIN = 24V
30 VIN = 36V
VIN = 48V
20
1
10
100
LOAD CURRENT (mA)
1000
1676 F04b
Figure 77. Minimum Component-Count Application
AN84-46
Figure 78. Efficiency of Figure 77’s Circuit
www.BDTIC.com/Linear
Application Note 84
+
C1
15µF
35V 1
C5
100pF
6
5
VIN
2
SHDN
VCC
3
VSW
LT1776
7
FB
8
VC
SYNC
GND
4
C1: AVX D CASE 15µF 35V
TPSD156M035R0300
(803) 946-0362
C2: AVX D CASE 100µF 10V
TPSD107M010R0080
C3: 2200pF, X7R
C4, C5: 100pF, X7R OR COG/NPO
C2
100µF
10V
L1
68µH
+
D1
MBRS1100
C3
2200pF
R3
22k
5%
C4
100pF
D1: MOTOROLA 100V, 1A, SMD SCHOTTKY
MBRS1100
(800) 441-2447
L1: COILCRAFT DO1608C-683
(847) 936-6400
Figure 79. Minimum PC Board Application
R1
36.5k
1%
90
VOUT
5V
0mA to
400mA
80
70
EFFICIENCY (%)
VIN
10V TO 30V
R2
12.1k
1%
60
50
40 VIN = 10V
1776 F07a
FOR 3.3V VOUT VERSION:
IOUT: 0mA TO 500mA
L1: 47µH, DO1608C-473
R1: 24.3k, R2: 14.7k
VIN = 20V
30
VIN = 30V
20
1
10
100
LOAD CURRENT (mA)
1000
1776 F07b
Figure 80. Efficiency of Figure 79’s Circuit
Minimum PC Board Area Application
The previous application example used the LT1676 to
demonstrate simultaneously the maximum input voltage
and output current capability. As such, the input bypass
capacitor choice was a high frequency aluminum electrolytic type, rated to 63V. Also, the 100kHz switching rate of
the LT1676 requires an inductor of about 220µH. The
DO3316 device size was chosen to support the output
current requirements. However, both of these components are physically large.
The application example in Figure 79 shows a circuit that
is much smaller physically than the previous minimum
component count application. The nominal 200kHz switching frequency of the LT1776 allows the use of a physically
smaller 68µH inductor—a Coilcraft DO1608C-683. This
inductor will support output current to 400mA at 5V.
However, the part is incapable of withstanding an indefinite
short circuit to ground. (Momentary shorts of a few
seconds or less can still be tolerated.) Additionally, the
bulky aluminum electrolytic capacitor previously on VIN
has been replaced by a compact 35V-rated tantalum type.
The result is a postage-stamp-sized circuit with efficiency
as shown in Figure 80.
Burst Mode Application
The minimum component count application demonstrates
that power supply efficiency degrades with lower output
load current. This is not surprising, as the LT1676 itself
represents a fixed power overhead. A possible way to
improve light load efficiency is to use Burst Mode™
operation.
Figure 81 shows the LT1676 configured for Burst Mode
operation. Output voltage regulation is now provided in a
“bang-bang” digital manner, via comparator U2, an
LTC1440. Resistor divider R4/R5 provides a scaled version of the output voltage, which is compared against U2’s
internal reference. Intentional hysteresis is set by the R6/
R7 divider. As the output voltage falls below the regulation
range, the LT1676 is turned on. The output voltage rises
and, as it climbs above the regulation range, the LT1676
is turned off. Efficiency is maximized as the LT1676 is only
powered up while it is providing heavy output current.
Figure 82 shows that efficiency is typically maintained at
75% or better down to a load current of 10mA. Even at a
load current of 2mA, efficiency is still a respectable 65%
to 75% (depending on VIN).
Resistor divider R1/R2 is still present, but does not
directly influence output voltage. It is chosen to ensure
that the LT1676 delivers high output current throughout
the voltage regulation range. Its presence is also required
to maintain proper short-circuit protection. Transistors
Q1 and Q2 and resistor R7 form a high VIN, low quiescent
current voltage regulator to power U2.
www.BDTIC.com/Linear
AN84-47
Application Note 84
VIN
12V TO 48V
+
R7
10M
C1
39µF
63V 6
Q1
PN2484
Q2
2N2369
1
5
VIN
2
SYNC
VCC
3
VSW
U1
LT1676
7
FB
8
SHDN
VC
GND
NC
4
L1
220µH
D1
MBRS1100
+
C3
100pF
C2
100µF
10V
VOUT
5V
R1
39k
5%
R2
10k
5%
R3
323k
1%
7
C1: PANASONIC HFQ
8
(201) 348-2552
C2: AVX D CASE
TPSD107M010R0080
(803) 946-0362)
C4, C5: X7R OR COG/NPO
D1: MOTOROLA 100V, 1A,
SMD SCHOTTKY
(800) 441-2447
L1: COILCRAFT DO3316-224
(847) 639-6400
V+
OUT
4
IN –
3
IN +
U2
LTC1440
6
REF
5
HYST
–
V
GND
2
1
R6
22k
R7
2.4M
R4
100k
1%
1676 F06
Figure 81. Burst Mode Operation Configuration
Battery Charger Application
Figure 83 shows the LT1776 configured as a constantcurrent/constant-voltage battery charger. An LT1620 railto-rail current sense amplifier (U2) monitors the differential voltage across current sense resistor R4. As this
equals and exceeds the voltage across resistor R5 in the
R5/R6 divider, the LT1620 responds by sinking current at
its IOUT pin. This is connected to the VC control node of the
LT1776 and therefore acts to reduce the amount of power
delivered to the load. The overall constant-current/constant-voltage behavior can be seen in Figure 84.
Target voltage and current limits are independently programmable. The output voltage of 7.2V, which corresponds to the charging voltage of a 3-cell lead-acid battery, is set by the R1/R2 divider and the internal reference
of the LT1776. Output current, presently 200mA, is set by
current sense resistor R4 and the R5/R6 divider. (A 16-pin
version of the LT1620 that implements end-of-cycle detection is also available. This is useful for implementing
lead-acid battery “top-off” charger behavior or the like.
See the LT1620 data sheet for further information.)
90
80
VIN = 12V
EFFICIENCY (%)
70
VIN = 48V
VIN = 36V
VIN = 24V
60
50
40
30
20
1
10
100
LOAD CURRENT (mA)
1000
The circuit as shown accommodates an input voltage
range of 11V to 30V. The upper input voltage limit of 30V
is determined not by the LT1776, but by the LT1121-5
regulator (U3). (A regulated 5V is required by the LT1620.)
This regulator was chosen for its micropower behavior,
which helps maintain good overall efficiency. However,
the basic catalog part is only rated to 30V. Substitution of
the industry standard LM317, for example, extends the
allowable input voltage to 40V (or more with the HV
version), but its greater quiescent current drain degrades
efficiency from that shown.
1676 F07b
Figure 82. Efficiency of Figure 81’s Circuit
AN84-48
www.BDTIC.com/Linear
Application Note 84
VIN
11V TO 30V
(SEE TEXT)
+
C1
39µF
63V
5
1
C5
100pF
6
VIN
SHDN
7
FB
2
VCC
U1
LT1776
3
VSW
8
VC
SYNC
GND
L1
100µH
C4
2200pF
C3
100pF
4
U3
LT1121-5
C6
0.33µF
C7
0.1µF
C8
1µF
R5
3k
R6
12k
R4
0.5Ω
7.2V
+
D1
MBRS1100
C2
100µF
10V
R3
22k
+
8
AVG
IOUT
3-CELL
LEAD-ACID
BATTERY
R2
12.1k
1%
6
VCC
R1
57.6k
1%
2
U2
LT1620
7
5
PROG
IN +
4
1
NC
SENSE
IN –
GND
1776 TA02
C1: PANASONIC HFQ
(201) 348-7522
C2: AVX TPSD107M010R0080
(803) 946-0362
L1: COILCRAFT DO3316P-104
(847) 639-6400
3
Figure 83. Wide VIN Range, High Efficiency Battery Charger
Dual Output SEPIC Converter
Positive-to-Negative Converter
All of the previous applications provide a single positive
output voltage. Real world situations often require dual
supply voltages. The SEPIC topology (single-ended primary inductance converter) offers a cost-effective way to
simultaneously generate a negative voltage with a single
piece of magnetics. The circuit in Figure 85 uses an
LT1776 to generate both positive and negative 5V. The two
inductors shown are actually just two windings on a
standard Coiltronics inductor. Capacitor C3 creates the
SEPIC topology, which improves regulation and reduces
ripple current in L1.
The previous example used a dual inductor to create a pair
of output voltages, one positive and the other negative.
The positive-to-negative converter topology illustrated in
Figure 86 generates a single negative output voltage from
a positive input voltage, using just an ordinary inductor.
The topology is somewhat similar to the original stepdown arrangement, but the inductor is grounded and the
LT1776 ground is now referred to the negative output
voltage. Note that the integrated circuit must now be rated
8
For the best negative supply voltage regulation, this output
should have a preload of at least 1% of the maximum
positive load. Total available current from both outputs is
limited to 500mA. Maximum negative supply current is
limited by the positive 5V load. A typical limit is one-half of
the positive current, but a more exact calculation includes
the input voltage. For this and further details of this
topology, see Linear Technology Design Note 100.
OUTPUT VOLTAGE (V)
7
6
5
4
3
2
1
0
0
50
100
150
200
OUTPUT CURRENT (mA)
250
1776 TA05
Figure 84. Battery Charger Output Voltage vs
Output Current for Figure 83’s Circuit
www.BDTIC.com/Linear
AN84-49
Application Note 84
for the worst case sum of the input voltage plus the
absolute value of the output voltage. The relatively high
input voltage rating of the LT1676/LT1776 parts along
with their good efficiency under such conditions make
them an excellent choice for implementing this topology.
The circuit as shown converts an input voltage in the range
of 10V to 28V to a –5V output. Available output current is
300mA at the worst case VIN of 10V.
actually more like a flyback topology, in that current is
delivered to the output in discrete pulses. The output
capacitor must supply the entire load current for at least
a portion of the switching cycle, so output capacitor ripple
current rating and ESR may be an issue. Maximum
available output current will usually be a strong function
of input voltage. Supporting low VIN-to-VOUT ratios may
require additional components for maintaining controlloop stability. A detailed theoretical analysis of this topology and its behavior can be found in Linear Technology
Application Note 44.
The user should exercise caution in modifying this circuit
for other applications. The positive-to-negative topology
is not as straightforward as the step-down topology. It is
VIN
10V TO 28V
+
C1
15µF
35V
5
VIN
1
VCC
SHDN
C7
100pF
LT1776
VSW
FB
6
VC
SYNC
2
L1* 100µH
3
7
8
D1
MBR1100
C6
100pF
C2
100µF
10V
+
GND
+
4
C1: AVX D CASE TPSD156M035R0300
(803) 946-0362
C2, C3, C4: AVX D CASE TPSD107M010R0080
C6, C7: X7R OR COG/NPO
D1, D2: MOTOROLA MBRS1100 100V, 1A, SMD SCHOTTKY
(800) 441-2447
*L1: COILTRONICS CTX100-3 SINGLE CORE WITH 2 WINDINGS
(561) 241-7876
R3
22k
5%
R2
12.1k
1%
C3
100µF
10V
C5
2200pF
X7R
VOUT 5V†
R1
36.5k
1%
VOUT –5V†
D2
MBR1100
+
L1*
100µH
C4
100µF
10V
†TOTAL AVAILABLE CURRENT IS LIMITED TO 500mA (SEE TEXT)
Figure 85. Dual-Output SEPIC Converter
VIN
10V TO 28V
+
C1
15µF
35V
5
VIN
1
C5
100pF
LT1776
6
C1: AVX D CASE TPSD156M035R0300
(803) 946-0362
C2: AVX D CASE TPSD107M010R0080
C4, C5: X7R OR COG/NPO
D1: MOTOROLA MBRS1100 100V, 1A, SMD SCHOTTKY
(800) 441-2447
L1: COILCRAFT D03316-104
(847) 639-6400
VCC
SHDN
VSW
FB
VC
SYNC
2
3
L1 100µH
7
8
GND
4
R3
22k
5%
C3
2200pF
X7R
C4
100pF
D1
MBRS1100
Figure 86. Positive-to-Negative Converter
AN84-50
R1
36.5k
1%
www.BDTIC.com/Linear
R2
12.1k
1%
+
C2
100µF
10V
VOUT
–5V
0mA TO 300mA
Application Note 84
Regulators—Switching (Boost)
The charge pump consists of two capacitors, two diodes
and a small inductor. When the power switch turns off, L1
also replenishes the charge on C5, forward biasing D3.
When the power switch turns on, the charge on C5 reverse
biases D3, forward biases D4 and supplies energy to C7
and the negative output load. L2 attenuates capacitive
current spikes. D2 was added so that the voltage drop
across both D1 and D2 would be approximately equal to
the sum of the voltage drops of D3, D4 and the saturation
voltage of the power switch in the LT1377. This makes
both output voltages approximately equal but opposite in
polarity. D1 and D2 can be replaced with a single Schottky
diode if equal outputs are not required.
±12 VOLT OUTPUT FROM THE LT1377
by John Seago
Many applications use positive and negative voltages,
with only one voltage requiring tight regulation. Often,
cost and board space are more important than regulation
of the second output. An equal output of opposite polarity
can be added to a boost configuration by means of a
negative charge pump. This two-output configuration is
shown in Figure 87. The 1MHz switching frequency of the
LT1377 decreases required board space, and the availability of both positive and negative feedback amplifiers
allows regulation of either positive or negative output.
Voltage and current waveforms of the internal power
switch are shown in Figure 88. These measurements were
taken at pin 8 of the LT1377 with the circuit powered from
a 5V supply. Figure 89 shows the ripple voltage from each
output. The high frequency spikes can be attenuated with
a small LC filter if necessary.
In the circuit of Figure 87, the LT1377 with L1, D1, D2 and
C6 make up a positive boost circuit. As the internal power
switch in the IC turns on, the voltage at pin 8 goes low and
energy is stored in inductor L1. When the power switch
turns off, L1 transfers energy through diodes D1 and D2
to capacitor C6 and the positive output load. C6 supplies
load current when the power switch is on. Resistors R2
and R3 provide feedback from the positive output. R1, C3
and C4 provide loop compensation. C1 is the input capacitor and C2 provides local decoupling for the IC.
The circuit of Figure 87 was intended to operate from a 5V
supply and provide ±12V outputs at 100mA each. It
operates over an input range of 4V to 10V and load current
variations from 15mA to 100mA. The regulated positive
output voltage remains constant for changes in the input
L1
10µH
4V TO 10V
INPUT
+
C1
10µF
25V
Y5U
D2
D1
MURS110 MURS110
12V
OUTPUT
5
4
VIN
VSW
S/S
U1
LT1377
C2
0.47µF
1
C3
0.0047µF
R1
2k
VC
FB
NFB
SGND
PGND
6
7
C4
0.047µF
C1 = C5 = 1E106ZY5U-C205-M, TOKIN (408) 432-8020
C6 = C7 = 1E225ZY5U-C203. TOKIN (408) 432-8020
L1 = CTX10-1P, COILTRONICS (407) 241-7876
L2 = PM20-R047M, GARRETT (805) 922-0594
R2
86.6k
8
+ C6
2
3
2.2µF
25V
Y5U
+ C5
10µF
25V
Y5U
R3
10k
D3
MBRS130L
D4
MBRS130L
+
L2
0.047µH
C7
2.2µF
25V
Y5U
–12V
OUTPUT
Figure 87. Positive Output Regulated Supply
www.BDTIC.com/Linear
AN84-51
Application Note 84
–13.00
–12.75
OUTPUT VOLTAGE
SWITCH VOLTAGE
5V/DIV
PIN 8
0
–12.50
±15mA LOAD
–12.25
±50mA LOAD
–12.00
±100mA LOAD
–11.75
–11.50
SWITCH CURRENT
0.5A/DIV
PIN 8
–11.25
0
–10.75
–11.00
0
0.5µs/DIV
1
2
3
4 5 6 7
INPUT VOLTAGE
8
9
10
Figure 90. Unregulated Negative Output
Voltage with Positive Output Voltage Regulated
Figure 88. Switch Voltage and Current Waveforms
13.00
12.75
12.50
OUTPUT VOLTAGE
12V OUTPUT
RIPPLE
0.1V/DIV
AC COUPLED
–12V OUTPUT
RIPPLE
0.1V/DIV
AC COUPLED
12.25
±100mA LOAD
12.00
±50mA LOAD
±15mA LOAD
11.75
11.50
11.25
11.00
10.75
0
1
2
3
4 5 6 7
INPUT VOLTAGE
8
9
10
0.5µs/DIV
Figure 91. Unregulated Positive Output Voltage
with Negative Output Voltage Regulated
Figure 89. Output Ripple Voltage
D1
MURS110
L1
10µH
4V TO 10V
INPUT
+
C1
10µF
25V
Y5U
D2
MURS110
12V
OUTPUT
5
4
VIN
VSW
S/S
U1
LT1377
C2
0.47µF
1
C3
0.0047µF
R1
2k
VC
FB
NFB
SGND
PGND
6
7
C4
0.047µF
C1 = C5 = 1E106ZY5U-C205-M, TOKIN (408) 432-8020
C6 = C7 = 1E225ZY5U-C203, TOKIN (408) 432-8020
L1 = CTX10-1P, COILTRONICS (407) 241-7876
L2 = PM20-R047M, GARRETT (805) 922-0594
8
3
+
C5
10µF
25V
Y5U
D3
MBRS130L
D4
MBRS130L
L2
0.047µH
R3
2.21k
R2
8.25k
Figure 92. Negative Output Regulated Dual Supply
AN84-52
C6
2.2µF
25V
Y5U
+
2
www.BDTIC.com/Linear
+
C7
2.2µF
25V
Y5U
–12V
OUTPUT
Application Note 84
voltage and load current, while the voltage of the unregulated negative output changes as shown in Figure 90. Line
and load regulation of the unregulated output will improve
with smaller changes of input voltage or load current.
A common requirement is for the positive output to
regulate the majority of power while the negative output
supplies a much smaller, unregulated bias current. Measurements taken on the test circuit of Figure 87 showed
the unregulated −12V output had less than ±1% variation
for a fixed 15mA load while the input voltage changed from
Introduction
The LT1370 is a 500kHz, 6A boost converter. At 65mΩ
on-resistance, 42V maximum switch voltage and 500kHz
switching frequency, the LT1370 can be used in a wide
range of output voltage and current applications.
The high efficiency switch is included on the die, along
with the oscillator, control and protection circuitry necessary for a complete switching regulator. This part combines the convenience and low parts count of a monolithic
solution with the switching capabilities of a discrete
power device and controller. The LT1370, features curL1*
D1
MBRD835L
VIN
OFF
ON
S/S
Occasionally, it is more important to regulate the negative
output than the positive output. The circuit in Figure 92 is
the same as that shown in Figure 87, except feedback
resistors R2 and R3 have different values and provide
feedback from the negative output to the negative feedback amplifier of the LT1377. Figure 91 shows the variation in unregulated positive output for input voltage and
load current variations.
rent mode operation, external synchronization and low
current shutdown mode (12µA typical). Only a few surface
mount components are needed to complete a small, high
efficiency DC/DC converter. The LT1370 will operate in all
the standard switching configurations, including boost,
buck, flyback, forward, inverting and SEPIC.
THE LT1370: NEW 500kHz, 6A
MONOLITHIC BOOST CONVERTER
by Karl Edwards
5V
4V to 10V with a load current change of 15mA to 200mA
on the regulated positive output.
VOUT†
12V
VSW
R1
53.6k
1%
FB
+
5V to 12V Boost Converter
Figure 93 shows a typical 5V to 12V boost application. The
feedback divider network has been selected to give the
desired output voltage. As long as R2 is less than 7k, FB
input bias current can be ignored. The inductor needs to be
chosen carefully to meet both peak and average current
values. The output capacitor can see high ripple currents—often, as in this application, higher than the ripple
rating of a single capacitor. This requires the use of two
surface mount tantalums in parallel; both capacitors should
be of the same value and manufacturer. The input capacitor does not have to endure such high ripple currents and
100
LT1370
C1**
22µF
25V
GND
C2
0.047µF
R3
2k
VC
R2
6.19k
1%
C3
0.0047µF
C4**
22µF
25V
×2
95
EFFICIENCY (%)
+
90
85
80
1370_02.EPS
*COILTRONICS
(561) 241-7876
UP2-4R7 (4.7µH)
UP4-100 (10µH)
**AVX TPSD226M025R0200
75
†
MAX IOUT
L1
IOUT
4.7µH 1.8A
10µH 2.0A
Figure 93. 5V to 12V Boost Converter
70
0
0.5
1.0
1.5
OUTPUT CURRENT (A)
2.0
1370_03
Figure 94. 12V Output Efficiency
www.BDTIC.com/Linear
AN84-53
Application Note 84
Positive-to-Negative Converter
VIN
2.7V TO 13V
+
C1
100µF
VIN
OFF
ON
VSW
S/S
LT1370
FB
2
D2
P6KE-15A
D3
1N4148 1 •
+
The NFB (negative feedback) pin allows negative output
regulators to be designed with direct feedback. In the
circuit shown in Figure 95, a 2.7V to 13V input, –5V output
converter, the output is monitored by the NFB pin and a
simple divider network. No complex level shifting or
unusual grounding techniques are required. The regular
FB pin is left open circuit and the divider network, R2, R3,
is calculated based on the –2.49V NFB reference voltage
and 30µA of input current. The switch-clamp diodes, D2
and D3, prevent the leakage spike from the transformer,
T1, from exceeding the switch’s absolute maximum voltage rating. The Zener voltage of D2 must be higher than the
output voltage, but low enough that the sum of input
voltage and clamp voltage does not exceed the switch
voltage rating.
C4
100µF
×2
3
R2
2.49k
1%
–VOUT†
–5V
R3
2.49k
1%
GND
C2
0.047µF
R1
2k
•4
D1
MBRD835L
NFB
VC
C3
0.0047µF
T1*
*PULSE PE-53719
(619) 674-8100
†MAX I
OUT
IOUT VIN
1.75A 3V
2.25A 5V
3.0A 9V
1370_04.EPS
Figure 95. Positive-to-Negative Converter with Direct Feedback
a single capacitor will normally suffice. The catch diode,
D1, must be rated for the output voltage and average
output current. The compensation capacitor, C2, normally
forms a pole with the internal gm of the part in the 2Hz to
20Hz range. It also creates a zero in conjunction with
series resistor R3, at 1kHz to 5kHz.
5V SEPIC Converter
Figure 96 shows a SEPIC converter. One of the advantages
of the SEPIC topology is that the input voltage can range
from below to above the output voltage. In Figure 96, the
input voltage range is from 4V to 9V, with a 5V output. The
magnetic coupling of inductors L1A and L1B is not critical
for operation, but generally they are wound on the same
core. C2 couples the inductors together and eliminates the
A second capacitor, C3, is sometimes required to prevent
erratic switching. Ripple current in the output capacitor’s
ESR causes voltage ripple. This feeds back through the
error amp to the VC pin, changing the current-trip threshold cycle-to-cycle. The problem appears as subharmonic
oscillation. Adding C3, typically one-tenth the value of
the main compensation capacitor, reduces the loop gain
at the switching frequency, preventing the oscillation.
VIN
4V TO 9V
L1A*
6.8µH
VIN
The ground return from the compensation network must
be separate from the high current switch ground. If drops
in the ground trace due to switch current cause the VC pin
to dip, premature switch-off will occur. This effect appears
as poor load regulation. A solution to this is to return the
compensation network to the FB pin. The S/S pin in this
example is driven by a logical on/off signal, a low input
forcing the LT1370 into its 12µA shutdown mode. Figure
94 shows the overall converter efficiency. Note that peak
efficiency is over 90%; efficiency stays above 86% at the
device’s maximum operating current.
OFF
ON
VSW
S/S
•
LT1370
+
C1
33µF
20V
FB
GND
D1
MBRD835L
R2
18.7k
1%
C2
4.7µF
•
VC
+
L1B*
6.8µH
R1
2k
C4
0.047µF
R3
6.19k
1%
C5
0.0047µF
1370_05.EPS
C1 = AVX TPSD 336M020R0200
C2 = TOKIN 1E475ZY5U-C304
C3 = AVX TPSD107M010R0100
* BH ELECTRONICS 501-0726
(612) 894-9590
** INPUT VOLTAGE MAY BE GREATER OR
LESS THAN OUTPUT VOLTAGE
†MAX I
OUT
IOUT VIN
1.8A 4V
2A 5V
2.6A 7V
2.9A 9V
Figure 96. Two Li-Ion Cells to SEPIC Converter
AN84-54
www.BDTIC.com/Linear
VOUT†
5V
C3
100µF
10V
×2
Application Note 84
need for a switch snubber network. C2 must have a very
low ESR, because the ripple current is equal to ISW/2. Its
capacitance value is not critical and has no significant
effect on loop stability. The voltage across C2 is equal to
the input. A 4.7µF, 50V ceramic will work in most SEPIC
applications. The S/S pin is used as a logical on/off signal.
In the off state, there is no leakage to the output, and only
12µA leakage from the input.
95
BOOTSTRAPPED SYNCHRONOUS BOOST CONVERTER
OPERATES AT 1.8V INPUT
by Tom Gross
ILOAD = 1A
EFFICIENCY (%)
90
Some applications, such as those powered by batteries or
solar cells, see their input voltage decrease as they operate. Many regulators that could operate with high input
voltages cease to function as the input voltage decreases.
The circuit in Figure 97 maintains the maximum load
current as the input voltage drops. The regulator boosts a
2.5V–4.2V input to 5V at a maximum load current of 2A
(10W of output power).
85
ILOAD = 0.25A
80
75
70
2.7
2.9
3.1
3.3
3.5 3.7
VIN (V)
3.9
4.1
4.3
1630_03.EPS
Figue 98. Efficiency of Figure 97’s Circuit
voltages allow the circuit to start up at low input voltages
(crucial for low series-cell-count, battery-powered applications). Diodes D3 and D4, along with capacitor C2,
form a charge-pump circuit, which the controller uses for
the MOSFETs’ gate drive. The switches are driven by an
LTC1266 synchronous regulator controller.
The circuit is a bootstrapped synchronous boost regulator
using an LTC1266 synchronous regulator controller.
Diodes D1 through D4 allow the circuit to start-up using
the (low) input voltage and then to be powered during
normal operation by the higher output voltage. The crucial
elements in this circuit are the switches: two IRF7401 Nchannel MOSFETs. These MOSFETs are fully enhanced at
very low gate-to-source voltages (at 2V of VGS, the peak
drain current is rated at 15A). The low enhancement
Because the circuit is powered from the 5V output, it will
still operate if the input supply voltage drops below the
minimum input voltage of the IC. This bootstrapping
R3 0.025Ω
VIN
2.5V TO 4.2V
C1
330µF
6.3V
TANT
ILOAD = 2A
R1
100Ω
1000pF
+
D1
D2
D3
D4
R2
100Ω
C2 1µF
9 LTC1266
SENSE+
2
D1 TO D4 = MBR0530T1
3
4
5
6
+
C5
10µF
16V
C7
220pF
C6
1µF
C8
0.012µF
7
8
SENSE–
PWR VIN BDRIVE
PINV
LBOUT
BINH
LBIN
VIN
TDRIVE
CT
VFB
ITH
SHDN
R8
SGND
6.2k
12
C9
0.033µF
L1
22µH/7A
COILCRAFT
DO-5022-223
16
R4 2Ω
Q2
IRF7401
14
13
1
10
11
R5 2Ω
Q1
IRF7401
R6
100k
1%
+
C4
1200pF
D5 MBRS120T3
PGND
15
VOUT
5V/2A
C3
3× 330µF
6.3V
TANT
R7
33.2K
1%
Figure 97. Bootstrapped Synchronous Boost Converter
www.BDTIC.com/Linear
AN84-55
Application Note 84
allows the circuit to start up even when the input voltage
is below the minimum input voltage of the IC (3.5V). With
a 1A load, the regulator operates down to 1.8V.
Figure 98 shows the efficiency of the regulator versus the
input voltage at three different load currents. At 2A of load
current, efficiency drops as the input voltage is decreased
due to the higher power losses in the inductor. A larger
inductor will increase efficiency and/or allow for larger
load currents. The efficiency with the indicated inductor is
good, averaging above 83% overall. Higher efficiency will
help to increase the run time of battery-powered
applications.
2000
Regulators (Switching)—Buck-Boost
500kHz BUCK-BOOST CONVERTER
NEEDS NO HEAT SINK
by Mitchell Lee
OUTPUT (mA)
1500
1000
Thanks to an efficient 0.25Ω switch, the LT1371 SEPIC
converter shown in Figure 99 operates at full power with
no heatsink. Up to 9W at 5V output is available, and the
circuit works over a wide range of input voltages extending
from the LT1371’s 2.7V minimum to 20V, limited by the
rating of the capacitors.
500
0
0
5
10
INPUT (V)
15
20
Figure 100. Maximum Avialable Output Current
A 1:1 bifilar-wound toroid is used as the magnetic element. A careful analysis showed that, in spite of the
500kHz operating frequency, a high permeability (mr =
125) Magnetics Inc. Kool Mµ® core exhibited the best
efficiency when compared to powdered iron materials.
Copper loss is minimized by the use of the high-perm Kool
Mµ material, with only a slight core-loss penalty.
2.7V TO 20V
INPUT
L1
HL-8798
100µF +
20V
OS-CON
+
33µF
20V
150µF
6.3V
OS-CON
+
5V
OUTPUT
Maximum available output current varies with input voltage, and is shown (for 3A peak switch current) in Figure
100. Efficiencies for several input voltages are shown in
Figure 101. At a 2.7V input, most of the loss is tied up in
the LT1371 switch, whereas the output diode is the
dominant source of loss with high inputs. Because these
losses are small, surface mount construction provides
adequate dissipation, eliminating the need for heat sinks.
In this application, the synchronization feature of the
LT1371 is not used. When driven with an external clock at
the shutdown/sync pin (S/S), the chip can be synchronized to any frequency between 600kHz and 800kHz.
OS-CON MBRS340T3
VIN
NC
3.6k
SW
VIN = 12V
S/S
80
LT1371
NFB
FB
GND
VC
20k
47nF
1.2k
4.7nF
L1 = HURRICANE ELECTRONICS LAB HL-8798
(801) 635-2003, FAX (801) 635-2495
COILTRONICS CTX10-4
(561) 241-7876, FAX (561) 241-9339
EFFICIENCY (%)
OFF ON
90
VIN = 5V
70
60
VIN = 2.7V
50
0
Figure 99. 5V, 9W Converter Operates Over a Wide Input Range
with Good Efficiency
AN84-56
500
1000
LOAD (mA)
1500
2000
Figure 101. Efficiency of Figure 99’s Circuit
www.BDTIC.com/Linear
Application Note 84
R1 100k
BATTERY-POWERED BUCK-BOOST CONVERTER
REQUIRES NO MAGNETICS
by John Seago
LTC1515CS8-3.3/5
1
ON OFF
The LTC1515 switched capacitor DC/DC converter, can
provide this buck-boost function for load currents up to
50mA with only three external capacitors. The circuit
shown in Figure 102 will provide a regulated 3.3V output
from a 3-cell input or a 5V output from a 4-cell input.
Connecting the 5/3 pin to VIN will program the output to
Regulators—Switching (Inverting)
MAKING –5V 14-BIT QUIET
by Kevin R. Hoskins
Many high performance data acquisition systems reap
multiple benefits when using ±5V supplies rather than a
single 5V supply. These benefits include the ability to
handle larger signal magnitudes than is possible with a
single 5V supply. This increases a system’s dynamic range
and helps improve the signal-to-noise ratio. Operating on
±5V also increases headroom, which is important for
signal conditioning. Compared to operating on 5V, conditioning circuitry operating on ±5V has twice the headroom,
allowing it to easily handle ±2.5V signals without clipping.
Additionally, the greater headroom avoids the limitations
of rail-to-rail operation and widens the selection of high
performance operational amplifiers and analog-to-digital
converters, such as the LTC1419.
2
RESET
3
5V 3.3V
7
C2
10µF
10V
(1206)
3 OR 4
CELLS
VOUT
POR
GND
3.3V/50mA
OR
5V/50mA
8
4
5/3
6
CI+
VIN
CI–
C3
10µF
10V
(1206)
5
C1
0.1µF
(0603)
C1 = AVX 0603YC104MAT2A
C2, C3 = TAIYO YUDEN LMK316F106ZL
Figure 102. Battery-Powered Buck-Boost Converter
5V, whereas grounding the 5/3 pin programs the output to
3.3V.
The absence of bulky magnetics provides another benefit:
this circuit requires only 0.07 square inches of board
space in those applications where components can be
mounted on both sides of the board. The addition of R1
provides a power-on-reset flag that goes high 200ms after
the output reaches 93.5% of its programmed value. The
SHDN pin allows the output to be turned on or off with a
3V logic signal.
supply, they are not generally recommended for use with
ADCs. Typical ADCs have inadequate PSRR, which decreases with increasing frequency. This poor PSRR performance cannot sufficiently attenuate the noise created
by switching or charge-pump supplies. However, LTC’s
AMPLITUDE OF POWER SUPPLY FEEDTHROUGH (dB)
One of the problems that designers of portable equipment
face is generating a regulated voltage that is between the
charged and discharged voltage of a battery pack. As an
example, when generating a 3.3V output from a 3-cell
battery pack, the regulator input voltage changes from
about 4.5V at full charge to about 2.7V when discharged.
At full charge, the regulator must step down the input
voltage, and when the battery voltage drops below 3.3V,
the regulator must step up the voltage. The same problem
occurs when a 5V output is required from a 4-cell input
voltage that varies from about 3.6V to 6V. Ordinarily, a
flyback or SEPIC configuration is required to solve this
problem.
SHDN
0
VRIPPLE = 0.1V
–20
–40
–60
VSS
–80
VDD
DGND
–100
–120
1k
10k
100k
1M
RIPPLE FREQUENCY (Hz)
10M
1410 G08
Although a switching or charge-pump power supply is an
efficient way to create a –5V supply from a single 5V
Figure 103. The LTC1419’s Positive Supply PSRR of 90dB to
200kHz is a Significant Contributor to this ADC’s Wideband
Conversion Performance and 80dB SINAD
www.BDTIC.com/Linear
AN84-57
Application Note 84
new family of ADCs, here represented by the LTC1419,
has excellent PSRR. This family makes it easy to achieve
high performance data conversion, even at 14 bits, using
a switch-mode regulator for a –5V supply.
ondary inductor. This current waveform is continuous,
producing much less harmonic content than is created by
a typical positive-to-negative voltage converter, with its
rectangular switching current waveform. With the components shown, the LT1373 operates continuously with load
currents above 10mA. Because the LTC1419 typically
draws 18mA of negative supply current, the LT1373 will
always operate in the quiet continuous mode.
The LTC1419’s high PSRR is shown in Figure 103. It
shows that when operating on ±5V, the negative and
positive PSRR are typically 80dB and 90dB, respectively,
up to 200kHz for a 100mV ripple voltage. Combined with
proper layout, the LTC1419’s high PSRR allows it to
convert signals without signal degradation while using
switching regulators and charge pumps to generate its
–5V supply. Applications including high speed communications, high resolution signal processing and wideband
multiplexing benefit from the LTC1419’s advantages—its
20MHz S/H bandwidth, 800ksps conversion rate and 14bit resolution. This article shows two supply designs that
are quiet enough to use with the LTC1419.
Regulated Charge Pump Converter
The LTC1419’s negative PSRR also allows the use of
charge pumps to create –5V. The circuit shown in Figure
105 uses the LT1054 regulated charge pump. This circuit
has the advantage of reduced board space, since it lacks an
inductor and requires fewer passive components. However, the LT1373 circuit can supply more current (150mA)
than the LT1054 circuit (100mA).
Low Noise Inverting Converter
Performance Results
The LT1373 switching regulator shown in Figure 104 is
configured as an inverting converter, creating –5V from
5V. This configuration has the advantage of a small
triangular switching-current waveform through the sec-
What is the effect of using either of these switch-based
supplies on the LTC1419’s conversion performance? The
FFTs in Figures 106–108 show the excellent results. Figure
ANALOG INPUT
5V
L1
2
3
C6
1
2
INVERTING
CONVERTER
5
C8
22µF
10V
TANT
+
4
7
6
S/S
C10
10µF
CER
8
VSW
VIN
U2
LT1373
GND
GND S
4
C11
100µF
10V
TANT
R4
4.99k
1%
3
NFB
1
VC
1µF CER
3
4
C5
5
6
C12
0.1µF
7
D1
R3
4.99k
C9
0.01µF
+
1
8
R5
4.99k
1%
R6
499Ω
1%
9
10
11
12
13
C5, C6, C7 = 10µF CERAMIC
L1 = OCTAPAC CTX-100-1
D1 = 1N5818
14
+AIN
AVDD
–AIN
DVDD
VREF
VSS
28
27
C7
26
25
U1
BUSY
LTC1419
24
AGND
CS
23
D13 (MSB)
CONVST
22
RD
D12
21
D11
SHDN
20
D10
D0
19
D9
D1
18
D8
D2
17
D7
D3
16
D6
D4
15
DGND
D5
COMP
DI_1419_01.eps
Figure 104. The LTC1419’s 80dB PSRR Allows the LTC1373 to Generate
the –5V and Power the ADC without Signal-Coversion Degradation
AN84-58
www.BDTIC.com/Linear
MICROPROCESSOR/
MICROCONTROLLER
INTERFACE
Application Note 84
5V
C4
100µF
TANT
C2
2µF
2
C1
10µF
TANT
FB/SHDN
CAP+
V+
8
7
OSC
U1
LT1054
6
VREF
5
4
CAP –
VOUT
+
3
R1, 30.1k
ANALOG
INPUT
3
4
C3
0.002µF
GND
2
1µF CER
+
1
C6
1
C5
R2, 120k
5
6
7
C5, C6, C7 = 10µF CERAMIC
8
9
10
11
Figure 105. The LTC1419’s High Negative Supply PSRR also
Allows the Use of the LT1054 Regulated Charge Pump to
Generate –5V without Loss of Performance
12
13
14
–AIN
DVDD
VREF
VSS
28
27
C7
26
25
U2
BUSY
LTC1419
24
AGND
CS
23
D13 (MSB)
CONVST
22
RD
D12
21
D11
SHDN
20
D10
D0
19
D9
D1
18
D8
D2
17
D7
D3
16
D6
D4
15
DGND
D5
COMP
MICROPROCESSOR/
MICROCONTROLLER
INTERFACE
DI_1419_02.eps
LTC1419
5V LAB SUPPLY
–5V LT1373
fSAMPLE =800kHz
fIN = 91kHz
S/N = 80.5dB
–20
–40
–60
–80
–100
–120
–140
–160
0
50
100
150
200
250
FREQUENCY (kHz)
300
350
400
DI_1419_04.eps
Figure 107. When the –5V Supply is Generated by an LT1373
Switching Regulator, the SINAD, the Noise Floor, and the 91kHz
Fundamental’s Harmonics Remain Essentially the Same as
in Figure 106
0
0
LTC1419
±5V LAB SUPPLIES
fSAMPLE = 800kHz
fIN = 91kHz
S/N = 80.5dB
–40
–60
LTC1419
5V LAB SUPPLY
–5V LT1054
fSAMPLE = 800kHz
fIN = 91kHz
S/N = 80.8dB
–20
–40
AMPLITUDE (dB)
–20
AMPLITUDE (dB)
AVDD
0
AMPLITUDE (dB)
106 is an FFT of a typical LTC1419 operating on ±5V from
a lab supply and converting a full-scale 91kHz sine wave
at 800ksps. The noise floor is approximately 114dB below
full scale, the second harmonic’s amplitude is approximately 90dB below full scale and the SINAD is 80.5dB.
Figure 107 shows the FFT of the same LTC1419 operating
on a 5V lab supply and –5V from the LT1373 circuit. The
noise floor and the second harmonic’s amplitude remain
the same relative to full scale and the SINAD remains the
same at 80.5dB. Figure 108 shows the LTC1419’s response when its –5V is generated by the LT1054 circuit.
As with the LT1373 circuit, the noise floor and the amplitude of the harmonics remain the same and the SINAD is
80.8dB.
+AIN
–80
–100
–60
–80
–100
–120
–120
–140
–140
–160
–160
0
50
100
150
200
250
FREQUENCY (kHz)
300
350
400
Figure 106. This FFT of an LTC1419 Powered by a ±5V
Lab Supply Shows a SINAD of 80.5dB for a 91kHz Input
sampled at 800ksps
DI_1419_03.eps
0
50
100
150
200
250
FREQUENCY (kHz)
300
350
400
Figure 108. When the –5V Supply is Generated by an LT11054
Inverter, the SINAD, the Noise Floor, and the 91kHz Fundamental’s
Harmonics Are Again Unchanged from Those in in Figure 106
www.BDTIC.com/Linear
DI_1419_05.eps
AN84-59
Application Note 84
NEGATIVE-TO-POSITIVE
TELECOMMUNICATION SUPPLY
by Kurk Mathews
+VOUT
+
Many telecommunication circuits require a positive supply voltage derived from a –48V input. The traditional
approach to negative-to-positive conversion has been to
use a buck-boost converter (see Figure 109). Unfortunately, this topology suffers drawbacks as the power level
and input-to-output voltage difference increases.
+
–VIN
DI_1680_01.eps
Figure 109. Buck-Boost Converter
a bootstrap winding on the output inductor, L1. When input
voltage is first applied, R1 begins charging C1. As C1
charges, Q1 is held on by R2, shorting R3. R4 and R5 form
a voltage divider that holds the RUN/SHDN pin below its
1.25V threshold until the 12VIN pin reaches approximately
14V. Once out of standby, Q1 is turned off by Q2, reducing
the run threshold to approximately 9V and allowing C1 time
to discharge slightly before the overwinding on L1 takes
over. The only remaining issue is feedback. Q3 translates
the output voltage to a current, which flows to the VFB pin.
A more appropriate solution for –48V to 5V conversion is
shown in Figure 110. The LT1680 is used to implement a
forward converter with its output referenced to the input
common. Compared to the buck-boost converter, switch
current is reduced by a factor of two and output capacitor
ripple is reduced by a factor of five.
The LT1680 is referenced to –48V and requires a 12V bias
supply. The 12V is generated by using the RUN/SHDN and
T1 = COILTRONICS VP5-1200, 1:1:1:1:1:1
(SIX WINDINGS, EACH 77µH)
L1 = PHILIPS EFD20-3F3-E63-S CORE SET (Al = 63nH/T2)
OUTPUT 18T BIFILAR 22AWG,
BIAS 54T BIFILAR 32 AWG
* = SANYO CV-GX
INPUT
COM
33Ω
1.5nF
T1
L1, OUTPUT
20µH
5V/6A
+
220µF* 220µF*
35V + 35V
24k
1k
1µF
63V
MBR2045CT
300pF
24k
+
220µF* 220µF*
35V + 35V
33Ω
1.5nF
MBR0520LT1
+
50Ω
1W
0.015Ω
1W
IRF640
R1
24k
330µF
6.3V
SANYO
OSCON
10Ω
4.22k
–48V
Q3
2N5401
R4
78.7k
15
SYNC
9
10
13
S– GATE
S+
11 RUN/SHDN
1
SL_ADJ
BAV21
L1
BIAS
R2
1M
C1
220µF
35V
+
14 12V
IN
LT1680
VFB
0.1µF
R5
7.5k
5VREF
CT
IAVG
16
2
3
Q1
2N7000
20k
Q2
2N3904
16k
SS
4
7
VC SGND PGND VREF
5
6
12
8
2.2nF
0.22µF
0.1µF
R3
4.75k
1µF
1nF
20k
1.2k
0.1µF
1k
Figure 110. –48V to 5V Telecommunications Supply
AN84-60
www.BDTIC.com/Linear
Application Note 84
The LT1680’s unique differential current sense amplifier
has an input common mode range of –0.3V to 60V. If VIN
is expected to exceed 60V, the sense resistor could be
relocated in the main FET’s source and the input capaciPOSITIVE-TO-NEGATIVE CONVERTER POWERS
–48V TELECOM CIRCUITS
by Mitchell Lee
If you’re designing a system that interfaces to telecom
equipment, chances are you’ll need a –48V supply. The
circuit in Figure 111 supplies up to 6W at –48V and scales
to more than 12W with higher power components. Based
on the inverting topology, the converter exhibits excellent
efficiency over a wide range of loading conditions (see
Figure 112).
CTX02-13836†
3
12
2
10
5
11
4
8
6
9
7
220µF
16V
The combination of the LT1171 and the VP-2 series
VERSA-PAC™ coil (CTX02-13836) are suited for 120mA
output current as shown. For lighter loads of up to 60mA,
use the LT1172 and a VP-1-series equivalent to the coil
shown. For up to 15W, use the LT1171 and a VP-5
equivalent. High voltage versions of the LT1170 family
(-HV) allow inputs of up to 20V without exceeding the peak
switch-voltage rating.
150µF
63V
VSW
VIN
D3**
This converter starts working at 2.7V and will regulate
–48V at reduced power. You can add undervoltage lockout
by inserting a Zener diode (VZ = VLOCKOUT – 2.7V) between
the input supply and the LT1172’s VIN pin.
LT1172HV
VC
GND
The LT1171’s error amplifier is designed for positiveboost applications, and hence its gain and reference are of
the wrong phase and polarity for sensing an inverted
output. In this application, the error amplifier is simply
bypassed and feedback is applied at the compensation
(VC) pin. Zener diode D2 senses the output, pulling down
on Q1 and the VC pin, in response to small increases in
output voltage. Pulling down on the VC pin reduces peak
switch current, and constitutes negative feedback. If the
output is a little low, the Zener’s diminished feedback
signal is overcome by an internal 200µA current source at
the VC pin, thereby increasing peak switch current and
restoring the output voltage.
FB
100
1k
100nF
D1*
10nF
90
1µF
GND
Q1
2N3904
1k
+
*D1 = 1N4148
**D3 = MUR120
†COILTRONICS
(561) 241-7876
1k
D2
1N5261B
47V
Figure 111. 12V to –48V Converter Features Good
Efficiency Over a Wide Range of Loads
220µF
63V
EFFICIENCY (%)
+
1
+
VIN
12V
tors’ voltage increased. Because the forward converter is
fundamentally an isolated topology, an optocoupler and
reference could be added to provide isolation between the
input and output of the supply.
80
70
60
VOUT
–48V/
120mA
50
0
30
90
60
LOAD CURRENT (mA)
120
DI_1171_02. EPS
Figure 112. Converter Efficiency
Rises to 80% at Only 20mA Load
www.BDTIC.com/Linear
AN84-61
Application Note 84
LOW NOISE LT1614 DC/DC CONVERTER DELIVERS
–5V AT 200mA FROM 5V INPUT
by Steve Pietkiewicz
The inverting DC/DC converter function is traditionally
realized with a capacitor-based charge pump. Although
simple, the output impedances of the best charge pump
solutions are in the 5Ω to 10Ω range, resulting in significant
regulation issues when the load current increases beyond a few tens of milliamperes. The LT1614 inductorbased inverting DC/DC converter uses closed-loop regulation to obtain an output impedance of 0.1Ω, eliminating
output voltage droop under load.
series with both input and output, results in low output
noise and also in low reflected noise on the 5V input
supply. The output and switch nodes are shown in Figure
114. Output ripple voltage of 40mV is due to the ESR of the
tantalum output capacitor C2. Ripple voltage can be reduced substantially by replacing output capacitor C2 with
a 10µF ceramic unit, as pictured in Figure 115.
VOUT
100mV/DIV
AC COUPLED
Figure 113 details the 5V to –5V converter circuit. The
LT1614 contains an internal 0.6Ω switch rated at 30V,
allowing up to 28V differential between input and output.
Quiescent current is 1mA and the device contains a lowbattery detector with a 200mV reference voltage. The
device switches at 600kHz, allowing the use of small,
inexpensive external inductors and capacitors. In fact, the
total cost of the components specified in Figure 113
(excluding the LT1614) is approximately $0.70 in 10,000piece quantities.
VOUT
100mV/DIV
AC COUPLED
The LT1614 operates by driving its NFB pin to a voltage of
–1.24V, allowing direct regulation of the negative output.
This converter topology, which consists of inductors in
VSW
5V/DIV
C3
1µF
L1
22µH
VIN
5V
Figure 114. LT1614 Output and Switch Node with a 33µF
Tantalum Capacitor and 200mA Load Current
L2
22µH
500ns/DIV
Figure 115. LT1614 Output and Switch Node with a 10µF
Ceramic Output Capacitor and 200mA Load Current
D1
SHDN
VC
VOUT
–5V/200mA
R1
69.8k
LT1614
C1
33µF
500ns/DIV
SW
VIN
+
VSW
5V/DIV
VOUT
100mV/DIV
AC COUPLED
NFB
GND
R2
24.9k
+
100k
C2
33µF
1nF
VSW
5V/DIV
D1: MBR0520
(800) 441-2447
L1, L2: MURATA LQH3C220 (814) 237-1431
C1, C2: AVX TAJB336M010 (803) 946-0362
C3: AVX1206YC105KAT (CERAMIC, X7R)
Figure 113. 5V to –5V DC/DC Converter Uses an Inverting
Topology and Delivers 200mA.
AN84-62
500ns/DIV
Figure 116. Improper Placement of D1’s Cathode
Results in 60mV Switching Spikes at Output, Even
with a 10µF Ceramic Output Capacitor
www.BDTIC.com/Linear
Application Note 84
90
In layout, be sure to tie D1’s cathode directly to the
LT1614’s GND pin, as shown in Figure 113. This keeps the
switching current loops tight and prevents the introduction
of high frequency spikes on the output. The low noise that
can be achieved with a ceramic capacitor may be corrupted
by noise spikes if proper layout practice is not followed. To
illustrate this point, output and switch waveforms from
Figure 113’s circuit, with a 10µF ceramic output capacitor
and 200mA load, but with D1’s cathode arbitrarily connected to the ground plane, are shown in Figure 115. 60mV
switching spikes ruin an otherwise clean output.
EFFICIENCY (%)
80
70
60
50
40
3
10
30
100
LOAD CURRENT (mA)
300
Figure 117. 5V to –5V Converter Efficiency Reaches 73%
1610 TA02
Efficiency of the circuit is detailed in Figure 117. Efficiency
reaches 73% at a 50mA load, and is above 70% at a
200mA load. Larger inductors with less copper resistance
can be used to increase efficiency, although such inductors are more expensive than the Murata units specified.
–48V TO 5V DC/DC CONVERTER OPERATES
FROM THE TELEPHONE LINE
by Gary Shockey
power consumption of 25mW maximum. The DC/DC
converter circuit presented here is 70% efficient at an
input power of 25mW, providing 5V at 3.4mA. Controlled,
low peak switch current ensures that the –48V input line
does not experience excessive voltage drops during
switching.
DC/DC converters for use inside the telephone handset
require operation from the high source-impedance phone
line. Additionally, the CCITT specifications call for on-hook
D1
T1
1N5817
10:1:1
+
L1
L3
VA
D3
1N4148
Q1
R4
2M
Q3
2N3904
6
7
C1
0.1µF
R2
1.30M
1%
R3
604k
1%
1
2
C3
47µF
D2
1N4148
L2
R1
1.3M
C2
0.022µF
VOUT
5V
VIN
SHDN
LB0
LT1316
+ C4
47µF
5
SW
FB
R7
Q2
MPSA92 432k, 1%
8
DI_48-5_01.eps
LBI
RSET
GND
3
4
R5
69.8k
1%
R6
121k
1%
– 48V
T1 =DALE LPE-4841-A313, LPRI = 2mH
Q1 =ZETEX ZVN 4424A
(605) 665-1627
(516) 543-7100
R6, Q2 AND R7 MUST BE PLACED
NEXT TO THE FB PIN
Figure 118. –48V to 5V Flyback Converter
www.BDTIC.com/Linear
AN84-63
Application Note 84
The circuit shown in Figure 118 operates as a flyback
regulator with an auxiliary winding to provide power for
the LT1316. To understand the operation of this circuit,
examine Figure 118. When power is first applied, the LBI
pin is low, causing the SHDN pin to be grounded through
LBO. This places the part in shutdown mode and only the
low-battery comparator remains active. During this state,
VIN rises at a rate determined by R1 and C1. The LT1316
draws only 6µA in shutdown mode; R1 needs to supply
only this current, the current through R2 and R4, and C1’s
charging current. When LBI reaches 1.17V (VIN ≈ 3.7V)
the LBO pin lets go of SHDN and the part enters the active
mode. Once this state is reached, switching action begins
and the output voltage begins to increase. As the device
switches, the LT1316 VIN pin draws current out of C1; VIN
then decreases sufficiently to trip the low-battery detector, stopping the switching. Start-up proceeds in this
irregular fashion until, eventually, the voltage at VA increases
to 5V. (VA is the same as VOUT, because L2 and L3 have the
same number of turns.) After start-up, current is supplied
to the LT1316 from VA rather than from the –48V rail,
increasing efficiency. VOUT must not be loaded until it
reaches 5V or the circuit will not start.
During each switch cycle, current in the transformer
primary ramps up until current limit is reached (See
Figures 119 and 120). This peak switch current can be set
by adjusting R5. The circuit shown uses a 69.8kΩ resistor
to give a peak switch current of 50mA. Increasing R5
decreases the current limit. Secondary peak current will be
approximately equal to the primary peak current multiplied by the transformer turns ratio. The FB pin has a sense
voltage of 1.23V and VOUT can be set by the following
formula:
VOUT = 1.23(R7/R6) + 0.6V.
Efficiency versus load current is detailed in Figure 121.
Note that for the range of 4mA to 80mA, 70% efficiency or
greater is achieved. Figure 122 shows input current versus
output power. Less than 80µA quiescent current flows
when the converter supplies 0.5mW over the 36V–72V
range.
VOUT
AC COUPLED
100mV/DIV
SWITCH PIN
VOLTAGE
10V/DIV
SECONDARY
CURRENT
200mA/DIV
SECONDARY
CURRENT
200mA/DIV
PRIMARY
CURRENT
50mA/DIV
PRIMARY
CURRENT
50mA/DIV
50µS/DIV
1µS/DIV
Figure 119. Switch Voltage and Current Waveforms
Figure 120. Output Ripple Voltage and Current Waveforms
90
0.3
VIN = 36V
VIN = 48V
INPUT CURRENT (mA)
EFFICIENCY (%)
80
70
VIN = 72V
60
VIN = 36V
0.2
VIN = 48V
VIN = 72V
0.1
50
40
1
10
LOAD CURRENT (mA)
100
DI_48-5_04.eps
Figure 121. Efficiency vs Load Current
AN84-64
0
0
1
2
3
POWER OUT (mW)
4
DI_48-5_05.eps
Figure 122. Input Current vs Power Out
www.BDTIC.com/Linear
5
Application Note 84
Regulators—Switching (Flyback)
Typical Applications
THE LT1425 ISOLATED FLYBACK CONTROLLER
by Kurk Mathews
Figure 123 shows a typical flyback LAN supply using the
LT1425. Figure 123 also includes details on an alternate
transformer for a complete PCMCIA type II height solution. The output voltage is within 1% of –9V for load
currents of 0mA–250mA. Input current is limited to 0.35
amps in the event the output is short circuited. The output
voltage droops only 300mV during a 50mA to 250mA load
transient (see Figure 124). The off-the-shelf transformers
provide 500VAC of isolation. The high switching frequency
allows the use of small case size, low cost, high value
ceramic capacitors on the input and output of the supply.
Introduction
Low voltage circuitry, such as local area networks (LAN),
isolation amplifiers and telephone interfaces, frequently
requires isolated power supplies. The flyback converter is
often the choice for these low power supplies because of
its simplicity, size and low parts count. Unfortunately,
designers are forced to add optocouplers and references in order to achieve the desired output regulation and
transient response.
The LT1425 provides a one-chip solution for these and
other applications. The LT1425 is a 275kHz current mode
controller with an integral 1.25A switch designed primarily
to provide well regulated, isolated voltages from 3V–20V
sources. The LT1425 is available in a 16-pin SO. Features
include a new error amplifier and load compensation
circuitry that eliminate the need for optocouplers while
maintaining output regulation typically within a few percent.
Figure 126 shows a ±15V supply with 1.5kV of isolation.
Output regulation remains within ±3% over the entire 5V
to 15V input voltage and ±60mA output current range,
200mV/DIV
100mA/DIV
5ms/DIV
Figure 124. Transient Response of LT1425 5V to –9V Converter
5V
C2
10µF
25V
C1
10µF
25V
1
0.1µF
22.1k
1%
2
3
4
5
100k
1000pF
47pF
3.01k
1%
6
7
8
INPUT
COM
GND
N/C
LT1425
GND
SD
RFB
ROCMP
VC
RCMPC
RREF
VIN
SYNC
VSW
SGND
PGND
GND
GND
16
D1
15
14
C5
13
D2
MBRS130LT3
R1
2
1
T1
C3
10µF
25V
4
11
3
C6
OUT
COM
7
R3
12
R2
C4
10µF
25V
1.8k
6
–9V
10
0.1µF
C1, C2, C3, C4 = MARCON THCS50E1E106Z CERAMIC
CAPACITOR, SIZE 1812. (847) 696-2000
9
1425_01.eps
TRANSFORMER T1
L PRI
TURNS
RATIO
ISOLATION
SIZE
(L × W × H)
IOUT
EFFICIENCY
D1
D2
R1, R2
C5, C6
R3
DALE
LPE-4841-A307
36µH
COILTRONICS
CTX02-13483
1:1:1
500VAC
10.7 × 11.5 × 6.3mm
250mA
76%
NOT USED
NOT USED
47Ω
330pF
13.3k
27µH
1:1
500VAC
14 × 14 × 2.2mm
200mA
70%
1N5248
MBR0540TL1
75Ω
220pF
5.9k
Figure 123. 5V to –9V/250mA Isolated LAN Supply
www.BDTIC.com/Linear
AN84-65
Application Note 84
A: VSW = 20V/DIV
B: ISW = 1A/DIV
C: VSW = 20V/DIV
D: ISW = 0.2A/DIV
1µs/DIV
Figure 125. Switch Voltage and Current for
Figure 123’s Circuit with Outputs of
–9V/250mA and –9V/30mA
even with one output fully loaded and the other unloaded
(±1.5% with input voltages of 10V–15V). The isolation
voltage is ultimately limited only by bobbin selection and
transformer construction.
Figure 127 implements a 12V to 5V/1A step-down regulator with off-the-shelf magnetics. The circuit uses an external, cascoded 100V MOSFET to extend the LT1425’s 35V
maximum switch voltage limit. D1 and Q1 ensure the
LT1425 does not start until almost 9V, guaranteeing
adequate gate voltage for the MOSFET. The MUR120
prevents the source from rising above the gate at turn-off.
The circuit in Figure 128 achieves even higher input
voltages, this time in the form of a –48V to 5V/2A isolated
telecom supply. The input voltage is too high to directly
run Q1 or the LT1425, so a bootstrap winding is used to
provide feedback and power for the IC after start-up. The
voltage to the VIN pin is controlled by D1, D2, Q2, Q3 and
associated components, which form the necessary startup circuitry with hysteresis. Nothing happens until C1
charges through R1 to 15V. At that point, Q2 turns on Q3,
pulling the shutdown pin high. Q3, in turn, latches Q2 on,
setting the turn-off voltage to approximately 11V. Switching begins and, before C1 has a chance to discharge to
11V, the bootstrap winding begins to supply power. If the
output is shorted, R2 prevents C1 from being charged by
the transformer’s leakage energy, causing the supply to
continually attempt to restart. This limits input and output
current during a short circuit. Feedback voltage is fed
directly through a resistor divider to the RREF pin. The
sampling error amplifier still works, but the load compensation circuitry is bypassed. This results in a ±5% load
regulation over line and load. A dedicated feedback winding referencing the feedback voltage to the VIN pin could
be used to include the load compensation function and
improve regulation.
330pF
130
MBRS1100T3
T1
5V TO
15V
2
+
+15V
+
6
1µF
22µF
35V
1N759
1
0.1µF
18.4k
0.1%
2
3
4
3.01k
1%
GND
N/C
SD
RFB
ROCMP
16
11
7
12
4
14
OUT
COM
MBR0540LT1
+
220pF
VC
1
9
15µF
35V
3k
–15V
7.32k
1%
0.1µF
T1: COILTRONICS CTX02-13498
INPUT
COM
1425_06.eps
Figure 126. Fully Isolated ±15V, ±60mA Supply
AN84-66
3k
MBRS1100T3
15
13
RCMPC
12
RREF LT1425 VIN
6
11
SYNC
VSW
7
10
PGND
SGND
9
8
GND
GND
5
1000pF
GND
75
15µF
35V
www.BDTIC.com/Linear
(516) 241-7876
Application Note 84
330pF
100Ω
4
12V
22µF
35V
+
3
7
8
MBRS340T3
9
1
2.4k
1
0.1µF
D1
1N755
7.5V
25.5k
1%
2
3
4
5
6
1000pF
7
3.01k
1%
8
GND
GND
N/C
SD
RFB
ROCMP
VC
16
220µF
10V
5
5V
+
220µF
10V
+
200Ω
15
14
2
6
10
11
12
OUT
COM
13
RCMPC
12
RREF LT1425 VIN
11
SYNC
VSW
10
PGND
SGND
9
GND
GND
10
MMFT1N10E
0.1µF
INPUT
COM
9.3k
1%
COILTRONIX
VP1-0190
TURNS RATIO 1 : 1 : 1 : 1 : 1 : 1
12µH PER WINDING
(561) 241-7876
MUR120
1425_07.eps
1.8k
Q1
2N3906
1000pF
Figure 127. 5V/1A Step-Down, Isolated Supply
470pF
18
MBR745
T1
INPUT
COM
2
BAV21
R1
24k
3.3µF
+
BAV21
1
R2
18
30.1k
1%
C1
27µF
35V
3
4
5
6
3.16k
1%
D1
7.5V
1N755
150µF
6.3V
6
2
0.1µF
5V
9, 10
7
8
1000pF
T1
GND
GND
N/C
SD
RFB
ROCMP
510
4
16
15
1
11, 12
+
150µF
6.3V
+
50Ω
1W
OUT
COM
150pF
14
13
RCMPC
12
RREF LT1425 VIN
11
SYNC
VSW
10
PGND
SGND
9
GND
GND
VC
10
Q1
IRF610
T1: COILTRONICS CTX 02-14143-X3
(561) 241-7876
MUR120
–36V TO
–72V
2.4k
Q2
2N3906
D2
7.5V
1N755
Q3
2N3904
5k
100k
10k
0.1µF
1425_08.eps
Figure 128. 5V/2A Telecommunications Supply
www.BDTIC.com/Linear
AN84-67
Application Note 84
30
HIGH ISOLATION CONVERTER USES
OFF-THE-SHELF MAGNETICS
by Mitchell Lee
OUTPUT VOLTAGE (V)
28
Isolated flyback converters usually evoke thoughts (or
bitter memories) of custom transformers, slipped delivery
schedules and agency approval problems. Off-the-shelf
flyback transformers are available from several vendors,
but these carry isolation ratings of only 300V–500V, and,
rarely, of up to 1kV. Flyback transformers with isolation
ratings of 3750VRMS are impossible to find, and if an
application requires this level of isolation, an expensive,
custom design is likely the only solution.
Gate-drive transformers, designed to couple switching
regulator controllers to MOSFET gates, are readily available from stock with high isolation ratings and low cost.
These are wound on ungapped cores and have very high
inductance (500µH to 2mH), and will quickly saturate in a
normal flyback converter circuit.
The transformer used in Figure 129’s circuit handles
significant current without saturating. The converter
operates from a 12V battery-backed input supply and
outputs 24V at 200mA. The key feature is that the second
coil is not a coil at all, but rather an off-the-shelf gate drive
20
D2
MUR120
C3
100µF
50V
50
150
100
TOTAL LOAD CURRENT (mA)
transformer. This component offers 3750VRMS isolation
and full VDE approval at a lower cost than a comparable
custom design.
Feedback is derived from the primary winding, through
D3. R1 acts to filter the leakage-inductance spike at switch
turn-off, and C4 smooths the recovered feedback voltage.
Note that the transformer is wound 1:1; C4 peak detects a
voltage roughly equal to the output. Sizing R1 and C4 is a
trade-off between minimum load and load regulation. As
shown, a minimum load of 3600Ω is recommended.
Output regulation is shown in Figure 130. Line regulation
from 10V to 20V input at full load is 0.13%/V.
D4
MUR120
+
PE63387
C5
100µF
50V
VOUT
24V/200mA
R4
3.6k
(6.7mA MIN LOAD)
D3
1N914
R1
200Ω
* L1 = PE53829
** T1 = PE63387
VIN
R2
18k
SW
VC
C2
100nF
PULSE ENGINEERING
(619) 674-8100
C4
10nF
LT1172 FB
GND
R3
1k
Figure 129. 24V/200mA Bulk Supply with 3750VRMS Isolation
AN84-68
200
DI_1068_04. EPS
L1*
100µH
+
+
0
Figure 130. Output Regulation for Figure 129’s Circuit
T1**
C1
100µF
50V
24
22
D1
P6KE36A
VIN
10V TO 15V
26
www.BDTIC.com/Linear
Application Note 84
WIDE-INPUT-RANGE, LOW VOLTAGE FLYBACK REGULATOR
by Kurk Mathews
Many new switching regulators are designed with a specific application or topology in mind. If your requirements
happen to fall within these parameters, all is well. Unfortunately, when faced with unusual requirements, the
designer is often forced to choose bare-bones, universal
regulators. The LTC1624 overcomes these issues by
providing a full featured regulator that can operate in the
step-down (buck), step-up (boost), buck-boost or flyback
mode.
This constant-frequency current mode controller includes
a high-side differential current sense amplifier and a
floating high current N-Channel MOSFET driver. In the
buck mode, an external bootstrap capacitor between the
BOOST and SW pins works in conjunction with the internal
5.6V regulator and diode to provide a regulated supply for
a high-side driver. In the boost, buck-boost or flyback
mode, the SW pin is grounded, providing drive for a lowside MOSFET.
An example of a wide-input-range flyback is shown in
Figure 131. The circuit provides ±50V at 75mA from a 4.75
to 24V source. The sum of line-, load- and cross-regulation is better than ±5%. The TG pin voltage is controlled by
the internal 5.6V regulator, allowing the input voltage to be
above Q1’s 16V maximum gate-to-source voltage rating.
200kHz fixed frequency operation minimizes the size of
T1. The R-C snubber formed by C1 and R1 in combination
with T1’s low leakage inductance keeps Q1’s drain voltage
well below its 100V rating. To improve cross-regulation,
Q2, R2 and R3 were included to disable Burst Mode™
operation (a feature that improves efficiency at light load
conditions by skipping switching cycles). The LTC1624’s
95% maximum duty cycle accommodates the 5-to-1 input
voltage range. Finally, by reconfiguring T1’s secondaries,
a variety of output configurations, such as 24V out (four
windings in parallel), single 50V/150mA or a single 100V
output, are possible with this same basic circuit.
220pF
47Ω
T1 12
50V/75mA
MURS120T3
18k
1
VIN
4.75V–24V
330µF
35V
SANYO
MV-GX
0.02Ω
+
R3
220Ω
1µF
8
1
SENSE–
VIN
BOOST
LTC1624
TG
SW
ITH/RUN GND
2
4
7
6
5
1µF
C1
220pF
Q2
MPS2222A
15k
11
10
4
2
8
9
5
10Ω
1µF
MURS120T3
18k
7
Q1
IRL540N
VFB
3
1k
0.01µF
1µF
R1
47Ω
3
1µF
6
–50V/75mA
620k
R2
43k
T1: COILTRONICS VP3-0138, 1:1:1:1:1:1
(SIX WINDINGS, EACH 11.2µH)
(561) 241-7876
Figure 131. Wide-Input-Range Flyback Regulator Provides ±50V at 75mA
www.BDTIC.com/Linear
AN84-69
Application Note 84
Regultors—Switching (Low Noise)
THE LT1533 HERALDS A NEW CLASS OF
LOW NOISE SWITCHING REGULATORS
by Jeff Witt
20mV/DIV
200µV/DIV
Introducing the LT1533 Low Noise Switcher
5µs/DIV
The LT1533 is a switching regulator that provides a
solution to EMI problems through two flexible approaches.
First, the slew rates of both the current through the power
switch and the voltage on it are easily programmed with
external resistors. Limiting these slew rates will remove
the highest harmonics from the switching waveforms.
Second, the LT1533, with two 1A power switches, is
designed to operate in push-pull circuits. Such circuits,
with their low input and output current ripple, are inherently quiet. The result is an integrated switching regulator
that provides very quiet output power and very low emissions. Figure 132 illustrates what can be achieved. The top
trace shows the output of a push-pull boost regulator
generating 120mA at 12V from an input of 5V. This trace
was measured using a 10MΩ oscilloscope probe with a
six-inch ground lead, demonstrating that there is no
significant inductively or capacitively coupled noise. Probing the output of the LT1533 circuit with a 50Ω low noise
1533_01.eps
Figure 132. Output Ripple of an LT1533 Switching Regulator
Producing 120mA at 12V from a 5V Input
amplifier reveals the real performance (second trace):
peak-to-peak output ripple of the low noise switcher is
only 150µV in a 10kHz to 100MHz bandwidth.
A Closer Look at the LT1533
The LT1533 is a fixed frequency current mode PWM
switching regulator. The output voltage is regulated by
controlling the peak switch current on each cycle of the
oscillator, resulting in good transient performance and
rapid current limiting. The oscillator drives a toggle flipflop, alternately enabling one of two 0.5Ω NPN power
switches, QA and QB. The switch current is monitored by
a sense resistor at the emitter of the switch. The output
5V
+
47µF
6.3V
14
1
VIN
NC
COL A
25nH*
PGND
8
11
4
1
4
1
4
NFB
DUTY
COL B
SHDN
RCSL
RVSL
SYNC
GND
9
RT
CT
6
18k
5
VC
FB
L2
10µH
L1
300µH
1.2k
16
LT1533
3
D1
T1
2
+
C1
22µF
20V
220pF
15
12
4k–68k
13
4k–68k
D2
7
21.5k
10
2.49k
15k
1000pF
1500pF 0.015µF
*BEAD OR PCB TRACE
T1 = COILTRONICS CTX02 13666-X1
(561) 241-7876
L1 = COILTRONICS CTX300-2
L2 = COILTRONICS DT1608C-103
D1, D2 = MOTOROLA MBRS1100T3
(800) 441-2447
1533_03.EPS
Figure 133. 5V to 12V Push-Pull PWM Converter
AN84-70
www.BDTIC.com/Linear
+
12V/200mA
C2
22µF
20V
Application Note 84
TRACE A
0.5A/DIV
TRACE A
0.5A/DIV
TRACE B
20mV/DIV
TRACE C
500µV/DIV
TRACE B
20mV/DIV
TRACE C
500µV/DIV
0.2µs/DIV
1533_04.eps
0.2µs/DIV
Figure 134. Lowering the Slew Rates of the Power Switches (Trace A) Eliminates High Frequency Ripple at the Output (Traces B and C)
voltage (either positive or negative) is compared with an
accurate internal 1.25V reference voltage by an error
amplifier whose current output, along with loop compensation components tied to the VC pin, determine the peak
switch current required for regulation; a comparator turns
off the switch when this current level is reached.
directly trade off quiet, low EMI operation with high
efficiency: low slew rates result in slowly changing stray
fields, which generate less interference, but increase the
conduction losses in the switches.
The LT1533 oscillator presents additional opportunities
for managing EMI. Its wide frequency range (20kHz to
250kHz) allows the designer to avoid sensitive frequencies. Operating frequency is set with a capacitor on the CT
pin and a resistor of nominally 17k on the RT pin. The
LT1533 can also be synchronized to an external clock,
allowing accurate placement of both switching frequency
and phase.
The slew-control circuitry monitors the collector voltages
and emitter currents of the power switches and adjusts
base drive to control both the voltage and current slew
rates. The desired rates are programmed by tying the RVSL
and RCSL pins to ground with resistors between 4k and
68k, corresponding to slew rates from ~80V/µs to 5V/µs
and 7A/µs to 0.4 A/µs. This allows the circuit designer to
5V
+
14
1
8
VIN
NC
COL A
PGND
+
16
LT1533
3
11
4
1
3.3
COL B
RCSL 12
13
R
SHDN
SYNC
VSL
GND
RT
CT
9
6
5
18k
3300pF
VC
FB
7
LT1121-CS8
1
3.3
+
L2
100µH
68k
4k–68k
47k
3
22µF
35V
1
332k
2
2 × BAT85
15
DUTY
8
T1
2
25nH*
NFB
L1
100µH
22µF
10V
+
150k
150k
22µF
35V
5 4
1, 2, 7, 8
12V
80mA
LT1175-CS8
332k
3
2.2µF
25V
+
2.2µF
25V
–12V
80mA
4 × 1N5819
10
10k
*BEAD OR PCB TRACE
T1 = COILTRONICS CTX02 13716-X1
(561) 241-7876
L1, L2 = COILCRAFT DT1608C-104
1533_05.EPS
(847) 639-6400
Figure 135. 5V to ±12V DC/DC Transformer
www.BDTIC.com/Linear
AN84-71
Application Note 84
VIN
3.3V
L1
100µH
+
C2
33µF
10V
1
3
8
11
4
D1
VOUT
5V/350mA
C1 +
100µF
10V
14
VIN
NC
COL A
COL B
DUTY
NFB
PGND
LT1533
SHDN
RCSL
SYNC
RVSL
FB
GND RT
9
CT
6
2
10Ω
15
*50nH
16
12
4k–68k
13
4k–68k
7.50k
7
VC
5
18k
10
2.49k
10k
1000pF
2200pF
0.01µF
*BEAD OR PCB TRACE
L1 = COILTRONICS CTX100-4
D1 = MOTOROLA MBRS120T3
C1 = AVX TPSD107M010R0100
C2 = AVX TPSC336M010R0375
10MΩ scope probe with a six-inch ground lead. The lower
trace is the output measured with a low noise amplifier. In
the left photo the switch slew rates are programmed to
their highest values with 3.9k resistors on the RCSL and
RVSL pins. The fast switch transients induce high frequency ripple on the output (the higher level of noise on
the middle trace is due to the inductance of the scope
probe’s ground lead). By lowering the slew rates (RCSL =
24k and RVSL = 8.2k) this potentially troublesome output
ripple is eliminated, as shown in the right photo. The
efficiency penalty is minor; the slower slew rates reduce
efficiency from 73% to 70%.
1533_06.EPS
(561) 241-7876
(800) 441-2447
(207) 222-5111
Figure 136. 3.3V to 5V Boost Converter
This combination of appropriate circuit topology and
controlled slew rates produces the exceptionally clean
output shown in Figure 132. This circuit is simply implemented with ordinary PCB construction, and can be placed
in close proximity to sensitive circuits without the need for
expensive electrostatic or magnetic shielding.
Push-Pull PWM Makes a Quiet Boost Converter
DC Transformer with Civilized Edges
The push-pull converter in Figure 133 produces 200mA at
12V from an input of 5V. The oscillator is set to 80kHz
(note that the circuit operates at half this frequency) and
the LT1533 applies a pulse-width modulated 5V to the
primary side of the transformer. The rectified secondary
voltage is filtered by L1 to generate 12V on C1. In this
circuit, L1 is the primary energy storage device, so the
transformer can be made fairly small. Additional output
filtering is provided by L2 and C2.
Grounding the Duty pin of the LT1533 disables the feedback loop and runs each switch at 50% duty cycle,
allowing the LT1533’s use in DC transformer circuits.
Such circuits are useful for generating bipolar or isolated
supplies; Figure 135 shows an example. The LT1533
switches 5V across a 3.3:1 transformer and a diode bridge
rectifies the secondary side voltages to produce nominally
16V bipolar outputs that are regulated to ±12V. Shortcircuit current limit at the output is provided by the
LT1533’s switch current limit; the 1A switch limit is
transformed to 0.3A on the secondary.
This topology is inherently quiet. Current through L1 into
the primary output capacitor C1 is a continuous triangle
wave with little high frequency content, resulting in low
conducted output noise. With an appropriate transformer
turns ratio, RMS input current is kept low, reducing the
potential for conducted noise on the input.
It is advantageous to start with a good topology, but high
frequency noise will still get around via stray capacitance
and mutual inductance; the best way to deal with this is to
eliminate fast edges. Figure 134 shows several waveforms
from the circuit as it delivers 120mA of output current. The
upper trace in each photo is the current in switch QA as it
turns off. Trace B is the output voltage probed with a
AN84-72
A common problem with isolated-output switchers is that
fast edges couple through stray capacitance between the
primary and secondary windings of the transformer to
create common mode noise on the outputs. Also, linear
regulators are incapable of rejecting high frequency noise
at their inputs. Both problems are greatly reduced by
limiting the switch slew rates. Shielding between the
windings can be eliminated, reducing transformer size
and cost. LC filters on the isolated side are unnecessary
with the linear regulators rejecting ripple at the operating
frequency and the controlled slew rates eliminating high
frequency ripple.
www.BDTIC.com/Linear
Application Note 84
TRACE A
5V/DIV
TRACE A
5V/DIV
TRACE B
0.5A/DIV
TRACE B
0.5A/DIV
TRACE C
20mV/DIV
TRACE C
20mV/DIV
5µs/DIV
5µs/DIV
1533_07.eps
Figure 137. Limiting Switch Slew Rates (Traces A and B) Lowers the High
Frequency Content of the Boost Regulator’s Output Ripple (Trace C)
3.3V to 5V Boost Converter
Simple switching topologies can also benefit from the
LT1533’s low noise features. In a boost regulator, for
example, the current into the output capacitor is a square
wave, which contains the high frequency harmonics generated by a fast power switch. Even when the rectifying
diode is off, fast voltage waveforms at the switch couple
through the Schottky diode’s capacitance. Fast switching
can also excite high frequency resonant circuits formed by
the diode’s capacitance and parasitic inductance due to
board traces. All of these effects can be reduced by
controlling the slew rate of the switch. Figure 136 shows
the LT1533 in a simple boost circuit generating 5.0V from
a 3.3V input, a typical requirement when interfacing 3.3V
logic systems to 5V high performance ADCs. The collectors of the two power switches are tied together and
alternately energize the boost inductor. Figure 137 shows
several waveforms at two different slew rate settings with
the circuit delivering 200mA of output current. Trace A is
the switch voltage, trace B is the current through the
output capacitor and trace C is the AC-coupled output
voltage in a 100MHz bandwidth. In the left photo, the slew
rates are set to their maximum values (RCSL = RVSL =
3.9kΩ). The rapidly switched current combined with the
finite series inductance of the output capacitor result in
large voltage spikes on the output. The right photo shows
the same waveforms with the slew rates lowered (RCSL =
RVSL = 22k), eliminating the troublesome transients. The
penalty is a drop in efficiency from 85%␣ to␣ 80%.
Conclusion: a Switcher for Sensitive Systems
With two 1A power switches, the ability to control positive
or negative outputs, and a wide input operating range (2.7
to 30V), the LT1533 is a highly flexible switching regulator. Thermal shutdown, in addition to switch-current limit,
provides circuit protection. The LT1533 is packaged in the
narrow 16-lead SO, and is available in commercial and
industrial grades.
The LT1533 allows the circuit designer to add a switching
regulator to sensitive analog systems without fear of
introducing uncontrollable noise and interference. The
programmable operating frequency and switch slew rates
allow final tuning to occur in the circuit, when the system
is running and interference problems may first become
apparent. In addition to providing a way to deal with
unforeseen problems, this flexibility means that sacrifices
in efficiency will be limited to those needed for proper
system performance. The LT1533 is the switching regulator of choice for high performance analog systems.
www.BDTIC.com/Linear
AN84-73
Application Note 84
High Voltage Input Regulator
LT1533 ULTRALOW NOISE SWITCHING REGULATOR
FOR HIGH VOLTAGE OR
HIGH CURRENT APPLICATIONS
by Jim Williams
The LT1533’s IC process limits collector breakdown to
30V. A complicating factor is that the transformer causes
the collectors to swing to twice the supply voltage. Thus,
15V represents the maximum allowable input supply.
Many applications require higher voltage inputs; the circuit in Figure 138 uses a cascoded3 output stage to
achieve such high voltage capability. This 24V to 5V (VIN
= 20V–50V) converter is reminiscent of previous LT1533
circuits, except for the presence of Q1 and Q2.4 These
devices, interposed between the IC and the transformer,
constitute a cascoded high voltage stage. They provide
voltage gain while isolating the IC from their large drain
voltage swings.
The LT1533 switching regulator1, 2 achieves 100µV output noise by using closed-loop control around its output
switches to tightly control switching transition time. Slowing down switch transitions eliminates high frequency
harmonics, greatly reducing conducted and radiated noise.
The part’s 30V, 1A output transistors limit available power.
It is possible to exceed these limits while maintaining low
noise performance by using suitably designed output
stages.
6
T1
7
5
8
24VIN
(20V TO 50V)
+
4
10µF
9
3
MBRS140
10
1
0.002µF
220Ω
10k
Q3
MPSA42
Q4
2N2222
10k
Q1
(
L3
OPTIONAL
100µH SEE TEXT
12
1k
2
+
4.7µF
2
14
4
3
1500pF
11
5
18k
L1, L3: COILTRONICS CTX100-3
(561) 241-7876
L2: 22nH TRACE INDUCTANCE, FERRITE BEAD
OR INDUCTOR
COILCRAFT B-07T TYPICAL
(847) 639-6400
Q1, Q2: ON SEMI MTD6N15
(800) 282-9855
T1: COILTRONICS VP4-0860
6
15
COL A
VIN
0.01µF
MBRS140
COL B
SYNC
DUTY
SHDN
CT
LT1533
L2
PGND
RT
NFB
10
11
VC
RVSL
FB
GND
RCSL
9
12
13
12k
10k
16
8
7
7.5k
1%
2.49k
1%
AN70 F40
Figure 138. A Low Noise 24V to 5V Converter (VIN = 20V–50V): Cascoded MOSFETs Withstand 100V Transformer Swings,
Permitting the LT1533 to Control 5V/2A
AN84-74
www.BDTIC.com/Linear
)
+
220µF
0.002µF
Q2
1k
5VOUT
+
10k
220Ω
L1
100µH
100µF
Application Note 84
A = 20V/DIV
B = 5V/DIV
(AC COUPLED)
A = 5mV/DIV
C = 100V/DIV
B = 100µV/
DIV
10µs/DIV
2µs/DIV
Figure 139. MOSFET-Based Cascode Permits the Regulator to
Control 100V Transformer Swings while Maintaining a Low
Noise 5V output. Trace A is Q1’s Source, Trace B is Q1’s Gate
and Trace C is the Drain. Waveform Fidelity through Cascode
Permits Proper Slew-Control Operation
Figure 141. Waveforms for Figure 139 at 10W Output: Trace A
Shows Fundamental Ripple with Higher Frequency Residue
Just Discernable. The Optional LC Section Results in Trace B’s
180µVP-P Wideband Noise Performance
Current Boosting
Figure 140 boosts the regulator’s 1A output capability to
over 5A. It does this with simple emitter followers (Q1–
Q2). Theoretically, the followers preserve T1’s voltage and
current waveform information, permitting the LT1533’s
slew-control circuitry to function. In practice, the transistors must be relatively low beta types. At 3A collector
current, their beta of 20 sources ≈150mA via the Q1–Q2
base paths, adequate for proper slew-loop operation.5 The
follower loss limits efficiency to about 68%. Higher input
voltages minimize follower-induced loss, permitting efficiencies in the low 70% range.
Normally, high voltage cascodes are designed simply for
supply isolation. Cascoding the LT1533 presents special
considerations because the transformer’s instantaneous
voltage and current information must be accurately transmitted, albeit at lower amplitude, to the LT1533. If this is not
done, the regulator’s slew-control loops will not function,
causing a dramatic output noise increase. The AC-compensated resistor dividers associated with the Q1–Q2 gatedrain biasing serve this purpose, preventing transformer
swings coupled via gate-channel capacitance from corrupting
the cascode’s waveform-transfer fidelity. Q3 and associated
components provide a stable DC termination for the dividers
while protecting the LT1533 from the high voltage input.
Figure 141 shows noise performance. Ripple measures
4mV (Trace A) using a single LC section, with high
frequency content just discernable. Adding the optional
second LC section reduces ripple to below 100µV (trace
B), and high frequency content is seen to be inside 180µV
(note ×50 vertical scale-factor change).
Figure 139 shows that the resultant cascode response is
faithful, even with 100V swings. Trace A is Q1’s source;
traces B and C are its gate and drain, respectively. Under
these conditions, at 2A output, noise is inside 400µV peak.
1N4148
330Ω
5V
1N5817
0.05Ω
T1
Q1
+
4.7µF
14
11
3
1500pF
4
5
18k
6
0.003µF
VIN
SHDN
COL A
DUTY
COL B
SYNC
CT
PGND
LT1533
RVSL
RT
RCSL
10
0.01µF
VC
GND
NFB
9
8
FB
+
2
4.7µF
15
Q2
0.05Ω
330Ω
16
L2
7
R2
2.49k
1%
L3
33µH
12V
+
(
OPTIONAL FOR
LOWEST RIPPLE
)
+
100µF
100µF
1N5817
1N4148
13 10k
12 10k
680Ω
L1
300µH
R1
21.5k
1%
AN70 F42
L1: COILTRONICS CTX300-4
(561) 241-8786
L2: 22nH TRACE INDUCTANCE, FERRITE BEAD OR
INDUCTOR. COILCRAFT B-07T TYPICAL
(847) 639-6400
L3: COILTRONICS CTX33-4
Q1, Q2: MOTOROLA D45C1
(800) 441-2447
T1: COILTRONICS CTX-02-13949-X1
: FERRONICS FERRITE BEAD 21-110J
Figure 140. A 10W, Low Noise, 5V to 12V Converter: Q1–Q2 Provide 5A Output Capacity while Preserving the LT1533’s Voltage/Current
Slew Control. Efficiency is 68%. Higher Input Voltages Minimize Follower Loss, Boosting Efficiency Above 71%
www.BDTIC.com/Linear
AN84-75
Application Note 84
Notes:
1 Witt, Jeff. The LT1533 Heralds a New Class of Low Noise Switching
Regulators. Linear Technology VII:3 (August 1997).
improvement or the like. Cascoding has been employed in op amps,
power supplies, oscilloscopes and other areas to obtain performance
enhancement.
2 Williams, Jim. LTC Application Note 70: A Monolithic Switching Regulator
with 100µV Output Noise. October 1997.
4 This circuit derives from a design by Jeff Witt of Linear Technology Corp.
3 The term “cascode,” derived from “cascade to cathode,” is applied to a
configuration that places active devices in series. The benefit may be
higher breakdown voltage, decreased input capacitance, bandwidth
5 Operating the slew loops from follower base current was suggested by
Bob Dobkin of Linear Technology Corp.
Regulators—Switching (Multioutput)
10Ω
+
0.1µF
100Ω
100Ω
1
2
CSS1
3
1000pF
1000pF
4
220pF
10kΩ
5
1000pF
6
56pF
7
8
470pF
10kΩ
9
1000pF
100pF
220k
10
11
390k
3pF
12
220pF
1000pF
0.1µF
13
10Ω
10Ω
24V
T1
15µH
14
BOOST1
TGL1
RUN/SS1
SW1
SENSE+1
VIN
SENSE–1
BG1
VPRGM1
ITH1
INT VCC
PGND
28
M1
SGND
EXT VCC
SFB1
SW2
ITH2
TGL2
VOSENS2
–
BOOST2
SENSE 2
AUXON
SENSE+2
AUXFB
RUN/SS2
AUXDR
3.3µF
35V
0.033Ω
VOUT1
5V/3A
27
26
+
CMDSH-3
25
24
MBRS140T3
M2
GND
23
22
M4
21
0.033Ω
19
17
MBRS140T3
100µF
10V
×2
VOUT2
3.3V/3.5A
L1
10µH
20
18
100µF
10V
×2
0.1µF 4.7µF, 16V
+
COSC
BG2
LTC1538CG-AUX
+
+
KEYBOARD
CONTROLLER
SIGNAL
0.1µF
VIN
5.2-28V
MBRS1100T3
22µF
35V
×2
+
22µF
35V
×2
M3
0.1µF
CMDSH-3
5V STANDBY
16
AUX ON/OFF
15
CSS2
47k
Q1
2N2907A
KEYBOARD
CONTROLLER
SIGNAL
AUX 12V OUT
1MΩ
+
VIN 5.2-28V; SWITCHING FREQUENCY = 200kHz
5V-3A / 3.3V-3.5A / 12V-120mA
M1-M4 = Si4412DY
T1 = DALE LPE-6562-A092; 15µH; 1:2.2
100k
L1 = SUMIDA CDRH125-100MC 10µH
INPUT AND OUTPUT CAPACITORS ARE AVX-TPS SERIES
HAVING A MAXIMUM ESR SPECIFICATION
Figure 142. LTC1538-AUX Provides 3.3V/3.5A, 5V/3A, 12V/120mA and 5V/20mA Standby Power
AN84-76
www.BDTIC.com/Linear
4.7µF
25V
Application Note 84
LTC1538-AUX: A NEW ADDITION TO LTC’S
ADAPTIVE POWER CONTROLLER FAMILY
by Steve Hobrecht
Notebook Computer Power Solution
The circuit shown in Figure 142 is a power solution for a
portable notebook computer. The switching controllers
provide 5V at 3A, 3.3V at 3.5A and a regulated 12V/120mA
output using the auxiliary regulator. See the LTC1538AUX/ LTC1539 data sheet for techniques illustrating how
to generate other voltage and current combinations using
the auxiliary regulator. The circuit provides a standby 5V
output to power a keyboard controller. The keyboard
controller has the ability to control the run/soft-start
HIGH EFFICIENCY, LOW POWER,
3-OUTPUT DC/DC CONVERTER
by John Seago
The recent proliferation of battery powered products has
created a lot of interest in low power, high efficiency DC/
DC converter designs. These products are small, lightweight and portable, so space for bulky batteries is limited.
Often, operating time between charges is a major selling
feature, making the efficient use of battery power very
important. Since many products cannot function with a
single regulated voltage, multiple-output DC/DC converters are required.
6V TO 20V
C1
68pF
+
C7
22µF
35V
U1
LTC1435CS
1
C2, 0.1µF
R1, 10k
C4, 47pF
2
3
C3
330pF
4
5
COSC
BOOST
ITH
SW
SFB
VIN
SGND
INT VCC
6
C6
0.001µF
T1
Q1
1/2 Si9936
16
15
D1, MBR0530
30µH
5V/0.1A
D3
MBR0540
+
C12
100µF
10V
14
3.3V/0.5A
C8
0.1µF
13
D2
MBRS130L
12
11
VOSENSE
BG
7 SENSE–
PGND 10
8 SENSE+ EXT V
9
CC
C5
100pF
Although developed for somewhat higher power levels,
the single output LTC1435 can be used in applications
requiring a very efficient, very small, low power, multipleoutput DC/DC converter (see Figure 143). This is
accomplished through the use of an overwound buck
inductor. With additional windings, the inductor can provide additional outputs, requiring only a diode and filter
capacitor for each output. As with the less efficient flyback
topology, the additional outputs are not as well regulated
as the primary output, but the regulation is suitable for
most applications.
R4
10Ω
TG
RUN/SS
(RUN/SS1 and RUN/SS2) pins of the LTC1538-AUX using
simple logic gates. The turn-on sequence is determined by
the ratio of Css1 to Css2. The secondary winding of
transformer T1 develops a somewhat unregulated voltage
due to the loading on VOUT1. The SFB1 control pin will keep
the minimum voltage of the secondary output at approximately 13V, but the peak voltage is affected by the loading
and leakage inductance of the transformer. The auxiliary
regulator will keep the 12V supply well within its normal
±5% specified tolerance. Short-circuit protection can be
added to this circuit if required, but it is assumed here that
the protection will only be required at the user PCMCIA
interface and will therefore be taken care of as part of the
interface and not duplicated here.
C9, 0.1µF
C10, 4.7µF
+
R5
0.1Ω
C11
220pF
R6
35.7k
+
R7
20k
Q2
1/2 Si9936
C13
100µF
10V
GND
T1
R3, 100Ω
+
R2, 100Ω
C14
100µF
10V
–5V/0.05A
DI1435_01.eps
R5 = IRC, LR2010-01-R100-J
C7 = AVX, TPSE226M035R0300
C12, 13, 14 = AVX, TPSD107M010R0100
T1 = COILTRINICS, CTX02-13299
Q1/Q2 = SILICONIX, Si9936DY
D4
MBR0540
Figure 143. High Efficiency, 3-Output DC/DC Converter
www.BDTIC.com/Linear
AN84-77
Application Note 84
The circuit of Figure 143 provides 3.3V at 0.5A, 5V at 0.1A
and –5V at 0.05A, and has greater than 93% efficiency for
test loads between 1.25W and 2.4W with a 6V input. Load
and line regulation of the positive outputs are quite good.
Each output voltage was measured with all output currents
varied independently between 20% and 100% of their full
load range, while the input voltage was varied from 6V to
20V. Table 3 shows the worst-case output voltages
measured.
Table 3. Worst-Case Output Voltages
Output
Minimum
Maximum
3.3V
3.307V
3.315V
5V
5.03V
5.24V
–5V
–4.98V
–5.51V
The buck regulator with an overwound inductor is a good
solution for those applications that do not have large load
current or line voltage variations. The smaller the load and
line variations, the smaller the voltage variations on the
overwound outputs. As a general rule, output voltage
regulation is suitable for most applications if the switch
duty cycle is kept between 15% and 50% and minimum
load current is kept above 20% of maximum. Since load
variation and line variation have an additive effect on
output voltage, applications with relatively constant load
current requirements can have a larger input voltage range
and vice versa. For zero output current requirements, a
small preload resistor can be used.
DUAL-OUTPUT VOLTAGE REGULATOR
by Peter Guan
The LTC1266-3.3 and LTC1263, as shown in the schematic of Figure 144, are perfect complements for one
another. The combination of these two parts provides two
regulated outputs of 3.3V/5A and 12V/60mA from an input
range of 4.75V to 5.5V. These two outputs are perfect for
notebook and palmtop computers with microprocessors
that burn several amps of current from a regulated 3.3V
supply, flash memories that consume milliamps of current
from a regulated 12V supply and interface and logic
components that still run off the 5V supply. In fact, this
quick and easy combination may well be the aspirin for
many of the headaches caused by the rigorous power
supply demands in today’s electronics.
The LTC1263, using only four external components (two
0.47µF charge capacitors, one 10µF bypass capacitor and
a 10µF output capacitor), generates the regulated 12V/
60mA output from a 5V input using a charge pump tripler.
During every period of the 300kHz oscillator, the two
charge capacitors are first charged to VCC and then stacked
in series, with the bottom plate of the bottom capacitor
shorted to VCC and the top plate of the top capacitor
connected to the output capacitor. As a result, the output
capacitor is slowly charged up from 5V to 12V. The 12V
output is regulated by a gated oscillator scheme that turns
AN84-78
the charge pump on when VOUT is below 12V and turns it
off when it exceeds 12V.
The LTC1266-3.3 then uses the 5V input along with the
12V output from the LTC1263 and various external components, including bypass capacitors, sense resistors and
Schottky diodes, to switch two external N-channel MOSFETs and a 5µH inductor to charge and regulate the 3.3V/
5A output. The charging scheme for this part, however, is
very different from that of the LTC1263. The LTC1266-3.3
first charges the output capacitor by turning on the top Nchannel MOSFET, allowing current to flow from the 5V
input supply and through the inductor. By monitoring the
amount of current flow in the inductor with a sense
resistor, the 3.3V output is regulated by turning on and off
the top and bottom N-channel MOSFETs to charge and
discharge the output capacitor.
If we replaced the top external N-channel MOSFET with a
P-channel, the LTC1266-3.3 could generate the same
3.3V/5A output without the help of the LTC1263. But, since
N-channel MOSFETs have lower gate capacitance and
lower RDS(ON), their higher efficiency at high currents
more than compensates for the extra complexity in bringing in another higher input voltage, especially if that
second input voltage is readily available.
www.BDTIC.com/Linear
Application Note 84
VCC
5V
FROM µP
1
C1–
SHDN
2
C1+
GND
3
C2–
VOUT
6
4
C2+
VCC
5
8
C1 = 0.47µF
7
LTC1263
C3 = 10µF
VOUT = 12V/60mA
C2 = 0.47µF
C4 = 10µF
VCC
Si9410DY
1
1µF
2
3
4
5
6
CT
180pF
7
CC
3300pF
RC
470Ω
8
BDRIVE
TDRIVE
PWR VIN
PGND
PINV
LBOUT
BINH LTC1266-3.3
LBIN
VIN
SGND
CT
SHDN
ITH
NC
SENSE–
SENSE+
16
+
D1
MBRS140T3
Si9410DY
15
CIN
100µF
20V
×2
14
13
12
11
10
9
1000pF
L*
5µH
+
RSENSE
0.02Ω
COUT
220µF
10V
×2
VOUT = 3.3V/5A
*COILTRONICS CTX0212801
DI1263_01.eps
Figure 144. 5V to 3.3V/5A and 12V/60mA Supply
Since both of these devices are very stingy on quiescent
current, their combination is also very gentle to the main
power supply, especially if that power supply is a battery.
In standby mode, the LTC1263 and the LTC1266-3.3 have
a total quiescent current of about 500µA. To conserve even
more current, both of these parts can be put into shutdown
mode by floating their shutdown pins or pulling them high.
The total shutdown current is less than 40µA. When
loaded, the LTC1263 has a 76% efficiency, whereas the
LTC1266-3.3 can squeeze out more than 90%. Together,
with a 60mA load at the 12V output and a 5A load at the
3.3V output, the overall efficiency is 87%.
The LTC1266-3.3 is available in the 16-pin SO package and
the LTC1263 is available in the 8-pin SO package. Together,
these two parts provide an easy and efficient solution for
multiple power supply demands.
www.BDTIC.com/Linear
AN84-79
Application Note 84
SWITCHER GENERATES TWO BIAS VOLTAGES
WITHOUT TRANSFORMER
by Jeff Witt
LCD displays and CCD imaging circuits in today’s portable
products require several bias voltages of 10V to 20V at a
few mA. When symmetric bipolar bias supplies are needed,
the negative supply can be generated with a discrete
charge pump operating from the power switch of the
boost regulator that generates the positive supply. However, an asymmetric bipolar supply is typically required:
for example 20V and –10V for LCD displays or 15V and
–7.5V for CCDs. One possible solution is to add a linear
regulator to the negative output; this adds cost and greatly
reduces the efficiency of the switcher. Another possibility
is a 2-output flyback circuit, but the added cost and bulk
of a transformer make this solution unappealing. The
circuit in Figure 145 avoids these penalties, producing 20V
at 5mA and –10V at 5mA from 3.3V with 73% efficiency.
The circuit uses standard surface mount parts.
The LT1316, a micropower Burst Mode switching regulator with an integrated 0.6A power switch, operates in an
ordinary boost circuit to generate the 20V (VOUT1) set by
resistor divider R1 and R2. An internal comparator at the
L1
47µF
VIN
3.3V
7
+
FB
SHDN
LT1316
C1
33µF
10V
LBO
LBI
RSET
3
R5
10k
8
C3
3.3µF
35V
+
R1
1M
150k
R3
1M
VOUT1 200mV/DIV
(AC COUPLED)
2
R2
64.9k
GND
4
L1 = COILCRAFT DO1608C-473
C1 = AVX TAJB336M035R
C2 = AVX TAJA105M035R
C3, C4 = AVX TAJB335M035R
VOUT1
20V/5mA
150pF
1
82k
SW PIN 20V/DIV
BAT54
+
C2
1µF
35V
VOUT2 1V/DIV
(AC COUPLED)
BAT54
BAT54
Q1
2N7002
+
R4
590k
C4
3.3µF 35V
Figure 145. By Gating the Charge Pump, this Circuit
Generates a Regulated Negative Output with a Magnitude
Different from that of the Positive Output
AN84-80
This circuit can also operate directly from two alkaline or
NiCd cells. Slightly higher peak currents are necessary;
change R5, which determines the peak switch current of
the LT1316, to 6.8kW and change L1 to 15mH.
BAT54
5
SW
6
VIN
FB pin regulates the output by gating the LT1316’s oscillator. A charge pump (C2 and associated diodes) coupled
to the LT1316’s switch pin generates the negative output
voltage. This negative output (VOUT2) is monitored by the
LT1316’s low-battery detector through the resistor divider
R3 and R4, using the positive 20V output as a reference.
When the negative output falls below 10V, the low-battery
detector output (LBO pin and lowest trace in Figure 146)
turns Q1 on, enabling the charge pump and charging
output capacitor C4. Note that the switch pin jumps
between ground and ~10V during this period. Once the
negative output has been charged enough to overcome the
low-battery detector’s hysteresis, Q1 turns off and the
switch pin is free to fly to 20V, charging the positive
output.
LBO PIN 5V/DIV
VOUT2
–10V/5mA
0.1ms/DIV
Figure 146. Voltage Waveforms of
Figure 145’s circuit
www.BDTIC.com/Linear
Application Note 84
NEW IC FEATURES REDUCE EMI
FROM SWITCHING REGULATOR CIRCUITS
by John Seago
tion. Also, transistor Q3 ensures constant frequency at
very low output current levels, thus eliminating audio
frequencies and maintaining high efficiency using the
internal Adaptive Power™ circuitry.
One disadvantage of using a switching regulator is that it
generates electronic noise, known as EMI (electromagnetic interference). This noise can be conducted or radiated, and it can affect other circuits in your product or
interfere with the operation of nearby products. The
LTC1436-PLL, LTC1437, LTC1439 and LTC1539 have
features that can be used to suppress this interference.
Switch-Frequency Synchronization
Switching regulator noise results from switching high
currents on and off. This creates high energy levels at the
switching frequency and all of its harmonics. A common
EMI-control technique is to synchronize the switching
frequency to an external clock so that all harmonic frequencies can be controlled. The LTC1436-PLL uses a
phase-locked loop for synchronization to avoid the loss of
slope compensation common to other synchronizing techniques. In addition, the input to the VCO in the phaselocked loop is available at the PLL LPF (phase-locked loop
lowpass filter) pin, so that a lowpass filter can be used to
control how fast the loop acquires lock.
Frequently, EMI problems don’t show up until the integration phase of product development. By using this EMI
suppression capability, a resistor or capacitor value change
may be all that is required to solve an interference problem. The LTC1436-PLL shown in the circuit of Figure 147
produces a switched 5V, 3A output and a 3.3V, 0.1A linear
output. The circuit is configured to provide either switchfrequency synchronization or switch-frequency modula5.5V TO
24V
MOD
R7
10Ω
PLL
SWITCHFREQUENCY
MODULATOR*
R2, 10k
1
C1
C2, 47pF
PLL
C3, 0.1µF
2
3
R3, 10k
4
C5, 47pF
C6
100pF
C11
22µF
35V
+
C12
22µF
35V
C11, C12: KEMET T495X226M035AS
C13, C14: AVX TPSD107M010R0065
L1: SUMIDA CDRH125-10
Q1 + Q2: SILICONIX Si4936DY (DUAL FET)
Q3: INTERNATIONAL RECTIFIER IRLML2803
R8: IRC LR2010-01-R033-J
R6
47k
MOD
0.01µF
+
C4
330pF
INTVCC
5
6
7
8
R4, 100Ω
C7, 0.001µF
R5
100Ω
PLLIN
COSC
POR
RUN/SS
BOOST
ITH
TGL
SFB
SW
SGND
VPROG
TGS
LTC1436-PLL
VIN
VOSENSE
INTVCC
9
SENSE –
BG
10
SENSE +
11
12
R9
20k
PLL LPF
PGND
AUXON
EXTVCC
AUXFB
AUXDR
* SEE FIGURE 151
24
23
POR
22
21
Q1
20
19
18
17
16
+
R8
0.033Ω
5V
3A
C10
0.1µF
Q3
C8
0.1µF
L1
10µH
+
D1
MBRS0530
C9
4.7µF
D2
MBRS130L
C13, C14
100µF
10V
×2
Q2
15
GND
14
13
R10
35.7k
R11
47k
+
C15
3.3µF
Q4
MMBT2907ALT1
DI1436_01.eps
3.3V
0.1A
Figure 147. 2-Output LTC1436-PLL Test Circuit
www.BDTIC.com/Linear
AN84-81
Application Note 84
–20dBm
VIN = 10V
VO = 5V AT 3A
BW = 100Hz
–40dBm
–60dBm
–80dBm
–100dBm
–120dBm
1kHz
500kHz
1MHz
DI1436_02.eps
Figure 148. Output Noise Before and After Switch-Frequency Modulation
–55dBm
–65dBm
–75dBm
–85dBm
VIN = 10V
VO = 5V AT 3A
BW = 300Hz
–95dBm
–105dBm
1MHz
15MHz
30MHz
DI1436_03.eps
Figure 149. Output High Frequency Noise Before Switch-Frequency Modulation
Switch-Frequency Modulation
Access to the VCO input also makes it possible to modulate the regulator’s switching frequency. Through frequency modulation, the peak energy of the fundamental is
spread over the frequency range of modulation, thus
decreasing the peak energy level at any one frequency.
This frequency spreading action increases with each har-
AN84-82
monic, so that the second harmonic has twice the bandwidth and the third harmonic has three times the bandwidth until all the harmonics blend together, decreasing
the signal strength at all frequencies. This can be seen in
the spectrum analyzer plots shown in Figures 148–150.
www.BDTIC.com/Linear
Application Note 84
–55dBm
–65dBm
–75dBm
–85dBm
VIN = 10V
VO = 5V AT 3A
BW = 300Hz
–95dBm
–105dBm
1MHz
30MHz
15MHz
DI1436_04.eps
Figure 150. Output High Frequency Noise After Switch-Frequency Modulation
The VCO in the LTC1436-PLL has an input range from 0V
to 2.4V. As shown in Figure 151, the switch frequency can
be modulated at least ±30% around the center frequency
fO. The ideal modulating signal varies an equal amount
above and below the center frequency voltage of 1.2V,
with a constant slope. The reference circuit of Figure 152
develops a 100Hz sawtooth voltage from 0.9V to 1.5V that
modulates the LTC1436-PLL in Figure 147 to generate the
plots shown in Figures 148–150. Modulator circuit complexity is largely determined by functional requirements.
For most applications, a precision modulating signal is not
required, because high order harmonics blend together.
Consequently, modulating frequency, slope and peak-topeak voltage are not critical.
Figure 148 shows the full load output noise level from the
circuit of Figure 147, before and after switch-frequency
modulation. The black trace shows the normal output
noise from 1kHz to 1MHz with the VCO at minimum
frequency, whereas the colored trace shows output noise
after modulation around the center frequency. The 228kHz
unmodulated switch-frequency output noise decreased
more than 30dB through modulation between 270kHz and
370kHz. Figures 149 and 150 show a 10dB to 15dB
attenuation in full-load output voltage noise from 1MHz to
30MHz after modulation.
1.5V
LTC1436-PLL
PIN 17
(5V)
COSC = 100pF
~10ms
8
1.2M
TLC555
7
DISCH
6
OUT
THRESH
4
5
RESET
CONT
220Ω
2
1.3f0
FREQUENCY
0.9V
0.1µF
1
GND
+VCC
0V
510k
TRIG
3
COSC = 47pF
f0
–
2N3904
0.7f0
+
LT1077
LTC1436-PLL
PIN 1
(MOD)
150k
0.1µF
100k
DI1436_06.eps
0
0.5
1.0
1.5
VPLL LPF (V)
2.0
2.5
DI1436_05.eps
Figure 151. Operating Frequency vs VPLL LPF
LTC1436-PLL
PIN 6
(GND)
Figure 152. Switch-Frequency Modulator
www.BDTIC.com/Linear
AN84-83
Application Note 84
–20dBm
–20dBm
– 40dBm
–40dBm
– 60dBm
–60dBm
– 80dBm
–80dBm
–100dBm
–100dBm
10VIN
5VOUT AT 3mA
BW = 100Hz
–120dBm
10Hz
10kHz
20kHz
10VIN
5VOUT AT 3mA
BW = 100Hz
–120dBm
10Hz
10kHz
20kHz
DI1436_08.eps
DI1436_07.eps
Figure 153. Audio Frequencies in Output Noise during
Cycle-Skipping Operation
Figure 154. Output Noise with Adaptive Power Operation
Audio Frequency Suppression
The Adaptive Power feature of the LTC1436-PLL
significantly reduces audio frequency generation, while
maintaining good efficiency under very light load conditions. Figure 153 shows the audio frequencies generated
by the highly efficient cycle skipping mode of the LTC1436PLL. Figure 154 shows the decrease in audio frequencies
resulting from Adaptive Power operation. Figure 155
shows efficiency curves of both the cycle skipping and
Adaptive Power modes along with the traditional, forced
continuous mode of operation.
the peak-to-peak inductor current to flow, even under no
load conditions. The synchronous buck topology allows
the top switch, Q1, to put current into the output capacitor,
followed by the bottom switch, Q2, taking current out of
the output capacitor while regulating the output voltage
under no-load conditions. Although constant frequency is
maintained, high current I2R losses and high gate charge
losses continue under light load conditions. Forced-current operation is useful for fast transient response required
for high di/dt loads like the Intel Pentium® processor.
Cycle skipping is the most efficient mode during light-load
operation, where the output capacitor supplies load current most of the time and is replenished by bursts of
energy at a rate determined by the load. When load current
is low enough, the burst rate falls into the audio-frequency
range, which can cause problems. With the addition of Q3,
an inexpensive SOT-23 size MOSFET, the Adaptive Power
circuitry inside the LTC1436-PLL takes control during
light load conditions, turning off high current MOSFETs
Q1 and Q2. Q3 and D2 are then used in a conventional
constant frequency buck mode, eliminating the power
loss caused by charging and discharging the large input
capacitance of both power MOSFETs.
Cycle skipping, Adaptive Power and forced current operation are all available on the LTC1436-PLL, so that the best
operating mode can be selected for each application.
The conventional way of avoiding audio-frequency interference is the forced current mode, where both high
current MOSFETs continue to operate at full frequency and
normal duty cycle under all load conditions. This causes
AN84-84
100
1. CYCLE SKIPPING OPERATION:
VARIABLE FREQUENCY
COMPONENTS AT LOWER
OUTPUT CURRENTS
3. FORCED CONTINUOUS
OPERATION: CONSTANT
FREQUENCY USING
LARGE MOSFETS Q1 AND Q2
EFFICIENCY (%)
2. Adaptive Power OPERATION:
CONSTANT FREQUENCY WITH
AUTOMATIC SWITCHOVER TO
SMALL MOSFET Q3
90
(1)
10V IN
5V OUT
80
70
60
50
1mA
(2)
(3)
10mA
100mA
1A
OUTPUT CURRENT
Figure 155. Efficiency Curves for Light Load Currents
www.BDTIC.com/Linear
10A
Application Note 84
For this circuit (Figure 156), power-up is initiated by a low
level signal on the NAND gate. This signal could come from
any front-panel switch or from an external interrupt signal.
The system power is turned off by means of a low level
signal from a controller/logic device. In either case, the
control signal to the LTC1174 must be latched. (A latched
turn-off signal ensures a known state on the LTC1174
shutdown pin during the collapse of the 5V supply.)
Regulators—Switching (Micropower)
POWER MANAGEMENT AND HIGH EFFICIENCY
SWITCHER MAXIMIZE NINE-VOLT BATTERY LIFE
by LTC Applications Staff
The LTC1174 (3.3V, 5V and adjustable versions) can
convert a 9V battery source to system power with very
high efficiency. Efficiency is over 90% at load currents
from 20mA to 425mA and over 85% at a load current of
4mA. For a given load, maximum battery life can be
obtained by minimizing shutdown current during system
shutdown and maximizing converter efficiency during
operation. A single control line to the LTC1174 can be used
to select shutdown mode or operational mode, as required.
3
6
7
LBIN VIN IPGM
5
SW
+
9V
The CD4012 and CD4013 are powered from the battery;
the 2N2222 provides simple level shifting to the battery
rail. R1 and C1 ensure that the circuit remains in powerdown mode during battery replacement. The circuit shown
here provides approximately 90% efficiency at 250mA
load current, and consumes less than 1µA shutdown
current. Turn-on and turn-off transitions are very clean.
22µF*
0.22µF
LTC1174-5
VOUT
GND SD
4
8
9V
100k
100k
1/2 CD4012
1
L1**
50µH
5V
+
D1
1N5818
0.1µF
TO CONTROLS,
ETC.
100µF*
* AVX TPS
** COILTRONICS CTX50-4
(561) 241-7876
9V
9V
9V
D
S
1/2 CD4013
Q
100k
5V
100k
93.1k
R
9V
100k
9V
TO
CONTROLLER
5V
C1
0.1µF
R1
200k
ANY FRONT PANEL SWITCH
0.0068µF
TO PIN 1
OF LTC1174
30.9k
100k
2N2222
FROM OPEN
COLLECTOR OUTPUT
OF CONTROLLER
1 = ON, 0 = OFF
FOR MINIMUM RF NOISE
USE LTC1174 - ADJUSTABLE
WITH ABOVE NETWORK
Figure 156. Schematic Diagram of High Efficiency DC/DC Converter
0.250A
RUN
STANDBY*
RUN
STANDBY
0
5 SEC
TIME
*STANDBY TIME IS LONG
IAVG < 5mA
Figure 157. Load Profile
www.BDTIC.com/Linear
AN84-85
Application Note 84
90
LT1307 MICROPOWER DC/DC CONVERTER
ELIMINATES ELECTROLYTIC CAPACITORS
by Steve Pietkiewicz
The relentless push towards increasing miniaturization in
portable electronic products has created the need for
small, high speed, low voltage DC/DC converter ICs. The
LT1307 combines a current-mode, fixed frequency PWM
architecture with Burst Mode™ micropower operation to
maintain high efficiency at light loads. It uses small, low
cost ceramic capacitors for both input and output, minimizing board area. By employing fixed frequency 575kHz
switching the LT1307 keeps spectral energy out of the
455kHz band. Dense, high speed bipolar process technology enables the LT1307 to fit in the MSOP package, and
micropower circuitry results in just 60µA quiescent current at no load. Conversion efficiency exceeds 80%, and
the device also includes a low battery detector.
Single-Cell Boost Converter
A complete single-cell to 3.3V converter is shown in Figure
158. The circuit generates 3.3V at up to 75mA from a 1.0V
input. The 10µF ceramic output capacitor can be obtained
from several vendors. Efficiency, detailed in Figure 159,
peaks at 80% and exceeds 70% over the 1:500 load range
of 200µA to 100mA at a 1.25V input. Changing the value
of R1 to 1.87MΩ moves the output to 5V. Efficiency of the
5V output converter is depicted in Figure 160. Figure 161’s
L1
10µH
1.5V
CELL
SHUTDOWN
LT1307
SHDN
LBO
GND
VC
C1 = MURATA-ERIE GRM235Y5V105Z01
MARCON THCS50E1E105Z
TOKIN 1E105ZY5U-C103-F
C2 = MURATA-ERIE GRM235Y5V106Z01
MARCON THCS50E1E105Z
TOKIN 1E106ZY5U-C304-F
VIN = 1.5V
50
0.1
1
10
LOAD CURRENT (mA)
100
300
Figure 159. 3.3V Efficiency
oscillograph shows output voltage and inductor current as
the load current is stepped from 5mA to 55mA, revealing
substantial detail about the operation of the LT1307. With
a 5mA load, VOUT (top trace) exhibits a ripple voltage of
60mV at 4kHz. The device is in Burst Mode at this output
current level. Burst Mode operation enables the converter
to maintain high efficiency at light loads by turning off all
circuitry inside the LT1307 except the reference and error
amplifier. When the LT1307 is not switching, quiescent
current decreases to 60µA. When switching, inductor
current (middle trace) is limited to approximately 100mA.
Switching frequency inside the “bursts” is 575kHz. As the
load is stepped to 55mA, the device shifts from Burst
Mode to constant switching mode. Inductor current increases to about 300mA peak and the low frequency Burst
Mode ripple goes away. R3 and C3 stabilize the loop.
90
R1
1.02M
1%
R2
604k
1%
3.3V
75mA
C2
10µF
80
VIN = 1.00V
70
VIN = 1.25V
VIN = 1.5V
60
D1 = MOTOROLA MBR0520L
L1 = SUMIDA CD43-100
Figure 158. Single Cell to 3.3V Boost Converter Delivers 75mA
at 1.0V Input. Changing R1 to 1.87M Moves the Output to 5V
AN84-86
VIN = 1.25V
D1
SW
FB
R3
100k
C3
680pF
VIN = 1.00V
70
60
EFFICIENCY (%)
C1
1µF
VIN
LBI
EFFICIENCY (%)
80
50
0.1
10
1
LOAD CURRENT (mA)
100 200
Figure 160. Efficiency at 5V Output
www.BDTIC.com/Linear
Application Note 84
IL
200mA/DIV
ILOAD 55mA
5mA
VIN = 1.25V
500µs/DIV
Figure 161. Transient Response with 5mA to 55mA Load Step
DC/DC Converter Noise Considerations
OUTPUT NOISE VOLTAGE (dBmVRMS)
–20
VOUT
200mV/DIV
AC COUPLED
–25
RBW = 100Hz
–30
–35
–40
–45
–50
–55
–60
–65
–70
255
Switching regulator noise is a significant concern in many
communications systems. The LT1307 is designed to
keep noise energy out of the 455kHz band at all load levels
while consuming only 60µW–100µW at no load. At light
load levels, the device is in Burst Mode, causing low
frequency ripple to appear at the output. Figure 162 details
spectral noise directly at the output of Figure 158’s circuit
in a 1kHz to 1MHz bandwidth. The converter supplies a
5mA load from a 1.25V input. The Burst Mode fundamental at 5.1kHz and its harmonics are quite evident, as is the
575kHz switching frequency. Note, however, the absence
of significant energy at 455kHz. Figure 163’s plot reduces
the frequency span from 255kHz to 655kHz with a 455kHz
center. Burst Mode low frequency ripple creates sidebands around the 575kHz switching fundamental. These
sidebands have low signal amplitude at 455kHz, measuring –55dBmVRMS. As load current is further reduced, the
Burst Mode frequency decreases. This spaces the sidebands around the switching frequency closer together,
moving spectral energy further away from 455kHz. Figure
164 shows the noise spectrum of the converter with the
Output filtering can reduce output conducted noise. Figure
158’s circuit, supplying a 50mA load at 3.3V from a 1.3V
source, is shown with an output filter (R4 and C4) in Figure
165. The lowpass filter created by R4 and C4 places a pole
at 34kHz, reducing high frequency spikes considerably.
0
20
10
0
–10
–20
–30
–40
–50
–60
load increased to 20mA. The LT1307 shifts out of Burst
Mode, eliminating low frequency ripple. Spectral energy is
present only at the switching fundamental and its harmonics. Noise voltage measures –5dBmVRMS or 560µVRMS at
the 575kHz switching frequency, and is below –60dBmVRMS
for all other frequencies in the range. By combining Burst
Mode with fixed frequency operation, the LT1307 keeps
noise away from 455kHz, making the device ideal for RF
applications where the absence of noise in the this band is
critical.
RBW = 100Hz
30
1
10
100
FREQUENCY (kHz)
1000
Figure 162. Spectral Noise Plot of 3.3V Converter Delivering
5mA Load; Burst Mode Fundamental at 5.1kHz is 23dBmVRMS
or 14mVRMS
655
Figure 163. Span Centered at 455kHz Shows –55dBmVRMS
(1.8µVRMS) at 455kHz. Burst Mode Creates Sidebands 5.1kHz
Apart around the Switching Frequency Fundamental of 575kHz
OUTPUT NOISE VOLTAGE (dBmVRMS)
OUTPUT NOISE VOLTAGE (dBmVRMS)
40
455
FREQUENCY (kHz)
RBW = 100Hz
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100
255
455
FREQUENCY (kHz)
655
Figure 164. With the Converter Delivering 20mA,
Low Frequency Sidebands Disappear. Noise is
Present Only at the 575kHz Switching Frequency
www.BDTIC.com/Linear
AN84-87
Application Note 84
L1
10µH
LED Driver
D1
VOUT1
VIN
1.5V
CELL
R2
1.02M
1%
SW
R4
4.7Ω
VOUT2
FB
LT1307
C1
1µF
R3
604k
1%
GND
VC
R1
100k
C2
10µF
C4
1µF
C3
680pF
Figure 165. Figure 158’s Circuit with Output Filter R4/C4
Viewed in a 50MHz bandwidth, the filter reduces switching
spikes from about 10mVP–P to about 1mVP–P, as detailed
in Figure 166. Beware, though; the oscilloscope used in
Figure 166’s oscillograph (a Tektronix Type 547) is helping with the filtering by attenuating frequencies above
50MHz. Figure 167 shows the same circuit viewed on a
400MHz oscilloscope. The filter still attenuates but the
magnitude of switching noise is far higher (140mVP–P
unattenuated). A small amount of copper trace can be used
in place of the resistor if the attendant voltage drop is
unacceptable. A surprisingly small amount of trace is
needed to create an effective filter; a PC trace of 1 oz.
copper, 1 inch long by 10 mils wide, has an inductance of
29nH. Inductive reactance at 50MHz (2πfL) is 9.1Ω. A
combination of copper trace and 0.1µF ceramic capacitors
will reduce high frequency spikes to acceptable levels in
most systems.
LEDs require current source drive. Typically, a 5V supply
with a series resistor to limit current is used to power the
LED. Although simple, this approach has poor efficiency
and requires a voltage source higher than the 2V–3V
forward drop of most LEDs. Additionally, each LED requires its own ballast resistor. Figure 168’s circuit uses the
LT1307 configured as a current source to drive a seriesconnected pair of LEDs from a single-cell input. The IC’s
low battery detector monitors the voltage across sense
resistor R1. LBO drives Q1; this provides correct phasing
to the VC pin. Q1 and R2 drive the VC pin, overriding the
internal error amplifier. With 200mV across R1, 25mA
flows through the LED pair. C3 provides frequency compensation. For proper operation, the circuit must always
supply enough power so as to not enter Burst Mode
operation. This precludes driving most single LEDs (high
brightness blue LEDs have a forward drop of 3.4V and can
be driven singly). In shutdown mode, the circuit draws a
only few microamperes. Start-up sequencing is detailed in
Figure 169. The voltage at LBI stabilizes in about 200µs
with minimal overshoot and ringing. The Lumex “MegaBrite” red LEDs specified in Figure 168 provide enough
light to act as a flashlight, providing young children with a
high technology toy. Mounted on a small PC board with a
push-button switch, the circuit entertained my two children for hours. They are both satisfied LT1307 customers.
VOUT1
5mV/DIV
AC COUPLED
VOUT1
50mV/DIV
AC COUPLED
VOUT2
5mV/DIV
AC COUPLED
VOUT2
5mV/DIV
AC COUPLED
200ns/DIV
Figure 166. VOUT1 is Output Voltage at 10µF Capacitor
C2; VOUT2 is After 4.7Ω/1µF Output Filter. Circuit
Supplies 50mA; Oscilloscope Bandwidth is 50MHz.
AN84-88
VIN = 1.3V
VIN = 1.3V
200ns/DIV
Figure 167. A Faster Oscilloscope Shows More High Frequency
Content at Both Outputs. Scope Bandwidth is 400MHz.
www.BDTIC.com/Linear
Application Note 84
L1
10µH
VIN
D2
100k
VIN
Q1
2N3906
AA
CELL
C2
1µF
+
D1
SW
LB0
FB
NC
LT1307
VC
C3
22µF
VLBI
100mV/DIV
LBI
SHDN
R2
22k
D3
R1
8Ω
GND
C1
1µF
ON
OFF
100k
L1 = MURATA-ERIE LQH3C100K04
D1 = MOTOROLA MBR0520L
C1, C2 = CERAMIC
ISW
100mA/DIV
D2, D3 = LUMEX SSL-X100133SRC/4
"MEGA-BRITE" RED LED
OR PANASONIC LNG992CF9
HIGH BRIGHTNESS BLUE LED
ON/OFF
VIN
Figure 169. Start-Up Response of LED
Circuit. Many Switching Cycles Elapse
before Current Flows in LEDs Because
of C1 Charging
Figure 168. Single-Cell LED Driver Supplies 25mA to LED String.
Two Red LEDs Can Be Replaced by One Blue LED
AN ULTRALOW QUIESCENT CURRENT,
5V BOOST REGULATOR
by Sam Nork
Many battery-powered applications require an auxiliary
5V supply to power infrequently used circuitry, such as
smart card readers, wireless i.d. tags, or the like. Keeping
the 5V supply permanently active is desirable, since this
eliminates timing delays and inrush currents due to supply
start-up. The downside is that most 5V boost converters
consume an unacceptable amount of quiescent current
under no-load conditions. This problem is addressed by
the SHDN features of the LTC1516 micropower, chargepump DC/DC converter. Toggling the SHDN pin of the
LTC1516 allows the 5V supply to remain in regulation with
a typical no-load input current of less than 5µA. When the
5V output load is enabled, the part can supply up to 50mA
of load current.
The LTC1516 produces a regulated 5V output from a 2V to
5V input In shutdown mode, the output load is disconnected from VIN and the quiescent current drops below
1µA.
When the output is in regulation, the internal sense resistor
draws only 1.5µA (typical) from VOUT. During no-load
conditions, this internal load causes a droop rate of only
150mV per second on VOUT with COUT = 10µF. Applying a
5Hz–100Hz, 95%–98% duty-cycle signal to the SHDN pin
ensures that the circuit in Figure 170 comes out of shutdown frequently enough to maintain regulation during noload (or low-load) conditions. Since the part is kept in
shutdown mode for the majority of the time, the no-load
quiescent current (see Figure 171) is approximately equal
to (VOUT␣ × (1.5µA + ILOAD))/(VIN × efficiency).
0.22µF
1
VIN = 2V TO 5V
2
+
10µF
3
+
10µF 4
C1+
C1–
VIN
SHDN
LTC1516
VOUT
GND
C2+
C2 –
0.22µF
8
7
FROM MPU
SHDN PIN WAVEFORMS:
6
5
LOW IQ MODE (5Hz TO 100Hz, 95% TO 98% DUTY CYCLE) VOUT LOAD ENABLE MODE
IOUT ≤ 100µA
(IOUT = 100µA TO 50mA)
VOUT = 5V ±4%
Figure 170. Ultralow Quiescent Current (<5µA) Regulated Supply
www.BDTIC.com/Linear
AN84-89
Application Note 84
The LTC1516 must be taken out of shutdown mode for a
minimum of 200µs to allow the internal sense circuitry to
start up and keep the output in regulation. As the VOUT load
current increases, the frequency with which the part is
taken out of shutdown must also be increased to prevent
VOUT from drooping below 4.8V during the OFF phase (see
Figure 172). A 100Hz, 98% duty cycle signal on the SHDN
pin ensures proper regulation with load currents as high
as 100µA. When load current greater than 100µA is
needed, the SHDN pin must be forced low, as in normal
operation. The typical no-load supply current for this
circuit with VIN = 3V is only 3.2µA.
1000
6.0
MAX SHDN OFF TIME (ms)
SHDN ON PULSE WIDTH = 200µs
COUT = 10µF
ICC (µA)
4.0
2.0
100
10
1
0.0
2.0
3.0
4.0
5.0
1
10
100
1000
IOUT (µA)
VIN (V)
Figure 171. No-Load ICC vs Input Voltage for Figure 170’s Circuit
CAPACITIVE CHARGE PUMP
POWERS 12V VPP FROM 5V SOURCE
by Mitchell Lee
The LTC1263, a regulating charge pump tripler, converts
a 5V input to a regulated 12V, 60mA output. No inductors
are required; charge pumps operate with capacitors only.
Figure 173 shows the LTC1263 configured to provide VPP
4.75V TO 5.5V
for two flash memory chips. The “flying” capacitors in the
charge pump, C1 and C2, are sized well within the surface
mount ceramic range. CIN and COUT, as shown, are surface
mount tantalum capacitors, such as Sprague 595D series.
In the 10µF capacitance range, tantalum capacitors cost
less than ceramic units. The chip operates by charging C1
and C2 in parallel across 5V and ground and then discharging them in series across 5V and the output. In theory, the
output could reach 15V, but an internal regulation loop
maintains the output at a constant 12V.
+
CIN
10µF
Figure 172. Maximum SHDN OFF-Time vs Output Load Current
for Ultalow IQ Operation
C1+
C1
470nF
VCC
SHDN
OFF
ON
C1–
LTC1263
C2+
C2
470nF
C2–
12V AT 60mA
VOUT
+
COUT
10µF
VPP
FLASH
MEMORY
Figure 173. Programming Two Flash Chips with the LTC1263
Charge Pump: In Shutdown Mode, the Output is Held at 5V
AN84-90
SHUTDOWN reduces the quiescent current of the LTC1263
to less than 1µA under logic control. In shutdown mode,
the output is held at 5V by an internal 500Ω, VCC-to-VOUT
switch. Output-voltage fall time is guaranteed to be less
than 15ms for the component values shown. Output rise
time coming out of shutdown is guaranteed to be less than
800µs.
Designing a circuit to generate a split supply from a single
5V source is usually an unpleasant chore; one to be
avoided at all costs. If load current requirements are
modest, the LTC1263 can generate both 12V and –7V for
op amps and biasing needs. Figure 174 shows how. The
www.BDTIC.com/Linear
Application Note 84
8
4.75V TO 5.5V
VCC = 5V
12V LOAD = 3mA
10µF
VCC
–7V OUTPUT (V)
+
C1+
SHDN
470nF
C1–
7
LTC1263
C2+
470nF
C2–
GND
VOUT
12V OUTPUT
6
1
+
10µF
1µF
10
–7V LOAD (mA)
100
Figure 175. Cross Regulation with a Constant 12V Load
16
MBR0520L
VCC = 5V
MBR0520L
12
+12V
10µF
OUTPUT (V)
–7V OUTPUT
8
–7V
+
4
Figure 174. Split-Supply Generator: Cross Regulation is
Improved by Driving the Inverting Charge Pump from C2+.
0
0
20
40
60
COMMON LOAD CURRENT (mA)
80
Figure 176. Output Regulation with a Common Load
LTC1263 is connected in the usual way to produce a
regulated, 12V output, but a 2 diode, 2-capacitor charge
pump is added to the C2+ pin. This pin switches between
VCC and VOUT, swinging approximately 7VP–P. The result
is an outboard charge pump inverter with a –7V output.
Schemes like this one often suffer from poor cross regulation. Although the inverting output is not directly regulated, the –7V load does affect the 12V output, thereby
improving cross regulation (see Figure 175). The regulation with a common load (such as op amps) is shown in
Figure 176.
LTC1474 AND LTC1475 HIGH EFFICIENCY SWITCHING
REGULATORS DRAW ONLY 10µA SUPPLY CURRENT
by Greg Dittmer
Introduction
Maximizing battery life, one of the key design requirements for all battery-powered products, is now easier with
Linear Technology’s new family of ultralow quiescent
current, high efficiency step-down regulator ICs, the
LTC1474 and LTC1475. The LTC1474/LTC1475 are step-
down regulators with on-chip P-channel MOSFET power
switches. These regulators draw only 10µA supply current
at no load while maintaining the output voltage. Wide
supply voltage range (3V–18V) and 100% duty cycle
capability for low dropout allow maximum energy to be
extracted from the battery, making the LTC1474/LTC1475
ideal for moderate current (up to 300mA) battery-powered
applications.
www.BDTIC.com/Linear
AN84-91
Application Note 84
Other features include Burst Mode™ operation to maintain
high efficiency over almost four decades of load current,
an on-chip low-battery comparator and a shutdown mode
to further reduce supply current to 6µA. The LTC1475
provides on/off control with push-button switches for use
in handheld products.
The LTC1474/LTC1475 are available in adjustable and
fixed 3.3V/5V output voltage versions, in 8-pin MSOP and
SO packages.
3.3V/200mA Step-Down Regulator
A typical application circuit using the LTC1474 is shown in
Figure 177. This circuit supplies a 200mA load at 3.3V with
an input supply range of 4V–18V (3.3V at no load). The
0.1Ω sense resistor reduces the peak current to about
285mA, which is the minimum level necessary to meet the
200mA load current requirement with a 100µH inductor.
The peak current can be reduced further if a higher value
inductor is used. Since the output capacitor dominates the
output voltage ripple, an AVX TPS series low ESR (150mΩ)
output capacitor is used to provide a good compromise
between size and low ESR. With this capacitor the output
ripple is less than 50mV.
Efficiency Considerations
The efficiency curves for the 3.3V/200mA regulator at
various supply voltages are shown in Figure 178. Note the
flatness of the curves over the upper three decades of load
current and that the efficiency remains high down to
extremely light loads. Efficiency at light loads depends on
low quiescent current. The curves are flat because all
significant sources of loss except for the 10µA standby
current—I2R losses in the switch, catch diode losses, gate
charge losses to turn on the switch and burst cycle DC
supply current losses—are identical during each burst
cycle. The only variable is the rate at which the burst cycles
occur. Since burst frequency is proportional to load, the
loss as a percentage of load remains relatively constant.
The efficiency drops off as the load decreases below about
1mA because the non-load-dependent 10µA standby current loss then constitutes a more significant percentage of
the output power. This loss is proportional to VIN and thus
its effect is more pronounced at higher VIN.
LTC1475 Push-Button On/Off Operation
The LTC1475 provides the option of push-button control
of run and shutdown modes for handheld products. In
contrast to the LTC1474’s run/shutdown mode, which is
controlled by a voltage level at the RUN pin (ground =
shutdown, open/high = run), the LTC1475 run/shutdown
mode is controlled by an internal S/R flip-flop that is set
(run mode) by momentarily shorting the ON pin to ground
and reset (shutdown mode) by a momentary ground at the
LBI pin (see Figure 179). This provides simple on/off
control with two push-button switches. The simplest
implementation of this function is shown in Figure 180,
with normally open push-button switches connected to
the ON and LBI pins. Note that because the switch on LBI
is normally open, it doesn’t affect the normal operation of
this input to the low-battery comparator. With a resistor
100
2
LBO
3
LBI
4
VFB
RUN
LBO
VIN
LBI
SENSE
SW
GND
8
100k
6
1000pF
0.1µF
5
0.1Ω
VOUT
3.3V/200mA
COUT
100µF
6.3V
L1 100µH
+
1.69M
10pF
MBR0530
90
RUN
7
D1:
L1:
COUT:
CIN:
MBR0530
SUMIDA CDRH74
TPSC107006R0150
THC50EIE106Z
1M
VIN
4V TO 18V
10µF
25V
EFFICIENCY (%)
LTC1474
1
VIN = 5V
VIN = 10V
80
VIN = 15V
70
L = 100µH
VOUT = 3.3V
RSENSE = 0.1Ω
60
50
0.02
0.2
2
20
LOAD CURRENT (mA)
200
1474_04.eps
Figure 177. LTC1474 3.3V/200mA Step-Down Regulator
AN84-92
Figure 178. Efficiency vs Load
Current for Figure 177’s Circuit
www.BDTIC.com/Linear
Application Note 84
RUN
100k
U1
LTC1475
LTC1474
ON
MODE
RUN
SHUTDOWN
RUN
1
VFB
2
3
ON OVERRIDES SHUTDOWN
WHILE RUN IS LOW
4
VFB
ON
LBO
VIN
LBI/OFF SENSE
SW
GND
8
7
6
10µF
5
100µH
ON
OFF
VOUT
VBATT
+
2.2M
1M
LTC1475
100µF
VBATT
LBI
VFB
1474_05.eps
MODE
RUN
SHUTDOWN
RUN
Figure 179. Comparison of RUN/SHUTDOWN
Operation for the LTC1474 and LTC1475
Figure 180. LTC1475 Step-Down Regulator
with Push-Button On/Off Control
divider network connected to the LBI to monitor the input
supply voltage level, the voltage at this pin will normally be
above the low-battery trip threshold of 1.23V. When this
pin is pulled below 0.7V by depressing the switch, the
internal flip-flop is reset to invoke shutdown.
can force the LTC1475 off when it detects a depressed
push button. Because the LTC1475 supplies power to the
microcontroller, once the microcontroller is off, it can no
longer turn the LTC1475 back on. However, since the push
button is also connected directly to the ON pin, the
LTC1475 can be turned back on directly from the push
button without the microcontroller. The LTC1475 then
powers up the microcontroller. The discrete inputs of
most microcontrollers have a reverse biased diode, D2,
between the input and supply; thus a blocking diode with
less than 1µA leakage is necessary to prevent the powered down microcontroller from pulling down on the ON
pin.
Figure 181 shows an example of push-button on/off
control of a LTC1475 microcontroller application with a
single push button. The push button is connected to the
microcontroller as a discrete input so that the
microcontroller can monitor the state of the push button.
The LTC1475 LBI pin is connected to one of the
microcontroller’s open-drain discrete outputs so that it
µC
MMBD914LT1
VCC
100k
LTC1475
ON/OFF
VFB
1
2
3
4
VFB
ON
LBO
VIN
LBI/OFF SENSE
GND
SW
8
7
6
100µH
1M
VBATT
10µF
5
VOUT
+
100µF
2.2M
VBATT
VFB
1474_06.eps
Figure 181. A Single Push-Button Controls On/Off for the LTC1475 Regulator and Microcontroller
www.BDTIC.com/Linear
AN84-93
Application Note 84
FREE DIGITAL PANEL METERS FROM THE
OPPRESSIVE YOKE OF BATTERIES
by Mitchell Lee
Digital panel meters (DPMs) have dropped in price to well
under $10 for 3-1/2 digit models, even in single-piece
quantities. These make excellent displays for many instruments, but suffer from one major flaw: they require a
floating power supply, usually in the form of a 9V battery.
This renders inexpensive meters useless for most applications because no one wants multiple 9V batteries in their
product.
The circuit shown in Figure 182 powers up to five meters
from a single 1.8V to 6V source. The source need not be
floating, yet all five outputs are fully floating, isolated and
independent in every respect. The circuit consists of an
LT1303 micropower, high efficiency DC/DC converter
driving a 5-output flyback converter. An off-the-shelf
surface mount coil, Coiltronics’ VERSA-PAC™ VP1-0190,
is used as the transformer. This device is hipot tested to
500VRMS—more than adequate for most applications.
Feedback is extracted from the primary by Q1, which
samples the flyback pedestal during the switch off time.
Typical DPMs draw approximately 1mA supply current.
The primary is also loaded with 1mA for optimum regulation and ripple. Primary snubbing components, a necessity in most flyback circuits, are obviated by the primary
feedback rectifier and smoothing capacitor. Although this
circuit has been set up for 9V output (9.3V, to be exact),
some DPMs need 5V or 7V. Use a 4.3kΩ or 6.2kΩ resistor
in place of R1 for these voltages. The output voltage is set
by
Output ripple measures 200mVP–P and can be proportionately reduced by increasing the output capacitance. If
more ripple is acceptable, the output capacitors can be
reduced in value. A shutdown feature is available on the
LT1303, useful where a “sleep” function is included to
save power.
With each output loaded at 1mA, the input current is
16.5mA on a 5V supply. This figure rises to about 45mA on
a 1.8V (2-cell) input. If the system is battery operated and
if the battery voltage does not exceed 7V, operate the
circuit directly from the battery for best efficiency. In lineoperated equipment, use a regulated 5VDC or 3.3VDC
supply.
5 × MBR0520L
10µF
25V
MBR0520L
+
1.8VDC–
6VDC
10µF
25V
OFF
10µF
25V
+
VIN
ON
LT1303 SW
SHDN
FB
GND
PGND
R1
8.2k
Q1
2N3906
10µF
25V
+
+
MBR0520L
R2
1.2k
10µF
25V
10µF
25V
10.7µH
COILTRONICS
VP1-0190
R1 = (VOUT – 0.7)/1mA.
+
+
DIGITAL
PANEL
METERS
DIDPM_01.eps
Do not attempt to regulate the output beyond 10V or you
will exceed the maximum switch rating of the LT1303. The
LT1111 is better suited for higher voltage applications.
AN84-94
10µF
25V
+
Figure 182. LT1303 Flyback Regulator Provides Fully Floating
and Isolated 9V Supplies to Five Independent Digital Panel
Meters. Substitute 4.3k for R1 if 5V Meters are Used.
www.BDTIC.com/Linear
Application Note 84
THE LTC1514/LTC1515 PROVIDE LOW POWER
STEP-UP/STEP-DOWN DC/DC CONVERSION
WITHOUT INDUCTORS
by Sam Nork
Dual Output Supply from a 2.7V to 10V Input
Introduction
Many applications must generate a regulated supply from
an input source that may be above or below the desired
regulated output voltage. Such applications place unique
constraints on the DC/DC converter and, as a general rule,
add complexity (and cost) to the power supply. A typical
example is generating 5V from a 4-cell NiCd battery. When
the batteries are fully charged, the input voltage is around
6V; when the batteries are near end of life, the input voltage
may be as low as 3.6V. Maintaining a regulated 5V output
for the life of the batteries typically requires an inductorbased DC/DC converter (for example, a SEPIC converter)
or a complex hybrid step-up/step-down solution. The
LTC1514/LTC1515 family of switched capacitor DC/DC
converters handles this task with only three external
capacitors (Figure 183).
A unique architecture allows the parts to accommodate a
wide input voltage range (2.0V to 10V) and adjust the
operating mode as needed to maintain regulation. Hence,
the parts can be used with a wide variety of battery and/
or adapter voltages. Low power consumption (IQ = 60µA
typ) and low parts count make the parts well suited for
space-conscious low power applications, such as cellular
phones, PDAs and portable instruments. The parts come
in adjustable and fixed output-voltage versions and include additional features such as power-on reset capability (LTC1515 family) and an uncommitted comparator that
is kept alive in shutdown (LTC1514 family).
The circuit shown in Figure 185 uses the low-battery
comparator as a feedback comparator to produce an
auxiliary 3.3V regulated output from the VOUT of the
LTC1514-5. A feedback voltage divider formed by R2 and
R3 connected to the comparator input (LBI) establishes
the output voltage. The output of the comparator (LBO)
enables the current source formed by Q1, Q2, R1 and R4.
When the LBO pin is low, Q1 is turned on, allowing current
to charge output capacitor C4. Local feedback formed by
R4, Q1 and Q2 creates a constant current source from the
5V output to C4. Peak charging current is set by R4 and the
VBE of Q2, which also provides current limiting in the case
of an output short to ground. R5 pulls the gate of Q1 high
when the auxiliary output is in regulation. C5 is used to
reduce output ripple. The combined output current from
the 5V and 3.3V supplies is limited to 50mA. Since the
regulator implements a hysteretic feedback loop in place
of the traditional linear feedback loop, no compensation is
needed for loop stability. Furthermore, the high gain of the
comparator provides excellent load regulation and transient response.
Conclusion
With low operating current, low external parts count and
robust protection features, the LTC1514 and LTC1515 are
well-suited to low power step-up/step-down DC/DC conversion. The shutdown, POR and low-battery detect features provide additional value and functionality. The simplicity and versatility of these parts make them ideal for
low power DC/DC conversion applications.
5.2
100k
1
OFF
2
RESET
5V
3.3V
3
4
SHDN
VOUT
POR
VIN
LTC1515-3.5
5/3
C1+
GND
C1–
8
VOUT = 5V ± 4% OR 3.3V ± 4%
IOUT = 0 TO 50mA
7
5.0
4.9
6
5
VOUT (VOLTS)
ON
5.1
+
0.22µF
+
10µF
10µF
VIN = 4 CELLS
4.8
3
1514_01.eps
Figure 183. Programmable 5V/3V Power-Supply with Power-On Reset
4
5
VIN (VOLTS)
6
1514_XX.eps
Figure 184. VOUT vs VIN for Figure 183’s Circuit
www.BDTIC.com/Linear
AN84-95
Application Note 84
ON
OFF
R4
10Ω
Q2
R5
220k
Q1
VOUT =
3.3V
+
C4
10µF
1
R1
47k
R3
750k,
1%
SHDN
2
C5
2.2nF
VIN
LBO
LTC1514-5
3
R2
402k,
1%
VOUT
C1+
LBI
4
C1–
GND
8
VOUT = 5V
7
VIN = 2.7V TO 10V
6
5
+
C1
0.22µF
+
C3
22µF
C2
10µF
1514_04.eps
Q1 = TP0610T
Q2 = MMBT3906LT1
Figure 185. Using the Low-Battery Comparator as a Feedback Comparator to Produce
an Auxiliary 3.3V Regulated Output from the VOUT of the LTC1514-5
LTC1626 LOW VOLTAGE MONOLITHIC STEP-DOWN
CONVERTER OPERATES FROM A SINGLE Li-Ion CELL
by Tim Skovmand
output current. The maximum peak inductor current is
externally programmable to minimize component size in
lower current applications.
Introduction
The LTC1626 incorporates automatic power saving Burst
Mode operation to reduce gate-charge losses when the
load current drops below the level required for continuous
operation. With no load, the converter draws only 160µA;
in shutdown it draws a mere 1µA—making it ideal for
current-sensitive applications.
The LTC1626 is a monolithic, low voltage, step-down
current mode DC/DC converter with an input supply voltage range of 2.5V to 6V, making it ideal for single-cell LiIon or 3- to 4-cell NiCd/NiMH applications. A built-in
0.32Ω P-channel switch (VIN = 4.5V) allows up to 0.6A of
(VIN = 2.7V TO 4.5V)
SINGLE
Li-Ion
CELL
5.0
Li-Ion CELL VOLTAGE (V)
4.5
+
0.1µF
VIN
PWR VIN
LBIN
4.0
+
CIN††
47µF
16V
L1*
22µH
RSENSE**
0.1Ω
VOUT
(2.5V/0.25A)
SW
LTC1626
LBOUT
3.5
D1
MBR0520LT1
PGND
3.0
SHDN
10k
1%
SHUTDOWN
SENSE+
2.5
ITH
1k
2.0
CT
+
1000pF
SENSE–
SGND
VFB
3900pF
1.5
0
1
2
3
4
5
6
DISCHARGE TIME (HOURS)
Figure 186. Typical Single-Cell
Li-Ion Discharge Curve
AN84-96
7
100pF
CT
270pF
* SUMIDA CDRH62-220
** IRC 1206-R100F
10k
1%
† AVX TPSD107K010
†† AVX TPSD476K016
Figure 187. Single-Cell Li-Ion Battery to 2.5V Converter
www.BDTIC.com/Linear
COUT†
100µF
10V
Application Note 84
100
VIN (2.7V TO 6V)
+
95
0.1µF
VIN
PWR VIN
SHUTDOWN
VOUT
2.5V/0.25A
D1†
+
P GND
CT
CT
270pF
RSENSE**
0.1Ω
SW
LTC1626
ITH
470Ω
L*
33µH
COUT††
100µF
6.3V
EFFICIENCY (%)
CIN†††
47µF
16V
VIN = 3.5V
90
85
80
L1 = 33µH
VOUT =2.5V
RSENSE =0.1Ω
CT = 270pF
SENSE+
3900pF
1000pF
10k
1%
SENSE–
SGND
* COILTRONICS CTX33-4
** IRC 1206-R100F
† MBRS130LT
†† AVX TPSC107M006R0150
††† AVX TPSD476K016
75
70
0.01
VFB
100pF
10k
1%
1.00
0.10
OUTPUT CURRENT (A)
Figure 189. Efficiency vs Output Load Current
Figure 188. High Efficiency 2.5V Step-Down Converter
Single-Cell Li-Ion Operation
As shown in Figure 186, a fully charged single-cell Li-Ion
battery begins the discharge cycle at either 4.1V or 4.2V
(depending upon the manufacturer’s charge voltage specifications). During the bulk of the discharge time, the cell
produces between 3.5V and 4.0V. Finally, toward the end
of discharge, the cell voltage drops fairly quickly below 3V.
It is recommended that the discharge be terminated somewhere between 2.2V and 2.8V (again, depending upon the
manufacturer’s specifications).
The LTC1626 is specifically designed to accommodate a
single-cell Li-Ion discharge curve. For example, using the
circuit shown in Figure 187, it is possible to produce a
stable 2.5V/0.25A regulated output voltage with as little as
a 2.7V from the battery—thus obtaining the maximum
possible run time.
High Efficiency Operation
Using the circuit shown in Figure 188, efficiencies of
greater than 90% are maintained from 20mA to 250mA of
load current with a 3.5V input supply voltage, as shown in
Figure 189.
(VIN = 2.7V TO 6V)
+
3 OR 4
CELL
NiCD OR
NiMH
0.1µF
VIN
PWR VIN
LBIN
* SUMIDA CDRH62-220
** IRC 1206-R100F
† AVX TPSD107K010
†† AVX TPSD476K016
CIN††
47µF
16V
L1*
22µH
RSENSE**
0.1Ω
LTC1626
LBOUT
SHDN
SHUTDOWN
R1
10k
1%
D1
MBR0520LT1
PGND
SENSE+
SGND
CT
270pF
+
1000pF
CT
1k
VOUT
2.5V/0.25A
SW
ITH
3900pF
+
SENSE–
VFB
100pF
R2
10k
1%
COUT†
100µF
10V
FOR 3.3V:
R1 = 15k 1%
R2 = 9.09k 1%
Figure 190. 3- or 4-Cell NiCd/NiMH to 2.5V Converter
www.BDTIC.com/Linear
AN84-97
Application Note 84
Typical Applications
Single Li-Ion 3.3V Buck/Boost Converter
3- or 4-Cell NiCd/NiMH DC/DC Converter
The circuit shown in Figure 191 produces 3.3V from an
input voltage ranging from 2.5V to 4.5V. The two windings
of a common inductor core are used to implement this
circuit. Note that the current sense resistor is connected to
ground. The table in Figure 191 shows the output current
capability as a function of battery voltage.
Figure 190 is a schematic diagram that shows the LTC1626
being powered from a 3- or 4-cell NiCd or NiMH battery
pack. (This circuit is also suitable for operation from three
or four alkaline cells.) All the components shown in this
schematic are surface mount and have been selected to
minimize the board space and height. The output voltage
is set at 2.5V, but is easily programmed to 3.3V for 4-cell
applications. Simply modify the two output ladder resistors, R1 and R2, from 10k each to 15k and 9.09k, respectively, as shown in Figure 190.
Conclusion
The LTC1626 is specifically designed to operate from a
single-cell Li-Ion battery. With its low dropout, high efficiency and micropower operating modes, it is ideal for
battery operated products and efficiency-sensitive devices such as cellular phones and handheld industrial and
medical instruments.
(2.5V TO 4.2V)
L1B
L1A
3
2
+
0.1µF
VIN
PWR VIN
LBIN
L1B
+
CIN††
100µF
16V
33µF
10V*
SW
1
LTC1626
LBOUT
L1A
PGND
MANUFACTURER
PART NO.
COILTRONICS
DALE
CTX33-4
LPT4545-330LA
L1B
33µH
3300pF
SENSE–
CT
75pF
1000pF
RSENSE**
0.1Ω
Figure 191. Single-Cell Li-Ion to 3.3V Buck/Boost Converter
AN84-98
VOUT
3.3V
15k
1%
SGND
SENSE+
100pF
* SANYO OS-CON CAPACITOR
** IRC 1206-R100F
† AVX TPSD107M010R0100
††AVX TPSE107M016R0100
* DESIGN LIMIT
VFB
CT
1k
200
350
0500*
0500*
0500*
+
3
ITH
2.5
3.0
3.5
4.0
4.2
2
D1
MBRS130LT1
4
SHUTDOWN
SHDN
L1A
33µH
IOUT (mA)
+
TOP VIEW
1
4
Li-Ion
SINGLE
CELL
VIN (V)
www.BDTIC.com/Linear
9.09k
1%
COUT †
100µF
10V
Application Note 84
90
12V WALL CUBE TO 5V/400mA DC/DC CONVERTER
IS 85% EFFICIENT
by Steve Pietkiewicz
88
EFFICIENCY (%)
84
The ubiquitous 12V wall cube, power source of countless
electronic products, generates an unregulated DC voltage
between 8V and 18V, depending on line voltage and load.
If you use a linear regulator to drop the voltage to 5V, a
400mA load means the linear regulator must dissipate 5W
under worst-case conditions. To deal with this heat, you
must provide adequate heat sinking, increasing your
product’s size and weight. Additionally, the heat is sometimes objectionable to customers. These factors can negate the cost advantage of a linear regulator. Figure 192’s
circuit, a negative buck converter, delivers 5V at loads up
to 400mA from a 7V–25V input with peak efficiency of
85%, eliminating the need for a heat sink. Since the
LT1307B (U1) is intended for use with a low input voltage,
Q1 and Q2 are used to make a simple preregulator,
providing 1.9V for U1’s VIN pin. The IC switches at 600kHz,
allowing a low cost 22µH inductor and 10µF ceramic
output capacitor to be used. Q3 is needed to level shift the
output voltage because U1’s feedback pin is referenced to
the negative input. Output ripple measures 10mVP-P at a
load of 400mA. The circuit’s efficiency is detailed in Figure
193, and response to a load step from 150mA to 300mA
is shown in Figure 194. Input bypass capacitor C1 sees
worst-case RMS ripple current equal to one-half the
output current and should have an ESR of less than 0.5Ω.
Take care during construction to keep R1–R3 and Q3 close
to U1’s FB pin and away from the SW pin to prevent
unwanted coupling. Use a ground plane and keep traces
for the power components short and direct.
+
12V
UNREGULATED
SUPPLY
VIN = 8V
86
VIN = 12V
82
VIN = 18V
80
78
76
74
72
70
50
100
150 200 250 300
LOAD CURRENT (mA)
350
400
DI-ADAP_02.EPS
Figure 193. Efficiency Peaks at 85%; It Is Above
80% Over an Input Range of 8V–18V
Although it might seem unsettling that the negative side of
the wall cube is not grounded, remember that the 9V wall
cube floats. The circuit merely regulates the negative side,
rather than the more conventional positive side.
VOUT
0.5V/DIV
IL1
0.5A/DIV
ILOAD
0.3A/0.15A
50µs/DIV
DI_ADAP_03.eps
Figure 194. Load-Step Response; the Load
Changes from 150mA to 300mA
5V
400mA
C2
10µF
CERAMIC
10k
Q2
2N3904
1N5818
–
3
6
SW
U1
LT1307B
FB
30k
Q1
2N3904
VC
1
1µF
CERAMIC
10k
L1
22µH
VIN
SHDN
5
2
Q3
2N3906
GND
4
30k
1000pF
R1
12.1k
1%
R2
42.2k, 1%
R3
100k
+
C1
33µF
25V
L1 = SUMIDA CD54-220
DI_ADAP_01.EPS
Figure 192. This Negative Buck Converter Delivers 5V at 400mA from a 7V–25V Input
www.BDTIC.com/Linear
AN84-99
Application Note 84
MICROPOWER 600kHz FIXED-FREQUENCY
DC/DC CONVERTERS STEP UP FROM A 1-CELL
OR 2-CELL BATTERY
by Steve Pietkiewicz
VOUT
200mV/DIV
AC COUPLED
Linear Technology introduces two new micropower DC/
DC converters designed to provide power from a singlecell or higher input voltage. The LT1308 features an
onboard switch capable of handling 2A with a voltage drop
of 300mV and operates from an input voltage as low as 1V.
The LT1317, intended for lower power requirements,
operates from an input voltage as low as 1.5V. Its internal
switch handles 600mA with a drop of 360mV. Both devices
feature Burst Mode operation at light load; efficiencies are
above 70% for load currents of 1mA. Both devices switch
at 600kHz; this high frequency keeps associated power
components small and flat; additionally, troublesome interference problems in the sensitive 455kHz IF band are
INDUCTOR
CURRENT
1A/DIV
1ms/DIV
Figure 197. Transient response of DC/DC
converter: VIN = 3V, 0A–1A load step
avoided. The LT1308 is intended for generating power on
the order of 2W–5W. This is sufficient for RF power
amplifiers in GSM or DECT terminals or for digital-camera
power supplies. The LT1317, with its smaller switch, can
generate 100mW to 2W of power. The LT1317 is available
in LTC’s smallest 8-lead package, the MSOP. This package
is approximately one-half the size of a standard 8-lead SO
package. The LT1308 is available in the 8-lead SO package.
3V TO 4.2V
L1
4.7µH
VIN
SHDN
SW
LBI
R1
301k
LT1308
Li-Ion
CELL
C1
100µF
LBO
5V
1A
FB
GND
VC
R2
100k
RC
47k
CC
22nF
+
C2
100µF
NiCD
CELL
C1
10µF
Figure 195. Single Li-Ion Cell to 5V/1A DC/DC Converter
C1: CERAMIC
C2: AVX TPS SERIES
D1: IR 10BQ015
D1
3.3V
400mA
FB
GND
RC
47k
CC
22nF
R2
100k
+
C2
100µF
1308_04.eps
L1: COILTRONICS CTX5-1
COILCRAFT DO3316-472
Figure 198. Single NiCd Cell to 3.3V 400mA DC/DC Converter
90
95
V IN = 3.6V
90
VIN = 1.2V
VOUT = 3.3V
R1 = 169k
85
V IN = 4.2V
80
85
EFFICIENCY (%)
EFFICIENCY (%)
LBO
VC
2200µF
1308_01,eps
R1
169k
LT1308
+
L1: COILTRONICS CTX5-1
COILCRAFT DO3316-472
C1,C2: AVX TPS SERIES
D1: INTERNATIONAL RECTIFIER 10BQ015
SW
LBI
D1
L1
4.7µH
VIN
SHDN
80
V IN = 3V
75
75
70
65
60
70
65
55
50
1
10
100
LOAD CURRENT (mA)
1000
1
10
100
LOAD CURRENT (mA)
1000
1308 G01
1308 F01a
Figure 196. Efficiency of Figure 195’s Circuit
AN84-100
Figure 199. Efficiency of Figure 198’s Circuit Reaches 81%
www.BDTIC.com/Linear
Application Note 84
VOUT
200mV/DIV
AC COUPLED
VOUT
200mV/DIV
AC COUPLED
IL1
1A/DIV
ILOAD
400mA
50mA
ILOAD
400mA
50mA
2ms/DIV
100µs/DIV
Figure 200. DECT Load Transient Response:
with a Single NiCd Cell, the LT1308 Provides 3.3V
with a 400mA Pulsed Load. The Pulse Width = 416µs
Figure 201. DECT Load Transient Response:
Faster Sweep Speed (100µs/DIV) Details VOUT and
Inductor Current of a Single DECT Transmit Pulse
(bottom trace) increases to 1.7A peak; the input capacitor
supplies some of this current, with the remainder drawn
from the Li-Ion cell.
Single Li-Ion Cell to 5V/1A DC/DC Converter for GSM
GSM terminals have emerged as a worldwide standard. A
common requirement for these products is an efficient,
compact, step-up converter to develop 5V from a single LiIon cell to power the RF amplifier. The LT1308 performs
this function with a minimum of external components. The
circuit is detailed in Figure 195. Many designs use a large
aluminum electrolytic capacitor (1000µF to 3300µF) at the
DC/DC converter output to hold up the output voltage
during the transmit time slice, since the amplifier can
require more than 1A. The output capacitor, along with the
LT1308 compensation network, serves to smooth out the
input current demanded from the Li-Ion cell. Efficiency,
which reaches 90%, is shown in Figure 196. Transient
response of a 0A to 1A load step with typical GSM profiling
(1:8 duty cycle, 577µs pulse duration) is depicted in Figure
197. Voltage droop (top trace) is 200mV. Inductor current
VIN
1.6V
TO 6V
8
L1A
N=1
10µH 1
VIN
C1 +
VC
2
SW
C8
1nF
R4
47k
C7
22nF
LT1308
Only minor changes are required in Figure 195’s circuit to
construct a single-cell NiCd to 3.3V converter. The large
output capacitor is no longer required as the output
current can be handled directly by the LT1308. Figure 198
shows the DECT DC/DC converter circuit. Efficiency, reaching 81% from a 1.2V input, is pictured in Figure 199.
Transient response of a typical DECT load of 50mA to
400mA is detailed in Figure 200. Output voltage droop (top
trace) is under 200mV. Figure 201 zooms in on a single
pulse to show the output voltage and inductor current
responses more clearly.
C6
10µF
3
L1C 3
N = 0.3
R3
340k
SHDN
100µF
Single NiCd Cell to 3.3V/400mA Supply for DECT
L1B
N = 0.7
D1
D2
4
5V
200mA
FB
GND
R1
100k
R2
2.01M
+
C2
100µF
+
3.3V
200mA
C3
100µF
D3
7
L1D
N = 3.5
+
6
6
C1, C2, C3 = AVX TPS
C4, C5 = AVX TAJ
C6 = CERAMIC
CCD BIAS
18V
10mA
D1, D2 = IR 10BQ015
D3, D4 = BAT-85
L1 = COILTRONICS CTX02-13973
+
L1E
N=2
5
1308_08.eps
D4
C4
10µF
C5
10µF
CCD BIAS
–10V
10mA
Figure 202. This Digital Camera Power Supply Delivers 5V/200mA, 3.3V/200mA, 18V/10mA and –10V/10mA from 2 AA Cells
www.BDTIC.com/Linear
AN84-101
Application Note 84
2-Cell Digital Camera Supply
Produces 3.3V, 5V, 18V and –10V
SHUTDOWN
Power supplies for digital cameras must be small and
efficient while generating several voltages. The DSP and
logic need 3.3V, the ADC and LCD display need 5V and
biasing for the CCD element requires 18V and –10V. The
power supplies must also be free of low frequency noise,
so that postfiltering can be done easily. The obvious
approach, to use a separate DC/DC converter IC for each
output voltage, is not cost-effective. A single LT1308,
along with an inexpensive transformer, generates 3.3V/
200mA, 5V/200mA, 18V/10mA and –10V/10mA from a
pair of AA or AAA cells. Figure 202 shows the circuit. A
coupled-flyback scheme is used, actually an extension of
the SEPIC (single ended primary inductance converter)
topology. The addition of capacitor C6 clamps the SW pin,
eliminating a snubber network. Both the 3.3V and 5V
outputs are fed back to the LT1308 FB pin, a technique
known as split feedback. This compromise results in
better overall line and load regulation. The 5V output has
more influence than the 3.3V output, as can be seen from
the relative values of R2 and R3. Transformer T1 is
available from Coiltronics, Inc. (561-241-7876). Efficiency
vs input voltage for several load currents on both 3.3V and
5V outputs is pictured in Figure 203. The CCD bias
voltages are loaded with 10mA in all cases.
LT1317 2-Cell to 5V DC/DC Converter
Figure 204 shows a simple 2-cell to 5V DC/DC converter
using the LT1317. This device generates a clean, low
ripple output from an input voltage as low as 1.5V.
Designed for 2-cell applications, it offers better perfor90
85
EFFICIENCY (%)
80
100mA LOADS
70
150mA
LOADS
60
1.5
2
200mA LOADS
2.5 3
3.5 4
INPUT VOLTAGE (V)
4.5
5
Figure 203. Camera Power Supply Efficiency Reaches 78%
1308_09.EPS
AN84-102
D1
5V
200mA
FB
GND
VC
RC
100k
CC
680pF
R2
324k
1%
+
C2
33µF
C1: CERAMIC
D1: MOTOROLA MBRO520L
L1: 22µH SUMIDA CD43-220
1308_10.eps
Figure 204. 2-Cell to 5V Boost Converter Using the LT1317
mance than its 1-cell predecessor, the LT1307. More gain
in the error amplifier results in lower Burst Mode ripple,
and an internal preregulator eliminates oscillator variation
with input voltage. For comparison, Figure 205 details
transient responses of both the LT1307 and the LT1317
generating 5V from a 3V input. The load step is 5mA to
200mA. Output capacitance in both cases is 33µF. The
LT1307 has low frequency ripple of 100mV, whereas the
LT1317 Burst Mode ripple of 20mV is the same as the
600kHz ripple resulting from the output capacitor’s ESR
with a 200mA load.
Single Li-Ion Cell to ±4V DC/DC Converter
By again employing the SEPIC topology, a ±4V supply can
be designed with one IC. Figure 206’s circuit generates 4V
at 70mA and –4V at 10mA from an input voltage ranging
from 2.5V to over 5V. Maximum component height is
2mm. This converter uses two separate inductors (L1 and
L2), so it is an uncoupled SEPIC converter. This reduces
the overall cost, but requires that all output current pass
ILOAD
1
LBO
VOUT
LT1317
100mV/DIV
5V OFFSET
55
50
2 CELLS
R1
1M
LT1317
VOUT
LT1307
100mV/DIV
5V OFFSET
75
65
SW
LBI
C1
10µF
10V
L1
22µH
VIN
SHDN
200mA
5mA
500µs/DIV
Figure 205. The LT1317 Has Reduced
Burst Mode Ripple Compared to the LT1307
www.BDTIC.com/Linear
Application Note 84
through C1. Since C1 is ceramic, its ESR is low and there
is no appreciable efficiency loss. C5 is charged to –VOUT
when the switch is off, then its bottom plate is grounded
when the switch turns on. The negative output is fairly well
regulated, since the diode drops tend to cancel. The circuit
is switching continuously at rated load, where efficiency is
75%. Output ripple is under 40mV and can be reduced
further with conventional postfiltering techniques.
Conclusion
The LT1308 and LT1317 provide low noise compact
solutions for contemporary portable-product power
supplies.
D2A
D2B
–VOUT
–4V/10mA
VIN
2.5V–5V
C5
1µF
SHDN
SHUTDOWN
C1
10µF
SW
VIN
C3
15µF
LB1
LT1317
D1
R1 1M
+VOUT
4V/70mA
FB
LB0
VC
C4
1µF
+
L1
22µH
GND
+
R3
47k
R2
442k
C2
33µF
L2
22µH
C6
680pF
L1, L2 = MURATA LQH3C220
C1 = MURATA GRM235Y5V106Z01
D1 = MBR0520
D2 = BAT54S (DUAL DIODE)
C2 = AVX TAJB33M6010
C3 = AVX TAJA156MO1O
C4, C5 = CERAMIC
Figure 206. This SIngle Li-Ion Cell to ±4V DC/DC Converter Has a Maximum Height of 2mm
LT1610 MICROPOWER STEP-UP DC/DC CONVERTER
RUNS AT 1.7MHZ
by Steve Pietkiewicz
The LT1610’s input voltage ranges from 1V to 8V, and the
30V, 300mA switch allows several different configurations, such as boost, SEPIC and flyback, to be successfully
L1
4.7µH
D1
VOUT
3V
30mA
Introduction
The LT1610, a micropower DC/DC converter IC, addresses
the issue of footprint in several ways. First, the switching
frequency is 1.7MHz, allowing the use of small, inexpensive, minimal-height inductors and capacitors. Second,
the frequency-compensation components have been integrated, eliminating the requirement for an external RC
network in most applications. Finally, the device comes in
LTC’s 8-lead MSOP package, one-half the size of the 8lead SO package.
6
VIN
3
+
1 CELL
C1
22µF
5
SW
FB
SHDN
R1
1M
2
LT1610
8
COMP
VC
1
GND
7
R2
681k
+
C2
22µF
PGND
4
C1, C2: AVX TAJA226M010R
D1: MOTOROLA MBR0520
L1: MURATA LQH3C4R7M24
1610 TA01
Figure 207. This Single Cell to 3V Converter Delivers 30mA
www.BDTIC.com/Linear
AN84-103
Application Note 84
85
L1
4.7µH
VOUT = 3V
80
VIN = 1.25V
VIN = 1.5V
6
75
EFFICIENCY (%)
D1
VIN
3
70
VIN = 1V
+
65
C1
15µF
2 CELLS
60
VOUT
5V/100mA
5
SW
FB
SHDN
1M
2
332k
LT1610
8
COMP
GND
VC
55
+
7
C2
15µF
PGND
4
1
50
1
10
LOAD CURRENT (mA)
0.1
100
1610 TA02
Figure 208. Single-Cell Converter Efficiency Reaches 77%
implemented. Output voltage can be up to 28V in boost
mode. Operating quiescent current is 50µA unloaded;
grounding the shutdown pin reduces the current to 0.5µA.
The device can generate 3V at 30mA from a single (1V)
cell, or 5V at 100mA from two cells (2V). Configured as a
Li-Ion cell to 3.3V SEPIC converter, the LT1610 can deliver
100mA. In boost mode, efficiency ranges from 60% at a
100µA load to 83% at full load.
Single-Cell to 3V DC/DC Converter
A 1V to 3V boost converter is shown in Figure 207. The
specified components take up very little board space. The
4.7µH Murata inductor specified measures 2.5mm by
3.2mm and is only 2mm high. The 22µF AVX “A” case
tantalum capacitors measure 1.6mm by 3.2mm and are
1.6mm tall. Circuit efficiency, which reaches 77%, is
detailed in Figure 208. Transient response to a 1mA to
31mA load step is pictured in Figure 209. The device
features Burst Mode operation at light loads. This can be
seen at a load of 1mA. When the load is increased to 31mA,
the device shifts to constant-frequency switching and peak
switch current is controlled to achieve output regulation.
C1, C2: AVX TAJA156M010R
D1: MOTOROLA MBR0520
L1: SUMIDA CD43-4R7
MURATA LQH3C4R7M24
Figure 210. 2 Cell to 5V Converter Delivers 100mA at 2V Input
2-Cell to 5V DC/DC Converter
By simply changing the feedback resistor values, the
LT1610 can generate 5V. Figure 210’s circuit generates 5V
at a load of up to 100mA from a 2-cell input. Figure 211’s
graph shows efficiency the of the circuit, which reaches
83%. This circuit is also suitable for 3.3V to 5V conversion,
supplying over 200mA.
Li-Ion to 3.3V SEPIC Converter
Figure 212 employs the SEPIC (single ended primary
inductance converter) topology to provide a regulated
3.3V output from an input that can range above or below
the output voltage. Although the circuit requires two
inductors and a ceramic coupling capacitor, the total
footprint of this solution is still attractive compared with
alternative methods of generating 3.3V, such as a boost
converter followed by a linear regulator. The circuit can
90
VIN = 3V
EFFICIENCY (%)
80
VOUT
50mV/DIV
AC COUPLED
IL1
100mA/DIV
1610 TA04
VIN = 2V
VIN = 1.5V
70
60
31mA
ILOAD
1mA
VIN = 1.25V
VOUT = 3V
500µs/DIV
Figure 209. Transient Load Response of Single-Cell
Converter, Load Stepped from 1mA to 31mA
AN84-104
50
0.1
1
10
100
LOAD CURRENT (mA)
1000
Figure 211. 2-Cell Converter Efficiency Reaches 83%
www.BDTIC.com/Linear
1610 TA05
Application Note 84
from the battery in shutdown mode, preventing inadvertent battery discharge through the load. The LT1610’s subµA shutdown current reduces standby losses, increasing
battery life.
supply up to 100mA. Efficiency, while lower than that of a
standard boost converter, reaches approximately 73%.
Unlike a boost converter, this topology provides input-tooutput isolation. The output is completely disconnected
C3
4.7µF
CERAMIC
L1
4.7µH
INPUT
Li-ION
3V to 4.2V
6
VIN
1
+
C1
22µF
6.3V
5
SW
2
L2
4.7µH
604k
LT1610
8
COMP
SHDN
GND
VOUT
3.3V
100mA
1M
FB
VC
D1
+
3
C2
22µF
6.3V
PGND
4
7
C1, C2: AVX TAJA226M010R
C3: AVX 1206YG475
D1: MOTOROLA MBR0520
L1, L2: MURATA LQH3C4R7M24
1610 TA06
SHUTDOWN
Figure 212. Li-Ion to 3.3V SEPIC Converter Delivers 100mA
LOW NOISE 33V VARACTOR BIAS SUPPLY
by Jeff Witt
Wideband tuning circuits, such as those used in cable
television systems, require a power supply for driving a
varactor. This bias supply is usually at a voltage higher
than the system supply voltage, allowing a large tuning
range. The supply must have very little noise; voltage
ripple, for example, can appear as sidebands on a local
oscillator. This circuit takes advantage of the fixed operating frequency of the LT1317B boost regulator to generate
a low noise 33V bias voltage.
D3
680Ω
150pF
D2
C3 0.1µF
VIN
C1
15µF
10V
L1 22µH
+
VIN
D1
B
VOUT
33V
0mA TO 10mA
SW
+
SHDN
LT1317B
33k
A 47Ω
150k
VC
0.1µF
C2
10µF
35V
0.1µF
0.1µF
FB
GND
3300pF
5.90k
D1 TO D3: MOTOROLA MMBD914LT1
C1: AVX TAJ156M010
C2: SANYO 35CV33GX
L1: MURATA LQH3C220
Figure 213. This Circuit Generates a Low Noise Bais Supply for Varactor-Based Tuning Circuits
www.BDTIC.com/Linear
AN84-105
Application Note 84
The circuit (Figure 213) is a simple boost regulator with its
output voltage doubled by diodes D2 and D3 and capacitor
C3. With this doubler, the circuit can generate an output
voltage greater than the voltage rating of the LT1317B’s
internal power switch. This supply can deliver 10mA at
33V from a 3V to 6V input, allowing operation from either
3.3V or 5V logic rails. The high operating frequency
(600kHz) results in low, easily filtered output ripple, as
shown in Figure 214. The high frequency also allows the
use of small, low cost external components.
NODE A
OUTPUT RIPPLE
20mV/DIV
NODE B
OUTPUT RIPPLE
20mV/DIV
1µs/DIV
Figure 214. The Output Ripple of Figure 213’s Supply as it
Delivers 5mA at 33V from a 5V Input; Traces A and B Show
Ripple Before and After the RC Output Filter, Respectively
THE LTC1516 CONVERTS TWO CELLS TO 5V WITH
HIGH EFFICIENCY AT EXTREMELY LIGHT LOADS
by Sam Nork
Many battery-powered applications require very small
amounts of load current from the regulated supply over
long periods of time, followed by moderate load currents
for short periods of time. In these types of applications (for
example, remote data-acquisition systems, hand-held remote controls, and the like), the discharge rate of the
battery is dominated by the overall current demands under
low load conditions. In such low load systems, a primary
source of battery drain is the DC/DC converter that converts the battery voltage to a regulated supply.
The circuit shown in Figure 215 converts an input voltage
from two cells to 5V using a switched-capacitor chargepump technique. An integral comparator on the LT1516
senses the output voltage and enables the charge pump as
LTC1516
1
C1
8
ON/OFF
7
2
2 CELLS
C1+
C2+
C1–
C2–
SHDN
GND
VIN
VOUT
+
4
C2
5
6
3
+
10µF
VOUT =
5V ±4%
IOUT =
0mA TO 20mA
10µF
C1 = C2 = 0.22µF
Figure 215. 2 Cell to 5V Converter
AN84-106
www.BDTIC.com/Linear
Application Note 84
the output begins to droop. The charge pump’s 2-phase
clock controls the internal switching of flying caps C1 and
C2. (See Figure 216.) On phase one of the clock, the flying
caps are connected between VIN and GND. On phase two,
the negative plate of C1 is connected to VIN, the negative
plate of C2 is connected to the positive plate of C1, and the
positive plate of C2 is connected to the output. During this
phase of the clock, the potential on the top plate of C2 is
approximately 3 • VIN and the charge is dumped from C2
onto the output cap to raise the output voltage. The
repeated charging and discharging of C1 and C2 continues
at a nominal frequency of 600kHz until the output voltage
has risen above the internal comparator’s trip point.
VIN
10µF
S2A
When the battery cells are fully charged (approximately
1.5V per cell, for a nominal 3V VIN), the circuit operates as
a voltage doubler to maintain regulation. In doubler mode,
only C2 is charged to VIN and discharged onto VOUT when
the charge pump is enabled. As the batteries discharge
and/or the load increases, the circuit will change from
doubler mode to tripler mode. Under light load conditions,
the part will remain in doubler mode until VIN has dropped
below 2.55V. Under heavier loads, the part will go into
tripler mode at a higher VIN to maintain regulation. By
switching operating modes as the VIN and the load conditions change, the LTC1516 optimizes overall efficiency for
the life of the batteries. As shown in Figure 217, Figure
215’s circuit achieves better than 70% efficiency with load
currents from 50µA to 20mA for almost the entire life of the
batteries.
VOUT
S1A
S2B
0.22µF
C2 –
C1 +
90
C1
VIN = 3V
VIN = 2.75V
VIN = 2V
S1B
S1C
0.22µF
–
100
10µF
EFFICIENCY (%)
C2 +
80
70
VIN = 2.5V
S2C
60
VIN = 2.25V
S1D
50
0.01
CHARGE PUMP
Figure 216. LT1516 Charge Pump
in Trippler Mode, Discharge Cycle
0.1
1.0
IOUT (mA)
10
100
Figure 217. Efficiency vs VOUT for Figure 215’s Circuit
Regulators—Linear
LOW DROPOUT REGULATOR DRIVER HANDLES
FAST LOAD TRANSIENTS AND OPERATES ON
A SINGLE 3V–10V INPUT
by Lenny Hsiu
Introduction
The LT1573 is designed to provide a low cost solution to
applications requiring high current, low dropout and fast
transient response. When combined with an external PNP
power transistor, this device provides up to 5A of load
current with dropout voltages as low as 0.35V. The LT1573’s
circuitry is designed for extremely fast transient response.
This greatly reduces the bulk storage capacitance required
when the regulator is used in applications with fast, high
current load transients.
Base-drive current to the external PNP is limited for
instantaneous protection and a time-delayed latch protects the regulator from continuous short circuits. The
latch time-out period can be varied by an external capacitor. Guaranteed minimum available base-drive current to
the external PNP is 250mA. The LT1573 is equipped with
an active-high shutdown and a thermal shutdown function. The shutdown function can be used to reset the
www.BDTIC.com/Linear
AN84-107
Application Note 84
VIN
4.5V–5.5V
CIN
100µF
TANT
VIN
LT1573
CTIME***
RC 1k
CC 100pF
GND
QOUT
MOTOROLA
D45H11
VOUT = 1.265 • (1 + R1/R2)
VOUT
COMP
LATCH
+
RB
50Ω
DRIVE
+
SHDN†
RD
24Ω
R1
1.6k
+
COUT1*
FB
LOAD
R2
1k
+
COUT2**
* FOR T <45˚C, COUT1 = 24 × 1µF Y5V CERAMIC SURFACE MOUNT CAPACITORS
FOR T >45˚C, COUT1 = 24 × 1µF X7R CERAMIC SURFACE MOUNT CAPACITORS
PLACE COUT1 IN THE MICROPROCESSOR SOCKET CAVITY
** COUT2 = 220µF CHIP TANTALUM
*** CTIME = 0.5µF FOR 100ms LATCH-OFF TIME AT ROOM TEMPERATURE
†
SHDN (ACTIVE HIGH) SHOULD BE TIED TO GROUND IF NOT USED
Figure 218. 3.3V, 5A Microprocessor Supply
overcurrent latch. The thermal shutdown function can be
used to protect the PNP power transistor if it is thermally
coupled to the LT1573.
Basic Regulator Circuit
The adjustable-output LT1573 circuit shown in Figure 218
senses the regulator output voltage from its feedback pin
via the output voltage divider and drives the base of the
external PNP transistor to maintain the regulator output at
the specified value. For fixed-output versions of the LT1573,
the regulator output voltage is sensed from the feedback
pin via an internal voltage divider. In this case, the FB pin
is left unconnected. The resistor RD is required for the
overcurrent latch-off function. RD is also used to limit the
drive current available to the external PNP transistor and
to limit the power dissipation in the LT1573. Limiting the
drive current to the external PNP transistor will limit the
output current of the regulator, thereby minimizing the
stress on the regulator circuit under overload conditions.
See the LT1573 Data Sheet for additional design details.
THE LT1575/LT1577 UltraFast LINEAR REGULATOR
CONTROLLERS ELIMINATE BULK TANTALUM/
ELECTROLYTIC OUTPUT CAPACITORS
by Anthony Bonte
requires for the microprocessor. Users realize significant
savings because all additional bulk capacitance is removed.
The additional savings of insertion cost, inventory cost
and board space are readily apparent.
Introduction
Precision-trimmed adjustable and fixed-output voltage
versions accommodate any required microprocessor
power supply voltage. Dropout voltage can be user defined
via selection of the N-channel MOSFET RDS(ON). The only
output capacitors required are the high frequency ceramic
decoupling capacitors. The regulator responds to transient load changes in a few hundred nanoseconds—a
great improvement over regulators that respond in many
microseconds. The ceramic capacitor network generally
consists of ten to twenty-four 1µF capacitors, depending
on individual microprocessor requirements. The LT1575/
LT1577 family also incorporates current limiting at no
The LT1575/LT1577 family of single/dual controller ICs
are new, easy-to-use devices that drive discrete N-channel
MOSFETs as source followers to produce extremely low
dropout, UltraFast™ transient response regulators. These
circuits achieve superior regulator bandwidth and transient load performance, and completely eliminate expensive tantalum or bulk electrolytic capacitors in the most
demanding microprocessor applications. For example, a
200MHz Pentium® processor can operate with only the
twenty-four 1µF ceramic capacitors that Intel already
AN84-108
www.BDTIC.com/Linear
Application Note 84
VIN 12V
* FOR T ≤ 45°C:
C6 = 24 × 1µF Y5V
CERAMIC SURFACE
MOUNT CAPACITORS.
FOR T > 45°C:
C6 = 24 × 1µF X7R
CERAMIC SURFACE
MOUNT CAPACITORS.
PLACE C6 IN THE
MICROPROCESSOR
SOCKET CAVITY
LT1575-3.5
1
SHDN
2
VIN
C2
1µF 3
GND
4
OUT
IPOS
INEG
GATE
COMP
C3
10pF
VIN
5V
C5
330µF
8
+
7
6
5
R2
5Ω
R1
7.5k
C4
1000pF
Q1
IRFZ24
VOUT
3.5V
5A
50mV/DIV
2A/DIV
0
C6*
24µF
GND
I = 0.2A to 5A
200µs/DIV
1575/77 TA01
Figure 219. UltraFast Transient Response 5V to 3.3V,
Low Dropout Regulator
additional system cost, provides on/off control and can
provide overvoltage protection or thermal shutdown with
the addition of a few simple external components. The
LT1575 is available in 8-pin SO or PDIP and the LT1577 is
available in 16-pin narrow-body SO.
UltraFast 5V to 3.3V Low Dropout Regulator
Figure 219 shows the basic regulator control circuit. The
input voltage is a standard 5V “silver box” and the output
voltage is set to 3.5V, the Pentium P54 VRE microprocessor supply voltage. The typical maximum output current is
about 5A in most Pentium microprocessor applications.
The output capacitor network consists of only twenty-four
inexpensive 1µF ceramic, surface mount capacitors. Proper
layout of this decoupling network is critical to proper
operation of this circuit. Consult Linear Technology Application Note 69: LT1575 UltraFast Linear Controller Makes
Fast Transient Response Power Supplies, for details on
board layout.
The photo in Figure 220 shows the transient response
performance for an output load current step of 0.2A to 5A.
The main loop compensation in Figure 219’s regulator
circuit is provided by R1 and C4 at the COMP pin. Capacitor C3 introduces a high frequency pole and provides
adequate gain margin beyond the unity-gain crossover
frequency of 1MHz. This compensation network limits
overshoot/undershoot to 50mV under worst-case load
transient conditions. With a 1% specified worst-case
output voltage tolerance, the 100mV output voltage error
Figure 220. Transient Response for
0.2A–5A Output Load Step
budget for a P54 VRE microprocessor is easily met with
production margin to spare. All bulk tantalum/electrolytic
capacitors are completely eliminated.
The discrete N-channel MOSFET chosen is a low cost
International Rectifier IRFZ24 or equivalent. The input
capacitance is approximately 1000pF with VDS = 1V. The
specified on-resistance is 0.1Ω at room temperature and
about 0.15Ω at 125°C. At 7A output current, the dropout
voltage is only 1.05V. This eases the restriction on local
input decoupling capacitor requirements because significant droop in the typical 5V input supply voltage is
permitted before dropout voltage operation is reached.
(Note that 5V supply tolerance restrictions are typically
limited by a ±5% tolerance so that 5V logic systems will
operate correctly.) However, a simple LC input filter can
eliminate the need for large input bulk capacitance at the
regulator 5V supply for additional system cost savings.
Figure 221 shows a more complete system configuration
that incorporates current limiting and current limit timeout with latch-off. Current limit is incorporated for no
additional system cost by manufacturing the current limit
resistor from a Kelvin-sensed section of PC board trace. In
this example, current limit is set to 7A. A capacitor from the
SHDN pin to ground sets a fault condition time-out period
that latches off the drive to the external MOSFET if the
time-out period is exceeded. The regulator is reset by
pulling the SHDN pin low. The output voltage in this
application is set to 3.3V. The ±5% tolerance permitted in
3.3V systems translates to a ±165mV output-voltage
www.BDTIC.com/Linear
AN84-109
Application Note 84
tolerance. This permits a 50% reduction in the number of
ceramic capacitors required from twenty-four to twelve.
Loop compensation is adjusted accordingly.
2.8V output. This circuit provides all the power requirements for a split-plane system: 3.3V for the logic supply
and 2.8V for the processor-core supply. Note that both
SHDN pins are tied to a common time-out capacitor. If
either or both regulators encounter a fault condition, both
regulator sections are latched off after the time-out period
is exceeded.
Figure 222 shows an application circuit using the LT1577,
a dual regulator. All functions for each regulator are
identical to those of the LT1575. One section is configured
for a 3.3V output and the other section is configured for a
LT1575-3.3
12V
1
Q2
VN2222L
RESET
C1
1µF
2
SHDN
C2
VIN
1µF 3
GND
4
OUT
IPOS
INEG
GATE
COMP
8
5V
R3*
0.007Ω
7
6
5
+
R2
5Ω
*R3 IS MADE FROM
“FREE” PC BOARD
TRACE
**C6 = 12 × 1µF X7R
CERAMIC SURFACE
MOUNT CAPACITORS.
C3
10pF
PLACE C6 IN THE
MICROPROCESSOR
SOCKET CAVITY
C5
330µF
Q1
IRFZ24
R1
3.9k
C4
1500pF
VOUT
3.3V
5A
C6**
12µF
GND
1575/77 TA12
Figure 221. 5V to 3.3V Regulator
FAULT RESET
INPUT
5V
C1
330µF
6.3V
+
+
1/2 LT1577
1/2 LT1577
1
2
12V
3
C3
0.33µF
4
SHDN
IPOS
VIN
INEG
GND
GATE
FB
COMP
16
5
15
6
14
13
C5
10pF
C4
0.1µF
R7
R3
2.1k
1.21k
R1
3.9Ω
R2
3.9k
C6
1500pF
7
Q1
IRFZ24
8
SHDN
IPOS
VIN
INEG
GND
GATE
FB
COMP
VI/O
3.3V
12
11
10
R5
3.9Ω
9
R6
7.5k
C7
10pF
C9 TO
C20*
1µF
R4
1.21k
R8
1.6k
C8
1000pF
*X7R CERAMIC 0805 CASE
Figure 222. LT1577 Dual Regulator for Split-Plane Systems
AN84-110
C2
330µF
6.3V
www.BDTIC.com/Linear
Q2
IRFZ24
VCORE
2.8V
C21 TO
C44*
1µF
AN69 F06
Application Note 84
LT1579 BATTERY-BACKUP REGULATOR PROVIDES
UNINTERRUPTIBLE POWER
by Todd Owen
A B
C
D
E
6V
VIN1
Introduction
5V
6V
Designed for a multitude of applications, the LT1579 is a
dual input, single output, low dropout regulator that
provides an uninterruptible output voltage from two independent input voltage sources on a priority basis. All
power supplied to the load is drawn from the primary input
(VIN1) until the device senses that the primary source is
failing. At this point, the LT1579 smoothly switches from
the primary input to the secondary input (VIN2) to maintain
output regulation. The LT1579 is capable of providing
300mA from either input at a dropout voltage of 0.4V. Total
quiescent current is 50µA: 45µA from the primary input
source, 2µA from the secondary input source, and an
additional 3µA from the higher voltage of the two.
VIN2
5V
VOUT 5V
4.8V
100mA
IIN1
0
100mA
IIN2
0
Circuit Examples
1
LB01
The basic application of the LT1579 is shown in Figure
223. It uses two independent voltage sources for the
inputs. These voltage sources may be batteries, wall
adapters or any other DC source. The low-battery comparators are configured to give a low output if either input
voltage drops below 5.5V. The trip points can be adjusted
by changing the values of the divider resistors (R1 and R2
for LB1, R3 and R4 for LB2). All logic outputs (LBO1,
0
1
BACKUP
0
1
LB02
0
1
DROPOUT
0
1579_04.eps
Figure 224. Basic Application Timing Diagram
+
1µF
6V
R1
2.7M
R2
1M
6V
100k
100k
5V
300mA
+
4.7µF
LBO1
LBO2
R3
2.7M
DROPOUT
SS
LBI2
R4
1M
100k
BACKUP
VIN2
1µF
100k
LT1579-5
LBI1
+
OUT
VIN1
SHDN
BIASCOMP
GND
TO POWER MANAGEMENT
0.01µF
1579_03.eps
Figure 223. LT1579 Basic Application
www.BDTIC.com/Linear
AN84-111
Application Note 84
LBO2, BACKUP and DROPOUT) are open-collector outputs that require an external pull-up resistor. They are
capable of sinking 20µA at a maximum output voltage of
0.32V, which is useful for driving both CMOS and TTL
logic families. For driving LED’s, all logic outputs can sink
5mA at a maximum output voltage of 1.2V.
Figure 224 is the timing diagram for the basic circuit. No
time scale is shown for the timing diagram because actual
discharge rates are a function of the load current and the
type of batteries used. The timing diagram is meant as a
tool to help in understanding the LT1579’s basic operation.
Five milestones are noted on the timing diagram. Time A
is where the primary input voltage drops enough to trip the
low-battery detector, LB1. The trip threshold for LB1 is set
at 5.5V, slightly above the dropout voltage of the primary
input. At time B, the BACKUP flag goes low, signaling the
beginning of the transition from the primary source to the
secondary source. Between times B and C, the input
current makes a smooth transition from VIN1 to VIN2. By
time C, the primary battery has exhausted most of its
useful charge. The primary input will still deliver a small
amount of current to the load, diminishing as the primary
input voltage drops. By time D, the secondary battery has
dropped to a low enough voltage to trip the second lowbattery detector, LB2. The trip threshold for LB2 is also set
at 5.5V, slightly above where the secondary input reaches
dropout. At time E, both inputs are low enough to cause the
LT1579 to enter dropout, with the DROPOUT flag signaling the impending loss of output regulation.
Some interesting things can be noted on the timing
diagram. The amount of current available from a given
input is determined by the input/output voltage differential. As the primary voltage drops, the amount of current
drawn from the input also drops, slowing discharge of the
battery. Dropout-detection circuitry will maintain the maximum current draw from the input for the given input/
output voltage differential, based on the impedance of the
pass transistor. In the case shown, this causes the current
drawn from the primary to approach zero, although it
OUT
VIN1
C1
1µF
IN1
R2
2.7M
R1
1M
R10
1M
LBI1
D2
D1
R3
1M
BACKUP
LBO1
R4
10M
C3
4.7µF
DROPOUT
VOUT
5V/300mA
MAIN GOOD
NC
SS
VIN2
D3
IN2
C2
1µF
R5
1M
R6
2.7M
LT1579-5
LBI2
R7
1M
R8
330k
BIASCOMP
C5
0.1µF
D4
5.1V
1N751A
C4
0.01µF
LBO2
VCC
1/4
74C02
1/4
74C02
SHDN
GND
GND
1/4
74C02
RESET
R9
1.5M
D1 TO D3 = 1N4148
Figure 225. Added SR Latch Shuts the LT1579 Off when Both Low-Battery Detectors are Tripped
AN84-112
www.BDTIC.com/Linear
1579_05.eps
Application Note 84
never reaches that point. Note that the primary begins to
supply significant current again when the secondary input
drops low enough to cause a loss in output regulation. This
occurs because the input/output voltage differential of the
primary input increases as the output voltage drops. The
LT1579 will automatically maximize the power drawn
from the inputs to maintain the highest possible output
voltage.
A final circuit example is shown in Figure 225. This circuit
has a few notable changes from the basic application.
First, the Secondary Select pin is connected directly to
LBO1. When the primary input voltage drops below the
threshold level for LB1, the comparator output will pull the
Secondary Select pin low. This forces the device to switch
completely over to the secondary input, limiting the discharge voltage of the cells. Second, the logic gates used
form an SR latch. When both batteries are below the
threshold level for their respective comparators, the latch
will be set, forcing the part into shutdown. The latch is
reset by pulling up on the RESET node, allowing the part
to come out of shutdown.
The series resistance of a battery can cause its terminal
voltage to rise as its current decreases. This effect can
reset the low-battery detector and cause the LT1579 to
oscillate between the primary and secondary inputs. To
combat this, the low-battery comparators have up to
18mV of built-in hysteresis at the input to the comparator
(LBI1, LBI2). The hysteresis is determined by the amount
of load current on the comparator output. At no load, the
comparator hysteresis is zero, increasing to a maximum
of 18mV for load currents above 20µA. For the pull-up
resistor shown, load current on the output of the comparator is 5µA, so hysteresis will be 5mV. With the values
shown for resistor divider R2/R3, this translates to 19mV
of hysteresis at the primary input of the LT1579. Additional
hysteresis can be added by connecting D1 and R4. The
values shown will give an additional 200mV of hysteresis.
When LBO1 and LBO2 are high impedance and either input
is greater than 6.5V, the logic-flag voltages can be above
the maximum voltage rating. Internal clamps on the logic
flags limit the output voltage to approximately 6.5V and
the pull-up resistor values shown will limit the current into
the logic flags to less than the maximum current rating.
Conclusion
The LT1579 can provide a continuous regulated output
voltage to critical circuits from any of a number of different
input sources. It will provide up to 300mA of output
current at a dropout voltage of 0.4V. Should the primary
input fail, the device switches seamlessly to the secondary
input, maintaining output regulation. A single error amplifier
controls both output stages so regulation remains tight
regardless of which input is providing power. The LT1579
can handle instantaneous removal of either one of its
inputs without losing regulation. System power management is aided by two status flags, which provide information about which input is providing power and signal the
loss of output regulation. Two independent low-battery
comparators can be used to monitor input voltages. Also,
an external pin can be used to force the switch to the
secondary input. Total quiescent current of the LT1579 is
50µA, dropping to a mere 7µA in its low power shutdown
state. Internal circuitry guards against a number of fault
conditions, including current limit, thermal limit and reverse voltages, protecting sensitive circuitry and inputs.
Whether the application is simple or complex, the LT1579
is truly a “smart” regulator.
www.BDTIC.com/Linear
AN84-113
Application Note 84
is reduced to keep the adapter current within specified
levels.
Battery Chargers
THE LT1511 3A BATTERY CHARGER CHARGES
ALL BATTERY TYPES, INCLUDING LITHIUM-ION
by Chiawei Liao
The LT1511 can charge batteries ranging from 1V to 20V.
Ground sensing of current is not required and the battery’s
negative terminal can be tied directly to ground.
The LT1511 current mode PWM battery charger is the
simplest, most efficient solution for fast charging modern
rechargeable batteries, including lithium-ion (Li-Ion),
nickel-metal-hydride (NiMH) and nickel-cadmium (NiCd)
that require constant-current and/or constant-voltage
charging. The internal switch is capable of delivering 3A
DC current (4A peak current). Full charging current can be
programmed by resistors or by a DAC to within 5%, and
the trickle charge current can be programmed to 10%
accuracy. With 0.5% reference voltage accuracy, the
LT1511 meets the critical constant-voltage charging
requirement for lithium cells.
LT1511 Applications
Lithium-Ion Charging
The 3A lithium battery charger (Figure 226) charges
lithium-ion batteries at a constant 3A until the battery
voltage reaches a limit set by R3 and R4. The charger will
then automatically go into a constant-voltage mode, with
the current decreasing to zero over time as the battery
reaches full charge. This is the normal regimen for lithiumion charging, with the charger holding the battery at “float”
voltage indefinitely. In this case no external sensing of full
charge is needed.
The LT1511 is equipped with a voltage-control loop to
control charging voltage and a current-control loop to
control charging current. A third control loop is provided
to regulate the current drawn from the AC adapter. This
allows simultaneous equipment operation and battery
charging without overloading the adapter. Charging current
Current though the R3/R4 divider is set at 15µA to minimize battery drain when the charger is off. The input
current to the OVP pin is 3nA and this error can be
neglected.
R7
500Ω
C1
1µF
CLN
VCC
SW
D1
MBR340
0.47µF
L1**
10µH
BOOST
LT1511
D2
1N4148
+
+
10µF
VIN (ADAPTER INPUT)
11V TO 25V
RS4
ADAPTER CURRENT SENSE
TO MAIN SYSTEM POWER
CIN*
10µF
R5†
UNDERVOLTAGE LOCKOUT
UV
COMP1
200pF
PROG
SPIN
OVP SENSE
NOTE: COMPLETE LITHIUM-ION CHARGER,
NO TERMINATION REQUIRED. RS4, R7
AND C1 ARE OPTIONAL FOR IIN LIMITING
*TOKIN 25V CERAMIC SURFACE MOUNT
**10µH COILTRONICS CTX10-4
†
CONSULT LT1511 DATA SHEET FOR R5 VALUE
DIN
CLP
GND
RS3
200Ω
1%
VC
BAT
RS2
200Ω
1%
RS1
0.033Ω
BATTERY CURRENT
SENSE
50pF
300Ω
CPROG
1µF
1k
RPROG
4.93k
1%
R6
5k
0.33µF
R3
390k
0.25%
BATTERY
VOLTAGE SENSE
+
COUT
22µF
TANT
R4
162k
0.25%
Figure 226. 3 Amp Lithium-Ion Battery Charger
AN84-114
www.BDTIC.com/Linear
+
4.2V
+
4.2V
VBAT
2 Li-Ion
Application Note 84
With divider current set at 15µA, R4 = 2.465/15mA = 162k
and
R1 =
VOUT – 1.245
1.245 + (3 × 10–7)
R2
where VOUT = battery float voltage
Lithium-ion batteries typically require float-voltage accuracy of 1% to 2%. The accuracy of the LT1511 OVP voltage
is ±0.5% at 25°C and ±1% over full temperature. This
leads to the possibility that very accurate (0.1%) resistors
might be needed for R3 and R4. Actually, the temperature
of the LT1511 will rarely exceed 50°C in float mode
because charging currents have tapered off to a low level,
so 0.25% accuracy resistors will normally provide the
required level of overall accuracy.
All battery chargers with fast charge rates require some
means to detect the full-charge state in the battery in order
to terminate the high charging current. NiCd batteries are
typically charged at high current until temperature rise or
battery voltage decrease is detected as an indication of
nearly full charge. The charging current is then reduced to
a much lower value and maintained as a constant trickle
charge. An intermediate “top off” current may be used for
a fixed time period to reduce 100% charge time.
Nickel-Cadmium and Nickel-Metal-Hydride Charging
NiMH batteries are similar in chemistry to NiCd but have
two differences related to charging. First, the inflection
characteristic in battery voltage as full charge is approached is not nearly as pronounced. This makes it more
difficult to use dV/dt as an indicator of full charge, and
temperature change is more often used, with a temperature sensor in the battery pack. Second, constant trickle
charge may not be recommended. Instead, a moderate
level of current is used on a pulse basis (1% to 5% duty
cycle) with the time-averaged value substituting for a
constant low trickle.
The circuit in the 3A lithium battery charger (Figure 226)
can be modified as shown in Figure 227 to charge NiCd or
NiMH batteries. Two-level charging is needed; 2A when Q1
is on and 200mA when Q1 is off. For 2A full current, the
current sense resistor (RS1) should be increased to 0.05Ω,
so that enough signal (10mV) will be across RS1at 0.2A
trickle charge to keep charging current accurate.
If overvoltage protection is needed, R3 and R4 should be
calculated according to the procedure described in lithiumion charging section. The OVP pin should be grounded if
not used. When a microprocessor DAC output is used to
control charging current, it must be capable of sinking
current at a compliance up to 2.5V if connected directly to
the PROG pin.
For a two-level charger, R1 and R2 are found from
R1 =
(2.465)(4000)
ILOW
R2 =
(2.465)(4000)
IHI − ILOW
LT1511
PROG
300Ω
RPROG
4.7k
5V
0V
CPROG
1µF
Q1
VN2222
PWM
IBAT = (DC)(3A)
Figure 227. 2-Step Charging
www.BDTIC.com/Linear
AN84-115
Application Note 84
LT1512/LT1513 BATTERY CHARGERS OPERATE
WITH INPUT VOLTAGES ABOVE OR BELOW
THE BATTERY VOLTAGE
by Bob Essaff
Applications
The LT1512 and LT1513 are specifically optimized to use
the SEPIC converter topology, which is shown in Figure
228’s typical application. The SEPIC (single-ended primary inductance converter) topology has several advantages for battery-charging applications. It will operate with
input voltages above or below the battery voltage, has no
path for battery discharge when turned off, and eliminates
the snubber losses of flyback designs. It also has a current
sense point that is ground referred and need not be
connected directly to the battery. The two inductors shown
are actually two identical windings on one inductor core,
although two separate inductors can be used.
Introduction
The LT1512 and LT1513 form a unique family of constantcurrent, constant-voltage battery chargers that can charge
batteries from input voltages above or below the battery
voltage. This feature can help simplify system design and
add product flexibility by allowing battery charging from
multiple sources, such as a wall adapter, a 12V automotive
system or a 5V power supply, all with the same circuit. The
constant-current, constant-voltage architecture makes the
LT1512 and LT1513 well suited for charging NiCd, NiMH,
lead-acid or lithium-ion batteries.
The topology is essentially identical to a 1:1 transformerflyback circuit except for the addition of capacitor C2,
which forces identical AC voltages across both windings.
This capacitor performs three tasks: it eliminates the
power loss and voltage spikes usually caused by a flybackconverter’s leakage inductance; it forces the input current
and the current in resistor R3 to be a triangle wave riding
on top of a DC component instead of forming a large
amplitude square wave; and it eliminates the voltage
spikes across the output diode when the switch turns on.
Both devices are current mode switching regulators that
operate at a fixed frequency of 500kHz. Product features
include a ±1% reference-voltage tolerance, 2.7V minimum
input voltage, easy external synchronization and 12µA
supply current in shutdown mode. The LT1512 and LT1513
also include low loss on-chip power switches rated for 1.5
amps and 3 amps respectively. High frequency switching
allows the use of small surface mount inductors and
capacitors, and the battery can be directly grounded.
L1A*
•
+
C2**
D1
1µF
×2
MBRS130LT3
5
C3
22µF
25V
VIN
CHARGE
VSW
8
0.5A
L1B*
LT1512
4
SHUTDOWN
S/S
GND
6
VFB
7
VC
IFB
1
3
C5
0.1µF
2
R1†
•
R2†
R4
24Ω
C4
0.1µF
2.4
R3
0.2Ω
+
C1
22µF
25V
CURRENT (A)
WALL
ADAPTER
INPUT
When the battery is below its float voltage, set by R1 and
R2, the charger is in the constant-current mode. The
suggested value for R2 is 12.4k. R1 is calculated from:
2.2
INDUCTOR = 33µH
2.0
1.8
SINGLE LITHIUM
CELL (4.1V)
1.6
LT1513
DOUBLE LITHIUM
CELL (8.2V)
SINGLE LITHIUM
CELL (4.1V)
1.4
1.2
1.0
0.8
0.6
0.4
LT1512
DOUBLE LITHIUM
CELL (8.2V)
0.2
*L1A, L1B ARE TWO 33µH WINDINGS ON A
COMMON CORE: COILTRONICS CTX33-3
**AVX1206Y2105KAT1A
†
TO CALCULATE R1, R2 VALUES, SEE TEXT
Figure 228. Battery Charger with 0.5A Output Current
AN84-116
0
0
5
10
15
20
INPUT VOLTAGE (V)
25
30
Figure 229. Maximum Charging Current
www.BDTIC.com/Linear
Application Note 84
Programming the Charge Current
VOUT – 1.245
R1 =
1.245 + (3 × 10–7)
R2
where VOUT = battery float voltage
Charging current in the battery, which also flows through
R3, develops a voltage on the IFB pin. The IFB pin’s 100mV
sense voltage sets the programmed charging current to
ICHG = 100mV/R3. The RC filter formed by R4 and C4
smoothes the signal presented to the IFB pin.
Charging current remains constant until the battery reaches
its float voltage, at which point the LT1512/LT1513 changes
to the constant-voltage mode. In this mode, the charging
current will taper off as required to keep the battery at its
float voltage. The circuit’s maximum input voltage is
partly determined by the battery voltage. When the switch
is off, the voltage on the VSW pin is equal to the input
voltage, which is stored across C2, plus the battery voltage. Both the LT1512 and LT1513 have a maximum input
voltage rating of 30V and a maximum rated switch voltage
of 35V, thereby limiting input voltage to 30V or 35V minus
the battery voltage, whichever is less.
Figure 229 shows the maximum available charging current for a single-cell or double-cell lithium battery pack.
Note that the actual programmed charging current will be
independent of the input voltage if it does not exceed the
values shown.
VIN
•
+
L1A
C2
1µF
×2
5
22µF
25V
VIN
CHARGE
VSW
SHUTDOWN
S/S
GND
6
VC
7
1
Off-State Leakage
Charging can be terminated by placing the LT1512/LT1513
into shutdown mode. If the battery remains connected to
the charger when in the off state, two leakage paths that
load the battery must be considered.
The first is the 100µA resistor-divider feedback current
that flows through R1 and R2. This current can be eliminated with the addition of a FET, Q1, between R1 and the
R2/VFB junction, as shown in Figure 231. In this example,
pulling the charge/shutdown input above 3.75V will activate charging and turn on Q1, whereas driving the charge/
shutdown input below 0.6V will shut down the LT1512/
LT1513 and turn off Q1.
MBRS130LT3
R1
L1B
VFB
IFB
2
0.1µF
0.47µF
R1
+
•
24Ω
3
BATTERY
CHARGE
8
LT1512
4
As mentioned earlier, charging current is set by R3, where
ICHG = 100mV/R3. The charge current is programmed by
changing the effective value of R3, as shown in Figure 230.
In the low charge mode, Q1 is off, setting charge current
to ICHG LOW = 100mV/R3A, or 100mV/2Ω = 50mA. In the
high-charge mode, Q1 is on, and charge current is ICHG HI
= 100mV/R3A + 100mV/(R3B + Q1’s RDS(ON)), or 100mV/
2Ω + 100mV/(0.24Ω + 0.04Ω)) = 50mA + 357mA =
407mA. Note that Q1’s RDS(ON) is a factor in the highcharge mode, requiring the use of a low RDS(ON) FET.
R3B
0.24Ω
HI CHARGE
C1
22µF
25V
SHUTDOWN
S/S
LT1512/LT1513
R2
Q1
VN2222
VFB
GND
R3A
2Ω
Q1
R2
LOW CHARGE
Q1 = SILICONIX Si9410DY
C2 = AVX1206Y2105KAT1A
Figure 230. 50mA/400mA Programmable Battery Charger
Figure 231. ShutdownControlled Disconnect
www.BDTIC.com/Linear
AN84-117
Application Note 84
The second leakage path to consider is in the output diode,
D1 (Figure 228). When the charger is in the off state, the
output diode sees a reverse voltage equal to the battery
voltage. Though the Schottky diode reverse leakage may
typically be only 10µA, its guaranteed specifications are
much worse, up to 1mA. One solution is to change the
output diode to an ultra-fast silicon diode, such as an
MUR-110. The higher forward voltage of the silicon diode
will decrease the circuit’s efficiency, but these diodes have
reverse leakage specifications below 5µA.
Li-Ion BATTERY CHARGER DOES NOT REQUIRE
PRECISION RESISTORS
by LTC Applications Staff
The charger selected for this example is the LT1510 and
the number of Li-Ion cells in the battery is three. Select a
value for R4 (20k) and calculate the values for resistors
R1, R2 and R3 using the equations in Figure 232. K is the
relative change required for a circuit with all its tolerances
in one direction. For example, in the case of a 0.5%
reference and two 1% resistors, the total tolerance is
2.5%. In order to bring it back to 1.2%, the percentage
change required is 2.5% – 1.2% = 1.3% and K = 0.013.
In constant-voltage mode charging, a Li-Ion cell requires
4.1V ±50mV. This 1.2% tolerance is tight. In a regulation
loop where a voltage divider is compared against a reference, the accuracy is achieved by selecting a 0.7% reference and a voltage divider with 0.25% tolerance resistors.
Unfortunately, 0.25% precision resistors cost three times
as much as 1% resistors and have very long lead times.
One solution for moderate volume production involves
adding two 1% resistors and two jumpers to the charger
circuit, as shown in Figure 232. The jumpers are removed
as necessary to bring the constant voltage to the required
accuracy of 1.2%.
The jumpers J1 or J2 need to be opened based on the
following:
If VOUT is K/2 below nominal, remove J1.
If VOUT is K/2 above nominal, remove J2.
D3
1N5819
C1
D1
0.22µF 1N5819
VIN
SW
LT1510
CONSTANT VOLTAGE/
CONSTANT CURRENT
BATTERY CHARGER
–
VREF
2.465V
+
–
+
BOOST
R1
R2
R3
L1**
33µH
J1
J2
+ 3-CELL
–
Li-ION
BATTERY
D2
1N914
PROG
1µF
LT1510
GND
VC
16V TO 28V
–+
CIN*
10µF
VOUT = 12.3V
BAT
OVP
VCC
0.1µF
300Ω
3.83k
1k
OVP
SENSE
BAT
+
R4
COUT
22µF
TANT
3-CELL
Li-ION
BATTERY
+
J1
R1
R2
R1 = R4 × VOUT – VREF
VREF
R2 = (R1+R2) × K
R3 =
R4 × K
V
1 – (1 – K) REF
VOUT
Q3
VN2222
*TOKIN OR MARCON CERAMIC SURFACE MOUNT
** COILTRONICS CTX33-2
AN84-118
J2
R3
R4
TO CALCULATE K, SEE TEXT
Figure 232. R2, R3, J1 and J2 Eliminate
the Need for Precision Resistors
R5
100k
Figure 233. 3-Cell Li-Ion Charger without Precision Resistors
www.BDTIC.com/Linear
Application Note 84
The following values were calculated: R1 = 20k, R2 =
324Ω, R3 = 80.6Ω and R4 = 4.99k.
The voltage below which J1 should be opened is 12.34V
– 1.3%/2 = 12.22V.
The complete schematic can be seen in Figure 234. Q3 is
off when the charger is not powered, preventing current
drain from the battery through the voltage divider. R5, a
100k resistor, isolates the OVP pin from any high frequency noise on VIN. The charger in Figure 233 is programmed for 1.3A constant current.
The voltage above which J2 should be opened is 12.34V
+ 1.3%/2 = 12.42V.
LT1510 CHARGER WITH –∆V TERMINATION
by LTC Applications Staff
Any portable equipment that requires fast charge needs
proper charge termination. Commonly, a LT1510 constant-voltage, constant-current type charger controlled by
a microcontroller is used. Sometimes, however, a microcontroller is not available or is not suitable for fast-charge
termination.
When fast charging NiCd batteries with constant current,
the internal battery temperature rises toward the end of the
charge. Since the temperature coefficient of NiCd is negative, the temperature rise causes the battery voltage to
drop. The drop can be detected and used for termination
(called –∆V termination). The circuit in Figure 234 is a
solution for a 3-cell (Panasonic P140-SCR) NiCd battery
charger with –∆V termination.
U1 in Figure 234 is programmed by resistor R2 for a
conservative charge current of 0.8A, which is 0.57C.
Typical fast-charge current is 1C. (The boldfaced C represents a normalization concept used in the battery industry.
A C rate of 1 is equal to the capacity of the cell in amperehours, divided by 1 hour. Since the capacity of the P140SCR is 1.4 ampere-hours, C is 1.4 amperes.)
To determine the voltage droop rate, the battery was
connected to an LT1510 charger circuit programmed for
a 0.8A constant-current. The data was plotted as voltage
versus time and the results are shown in Figure 235. The
voltage slope is calculated to be –0.6mV/s. After the
battery voltage dropped 300mV from the peak of 4.93V
(100mV per cell), the charger was disabled.
At the heart of the circuit in Figure 234 is U3, a sample-andhold IC (LF398). For every clock pulse at pin 8, the output
of U3 (pin 5) updates to the input level on pin 3. When the
battery voltage drops, the input to U3 also drops. If the
update step at the output of U3 is sufficiently negative, U2B
latches in the high state and Q1 turns on. Q1 terminates the
charge by pulling down the LT1510’s VC pin, and thereby
disabling it.
U2A and the associated passive components smooth,
amplify and level shift the battery voltage. The timer (U4)
updates the hold capacitor (C8) every fifteen seconds. The
timer signal stays high for 7ms, sufficient time for the hold
capacitor to be charged to the input level. U2B and the
associated parts form a latch that requires a momentary
negative voltage at pin 6 to change state. R15 supplies the
negative feedback and Q2, R16, R17 and C10 reset the
latch on turn-on.
U3’s output voltage droops at a rate proportional to the
hold capacitor’s internal leakage and the leakage current at
pin 6 (10pA typical). This droop is very low and does not
affect the operation of the circuit.
The minimum negative battery voltage slope required to
trigger termination (–dV/dT) is 0.3mV/s. It can be calculated from:
–dV/dT = VTRIG/(TCLK × GU2A) where:
VTRIG is the trigger voltage of U2B,
VTRIG = VREF × R12/(R11 + R12) = 5 × 1/101 = 49.5mV
VREF = 5V
TCLK is the clock period, 15 seconds,
GU2A is the gain of the first stage, = R8/(R4 || R5) = 11
www.BDTIC.com/Linear
AN84-119
Application Note 84
12V
2
15
VCC1
VCC2 14
13
PROG
SW
C1
0.22µF
3 BOOST
U1
LT1510
4
GND
CR2
1N5819
L1*
30µH
CR1
1N914
6
VC
SENSE
GND BAT
12
CR3
1N5819
C3 +
R1 1µF
300
R2
6.19k
C4
R3 0.1µF
1k
C7
0.1µF
R7
10
5VREF
N/C
U3
LF398
2
R4
10k
C2†
10µF
R8
100k
2
–
1
5VREF
C9
0.1µF
5
–
–
8
U2A
3 LT1013
+
4
R6
100k
R14
10k
30k
OFFSET
R5
100k
11
R9
30.1k
1
+
3
R11
100k
6
LT1029CZ
U2B
5 LT1013
OUTPUT
+
C10 +
22µF
–
+
INPUT
7
R15
100k
R16
100k
**1, 7, 8, 9, 10, 16
+
C5
22µF
25V
B1***
C6
0.1µF
CLK
CLK
R19
100k
C12
16
2
15
4
LOGIC
REFERENCE
U4
CD45368
4
13
0.01µF
5
R20
100k
6
11
7
10
8
9
HOLD
CAPACITOR
6
C8
1µF
ECQV1HIOSJL
PANASONIC
R10
30.1k
14
Q1
2N3904
12
Q2
2N3904
R17
100k
7
1
3
R13
10k
LOGIC
R18, 100k
C11
0.22µF
150Ω
8
R12
1k
R21
10k
NOTES: *
**
***
†
L1 IS COILTRONICS CTX 33-2
SOLDER TO GROUND PLANE FOR HEAT DISSIPATION
B1 IS A NiCd 3 CELL PANASONIC P140-SCR
C2 IS A TOKIN OR MARKON CERAMIC SURFACE MOUNT
Figure 234. Schematic Diagram: 3-Cell NiCd Charger with –∆V Termination
The circuit in Figure 234 was built and connected to a
system that discharges the battery to 3V after termination,
at constant current of 0.8A. Once the battery drops to 3V,
the system reenables charging, and thus the complete
system repeats charge/discharge cycles indefinitely. The
duration of 70 charge discharge cycles was recorded. The
following is condensed data from the test:
the time of termination is very consistent because the
discharge time at constant current is a better measure of
charge level than charge time. A secondary termination
method, such as time, battery temperature, or the like, is
also recommended.
5.0
NEGATIVE
VOLTAGE SLOPE
4.9
2:00:55 Hours
2. Standard Deviation of Charge Time:
3. Average Discharge Time:
5:37 Minutes
1:59:14 Hours
4. Standard Deviation of Discharge Time:
48 Seconds.
The ratio of standard deviation of charge time to average
charge time proves that the charger has good repeatability.
However, the ratio of standard deviation of discharge time
to average discharge time shows that the charge level at
AN84-120
BATTERY VOLTAGE (V)
1. Average Charge Time:
END OF
0.8A CHARGE
4.8
4.7
4.6
4.5
4.4
4.3
4.2
7:30
15:00
22:30
TIME (MIN)
30:00
Figure 235. Voltage-Droop Rate, 3-Cell NiCd Battery
www.BDTIC.com/Linear
Application Note 84
CONSTANT-VOLTAGE LOAD BOX
FOR BATTERY SIMULATION
by Jon Dutra
Linear Technology has developed many new switcherbased battery charger ICs. Testing accuracy, regulation
and efficiency in the lab with a battery load is inconvenient
because the terminal voltage of a battery constantly changes
as it is being charged. If much testing is to be done, a large
supply of dead batteries will be needed, since one set of
cells can quickly become overcharged. This article
describes an active load circuit that can be used to simulate a battery in any state of charge. The battery simulator
provides a constant-voltage load for a battery-charging
circuit, independent of applied charging current. The
simulator’s impedance is less than 500mΩ at all reasonable input frequencies. Best of all, the simulator can never
be overcharged, allowing long-term testing and debugging of a charger system without the possibility of battery
damage.
Circuit Operation
The simulator (Figure 236) uses an LT1211 high speed,
single-supply op amp to drive the base of a high gain PNP
transistor-stage active load. Power for the LT1211—a
portion of the charging current—is supplied through a
diode so the op amp and reference can survive brief
periods of zero charging current. The op amp is configured
for a DC gain of four, so the voltage on its noninverting
input is one fourth of the voltage that the load box is set to.
With S1 open, the load-voltage adjust range will be from
10V to 20V, and with S1 closed it will be approximately
3.5V–10V. Low voltage operation could be improved by
replacing the top LT1004-2.5 with an LT1004-1.2 and
reducing R1, the reference bias resistor, to 1k. The 510Ω
and 1.1k resistors are required for high frequency stability;
they suppress a 1MHz oscillation. The 1N5400 diode and
4-amp fuse protect the circuit from reverse voltages.
Results
The battery simulator circuit has been tested “swallowing”
currents from 30mA to 3A with the output voltage essentially unchanged. When simulating a battery, the voltage
adjust can be increased until the charger thinks the battery
is fully charged and reduces the current into the simulator.
Conversely, as the voltage is adjusted down, the battery
charger may think the battery is becoming discharged and
increase the current into the simulator.
Figure 237 shows the circuit’s capacity for current absorption at two voltages, 5V and 15V, from 50mA to 3 amps.
R1 10k
0.033µF
1N5817 OR
BAT-85
+
5.0V OR 2.5V
100kΩ
10 TURN
POT
–
1/2
LT1211
+
LT1004-2.5
0.5Ω
5W
30k, 1%
100µF
25V
510Ω
5.3
15.5
15.4
5.2
15.3
Q1*
2N6667
+
0.033µF
LT1004-2.5
IN+
1.1kΩ
270µF
25V
2.5V OR 0V
4A FUSE
VOLTAGE (V)
10k, 1%
5.1
15.2
5V
5
15.1
15
15V
1N5400
S1
4.8
14.8
IN–
4.7
14.7
0
S1 CLOSED ≥ 0 TO 10V RANGE
S1 OPEN ≥ 10V TO 20V RANGE
ALL RESISTORS 5% UNLESS NOTED
4.9
14.9
0.5
1
1.5
2
CURRENT (A)
2.5
3
* Q1 DISSIPATES MOST OF THE POWER,
MOUNT ON AN ADEQUATE HEAT SINK
Figure 236. Schematic Diagram of the Battery Simulator
Figure 237. Current Absorption Capacity
of the Battery Simulator at 5V and 15V
www.BDTIC.com/Linear
AN84-121
Application Note 84
100
HIGH EFFICIENCY, LOW DROPOUT
LITHIUM-ION BATTERY CHARGER CHARGES
UP TO FIVE CELLS AT 4 AMPS OR MORE
by Fran Hoffart
VBATT = 16.8V
EFFICIENCY (%)
95
Introduction
VBATT = 12.8V
90
85
Rechargeable lithium batteries feature higher energy density per volume, higher energy density per weight and
higher voltage per cell than any of the competing battery
chemistries. For these reasons, manufacturers of portable
equipment are adopting the lithium-ion rechargeable battery as the battery of choice for high performance portable
equipment. Lighter weight and increased operating time
between charges are important features that customers
want and need from portable products.
VIN = 24V
80
0
1
2
3
4
CHARGE CURRENT (A)
5
Figure 239. Charger Efficiency for 3- and 4-Cell Applications
1620_02.eps
Higher Charge Currents
Paralleling cells, regardless of cell chemistry, requires
relatively high charge currents to bring the battery up to
full charge in a short period of time. When charging needs
exceed the 3A maximum rating of the LT1511 or LT1513,
the circuit shown in Figure 238 can provide much higher
current solutions, and very high efficiency. This circuit
uses the LTC1435 and LT1620 in a charger that delivers 4A
or more with exceptional efficiency and low dropout
voltage (Figures 238 and 239).
Increased demands from laptop computers have forced
manufacturers to use multiple cells in a combination of
series and parallel configurations. Paralleling cells increases the amount of current that can be drawn from the
battery and/or increases the operating time between
charges, but it also increases the current requirements of
the charger.
+VIN
R7
1.5M
13
VIN
2
RUN/SS
SHUTDOWN INPUT
(SD = 0V)
C12
0.1µF
R5
1k
C14
1000pF
C11
56pF
2
3
7
IPROG
IOUT
SENSE
1
5
IN+
GND
LT1620CMS8
4
IN–
PROG AVG VCC
8
C15
0.1µF
R6
4k
RPROG = 21k
FOR 4A
C10
100pF
3
7
16
SW
LTC1435CG
12
INT VCC
COSC
BOOST
ITH
–
SENSE
BG
PGND
8
5
6
C16
0.33µF
C18
0.1µF
C17, 0.01µF
TG
C1, C2
22µF ×2
35V TANT
Q1
RSENSE
0.02Ω
14
1
C13
0.033
C4
0.1µF
6
D2
VOSENS
+
L1
27µH
C5, 0.1µF
15
+
C6, 0.33µF
11
Q2
10
C8
100pF
SENSE+
SGND
D1
SFB
4
C7, 4.7µF
C9, 100pF
R3
(SEE TEXT)
L1 = CTX27-4, COILTRONICS
Q1, Q2 = Si4412DY, SILICONIX
D1, D2 = CMDSH-3, CENTRAL
R2, 0.1%
(SEE TEXT)
C1, C2 = 22µF,35V, AVX TPS SERIES
C3 = 22µF, 25V, AVX TPS SERIES
1620_01.eps
Figure 238. Complete Schematic of the High Efficiency, 4A, Constant-Voltage/Constant-Current
Charger Using All Surface Mount Components, with a Circuit Board Area of 1.5in2
AN84-122
IBATT
www.BDTIC.com/Linear
C3
22µF
Application Note 84
DROPOUT VOLTAGE (V)
2.0
9V), programmable soft start, logic-controlled micropower shutdown and a secondary feedback control pin.
Because external MOSFET switches are used, the maximum output load current is determined by the current
capabilities of the selected FETs.
VBATT = 16.8V
IPROG = 200µA
CONSTANT CURRENT
PROGRAMMED FOR 4A
1.5
1.0
0.5
The LTC1435 as a Battery Charger
The low dropout voltage, high current capability and high
efficiency of the LTC1435 switching regulator would seem
to make it an appropriate choice for high current battery
chargers, but it has several limitations. The absolute
maximum output voltage of 10 volts allows only two
series-connected lithium cells to be charged and the
output current is not readily programmable.
0
0
1
2
3
4
CHARGE CURRENT (A)
5
1620_03.eps
Figure 240. Charger Dropout Voltage vs Charge Current
The LT1435 Switching Regulator Controller
The LTC1435 is a step-down current mode switching
regulator controller designed to drive two external Nchannel power MOSFETs. Operating from input voltages
between 3.5V and 36V, this device includes a programmable switching frequency, synchronous rectification,
Burst Mode™ operation and a 99% maximum duty cycle
for low dropout voltage. Additional features include a 1%
tolerance output voltage (adjustable between 1.2V and
Introducing the LT1620
The LT1620 is an IC designed to be used with a current
mode PWM controller (such as the LTC1435 and similar
products) to increase the output voltage range and optimize the circuit for battery charging applications. Used
together, these two products overcome the voltage and
+VIN
VCC
RUN/SS
SHUTDOWN INPUT
TG
Q1
CSS
L1
LTC1435
IBATT
RSENSE
+
COSC
+
BG
Q2
R2
GND
VOSENS
INT VCC
ITH
SENSE
R3
PROGRAM
CONSTANT
VOLTAGE
C15
VCC
IN
AVG
LT1620
SENSE
PROG
R6
IOUT
RPROG
PROGRAM
CONSTANT
CURRENT
1620_04.eps
Figure 241. Simplified Diagram of the Constant-Voltage/Constant Current Charger
www.BDTIC.com/Linear
AN84-123
Application Note 84
INDUCTOR
CURRENT
80mV
CHARGE
CURRENT
IBATT
AVG PIN
RSENSE
EXTERNAL AVERAGING
CAPACITOR, C15
IN+
IN–
+5V
X1
≈800mV
R6
800mV
CAVG
2.5k
4.2V PROG
IPROG
RPROG
X10
CURRENT
SENSE
AMPLIFIER
gm
LT1620
40mV OFFSET
VCC
IOUT
SENSE
+5V
ITH
INT VCC
SENSE+
SENSE–
LTC1435
1620_05.eps
Figure 242. Simplified Digram of Constant-Current Control Loop
current programming limitations previously mentioned,
to produce a high current, high performance constantvoltage/constant-current battery charger for lithium-ion
and other battery types.
How They Work Together
To understand how the two parts work together, a brief
review of the LTC1435 operation is necessary. See Figure
241. During each cycle of operation, the series MOSFET
switch Q1 is turned on by the LTC1435 oscillator (Q2 is
off). This causes a current to begin ramping up in inductor
L1. When the current in L1 reaches a peak level determined
by the voltage at the ITH pin, Q1 is turned off and the
synchronous MOSFET Q2 is turned on, causing the current in L1 to ramp down to the level at which it started.
Thus, a sawtooth of inductor ripple current is generated,
with a peak level set by the voltage on the ITH pin. This
inductor current is sensed via an external, low value sense
resistor in series with the inductor and is used to drive the
LTC1435 internal current sense amplifier as the current
mode feedback signal. This current sense amplifier has a
maximum common mode voltage limit of 10V, which
limits the maximum output voltage to 10V.
Enter the LT1620. The LT1620 also contains a current
sense amplifier, which has a common mode range that
AN84-124
extends up to 28V. This amplifier is used to level shift the
differential sense voltage, which is riding on the battery
voltage, and reference it to the internal 5V VCC voltage
generated by the LTC1435. This level-shifted signal is
used to drive the LTC1435 current sense pins, thus
providing current mode feedback for the constant-voltage
feedback loop. This signal is also used to control the
constant output current feedback loop, as explained below.
Constant Charge Current
The LT1620 also provides a simple method of accurately
programming the constant-current output. Sinking an
adjustable current from the PROG pin to ground controls
the charge current from zero current to maximum current.
This program current can be derived from a variety of
sources, such as a single resistor to ground or the output
of a DAC.
The constant-current feedback loop operates as follows.
With a discharged battery connected to the charger, and
assuming that the battery voltage is less than the float
voltage programmed by R2 and R3, the error amplifier in
the LTC1435 begins pulling up on the ITH pin. This increases the peak inductor current in an effort to force the
battery voltage to be equal to the programmed voltage. By
www.BDTIC.com/Linear
Application Note 84
D3
+VIN
1M
Q3
2N3906
SHUTDOWN
D4
47k
CSS
VCC
RUN/SS
TG
Q1
L1
LTC1435
+
D3, D4 = 1N4148
1620_06.eps
Figure 243. Circuitry that Shuts Down the Charger when Input Power is Removed, Minimizing Reverse Battery Current Drain
limiting the voltage on the ITH pin, the peak inductor
current and the average output current can be controlled.
The ITH pin has an internal 2.4V clamp that sets the peak
inductor to its maximum level. This 2.4V clamp provides
some degree of current regulation, but the average battery
current will vary considerably as a result of dependence on
inductor ripple current and LTC1435 parameter variations. By adding the LT1620 to the circuit, the constant
charging current control performance is considerably
improved. As shown in Figure 242, the signal from the
current sense amplifier in the LT1620 is amplified by 10,
averaged by CAVG (C15 in Figure 238) and compared to the
voltage drop across R6. This voltage is developed by a
current, IPROG, flowing through R6. When the voltage at
the LT1620 AVG pin approaches the voltage on the PROG
pin, IOUT begins to pull the ITH pin of the LTC1435 down,
limiting the peak inductor current and completing the
constant-current feedback loop.
Complete Charger Circuit
The circuit shown in Figure 238 can charge up to five
series-connected lithium-ion cells at currents up to 4A.
Using low RDS(ON) MOSFET switches for the switch and
synchronous rectifier results in efficiency exceeding 95%
and allows all surface mount components to be used,
resulting in a design that occupies less than 1.5 in2 of
board space. This circuit operates at a switching frequency of 200kHz and is capable of up to 99% duty cycle;
it can operate over a very wide input voltage range, from
a minimum input of only 600mV greater than the battery
charging voltage to a maximum of 28V (limited by the
MOSFETs).
Constant-voltage charging with better than 1.2% accuracy
and constant-current charging with 7.5% accuracy provides almost ideal lithium-ion battery charging
conditions.
In battery charger designs, an important issue is reverse
battery drain current caused by the charger when the input
power is removed or the charger is shut down, or both. If
the battery will remain connected to the charger for
extended periods of time, it is important to minimize this
reverse drain current to prevent discharging the battery.
The charger can be shut down by using the RUN/SS pin on
the LTC1435. This stops the charging current and results
in a reverse battery drain current in the tens of microamps.
The LTC1435 and LT1620 have been configured so that
the battery can remain connected to the charger when the
input power is removed, but because of the inherent body
diode in the Q1 MOSFET, current can flow from the battery,
through the Q1 body diode, to the LTC1435’s VIN pin,
keeping it powered up. In this situation, because the
charger is effectively powered by the battery, the reverse
battery drain can be several mA, which could discharge the
battery over an extended period. Figure 243 contains
circuitry that automatically shuts down the LTC1435 when
the input power is removed and puts it into a low quiescent
current condition. Because the LT1620 is powered from
the LTC1435 INT VCC pin, it is also turned off.
www.BDTIC.com/Linear
AN84-125
Application Note 84
When input power is applied, the charger can still be shut
down with an external signal to the RUN/SS pin. Shutdown
occurs by pulling this pin low; releasing it allows the
capacitor to charge up via the internal 3µA current source,
producing a soft start.
By substituting higher current MOSFETs and changing
some component values, much higher charging currents
can be obtained.
Selecting Battery Voltage Programming Resistors
The charging voltage of lithium-ion cells is either 4.1 or 4.2
volts per cell, depending on the battery chemistry. Contact
the battery manufacturer for the recommended charge
voltage. To program battery charging voltage (float voltage) use the following equation (for best accuracy and
stability, use 0.1% resistors).
( )
VBATT = VREF 1 + R2
R3
VREF = 1.19V; USE APPROXIMATELY
100kΩ FOR R3
R2 = R3
)
)
VBATT
–1
VREF
Selecting RSENSE
RSENSE is an external, low value resistor that is placed in
the inductor current path to develop a signal representative of the inductor or charge current (IBATT). This signal
is used as feedback to control the switching regulator
constant-voltage and constant-current loops. To minimize overall dropout voltage and power dissipation in the
sense resistor, a sense voltage of 80mV was chosen to
represent maximum charging current. Use the following
equation to select current sense resistor RSENSE. The
maximum battery charge current (MAX IBATT) must be
known.
AN84-126
RSENSE =
0.08V
MAX IBATT
Selecting IPROG
IPROG is a current from the PROG pin to ground that is used
to program the maximum charging current. IPROG can be
derived from a resistor to ground, from the output of a DAC
or by other methods. This program current is generated
using resistors and the 5V VCC available from the LTC1435.
Refer to the simplified diagram of the constant-current
control loop shown in Figure 242. The DC voltage across
CAVG is proportional to the average charge current. This
voltage drives one input of a transconductance (gm)
amplifier. A program voltage (relative to the 5V VCC line)
proportional to the desired, or programmed charge current is applied to the other input of the transconductance
amplifier. This voltage should be selected to be ten times
the average voltage dropped across RSENSE when the
charger is in a constant-current mode.
If the voltage across CAVG increases to a level equal to the
voltage at the PROG pin, the transconductance amplifier
begins pulling down on the ITH pin of the LTC1435, thereby
limiting the peak inductor current, and thus the average
charge current.
The program voltage needed on the program pin can easily
be generated by two resistors, as shown in Figure 242. A
current (IPROG) is generated by these resistors and the 5V
VCC voltage. This IPROG develops a voltage across R6,
which is used to set the maximum constant charge current
level. The circuit is designed for an approximate PROG
voltage of 800mV (don’t exceed the maximum spec of
1.25V), referenced to the LT1620 VCC pin. Because of the
gain-of-10 amplifier, this corresponds to a typical voltage
across RSENSE of 80mV (with a maximum of 125mV).
The recommended range of resistor values for R6 is
approximately 2kΩ to 10kΩ. With 0.8V across R6, this will
result in program currents (IPROG) between 400µA and
80µA.
www.BDTIC.com/Linear
Application Note 84
The LT1620 was designed to reduce the charging current
to zero under all conditions when the IPROG is set to zero.
To ensure that the charging current will always go to zero,
an offset was designed into the transconductance
amplifier. In the equations for R6 and RPROGRAM, this
offset is represented by using 840mV rather than 800mV.
PC Board Layout
Example:
Even with efficiency numbers in the mid 90s, under some
charging conditions power losses can be as high as 4
watts. These losses are primarily in the two MOSFETs, the
inductor and the current sensing resistor. Since these are
surface mount components, the major thermal paths are
through the pc board copper to the surrounding air.
Maximizing copper area around the heat producing components, increasing board area and using double-sided
board with feedthrough vias all contribute to heat dissipation. Remember, the pc board is the heat sink.
GIVEN: MAXIMUM IBATT = 4A
IPROG = 200µA (FOR MAXIMUM IBATT)
RSENSE =
R6 =
RPROG =
0.08V
0.08V
=
= 0.02Ω
MAX IBATT
4A
0.84V
0.84V
=
= 4.2kΩ
IPROG 200µA
5V – 0.84V
5V – 0.84V
=
= 20.8kΩ
200µA
IPROG
Once RPROG and R6 are known, the following equations
can be used to determine RPROG and IPROG for lower IBATT
currents:
RPROG =
R6 [5 – 10(IBATT)(RPROG)]
0.04 + 10(IBATT)(RPROG)
IPROG =
10(IBATT)(RPROG) + 0.04
R6
BATTERY CHARGER IC CAN ALSO SERVE
AS MAIN STEP-DOWN CONVERTER
by LTC Applications Staff
Using a power adapter with the highest feasible output
voltage is attractive to portable system designers for a
couple of reasons. Lower current is required to maintain
the same system power, which translates into a smaller
cable and input connector. If the adapter output voltage is
considerably higher than the battery voltage, the adapter
output voltage does not need to be regulated or well
filtered, resulting in lower adapter cost.
As with any high frequency switching regulator, layout is
important. Switching current paths and heat producing
thermal paths should be identified and the printed circuit
board designed using good layout practices.
One exception to the maximum copper area rule is the
switch node consisting of Q1’s source, Q2’s drain and the
left side of L1. This node switches between ground and VIN
at a 200kHz rate. To minimize radiation from this node, it
should be short and direct. Other copper traces related to
input and output capacitors and MOSFET connections
should also be as short as practical. See the LTC1435 data
sheet for information on good layout practices and additional applications information.
A portable system with a high output-voltage adapter,
however, requires that the system’s DC-to-DC converter
functions over a very wide range of input voltage: from fully
discharged battery voltage to the highest adapter output
voltage.
This problem can be resolved by using the LT1510 as both
the battery charger and the main step-down converter, as
shown in Figure 244. An important feature of the circuit in
Figure 244 is the glitch-free transfer from AC operation to
battery operation and back.
www.BDTIC.com/Linear
AN84-127
Application Note 84
VIN
CR3 1N5819
U1
LTC1510CS16
1
C1
0.22µF
VIN
C6
100µF
+
2
L1**
33µH
CR2
1N914
4
5
L2
2.2mH
6
CR4
1N5817
L3
10µH
6
3
CR1
1N5819
SYSTEM
ON/OFF
SWITCH
2
7
8
GND
GND
SW
VCC1
BOOST
VCC2
GND
PROG
OVP
VC
SENSE
BAT
GND
GND
GND
GND
16
C2†
10µF
15
14
13
12
11
10
R3
1.5k
9
7
SW
Q3
VN2222
4
SENSE
+
U2
LT1300CS8
3
5
C7
NC
SHDN
ILIM
NC 100µF
PGND
GND
8
1
SELECT
C8*
0.1µF
SYSTEM
LOAD
Q1*
VIN MPS3906
R1
100k
CR7*
1N914
CR6*
1N914
C9*
0.1µF
C3
Q2*
22µF
Si9433
25V
C5
1µF
R5
1k
R6
300Ω
R8
12.4k
R9
100k
VIN
R4
4.99k
CR5 1N5819
C4
0.1µF
R7*
100k
+
Q4
VN2222
BAT1
CHARGE/TRICKLE
R2*
1M
* SEE TEXT
** COILTRONICS CTX33-2
† TOKIN OR MARCOM CERAMIC SURFACE MOUNT
Figure 244. LT1510 Battery Charger/Main Step-Down Converter Provides Glitch-Free Transfer between AC and Battery Operation
In the circuit shown in Figure 244, the system’s DC-to-DC
converter is connected to the SENSE pin. This way, the
internal sense resistor is bypassed for the system load but
is active in regulating the charging current. The sum of the
charging current and system current should not exceed
the maximum output current allowed (limited by thermal
considerations or peak switch current). Since the DC-toDC converter circuit has a large input capacitor, it cannot
AN84-128
be connected directly to the SENSE pin. This is because the
internal sense resistor between SENSE and BAT pins will
see a large capacitance across it, which will cause instability. A 2.2mH inductor, such as the DT1608C-222 by
Coilcraft (L2), is used to isolate the input capacitance of
the DC-to-DC converter. CR5 limits the transient current
through the LT1510’s internal sense resistor when the
72.5
72.0
71.5
EFFICIENCY (%)
The LT1510 battery charger IC is capable of charging
currents up to 1.5A and output (battery) voltages up to
20V. High efficiency and small inductor size are achieved
by a saturating switch running at 200kHz. The LT1510 is
capable of charging lithium-ion and sealed-lead-acid
batteries in the constant-voltage/constant-current configuration, and nickel-cadmium and nickel-metal-hydride batteries in the constant-current configuration. The
LT1510 contains an internal switch and current sense
resistor. All the designer needs to do in order to program
the current and voltage is select the current-programming
resistor and the voltage-divider resistors.
71.0
70.5
70.0
69.5
69.0
68.5
68.0
67.5
8
13
18
23
INPUT VOLTAGE (V)
28
DI1510_02.eps
Figure 245. System Efficiency vs Input Voltage
www.BDTIC.com/Linear
Application Note 84
system is operating from the battery and turned on. Q2
(Si9433) is required if the series resistance of 0.2Ω
between the BAT pin and SENSE pin is too high. The
Si9433’s on resistance is 0.075Ω. The charge pump
comprising C8, C9, CR6, CR7 and R2 biases the gate of
Q2. Q1 and R1 turn Q2 off on AC operation (VIN active). R7
programs the trickle-charge current (maximum value is
about 100k) and the equivalent value of R7 and R8
programs the charge current. The Charge input must be
pulled low at the end of the charge.
LT1635 1A SHUNT CHARGER
by Mitchell Lee
Most battery chargers comprise nothing more than a
series-pass regulator with current limit. In solar-powered
systems, you can’t count on sufficient headroom to keep
a series regulator alive, so a shunt method is preferred. A
simple shunt battery charger is shown in Figure 246. It
consists of an op amp driving a shunt transistor and
ballast resistor, and is built around an LT1635. This device
contains both an op amp and a reference, making it
perfectly suited for regulator and charger applications.
Operation is straightforward: the battery voltage is sensed
by a feedback divider composed of two 1M resistors. The
The charger in Figure 244 is connected to a 2-cell NiCd
battery, BAT1. The system switching regulator is LT1300
(U2) based and powers a 5V/250mA load. The efficiency,
η, of the complete system is defined as:
η=
LT1300 Output Power + Battery Charger Power
LT1510 Input Power
The efficiency plot is shown in Figure 245. For the purpose
of measurement, the battery voltage was 3.2V, the charging current was 0.4A and the trickle charge was 40mA.
internal 200mV reference is amplified to 7.05V and compared against the feedback. RT1 introduces a TC that
accurately tracks the battery’s correct charging voltage
over a wide temperature range. Because RT1 is designed
to compensate for changes in battery temperature, it
should be located close to the battery and as far as
possible from the shunt elements. When the battery
charges to 14.1V, the op amp output voltage begins to rise,
turning on the Darlington shunt and resisting further
increases in voltage. Full panel power is divided equally
between the transistor and 7.5Ω resistor when the battery
is completely charged. Don’t forget to provide adequate
heat sinking and air flow for up to 15W dissipation.
14.1V
2A
100nF
1M
3
1A SOLAR
ARRAY
+
200mV
8
REF
1/2 LT1635
–
1 7.05V
2
64.5k
1M
+
7
OA
1/2 LT1635
–
220Ω
6
TIP121
12V, 5Ah
Gelcell
4
7.5Ω/10W
DALE HLM-10
105Ω
2.43k
RT1
7.5k
RT1 = THERM-O-DISC 1K752J
Figure 246. 1A Shunt Battery Charger (IDARK = 230µA; VFLOAT = 14.1V)
www.BDTIC.com/Linear
AN84-129
Application Note 84
The charger is designed to handle 1A continuous, which
is compatible with a “20W” panel. There is no need to
disconnect or diode isolate the charger during periods of
darkness, because the standby current is only 230µA—
less than 10% of the self discharge of even a small battery.
If a different or adjustable output is desired, the feedback
ratio can be easily modified at the 1M divider. 14.1V is a
compromise between an aggressive charge voltage and a
conservative float voltage. Given the cyclic nature of
800mA Li-Ion BATTERY CHARGER OCCUPIES
LESS VOLUME THAN TWO STACKED QUARTERS
by Fran Hoffart
insolation, allowing periodic charging at 14.1V is not
detrimental to Gelcell™ batteries. The circuit in Figure 246
will work with larger or smaller batteries than that shown.
As a rule of thumb, the panel should be sized from 1W per
10Ah battery capacity (a float charge under good conditions with a good battery) to 5W per 1Ah battery capacity
(1 day recharge of a completely discharged battery under
favorable conditions of insolation).
Gelcell is a trademark of Johnson Controls, Inc.
LT1510-5CGN High Efficiency 500kHz
Switch Mode Battery Charger IC
Each new generation of cell phones, PDAs, portable
instruments and other handheld devices is invariably more
powerful, smaller and, most likely, thinner than the last.
The circuit shown in Figure 247 is designed to charge one
or two Lithium-Ion cells at currents up to 800mA, with all
components equal to or less than 2.2mm (0.086 inches)
tall. Using 0.031 inch PC board material, the total circuit
thickness for this charger is 3.4mm (0.136in) or the
thickness of two quarters. The complete 800mA constantcurrent/constant-voltage charger, including the PC board,
occupies less volume than two quarters. This compact,
low profile construction is ideal for cell phones or other
applications where circuit height is restricted.
The charger consists of an LT1510 constant-voltage/
constant-current PWM IC, which includes an onboard
1.5A switch. The LT1510 is available in either 200kHz or
500kHz versions; the higher frequency version allows
lower value, smaller-sized inductors to be used. An internal 0.5% reference allows precision battery-voltage programming and a current programming pin allows a single
resistor, PWM signal or a programming current from a
DAC to control the charging current. Also included are
undervoltage lockout and a low quiescent current sleep
mode that is activated when input power is removed.
VIN = 12V–20V
D1
MBRM140T3
D2
MBRM140T3
1
2
C2
0.22µF
L1
TP3-100
10µH
3
4
5
6
D3
MMBD914LT1
7
8
(
R5 + R6
VBAT = 2.465V 1 +
R4
GND
GND
SW
VCC
BOOST
VCC
PROG
GND
OVP
LT1510-5
VC
NC
NC
SENSE
BAT
GND
GND
16
C1
15
14
10µF
IBAT = 2000
13
12
11
10
9
R2
300
R3
1k
2.465V
R1
)
R1
6.19k
1%
C3
1µF
C4
0.1µF
IBAT
)
+
TO VIN
R4
4.99k
0.5%
(
Q1
2N7002
R5
R6
11.0k
0.5%
1.02k
0.5%
+
VBAT = 8.4V
Li-ION
BATTERY
(2 CELLS)
C5
22µF
IBAT = 800mA
DI 1510 01.eps
Figure 247. Compact, Low Profile, Constant-Current/Constant-Voltage Charger for Li-Ion Batteries
AN84-130
www.BDTIC.com/Linear
Application Note 84
Fused-Lead Package Offers
Lower Thermal Resistance
The LT1510-5 is available in a specially constructed 16lead plastic SSOP package that has the die-attach paddle
connected (fused) directly to the four corner leads and fits
in the same area as an SO-8 package. This low profile
fused-lead package provides a lower thermal resistance
by conducting much of the heat generated by the die
through the copper leads to the PC board copper.
To take advantage of the improved thermal properties of
this fused-lead package, it is important to provide as much
PC board copper around the package leads as practical.
Back-side copper and internal copper layers interconnected by feed-through vias all contribute to the overall
effectiveness of the PC board as a heat sink.
changes to a constant-voltage charge, with the charging
current gradually decreasing to near 0mA as the battery
approaches full charge. If complete charge termination is
required, pulling the VC pin low or sinking zero current
from the program pin stops the charge current. These
signals could be supplied by an external timer or
microprocessor.
When the input power is removed, the LT1510-5 goes into
a low quiescent current (3µA) sleep mode, with this
current coming from the battery. This low battery drain
current allows the battery to remain connected to the
charger for an extended period of time without appreciably
discharging the battery. Additional battery-drain current
can result from reverse leakage current in the Schottky
catch diode D1. Many Schottky diodes have relatively high
leakage currents, so care must be exercised in their
selection.
Charger Operation
A typical charge profile for a discharged Li-Ion battery is
an initial constant-current charge at 800mA until the
battery voltage rises to the programmed voltage. It then
Refer to the LT1510 data sheet for complete product
specifications and to design notes DN111 and DN124 and
application note AN68 for additional application
information.
Table 4. Low Profile Components Used in Figure 247's Circuit
Reference
Designator
C1
C2
C3
C4
C5
D1, D2
D3
L1
Q1
R1
R2
R3
R4
R5
R6
U1
Quantity
1
1
1
1
1
2
1
1
1
1
1
1
1
1
1
1
Part Number
THCR50E1E106ZT
12063C224MAT1A
0805ZC105MAT
08055G104MAT1A
EEFCD1B220R
MBRM140T3
MMBD914LT1
TP3-100
2N7002
LT1510-5CGN
Description
10µF, 25V, 20% Y5U Ceramic
0.22µF, 25V, 20% X7R Ceramic
1µF, 10V, 20% X7R Ceramic
0.1µF, 50V, 20% X7R Ceramic
22µF, 12.5V, 20% Polymer Aluminum Electrolytic
1A, 40V Schottky
0.2A 100V Silicon
10µH Thin-Pac
SOT-23 N-Channel MOSFET
6.19k. 1% Chip Resistor
300Ω,, 5% Chip Resistor
1k, 5% Chip Resistor
4.99k, 0.5% Chip Resistor
11.0k, 0.5% Chip Resistor
1.02k, 0.5% Chip Resistor
Battery Charger IC
www.BDTIC.com/Linear
Vendor
Marcon
AVX
AVX
AVX
Panasonic
Motorola
Motorola
Coiltronics
Zetex
IRC
IRC
IRC
IRC
IRC
IRC
LTC
Phone
(847) 696-2000
(207) 282-5111
(207) 282-5111
(207) 282-5111
(408) 945-5660
(800) 441-2447
(800) 441-2447
(561) 241-7876
(516) 543-7100
(512) 992-7900
(512) 992-7900
(512) 992-7900
(512) 992-7900
(512) 992-7900
(512) 992-7900
(408) 432-1900
AN84-131
Application Note 84
erratic. Although Li-Ion battery use is becoming widespread, it is costly to damage the battery. The supervisory
circuit protects the battery from overcharging and/or
overdraining and prevents the battery voltage from falling
out of its operating region. The LT1496 operates down to
2.2V, ensuring that circuit operation is maintained when
the battery voltage falls below 3V.
SINGLE-CELL Li-Ion BATTERY SUPERVISOR
by Albert Lee
Recently introduced precision products from Linear Technology allow designers to implement high precision applications at supermicropower levels. Among these devices
are the LT1496 quad precision input/rail-to-rail output op
amp and the LT1634 precision shunt voltage reference,
which operate at only 1.5µA and 10µA, respectively. Even
at such low power levels, precision performance is not
compromised. The LT1496 features 475µV maximum
input offset voltage and 1nA maximum input bias current.
The LT1634 achieves 0.05% initial accuracy and 25ppm/°C
maximum temperature drift.
The Li-Ion battery is monitored via a voltage divider off the
battery voltage (node A). The divided voltage is fed into the
positive inputs of comparators A2 and A3 and compared
to the threshold voltages of 1.75V and 1.25V, respectively.
These voltages are selected so that the minimum battery
charge voltage is 3V and the maximum is 4.2V. The
LT1634 1.25V reference is buffered by op amp A1. The
constant 1.25V across R2 creates a 1µA constant current,
so that the output of A1 is amplified to 1.75V. This output
drives RS to provide constant bias current for the LT1634.
Figure 248 shows a single-cell Li-Ion battery supervisory
circuit. The building blocks of this circuit are the LT1496
precision op amp and LT1634 voltage reference. The
useful region of operation of a single-cell Li-Ion battery is
between 4.2V and 3V. The cell voltage drops fairly quickly
below 3V. System operation below this voltage can be
Depending on the battery voltage, the circuit is in one of
the three states, as shown in Table 5.
OFF
CHARGER
SW*
VBAT
1.75V
10µA
–
A2
1/4 LT1496
+
RS
175k
5%
TO LOAD
RH1
10M
5%
RSW
1M
5%
R3
1.75M
0.1%
D1
BATTERY
+
A
A1
1/4 LT1496
R1
500k
0.1%
–
LT1634
1.250V
1.25V
1µA
R2
1.25M
0.1%
VBAT
+
RH2
10M
5%
D2
–
A3
1/4 LT1496
A4
1/4 LT1496
–
+
R4
1.25M
0.1%
D1, D2 = 1N458
R1–R4 = CAR6 SERIES IRC (512) 992-7900
*TP0610L for 50mA LOAD
Figure 248. Single-Cell Li-Ion Battery Supervisor Circuit
AN84-132
www.BDTIC.com/Linear
Application Note 84
If node A were to bounce around at either threshold
voltage, the circuit would bounce between states. To avoid
this problem, hysteresis is added via the resistor and
diode networks connected between the outputs of A2 and
A3 and their positive inputs. Figure 249 shows the behavior of VBAT vs node A entering the trip points with hysteresis. When VBAT rises to 4.2V (node A increases to 1.75V),
op amp A2’s output will switch from low to high, causing
current to flow through RH1. The additional current will
raise node A by an amount ∆VAHYS1, which will clearly put
the circuit in state 3. The circuit will not exit state 3 until
VBAT falls to ∆VHYS1 (310mV for the circuit shown) below
4.2V, which will cause node A to fall back to the upper trip
point of 1.75V (point 1 of Figure 2). Similarly, when VBAT
drops below 3V (node A falls below 1.25V), op amp A3’s
output will switch low, causing current to conduct through
RH2. This will drag node A an amount ∆VAHYS2 below
1.25V, which will put the circuit in state 1. The circuit will
not exit state 1 until the battery voltage is charged to an
amount ∆VHYS2 (149mV for circuit shown) above 3V
(point 2). This will bring node A back up to the lower trip
point, 1.25V, bringing the circuit out of state 1. The
VBAT
Node A
Output
A2
Output
A3
Output
A4
1
< 3V
< 1.25V
Low
Low
High
2
3V< V
< 4.2V
1.25V < V
< 1.75V
Low
High
Low
3
> 4.2V
> 1.75V
High
High
Low
VBAT =
(
(
R3
R3
• (VOHMIN + VBE + 1.75V) + (1.75V •
) + 1.75
R4
RH1
Status
Load Off,
Charge State
Load On,
Charge State
Load On,
Charge
Terminated
R3
RH1
1+
∆VHYS1 = 4.2V – VBAT
Low Trip Point:
IRH2 = (1.25V – VOLMAX – VBE)/RH2
∆VHYS2 = IRH2 • R3
where:
VOHMIN = output voltage swing high (LT1496)
VOLMAX = output voltage swing low (LT1496)
VBE = diode voltage of 1N458
Using an automobile analogy, if the LT1496 op amp is the
transmission of the circuit (switching from one state to the
next), the LT1634 voltage reference is the engine. It not
only generates the threshold voltages, but also the amount
of error that the circuit will have. How much accuracy and
error you get depends on the car you drive. Maximum
input offset voltage and input bias current for the LT1496
are 475µV and 1nA, respectively. The LT1634 is a 0.05%
initial accuracy, 25ppm/°C tempco, 10µA precision shunt
reference. Its 1.250V output voltage will appear at the
input of A3 with an accuracy of 0.088% (initial accuracy +
input offset voltage). R1 and R2 being 0.1% resistors, the
worst-case ratio error will be 0.2%. The worst-case volt-
∆VAHYS1
1
1.75
1.25
∆VAHYS2
2
∆VHYS2
State
High Trip Point:
VA (V)
Table 5. Circuit States
amount of hysteresis desired can be calculated using the
following formulas:
∆VHYS1
The voltage at node A is compared to the two threshold
voltages to determine the state of the circuit. For instance,
when node A reaches or exceeds 1.75V (battery voltage
reaches 4.2V), the outputs of A2 and A3 will swing to the
positive rail, terminating the charger and connecting the
load to the battery. When node A falls between 1.25V and
1.75V (battery voltage between 3V and 4.2V), the output
of A2 swings low, turning the charger on, while the output
of A3 stays high, leaving the load connected. When node
A falls below 1.25V (battery voltage less than 3.0V), the
output of A2 stays low, keeping the charger on. The output
of A3 will also swing low, which, in turn, will cause the
output of A4 to go high, turning off the FET SW that
disconnects the load from the battery.
3.0
VBAT (V)
4.2
Figure 249. VBAT vs VA with Hysteresis
www.BDTIC.com/Linear
AN84-133
Application Note 84
age error across R1 will then be 0.2% or 1mV. This error
compared to the 1.75V threshold voltage is 0.057%.
Similarly, error at 1.75V due to worst-case 2nA input bias
current is 0.057%. Total worst-case error at 1.75V will be
0.202%.
battery voltage error at either trip points is better than
0.47%. Since only the ratios of R1 to R2 and R3 to R4 are
critical, precision matched resistors with ten times better
performance can be used to reduce the overall error by
33%.
VBAT error contributed by the voltage divider branch will
consist of three terms: resistor matching, op amp input
bias current and input offset voltage. The amount of error
is different at the two trip points when VBAT is 3V or 4.2V.
Similar calculations as above result in 0.328% when VBAT
= 3V and 0.268% when VBAT = 4.2V. Therefore, total
This supervisory circuit demonstrates unparalleled performance achievable only with Linear Technology’s
supermicropower precision devices. The supervisory circuit consumes only 20mA. Battery voltage monitoring and
control accuracy is better than 0.5%.
Power Management
Figure 250 is a conceptual block diagram that illustrates
the main features of an LTC1479 dual-battery power
management system, starting with the three main power
sources and ending at the input of the DC/DC switching
regulator.
LTC1479 PowerPath CONTROLLER SIMPLIFIES
PORTABLE POWER MANAGEMENT DESIGN
by Tim Skovmand
Introduction
The LTC1479 PowerPath™ controller drives low loss Nchannel MOSFET switches to direct power in the main
power path of a dual rechargeable battery system, the type
found in most notebook computers and other portable
equipment.
AC
ADAPTER
Switches SWA/B, SWC/D and SWE/F direct power from
either the AC adapter (DCIN) or one of the two battery
packs (BAT1 and BAT2) to the input of the DC/DC switching regulator. Switches SWG and SWH connect the desired battery pack to the battery charger. These five
switches are intelligently controlled by the LTC1479,
which interfaces directly with the power management
microprocessor.
SWA/B
DCIN
SWC/D
BAT1
SWE/F
INRUSH
CURRENT
LIMITING
+
SWG
BAT2
CIN
HIGH EFFICIENCY
DC/DC SWITCHING
REGULATOR
(LTC1435, ETC.)
SWH
BATTERY
CHARGER
(LT1510)
BACK-UP
REGULATOR
LTC1479
PowerPath™ CONTROLLER
POWER
MANAGEMENT
µP
LTC1479 - BLK1
Figure 250. Dual PowerPath Controller Conceptual Block Diagram
AN84-134
www.BDTIC.com/Linear
5V
Application Note 84
SWA
DCIN
SWB
12V AUX
SWC
RSENSE
0.033Ω
SWD
LTC1538-AUX
TRIPLE, HIGH EFFICIENCY,
SWITCHING REGULATOR
0.1µF
SWE
SWF
MBRS140T3
330Ω
GA SAB GB
GC SCD GD
GE SEF GF
SENSE+
3.3V
BACKUP
BATTERY
RDC2
DCIN
5.0V
SENSE-
DCDIV
Li-Ion
BATTERY
PACK #1
VBKUP
RDC1
BACK-UP
REGULATOR
BAT1
BAT2
Li-Ion
BATTERY
PACK #2
LTC1479
PowerPath™ CONTROLLER
VBAT
POWER
MANAGEMENT
µP
RB2
BDIV
RB1
CHGMON
VCC
VCCP
V+
SW VGG
GG SG
GH SH
DCIN
+
2.2µF
16V
+
0.1µF
1µF
50V
+
1mH *
1µF
50V
+
SWG
SWH
LT1510
Li-Ion BATTERY
CHARGER
* 1812LS-105 XKBC, COILCRAFT (708) 639-1469
LTC1479 - FIG03
Figure 251. Dual Li-Ion Battery Power-Management System (Simplified Schematic)
Typical Application Circuit
A typical dual Li-Ion battery power management system is
illustrated in Figure 251. If “good” power is available at the
DCIN input (from the AC adapter), both MOSFETs in switch
pair SWA/B are on—providing a low loss path for current
flow to the input of the LTC1538-AUX DC/DC converter.
Switch pairs SWC/D and SWE/F are turned off to block
current from flowing back into the two battery packs from
the DC input.
Battery Charging
The LTC1479 works equally well with both Li-Ion and
NiMH batteries and chargers. In this application, an LT1510
constant-voltage, constant-current (CC/CV) battery charger
circuit is used to alternately charge two Li-Ion battery
packs.
The power management microprocessor decides which
battery is in need of recharging by either querying a smart
battery pack directly or by more indirect means. After the
determination is made, switch pair SWG or SWH is turned
on by the LTC1479 to pass charger current to one of the
batteries. Simultaneously, the selected battery voltage is
returned to the voltage feedback input of the LT1510 CV/
CC battery charger via a built-in switch in the LTC1479.
After the first battery is charged, it is disconnected from
the charger circuit. The second battery is then connected
through the other switch pair and the second battery
charged. (The LTC1479 works equally well with the LT1511
3A CC/CV Battery Charger and LTC1435/LT1620 4A CC/
CV Battery Charger.)
Running on Batteries
When the AC adapter is removed, the LTC1479 instantly
informs the power management microprocessor that the
DC input is no longer “good” and the desired battery pack
is connected to the input of the LTC1538-AUX high
efficiency switching regulator through either switch pair
SWC/D or SWE/F.
www.BDTIC.com/Linear
AN84-135
Application Note 84
Back-Up Power and System Recovery
the three main input power sources. The power path diode
with the highest input voltage passes current through to
the input of the DC/DC converter to ensure that the system
cannot lock up regardless of how power is initially applied.
Backup power is provided by a standby switching regulator, which is typically powered from a small rechargeable
battery and ensures that the DC/DC input voltage does not
drop below a predetermined level (for example, 6V).
After “good” power is reconnected to one of the three
main inputs, the LTC1479 drives the appropriate switch
pair on fully as the other two are turned off, restoring
normal operation.
The “3-Diode Mode”
When the system is powered by the backup regulator, the
LTC1479 enters a unique operating state called the “3diode mode,” as illustrated in Figure 252. Under normal
operating conditions, both halves of each switch pair are
turned on and off simultaneously. For example, when the
input power source is switched from a good DC input (AC
adapter) to a good battery pack, BAT1, both gates of
switch pair SWA/B are turned off and both gates of switch
pair SWC/D are turned on. The back-to-back body diodes
in switch pair SWA/B block current flow in or out of the DC
input connector.
Interfacing to the Power Management Microprocessor
The LTC1479 takes logic level commands directly from
the microprocessor and makes changes at high current
and high voltage levels in the power path. Further, it
provides information directly to the microprocessor on
the status of the AC adapter, the batteries and the charging
system.
The LTC1479 logic inputs and outputs are TTL level
compatible and therefore interface directly with standard
power management microprocessor. Because of the direct interface via five logic inputs and two logic outputs,
there is virtually no latency (time delay) between the
microprocessor and the LTC1479. In this way, time-
In the 3-diode mode, only the first half of each power path
switch pair, that is, SWA, SWC and SWE, is turned on; and
the second half , that is, SWB, SWD and SWF, is turned off.
These three switch pairs now act as 3-diodes connected to
SWB
SWA
DCIN
ON
OFF
SWD
RSENSE
SWC
12V
BAT1
ON
OFF
SWF
SWE
CIN
+
BAT2
HIGH
EFFICIENCY
DC/DC
SWITCHING
REGULATOR
5V
3.3V
ON
OFF
LTC1479
POWER
MANAGEMENT
µP
LTC1479 - FIG04
Figure 252. LTC1479 PowerPath Controller in “3-Diode Mode”
AN84-136
www.BDTIC.com/Linear
Application Note 84
critical decisions can be made by the microprocessor
without the inherent delays associated with bus protocols
and the like. These delays are acceptable in certain portions of the power management system, but it is vital that
the power path switching control be made through a direct
connection to the power management microprocessor.
The remainder of the power management system can be
easily interfaced to the microprocessor through either
parallel or serial interfaces.
THE LTC1473 DUAL PowerPath SWITCH DRIVER
SIMPLIFIES PORTABLE POWER
MANAGEMENT DESIGN
by Jaime Tseng
The Power Management Microprocessor
The power management microprocessor provides intelligence for the overall power system, and is easily programmed to accommodate the custom requirements of
each system and to allow performance updates without
resorting to costly hardware changes. Many inexpensive
microprocessors are available that can easily fulfill these
requirements.
type found in most notebook computers and other portable equipment.
Overview
Introduction
The LTC1473 is the latest addition to Linear Technology’s
new family of power management controllers, which
simplify the design of circuitry for switching between two
batteries or a battery and an AC adapter.The LTC1473 dual
PowerPath™ switch driver drives low loss N-channel
MOSFET switches that direct power in the main power
path of a single or dual rechargeable battery system, the
The power management system in Figure 253 shows the
LTC1473 driving two sets of back-to-back N-channel
MOSFET switches connecting the two batteries to the
system DC/DC regulator. Each of the switches is controlled by a TTL/CMOS compatible input that interfaces
directly with a power management system microprocessor. An internal boost regulator provides the voltage to
fully enhance the logic-level N-channel MOSFET switches.
MBRD340
DCIN
Si9926
SWA1
RSENSE
SWB1
12V
BAT1
Si9926
SWA2
CIN
SWB2
+
BAT2
HIGH
EFFICIENCY
DC/DC
SWITCHING
REGULATOR
5V
3.3V
MB914LT1
C1
1µF
50V
+
L1
1mH
SW
VGG
C2
1µF
50V
V+
STEP-UP
SWITCHING
REGULATOR
GATE
DRIVER
GATE
DRIVER
LTC1473
IN1
INRUSH
CURRENT
SENSING
AND LIMITING
IN2
DIODE
POWER
MANAGEMENT
µP
+
1473_01.eps
TIMER
CTIMER
4700pF
Figure 253. Dual-Battery PowerPath Switch Driver: VGG Regulator, Inrush Limiting and Switch-Gate Drivers
www.BDTIC.com/Linear
AN84-137
Application Note 84
The LTC1473 uses a current sense loop to limit current
rushing in and out of the batteries and the system supply
capacitor during switch-over transitions or during a fault
condition. A user programmable timer monitors the time
during which the MOSFET switches are in current limit and
latches them off if the programmed time is exceeded. A
unique “2-diode logic mode” ensures system start-up,
regardless of which input receives power first.
microprocessor to select the appropriate battery. The
microprocessor monitors the presence of batteries and
the AC adapter through a supply monitor block, or, in the
case of some battery packs, through a thermistor sensor.
This block comprises a resistor divider and a comparator
for each supply. If the AC adapter is present, the two
switches are turned off by the microprocessor and the
power is delivered to the input of the system DC/DC
switching regulator via a Schottky diode.
Typical Application
A typical dual-battery system is shown in Figure 254. The
LTC1473 accepts commands from a power management
Si9926
MBRD340
BAT1
MMBD2823LT1
MMBD2823LT1
DCIN
SUPPLY
MONITOR
CTIMER
4700PF
GA1
LTC1473
SAB1
IN2
IN1
POWER
MANAGEMENT
µP
+
+
1µF
1mH
DIODE
GB1
TIMER
SENSE+
V+
SENSE-
VGG
GA2
SW
SAB2
GND
GB2
RSENSE
INPUT OF SYSTEM
HIGH EFFICIENCY DC/DC
SWITCHING REGULATOR
(LTC1435,ETC)
0.04Ω
+
COUT
1µF
1473_03.eps
MMBD914LT1
BAT2
Si9926
Figure 254. Dual-Battery Power-Management System
AN84-138
www.BDTIC.com/Linear
Application Note 84
SHORT-CIRCUIT-PROOF ISOLATED
HIGH-SIDE SWITCH
by Mitchell Lee
Figure 255 shows a MOSFET switch, driven by the
LTC1177–5 2.5kVRMS isolator. This device allows a logic
signal to control a power MOSFET and provides complete
galvanic isolation. The device includes an internal current
limiting circuit, but at higher voltages limiting the current
is just not enough for effective protection of the MOSFET.
Foldback (shown on the LTC1177 data sheet) helps, but
the part has trouble starting certain types of loads when
foldback current limiting is used. The circuit shown here
latches off in an overcurrent condition and is restarted by
cycling the logic input.
Q1 and Q2 form an SCR with a holding current of less than
100nA. If the load current exceeds approximately 1A, the
5V
SCR fires, shorting the MOSFET gate to source. The
LTC1177 output current (about 7µA) is more than adequate to hold the SCR on indefinitely. The circuit resets
when the logic input briefly cycles off.
Inductive loads present a special problem. If the load
creeps up on the overcurrent threshold and fires the SCR,
the load’s inductance will carry the MOSFET source far
below ground, which could destroy the MOSFET. Diode D1
clamps the gate at ground, turning the MOSFET back on,
and safely dissipates the stored magnetic energy in the
MOSFET.
As shown the output rise time is about 2ms, allowing the
circuit to successfully charge capacitors of up to 100µF.
Increase C1 proportionately to handle higher value load
capacitors.
2.5kV
ISOLATION BARRIER
24V
OFF
100Ω
ON
VIN
C1
10nF
OUT
LTC1177
G1
SENSE
G2
Q2
2N3904
10M
D1
1N914
Q1
2N3906
MTD3055EL
20M
10M
0.5Ω
1W
LOAD
Figure 255. Short-Circuit Protected, Isolated High-Side Switch
www.BDTIC.com/Linear
AN84-139
Application Note 84
Applications
TINY MSOP DUAL SWITCH DRIVER
IS SMBus CONTROLLED
by Peter Guan
The main application of the LTC1623 is to control two
external high-side N-channel switches (Figure 256). As
seen in the figure, a 0.1µF capacitor and a 1k resistor are
placed on each gate-drive output to respectively slow
down the turn-on time of the external switch and to
eliminate any oscillations caused by the parasitic capacitance of the external switch and the parasitic inductance
of the connecting wires.
Introduction
The LTC1623 SMBus switch controller offers an inexpensive, space-saving alternative for controlling peripherals
in today’s complex portable computer systems. Pin-to-pin
connections between the system controller and each
peripheral device not only result in complicated wiring, but
also limit the number and type of peripheral devices
connected to the system controller. Using the SMBus
architecture, the LTC1623 eliminates these problems by
requiring only two bus wires and allowing easy upgrades
and additions of new peripherals.
Tracking the growing popularity of portable communication systems, the LTC1623 makes a very handy single-slot
3.3V/5V PC Card switch matrix. As shown in Figure 257,
this circuit enables a system controller to switch either a
3.3V or a 5V supply to any of its SMBus-addressed
peripherals. Besides N-channel switches, the LTC1623
can also be used to control a P-channel switch, as shown
in Figure 258. As a result, the load connected to the Pchannel switch will be turned on upon power-up of the
LTC1623, whereas the other load must wait for a valid
address and command to be powered.
VCC
2.7V TO 5.5V
10µF
VCC
(FROM
SMBus)
1k
GA
CLK
DATA
Q1
0.1µF
(PROGRAMMABLE)
VCC
2.7V TO 5.5V
1k
Q2
LTC1623 GB
0.1µF
AD0
10µF
AD1
GND
VCC
LOAD 2
LOAD 1
1623 F02
1k
DATA
Q1, Q2: Si3442DV
Q1
GA
CLK
(FROM
SMBus)
0.1µF
1k
Q2
LTC1623 GB
0.1µF
AD0
(PROGRAMMABLE)
AD1
Figure 256. LTC1623 Controlling Two High-Side Switches
GND
LOAD 1
LOAD 2
Q1: Si3442DV
Q2: Si6433DQ
Figure 258. LTC1623 Controlling a P-Channel Switch (Q2)
5V
10µF
VCC
1k
GA
CLK
DATA
Q1
Si3442DV
TO PC CARD VCC
0V/3.3V/5V
0.1µF
Q2*
LTC1623
AD0
1µF
10k
1k
Q3*
GB
AD1
GND
0.1µF
3.3V
*1/2 Si6926DQ
Figure 257. PC Card 3.3V/5V Switch Matrix
AN84-140
www.BDTIC.com/Linear
1623 TA02
1623 F02
Application Note 84
LTC1710: TWO 0.4Ω SWITCHES WITH SMBus
CONTROL FIT INTO TINY MSOP-8 PACKAGE
Introduction
internal high-side N-channel switches, each capable of
delivering 300mA at an RDS(ON) of 0.4Ω, are available in
the tiny MSOP-8 package. With a low standby current of
14µA, the LTC1710 operates on an input voltage of 2.7V
to 5.5V while maintaining the SMBus-specified 0.6V VIL
and 1.4V VIH input thresholds.
The LTC1710 SMBus dual switch (Figure 259) is a complete solution for supplying power to portable-equipment
peripherals without the need for external switches. Two
Figure 260 shows a circuit using SMBus peripherals
requiring different input voltages can be simultaneously
switched by the LTC1710.
by Peter Guan
VCC
2.7V TO 5V
10µF
VCC
5V
SW0D
GND TO VCC
8
SW0D
2.7V
10µF
1
10µF
10µF
LTC1710
8
1
SW0
5
CLK
2
LTC1710
LOAD 1
5
FROM SMBus
6
DATA
CHARGE
PUMPS
6
SW1
3
3
7
AD1
CLK
OUT0
2
2.7V LOAD
FROM SMBus
LOAD 2
DATA
OUT1
AD1
7
5V LOAD
4
4
Figure 259. Typical Application: The
LTC1710 Switches Two SMBus Peripherals
Miscellaneous
VID VOLTAGE PROGRAMMER
FOR INTEL MOBILE PROCESSORS
by Peter Guan
Microprocessor manufacturers’ relentless push for higher
speed and lower power dissipation, especially in areas of
mobile laptop computer processors, is forcing supply
voltages to these processors to a level previously thought
impossible or impractical. In fact, the supply voltage has
become so critical that different microprocessors demand
different yet precise supply voltage levels in order to
function optimally.
Figure 260. LTC1710 Switches Two SMBus
Peripherals with Different Input Voltages
To accommodate this new generation of microprocessors, LTC introduces the LTC1706-19 VID (voltage
identification) voltage programmer. This device is a precision, digitally programmable resistive divider designed for
use with an entire family of LTC’s DC/DC converters with
onboard 1.19V references. These converters include the
LTC1433, LTC1434, LTC1435, LTC1435A, LTC1436,
LTC1438, LTC1439, LTC1538-AUX, LTC1539 and
LTC1624. (Consult the factory for future compatible DC/
DC converter products.) The LTC1706-19 is fully compliant with the Intel mobile VID specifications and comes in
a tiny SO-8 package. Four digital pins are provided to
program output voltages from 1.3V to 2.0V in 50mV steps
with an accuracy of ±0.25%.
www.BDTIC.com/Linear
AN84-141
Application Note 84
VIN
4.5V TO 22V
LTC1435A
COSC
43pF
1
CSS
0.1µF
2
3
CC2
220pF
COSC
VIN
RUN/SS
TG
ITH
SW
CC
1000pF
INTVCC
RC
10k
BOOST
5
51pF
6
SGND
BG
VOSENSE
PGND
RF
4.7Ω
13
CF
0.1µF
16
+
M1
Si4410DY
CIN
10µF, 30V
×2 R
SENSE
VOUT
1.30V TO
2.00V/7A
0.015Ω
14
L1 3.3µH
DB*
12
0.22µF
+
+
11
10
SENSE– SENSE+
7
8
SENSE
VCC
15
4.7µF
M2
Si4410DY
D1
MBRS
-140T3
FB
LTC1706-19
VID VID VID VID
0 1 2 3
COUT
820µF
4V
×2
GND
*DB = CMDSH-3
FROM µP
1000pF
Figure 261. Intel Mobil Pentium II Processor VID Power Converter
Applications
Figure 261 shows a VID-programmed DC/DC converter
for an Intel mobile processor that uses the LTC1435A and
LTC1706-19 to deliver 7A of output current with a programmable VOUT of 1.3V to 2.0V from a VIN of 4.5V to 22V.
Simply connecting the LTC1706-19’s FB and SENSE pins
to the LTC1435A’s VOSENSE and SENSE– pins, respectively, closes the loop between the output voltage sense
and the feedback inputs of the LTC1435A regulator with
the appropriate resistive divider network, which is controlled by the LTC1706-19’s four VID input pins.
Table 6 shows the VID inputs and their corresponding
output voltages. VID3 is the most significant bit (MSB) and
VID0 is the least significant bit (LSB). When all four inputs
are low, the LTC1706-19 sets the regulator output voltage
to 2.00V. Each increasing binary count is equivalent to
decreasing the output voltage by 50mV. Therefore, to
obtain a 1.30V output, the three MSBs are left floating
while only VID0 is grounded. In cases where all four VID
inputs are tied high or left floating, such as when no
processor is present in the system, a regulated 1.25V
output is generated at VSENSE.
AN84-142
Table 6. VID Inputs and Corresponding Output Voltages
Code
VID3
VID2
VID1
VID0
Output
0000
GND
GND
GND
GND
2.00V
0001
GND
GND
GND
Float
1.95V
0010
GND
GND
Float
GND
1.90V
0011
GND
GND
Float
Float
1.85V
0100
GND
Float
GND
GND
1.80V
0101
GND
Float
GND
Float
1.75V
0110
GND
Float
Float
GND
1.70V
0111
GND
Float
Float
Float
1.65V
1000
Float
GND
GND
GND
1.60V
1001
Float
GND
GND
Float
1.55V
1010
Float
GND
Float
GND
1.50V
1011
Float
GND
Float
Float
1.45V
1100
Float
Float
GND
GND
1.40V
1101
Float
Float
GND
Float
1.35V
1110
Float
Float
Float
GND
1.30V
www.BDTIC.com/Linear
Application Note 84
Figure 262 shows a combination of the LTC1624 and the
LTC1706-19 configured as a high efficiency step-down
switching regulator with a programmable output of 1.3V
to 2.0V from an input of 4.8V to 20V. Using only one
N-channel power MOSFET, the two SO-8 packaged LTC
parts offer an extremely versatile, efficient, compact regulated power supply.
Figure 263 shows the LTC1436A-PLL and the LTC170619, a combination that yields a high efficiency low noise
synchronous step-down switching regulator with programmable 1.3V to 2V outputs and external frequency
synchronization capability.
VIN
4.8V TO 20V
VCC
2.7V TO 5.5V
LTC1624 1000pF
LTC1706-19 3
VCC
7
6
VID0
SENSE
8
VID1
100pF
1
VID2
2
5
VID3
FB
470pF
GND
4
6.8k
10µF
1
2
3
4
SENSE–
VIN
ITH/RUN BOOST
VFB
GND
TG
SW
8
7 0.1µF
6
RSENSE
0.05Ω
+
Si4412DY
CIN
22µF
35V
×2
VOUT
1.3V TO 2.0V
5
10µH
+
MBRS340T3
COUT
100µF
10V
×2
Figure 262. High Efficiency SO-8, N-Channel Switching Regulator with Programmable Output
10k
EXTERNAL
FREQUENCY
SYNCHRONIZATION
0.1µF
COSC
39pF
2
CSS
0.1µF
3
4
COSC
VIN
RUN/SS
TGL
TGS
LTC1436A-PLL
ITH
SW
CC
510pF
INTVCC
RC
10k
BOOST
6
100pF
VIN
4.5V TO 22V
1
24
PLL LPF PLLIN
8
SGND
BGL
VOSENSE
PGND
18
+
M1
Si4412DY
21
19
M3
IRLML2803
20
L1
3.3µH
CIN
22µF, 35V
×2
RSENSE
0.02Ω
VOUT
1.30V TO
2.00V/5A
D B*
17
0.22µF
+
+
16
4.7µF
15
M2
Si4412DY
SENSE– SENSE+
9
10
1000pF
SENSE
VCC
22
D1
MBRS
-140T3
FB
LTC1706-19
VID VID VID VID
0 1 2 3
COUT
100µF
10V
×2
GND
*DB = CMDSH-3
FROM µP
Figure 263. High Efficiency, Low Noise, Synchronous Step-Down Switching
Regulator with Programmable Output and External Synchronization
www.BDTIC.com/Linear
AN84-143
Application Note 84
Besides the LTC family of 1.19V-referenced DC/DC converters, the LTC1706-19 can also be used to program the
output voltages of regulators with different onboard references. Figure 264 shows the LTC1706-19 programming
the output of the LT1575, an UltraFast™ transient response,
low dropout regulator that is ideal for today’s powerhungry desktop microprocessors. However, since the
LT1575 has a 1.21V reference instead of a 1.19V reference, the output will range from 1.27V to 2.03V in steps of
50.8mV.
VIN
12V
VCC 3.3V
LT1575
LTC1706-19 3
VCC
7
SENSE
VID0
8
VID1
1
VID2
2
FBK
VID3
GND
4
1
6
2
1µF
5
3
4
SHDN
IPOS
VIN
INEG
GND
GATE
FB
COMP
8
3.3V
7
6
5.1Ω
IRFZ24
+
5
220µF
VOUT
1.27V–2.03
IN 50.8mV STEPS
7.5k
24µF
10pF
1000pF
Figure 264. UltraFast Transient Response, Low Dropout Regulator with Adjustable Output Voltage
AN84-144
www.BDTIC.com/Linear
Application Note 84
BATTERY CHARGER IC
DOUBLES AS CURRENT SENSOR
by Craig Varga
It’s always fun to find applications for an IC that its
designer never intended. The circuit shown in Figure 264
is such a design. In many cases, a circuit is required to
provide a ground-referenced output voltage that is proportional to a measured current. Frequently, the current must
be measured with a shunt in the positive rail that may be
well above ground and, worse yet, may vary considerably
with time. The LT1620 was originally intended as a controller for a synchronous buck regulator in battery-charger
applications. The normal operating mode for this IC is to
mirror a current signal down to a 5V reference supply. By
adding a single small-signal MOSFET and a few resistors,
it is possible to again mirror this signal to provide a ground
referenced output.
Circuit operation is as follows: The LT1620 operates by
producing a voltage between the VCC pin and the AVG pin
that is 10× the voltage across sense resistor R5. C2 filters
this voltage. An internal op amp has its noninverting input
at the AVG pin (pin 8), its inverting input at the PROG pin
(pin 7) and its output at the IOUT pin (pin 2). With the circuit
connected as shown in Figure 265, this amplifier will force
enough current through R4 to make the voltage drop on R4
equal to the voltage across C2. This current is mirrored
through R3 and is filtered by C3, producing a clean,
ground-referenced, DC output voltage. Resistor R2 cancels a small built-in offset in the LT1620’s amplifiers. The
output voltage obeys the following relationship: VO = IL
(R5 • R3 • 10)/R4. Changing the value of R3 selects
different scale factors.
The circuit yields excellent linearity over a wide range of
loads and input voltages. The curve shown in Figure 266
was measured with the sense resistor referenced to a 5V
input source. The curve looks the same even at inputs over
25V, so only one curve is presented. Maximum input
voltage is 36V. There is a small offset at no load, but in a
typical microprocessor-based data acquisition system,
only a simple 2-point calibration is needed to obtain
absolute accuracy.
4.0
C2
0.33µF
IL
LOAD
R5
0.02Ω
INPUT
IL
C1
1000pF
LT1620
7
8
AVG
PROG
1
6
VCC
SENSE
2
4
IN–
IOUT
3
5
IN+
GND
3.5
R1
100k
R4
10k
1%
R2
1.2M
Q1
TP0610T
R3
31.6k
OUTPUT
C3
0.33µF
OUTPUT VOLTAGE (V)
5V
3.0
2.5
2.0
1.5
1.0
0.5
0.0
0
Figure 265. Current Sensor Schematic
1
2
3
4
LOAD CURRENT (A)
5
6
Figure 266. Transfer Function
www.BDTIC.com/Linear
AN84-145
Application Note 84
100V, 2A, CONSTANT-VOLTAGE/
CONSTANT-CURRENT BENCH SUPPLY
by Mitchell Lee and Jesus Rosales
The converter is designed to operate from an input of 40V
to 60V, supplied by a line transformer, diode bridge and
filter capacitor (not shown). Output voltage is linearly
adjustable from zero to 100V via potentiometer R20.
Most engineering labs are well stocked with low voltage,
moderate current power supplies, but higher voltage supplies capable of several amperes of output current are hard
to find. We solved this problem in our lab by building the
supply shown in Figure 267.
The current is limited by two independent loops. The first
current limit loop is user controlled over a range of zero to
8A by setting potentiometer R21. This setting does not
interact with changes in output voltage. A second current
limit loop limits the maximum available current as a
function of voltage (components R1–R5 and U2), minimizing component stress. Under any given operating
condition, the lower of the two loops takes control. Maximum available output current is highest at low output
voltage settings (about 8A), and decreases to 2A at 100V
output.
The circuit is based on U1, an LT1270 high efficiency
switching regulator configured in a SEPIC topology, which
allows the output to be adjusted higher or lower than the
input voltage. Operation is similar to that of a flyback
converter, but the primary and secondary windings are
coupled together by capacitor C1. This allows the primary
and secondary windings to share current, reducing copper
loss; it also eliminates the snubbing circuitry and losses
found in flyback converters.
VIN
40V TO
60V
T1
3.3k
2W
L1
20µH
MUR1560
+
MBR745
2N6387
10Ω
MUR120
IRF450
T1 : PRIMARY: 57 TURNS 20AWG
SECONDARY: 57 TURNS 20AWG
MPP 55076 MAG INC CORE
L1: 18 TURNS 18AWG
55380-A2 MAG INC CORE
2.2k 1000pF
+
C1
10µF, 100V
FILM CAP
10µF, 200V +
FILM CAP
10µF, 200V
FILM CAP
VOUT
0V TO
100V
0.03Ω, 2W
0.1µF
1k
4.7k
100k
1N5817
100Ω
1N5817
4
U1
LT1270
+
1N4148
1k
3
–
5
1/2
LT1413
7
2
10Ω
8
U2A
R21, 1k
3.9k
6
5
0.1µF
4
3.9k
1k
3.9k
R3
4.5k
R4
3.9k
R1
3.9k
R2
3.9k
2.2k
1
1N4148
0.1µF
15k
3
+
1/2
LT1413
2
–
U2B
1
10k
2N2907
7
680µF +
100V
×2
56µF
+ 35V
+
3.9V
1/2
LT1215
+
15V
0.33µF
1µF
2.2k
–
U3B
100Ω
6
U3A
20k
1
8
1/2
LT1215
+
100Ω
–
22V
2
3
R20
10k
4
5
15k
1µF
10k
LT1034CZ-2.5
0.01µF
Figure 267. Constant-Voltage/Constant-Current Bench Supply
AN84-146
www.BDTIC.com/Linear
R5
2.7M
Application Note 84
A COMPLETE BATTERY BACKUP SOLUTION
USING A RECHARGEABLE NiCd CELL
by L.Y. Lin and S.H. Lim
lithium battery. This solution requires low-battery detection, necessitates battery access and invites inadvertent
battery removal. The LTC1558 battery backup controller
eliminates these problems by permitting the use of a
single, low cost 1.2V rechargeable Nickel-Cadmium (NiCd)
cell. The LTC1558 has a built-in fast-/trickle-mode charger
that charges the NiCd cell when main power is present.
Battery-powered systems, including notebook computers, personal digital assistants (PDAs) and portable instruments, require backup systems to keep the memory
alive while the main battery is being replaced. The most
common solution is to use an expensive, nonrechargeable
FROM µP
OPEN DRAIN
SOFT RESET
L11†
22µH
BACKUP
BATTERY
NiCd††
1.2V
+
R14
10k
1
C11
47µF
6.3V
SW
VCC
VBAK
3
7
+ C12
8
1µF
CTL LTC1558-3.3
5
2
RESET
GND
PUSH-BUTTON
RESET
4
SW11
FB
BKUP
R15
12k
TO
µP
6
R13
100k
Q11
Si4431DY
R11
51k
1%
MAIN BATTERY
4.5V TO 10V
R12
21.2k
(20.0k 1% +
1.21k 1%)
C2
0.1µF
13
9
CSS
0.1µF
CC2
51pF
C1
100pF
RC
10k
16
TG
SW
14
C4
15
BOOST
0.1µF
LTC1435
12 D1***
6
INTVCC
VOSENSE
8
3
ITH
SENSE+
C5
7
2
1000pF
RUN/SS
SENSE–
Q2
11
1 C
BG
OSC
Si4412DY
+ C3
SGND PGND
4.7µF
10
16V
COSC 5
68pF
4
CC
330pF
VIN
EXTVCC
Q1
Si4412DY
SFB
*
**
***
†
††
+
CIN
100µF
16V
×2
L1*
10µH
RSENSE**
0.033Ω
+
D2
MBRS140T3
SUMIDA CDRH125-100
IRC LR2010-01-R033-F
CENTRAL CMDSH-3
SUMIDA CDRH73-220
SANYO CADNICA N-110AA
COUT
100µF
10V
×2
VOUT
3.3V
LOAD CURRENT
3A IN NORMAL MODE
30mA IN BACKUP MODE
R1
35.7k 1%
R5
20k
1%
C6
100pF
1558 01.eps
Figure 268. LTC1558 Backup System with an LTC1435 as the Main System Regulator
www.BDTIC.com/Linear
AN84-147
Application Note 84
180
300
VBAK = 4V
VOUT = 3.3V
VBAK = 4V
VOUT = 3.3V
250
140
BACKUP TIME (MINS)
OUTPUT POWER (mW)
160
120
100
80
60
200
150
100
40
50
20
0
1.00 1.05 1.10 1.15 1.20 1.25 1.30 1.35 1.40
BACKUP CELL VOLTAGE (V)
0
0
5
10
15
20
LOAD CURRENT (mA)
25
30
Figure 269. 3.3V Output Power vs Backup1558_02
Cell Voltage
Figure 270. Backup Time vs 3.3V Output Load Current
Figure 268 shows a typical application circuit with an
LTC1558-3.3 providing backup power to an LTC1435
synchronous step-down switching regulator. The backup
circuit components consist of the NiCd cell, R11–R14,
C11–C12, L11 and Q11. SW11 and R15 provide a soft or
hard reset function.
mode. In backup mode, the LTC1558’s internal switches
and L11 form a synchronous boost converter that generates a regulated 4V at VBAK. The LTC1435 operates from
this supply voltage to generate the 3.3V output voltage.
The BKUP pin is pulled high by R13 and Q11 turns off ,
leaving its body diode reverse biased. The BKUP pin also
alerts the system microprocessor. C11, a 47µF capacitor,
provides a low impedance bypass to handle the boost
converter’s transient load current; otherwise, the voltage
drop across the NiCd cell’s internal resistance would
activate the LTC1558’s undervoltage-lockout function.
Table 7 shows several values of VFB vs the VBAK voltage.
Figure 269 shows the maximum output power available at
the 3.3V output vs the NiCd cell voltage. Over 100mW of
output power is achieved for a NiCd cell voltage greater
than 1V. Figure 270 shows the backup time vs the 3.3V
load current using a Sanyo Cadnica N-110AA cell (standard series with a capacity of 110mAhrs). Over one hour
of backup time is realized for less than 80mW of 3.3V
output power.
Normal Mode (Operation from the Main Battery)
During normal operation, the LTC1435 is powered from
the main battery, which can range from 4.5V to 10V (for
example, a 2-series or 2-series × 2-parallel Li-Ion battery
pack, or the like) and generates the 3.3V system output.
The LTC1558 operates in standby mode. In standby mode,
the LTC1558 BKUP (backup) pin is pulled low and Pchannel MOSFET Q11 is on. The NiCd cell is fast charged
by a 15mA current source connected between the
LTC1558’s VCC and SW pins. Once the NiCd cell is fully
charged (according to the LTC1558’s gas-gauge counter),
the LTC1558 trickle charges the NiCd cell. R14 sets the
trickle-charge current according to the formula I(TRICKLE)
= 10 • (VNiCd – 0.5)/R14. The trickle-charge current is set
to overcome the NiCd cell’s self-discharge current, thereby
maintaining the cell’s full charge.
1558_03
Table 7. VFB and VBAK Voltages
Relative %
Below VREF
–0%
Backup Mode (Operation from the Backup Battery)
–6%
–7.5%
% of VREF
VFB
VBAK
100%
1.272V
4.325V
94%
1.196V
4.065V
92.5%
1.177V
4.000V
The main battery voltage is scaled down through resistor
divider R11–R12 and monitored by the LTC1558 via the FB
pin. If the voltage on the FB pin drops 7.5% below the
internal 1.272V reference voltage (due to discharging or
exchanging the main battery), the system enters backup
AN84-148
www.BDTIC.com/Linear
Application Note 84
Recovery from Backup Mode to Normal Mode
When a new main battery pack is inserted into the system,
Q11’s body diode forward biases. Once the voltage at the
FB pin increases to more than 6% below VREF, the boost
converter is disabled and the system returns to normal
C6
100µF
10V
Introduction
VOUT
3.3V
In switching regulators’ data sheets, there are always
efficiency curves that show how efficient the regulators
are in transforming one voltage to another. Although these
curves are useful in comparing one regulator to another,
they don’t allow a system designer to determine accurately
how long batteries will last before they need to be replaced
or recharged when they are used as the power source. This
complication arises because the type of batteries used to
power the system and the regulator load characteristic
strongly affect the lifetime of the batteries.
In this article, battery lifetime curves are obtained for the
LTC1174 and the LTC1433.
+
+
VOUT
3.3V
C6
100µF
10V
D1
MBRM5819
1
L1
22µH
2
3
4
5
6
C7
0.1µF
7
8
NC
BSW
LTC1433
NC
SVIN
COSC
SGND
POR
RUNSS
ITH
LB0
VOSENSE
LB1
VPROG
14
C4
0.1µF
9
L1
100µH
MBRM520LT1
C3
22µF, 20V
1
16
PWRVIN
PGND 15
2
L2
3
22µH
4
5
SSW
NC
BSW
LTC1433
SVIN
NC
COSC
SGND
POR
6
RUNSS
7
C7
LB0
0.1µF 8
LB1
ITH
VOSENSE
VPROG
VIN
C5
47pF
14
13
12
R1 5.1k
11
10
C1
6800pF
C2
680pF
9
DI_EFF_01b.eps
L1 = SUMIDA CD54-101
L2 = SUMIDA CD54-220
Figure 271b. LTC1433 Dual-Inductor Configuration
A Short Introduction to the LTC1174 and LTC1433
C5
47pF
6
VIN
8
LBIN
SHDN
2 LTC1174-3.3 1
VOUT
LBOUT
3
12
10
MBRM5819
VIN
13
11
+
The LTC1174 uses a constant off-time architecture to
switch its internal P-channel power MOSFET. The inputto-output voltage ratio sets the on time and requires the
inductor current to reach a preset limit. Even at low load
current, the LTC1174 still requires the inductor current to
reach the preset limit before it initiates the off-time cycle.
Burst Mode operation of the LTC1174 enhances efficiency
C3
33µF, 20V
16
PWRVIN
PGND 15
SSW
C4
0.1µF
+
WHAT EFFICIENCY CURVES DON’T TELL
by San-Hwa Chee
mode. The BKUP pin pulls low and turns Q11 back on. This
allows the new battery pack to supply input power to the
LTC1435. The LTC1558 now accurately replenishes the
amount of charge removed from the NiCd cell through the
internal charger and gas-gauge counter.
R1 5.1k
C1
6800pF
C2
680pF
7
IPGM
GND
SW
5
4
C3
0.1µF
VIN
C2
22µF
50V
+
L1
68µH
VOUT
+
C1
100µF
10V
D1
MBR0520LT1
DI_EFF_01a.eps
L1 = SUMIDA CD54-220
DI_EFF_01c.eps
L1 = SUMIDA CDRH74-680
Figure 271a. LTC1433 Single-Inductor Configuration
Figure 271c. LTC1174 Test Circuit
www.BDTIC.com/Linear
AN84-149
Application Note 84
100
100
FIGURE 270a
FIGURE 270b
FIGURE 271
90
85
90
85
80
80
75
75
4
5
6
7
INPUT VOLTAGE (V)
8
4
9
throughout the load-current range by switching only the
required number of cycles to bring the output into regulation and then stopping switching (going into sleep mode).
When the output voltage has dropped slightly, the switching sequence resumes. By doing this, switching losses are
5
OUTPUT
VOLTAGE
3
BATTERY OUTPUT
VOLTAGE VOLTAGE
LTC1433 WITH DUAL INDUCTORS
LTC1433 WITH SINGLE INDUCTOR
LTC1174HV
1
9
The LTC1433 is a constant-frequency, current mode,
monolithic switching regulator in which the inductor peak
BATTERY
VOLTAGE
5
4
OUTPUT
VOLTAGE
3
BATTERY OUTPUT
VOLTAGE VOLTAGE
LTC1433 WITH DUAL INDUCTORS
LTC1433 WITH SINGLE INDUCTOR
LTC1174HV
2
1
0
0
0
0.5
1.0
1.5
TIME (HOURS)
2.0
Figure 273. Lifetime at ILOAD =
400mA—Four AA Alkaline Batteries
0
2.5
0.5
DI_EFF_03.eps
1.0
1.5
TIME (HOURS)
2.0
Figure 274. Lifetime at ILOAD =
400mA—Four AA NiCd Batteries
2.5
DI_EFF_04.eps
6
6
BATTERY
VOLTAGE
5
4
BATTERY AND OUTPUT VOLTAGE (V)
BATTERY AND OUTPUT VOLTAGE (V)
8
6
BATTERY
VOLTAGE
4
6
7
INPUT VOLTAGE (V)
reduced and are minimized when the load current is low,
because the sleep duration is long.
BATTERY AND OUTPUT VOLTAGE (V)
BATTERY AND OUTPUT VOLTAGE (V)
6
5
DI_EFF_02b.eps
Figure 272b. Efficiency Curves
for
Figure 271’s Circuits, ILOAD = 10mA
DI_EFF_02a.eps
Figure 272a. Efficiency Curves for
Figure 271’s Circuits, ILOAD = 400mA
2
FIGURE 271a
FIGURE 271b
FIGURE 271c
95
EFFICIENCY (%)
EFFICIENCY (%)
95
OUTPUT
VOLTAGE
3
2
BATTERY OUTPUT
VOLTAGE VOLTAGE
LTC1433 WITH DUAL INDUCTORS
LTC1433 WITH SINGLE INDUCTOR
LTC1174HV
1
0
0
10
20
30
TIME (HOURS)
40
5
4
OUTPUT
VOLTAGE
3
BATTERY OUTPUT
VOLTAGE VOLTAGE
LTC1433 WITH DUAL INDUCTORS
LTC1433 WITH SINGLE INDUCTOR
LTC1174HV
2
1
0
50
60
DI_EFF_05.eps
Figure 275. Lifetime with Load Step from 10mA to 410mA,
10% Duty Cycle, TPERIOD = 20s—Four AA Alkaline Batteries
AN84-150
BATTERY
VOLTAGE
0
2
4
6
8
10
12
TIME (HOURS)
14
16
18
20
DI_EFF_06.eps
Figure 276. Lifetime with Load Step from 10mA to 410mA,
10% Duty Cycle, TPERIOD = 20s—Four AA NiCd Batteries
www.BDTIC.com/Linear
Application Note 84
9
BATTERY OUTPUT
VOLTAGE VOLTAGE
8
7
LTC1433 WITH DUAL INDUCTORS
LTC1433 WITH SINGLE INDUCTOR
LTC1174HV
BATTERY
VOLTAGE
6
5
OUTPUT
VOLTAGE
4
3
2
1
0
0
0.5
1.0
1.5
TIME (HOURS)
2.0
2.5
DI_EFF_07.eps
Figure 277. Lifetime at ILOAD = 400mA—on a 9V Alkaline Battery
current varies according to the load current. In place of
Burst Mode operation, the LTC1433 has an Adaptive
Power output stage to enhance its efficiency at low load
current. Under low load conditions, the LTC1433 uses
only a fraction of its power MOSFET, effectively reducing
switching losses without introducing low frequency noise
components.
For more information on both parts, consult the data
sheets.
The Setup
The circuits in Figures 271a, b and c were used to obtain
the lifetime data. All outputs were set at 3.3V and the power
was supplied by either four AA alkaline (Eveready No.
EN91) or four AA NiCd (Eveready No. CH15) cells or a
single 9V alkaline (Eveready No. EN22) battery. A currentsink load was set up to either draw a constant 400mA or
provide a load-step characteristic. The load stepping operated at 0.05Hz, going from 10mA to 410mA with a duty
cycle of 10%, providing an average load current of 50mA.
In Figure 271b, the LTC1433 was set up to optimize low
load current efficiency by configuring the Adaptive Power
output stage with separate inductors for low and high
current operation.
Efficiency curves for each circuit are shown in Figure 272a
and 272b. Figures 273 through 278 show the battery
voltage and regulator output voltage versus time for
various battery and load combinations.
BATTERY AND OUTPUT VOLTAGE (V)
BATTERY AND OUTPUT VOLTAGE (V)
9
BATTERY OUTPUT
VOLTAGE VOLTAGE
8
7
LTC1433 WITH DUAL INDUCTORS
LTC1433 WITH SINGLE INDUCTOR
LTC1174HV
BATTERY
VOLTAGE
6
5
4
OUTPUT
VOLTAGE
3
2
1
0
0
2
4
6
8
10
12
TIME (HOURS)
14
16
18
20
DI_EFF_08.eps
Figure 278. Lifetime with Load Step from 10mA to 410mA—
one 9V Alkaline Battery (10% Duty Cycle, TPERIOD = 20s)
4-Cell to 3.3V Configuration
Figures 273 and 274 were obtained with a load current of
400mA. For Figure 273, the input power to the regulator
was provided by four AA alkaline batteries, whereas four
AA NiCds were used in Figure 274. The alkaline batteries
lasted longer than the NiCds, due to their higher energy
capacity. From Figure 274, it is apparent when the NiCd
gives up, from the cliff-like shape of the output voltage.
For Figures 275 and 276, a step load was applied to the
regulators instead of a DC load. Figure 275 and 276 are the
data obtained for alkaline and NiCd AA cells, respectively.
With the average load one-eighth of the previous experiment, it would be expected that the lifetime of the alkaline
batteries would be eight times longer or approximately 18
hours, but Figure 275 shows a significantly better result.
The main reason for this improvement has to do with the
internal resistance of the alkaline cell. At high constant DC
load current, heat is dissipated by the internal resistance
of the alkaline batteries. The internal resistance increases
as the batteries voltage decreases, and hence causes more
heat to be dissipated, thus lowering the lifetime.
For the NiCd battery, internal resistance is low and remains
relatively constant over its life span. Therefore, the lifetime
of the NiCd batteries for the load step case comes out to
be approximately the expected eight times that of a constant DC load current.
www.BDTIC.com/Linear
AN84-151
Application Note 84
The above result indicates that if the load is intermittent in
nature, the user can operate the device much longer if the
power is provided by alkaline batteries. Again, the NiCd
exhibits a sudden “death” at the end of its life, whereas the
alkaline shows a much gentler decay. The gentle sloping
of the output voltage of Figure 275 towards the end of the
battery life can be attributed to the on-resistance of the
switch when the regulator is in dropout.
value of 600mA whether the load current is at 10mA or at
410mA. This high peak inductor current, combined with
the high internal resistance of the alkaline AA cells, shortens the lifetime. Figure 276 shows that the use time is
about the same for the LTC1174 and the LTC1433 because
of the low, constant internal resistance of the NiCd batteries.
For the above load characteristic, where the load is light
most of the time, making full use of the Adaptive Power
mode of the LTC1433 by means of the dual inductor
configuration helps to squeeze an additional 1.5 hours of
life compared to the single inductor LTC1433 configuration.
The lifetime graphs are shown in Figures 277 and 278.
Comparing the data between the 9V and the AA alkaline
cells, the lifetime of the AA cells is about 2.5 times longer.
This is because the energy capacity of the 9V alkaline is
much smaller than that of the AA cells. In addition, the
internal resistance of the 9V alkaline is much higher than
the AA cells, causing more energy to be dissipated as heat.
For the load step case, the battery lasted 13.8 times longer
than a constant 400mA load. The dual inductor
configuration of the LTC1433 lasted about an hour longer
than the single inductor one.
Another important point to note is that although the
efficiency for the LTC1174 is better than that of the single
inductor configuration of the LTC1433 at 10mA load
current, the LTC1433 lasted 2.9 hours longer than the
LTC1174 in Figure 275. The reason for this is that the
LTC1174 inductor’s current always ramps up to the preset
AN84-152
9V-to-3.3V
www.BDTIC.com/Linear
Application Note 84
APPENDIX A: COMPONENT VENDOR CONTACTS
The tables on this and the following pages list contact
information for vendors of non-LTC parts used in the
application circuits in this publication. In some cases,
components from other vendors may also be suitable. For
information on component selection, consult the text of
the respective articles and the appropriate LTC data sheets.
Capacitors
Vendor
Product
Phone
URL
AVX
Chip Capacitors
(843) 946-0362
AVX
Tantalum Capacitors
(207) 282-5111
Electronic Concepts
400V Film Capacitors
(908) 542-7880
Kemet
Tantalum Capacitors
(408) 986-0424
www.kemet.com
Marcon
High C/V Capacitors
(847) 696-2000
www.chemi-con.com/main/company/marcon.html
Murata Electronics
Capacitors
(814) 237-1431
www.iijnet.or.jp/murata/products/english
Nichicon
Electrolytic Capacitors
(847) 843-7500
www.nichicon-us.com
Panasonic
Poly Capacitors
(714) 373-7334
www.avxcorp.com/products/capacitors
www.eci-capacitors.com
Sanyo
Oscon Capacitors
(619) 661-6835
www.panasonic.com/industrial_oem/electronic_components/
electronic_components_capacitors_home.htm
www.sanyovideo.com
Sprague
Capacitors
(207) 324-4140
www.comsprague.com
Taiyo Yuden
Chip Capacitors
(408) 573-4150
http://www.t-yuden.com
Tokin
Capacitors
(408) 432-8020
www.tokin.com
United Chemicon
Electrolytic Capacitor
(847) 696-2000
www.chemi-con.com/main
Vitramon
Ceramic Chip Capacitor
(203) 268-6261
www.vishay.com
Wima
Paper/Film Capacitors
(914) 347-2474
www.wimausa.com
Diodes
Vendor
Product
Phone Number
URL
Agilent (formerly Hewlett
Packard)
IR LEDs
(800) 235-0312
www.semiconductor.agilent.com/ir
Central Semiconductor
Small Signal Discretes
(516) 435-1110
www.centralsemi.com
Chicago Miniature Lamp
LEDs
(201) 489-8989
www.sli-lighting.com/cml
Data Display Products
LEDs
(800) 421-6815
www.ddp-leds.com
Fuji
Schottky Diodes
(201) 712-0555
www.fujielectric/co/jp/eng/index-e.html
General Semiconductor
Diodes
(516) 847-3000
www.gensemi.com
Motorola*
Discretes
(800) 441-2447
www.mot-sps.com/products/index.html
ON Semiconductor*
Discretes
(408) 749-0510
www.onsemi.com/home
Panasonic
LEDs
(201) 348-5217
www.panasonic.com/industrial_oem/semiconductors/
semiconductor_home.htm
Temic
IR Photo Diodes
Zener/Small Signal
Diodes
(408) 970-5700
www.temic.com
(650) 665-9301
www.vishay.com
Small Signal Discretes
(516) 543-7100
Vishay
Zetex
www.zetex.com
*Discretes formerly manufactured by Motorola are now manufactured by ON Semiconductor.
Part numbers have not been chanaged as of January 2000
www.BDTIC.com/Linear
AN84-153
Application Note 84
Inductors and Transformers
Vendor
Product
Phone Number
URL
API Delevan
Inductors
(716) 652-3600
www.delevan.com
BH Electronics
Inductors
(612) 894-9590
www.bhelectronics.com
BI Technologies
Transformers
(714) 447-2656
www.bitechnologies.com
Coilcraft
Inductors
(847) 639-6400
www.coilcraft.com
Coiltronics
Inductors/ Transformers
(561) 241-7876
www.coiltronics.com
Dale
Inductors/ Transformers
(605) 665-1627
www.vishay.com/fp/fp.html#inductors
Gowanda
Inductors
(716) 532-2234
www.gowanda.com
Midcom
Inductors/ Transformers
(800) 643-2661
www.midcom-inc.com
Murata Electronics
Inductors,
(814) 237-1431
www.murata.com
Panasonic
Inductors/ Transformers
(714) 373-7334
www.panasonic.com/industrial_oem/electronic_components/
electronic_components_inductors_coils_and_transformers.htm
Philips
Inductors
(914) 246-2811
www.acm.components.philips.com
Philips
Planar Inductors
(914) 247-2036
www.acm.components.philips.com
Pulse
Inductors
(619) 674-8100
www.pulseeng.com
Sumida
Inductors
(847) 956-0667
www.japanlink.com/sumida
Tokin
Inductors
(408) 432-8020
www.tokin.com
Vendor
Product
Phone Number
URL
Logic
Fairchild
Logic
(207) 775-4502
www.fairchildsemi.com
Intersil (formerly Harris)
Logic
(800) 442-7747
www.intersil.com
*Motorola
Logic
(800) 441-2447
www.mot-sps.com/products/index.html
*ON Semiconductor
Logic
(408) 749-0510
www.onsemi.com/home
Toshiba
Logic
Single Gate Logic
(949) 455-2000/
(714) 455-2000
www.toshiba.com/taec
*Logic Devices formerly manufactured by by Motorola are now manufactured by ON Semiconductor; there have been no changes in part numbers as of January 2000
Resistors
Vendor
Product
Phone Number
URL
Allen Bradley
Carbon Resistors
(800) 592-4888
www.ab.com
AVX
Chip Resistors
(843) 946-0524
www.avxcorp.com/products/resistors/chiprstr.htm
BI Technologies
Resistors/Resistor
Networks
(714) 447-2345
www.bitechnologies.com
Bourns
Potentiometers, SIPs
(801) 750-7253
Dale
Sense Resistors
(605) 665-9301
IRC
Sense Resistors
(361) 992-7900
www.bourns.com
www.vishayfoil.com
or www.vishay.com
www.irctt.com
RG Allen
Metal Oxide Resistors
(818) 765-8300
www.rgaco.com
TAD
Chip Resistors
(800) 508-1521
www.tadcom.com
Taiyo Yuden
Chip Resistors
(408) 573-4150
www.t-yuden.com
Thin Film Technology
Thin Film Chip Resistors
(507) 625-8445
www.thin-film.com
Tocos
SMD Potentiometers
(847) 884-6664
www.tocos.com
AN84-154
www.BDTIC.com/Linear
Application Note 84
Transistors
Vendor
Product
Phone Number
URL
Central Semiconductor
Small Signal Discretes
(516) 435-1110
www.centralsemi.com
Fairchild
MOSFETs
(408) 822-2126
www.fairchildsemi.com
IR
MOSFETs
(310) 322-3331
www.irf.com
Motorola*
Discretes
(800) 441-2447
www.mot-sps.com/products/index.html
ON Semiconductor*
Discretes
(408) 749-0510
www.onsemi.com/home
Philips
Discretes
(401) 767-4427
www-us.semiconductors.philips.com
Siliconix
MOSFETs
(800) 554-5565
www.siliconix.com
Zetex
Small Signal Discretes
(516) 543-7100
www.zetex.com
*Discretes formerly manufactured by Motorola are now manufactured by ON Semiconductor; There are no changes in part numbers as of January 2000.
Miscellaneous
Vendor
Product
Phone Number
URL
Aavid
Heat Sinks
(714) 556-2665
www.aavid.com
Epson
Crystals
(310) 787-6300
www.eea.epson.com
Infineon
(formerly Siemens
Semiconductor)
Optoelectronics
(108) 257-7910
www.infineon.com/us/opto/content.htm
Magnetics, Inc.
Toroid Cores, etc.
(800) 245-3984
www.mag-inc.com
MF Electronics
Crystal Oscillators
(914) 576-6570
www.mfelec.com
Murata Electronics
RF Devices
(770) 433-5789
www.murata.com
QT Optoelectronics
RF Switches
(408) 720-1440
www.qtopto.com
Raychem
Fuses
(800) 227-4856
www.raychem.com
RF Micro Devices
RF Semiconductors
(336) 664-1233
www.rfmd.com
RTI/Ketema
Surge Suppressors
(714) 630-0081
www.rtie.rti-corp.com
Schurter
Fuses and Holders
(707) 778-6311
www.schurterinc.com
Thermalloy
Heat Sinks
(972) 243-4321
www.thermalloy.com
Toko
RF Products
(847) 699-3430
www.tokoam.com
www.BDTIC.com/Linear
AN84-155
Application Note 84
Index
B
L
Battery Backup
LTC1558 System with LTC1435 Main System Regulator 147
Battery Chargers 114–115
Additional Feature Circuits
LT1512/LT1513, Shutdown-Controlled Disconnect 117
LTC1435/LT1620, Shutdown when Input Power is
Removed 125
LTC1510, Doubles as Main System Regulator 128
General
LT1511, Mod for NiCd and NiMH Charging 115
LT1512, 0.5 Amp 116
LT1635, 1A Shunt 129
Lead-Acid
LT1776/LT1620 , Wide VIN Range, High Efficiency 49
Lithium-Ion
LT1510, 1–2 Cell 130
LT1510, 3-Cell, without Precision Resistors 118
LT1511, 3 Amp 114
LT1512, 50mA/400mA Programmable 117
LTC1435/LT1620, 3–5 Cell 122
NiCd
LT1510, 3-Cell with –∆V Termination 120
Testing
Constant-Voltage Battery Simulator 121
Battery Simulators 121
Battery Supervisor
Single Cell Li-Ion 132
Bench Supply
100V/2A Constant Voltage, Constant Current 146
Linear Regulators. See Regulators—Linear
C
Component Vendors
Capacitors 153
Diodes 153
Inductors 154
Logic 154
Miscellaneous 155
Resistors 154
Transformers 154
Current Sensor 145
AN84-156
M
Micropower Switching Regulators. See Regulators—Switching
(Micropower)
Miscellaneous 141–149
Modulator
Switch-Frequency
for LTC1436-PLL 83
P
Power Magagement 134–141
LTC1479 PowerPath Controller
3-Diode Mode 136
Block Digram 134
PowerPath Switch Driver
LTC1473, Dual-Battery 137, 138
SMBus
LTC1623, Controls P-Channel Switch 140
LTC1623, Controls Two High-Side Switches 140
LTC1710, Switches Two Peripherals 141
LTC1710, Switches Two Peripherals with Different
Voltages 141
System
Dual Li-Ion Battery 134
VID Controlled
LT1575/LTC1706, LDO with Adjustable Output Voltage 144
Power Supply. See Regulators—Linear; Regulators—
Switching; Regulators—Switching (Micropower)
R
Regulators—Linear 107–111
Adjustable
LT1575/LTC1706, LDO with Adjustable Output Voltage 144
Battery Backup
LT1579, 6V to 5V/300mA 111
LT1579, with Added Latch for Shutdown 112
Low Dropout
LT1573, 3.3V/5A Microprocessor Supply 108
LT1575, 1.27V–2.03V VID Controlled 34
www.BDTIC.com/Linear
Application Note 84
LT1575, 5V to 3.3V with Current Limit 110
LT1575, 5V to 3.3V/5A 109
Microprocessor Supply
LT1573, 3.3V/5A 108
LT1577, Dual Regulator for Split-Plane Systems 110
Multioutput
LT1577, Dual Regulator for Split-Plane Systems 110
Regulators—Switching. See also Regulators—Switching
(Micropower)
Boost 51–53
LT1339, 5V In, 28V/6A Out Synchronous 28
LT1370, 5V In, 12V/2A Out 53
LT1377, 4V–10V In, ±12V/100mA Out 51, 52
LT1533, 3.3V to 5V/350mA Boost Converter 72
LTC1266, 2.5V–4.2V In, 5V/2A Out 55
LTC1624, 5V In, 12V/1A Out 20
Buck 4–50
12V to 3.3V/9A Hybrid 17
LT1339, 10V–18V In, 5V/50A Out 26
LT1339, 48V In, 5V/50A Out 27
LT1374, 6V–25V In, 5V/4.25A Out 23
LT1425, 12V to 5V/1A Isolated Supply 67
LT1506, 5V In, 3.3V/4A Out 29
LT1676, 12V-48V In, 5V Out 48
LT1676, 12V-48V In, 5V/0.5A Out 46
LT1676, Minimum Component-Count 46
LT1676/LTC1440, Burst Mode Configuration 48
LT1776, 10V–30V In, 5V/0.4A Out 47
LT1776, Minimum PC Board Area 47
LTC1266, 12V In, 3.3V/12A Out 7
LTC1266, 24V In, 3.3V/12A Out 8
LTC1430, 3.3V In, 1.9V/6A Out 22
LTC1430, Dual, Synchronized 17
LTC1430A, 2.5V/30A, 2-Phase Synchronous 40
LTC1433, 3.6V–12V In, 3.3V/600mA Out 11
LTC1435, 18V–28V In, 14V/15A Out 12
LTC1435, 5.5V–28V In, 2.9V/2.65V Out 9
LTC1435A 4.5V–22V In, 1.3V–2V/7A Out 36
LTC1435A, 4.5V–22V In, 1.3V–2V/7A Out 32
LTC1435A, 4.5V–22V In, 1.6V/3A Out 35
LTC1436A-PLL, 4.5V–22V In, 1.3V–2V/5A Out 33
LTC1436A-PLL/LTC1706, 4.5V–22V to 1.3V–2V/5A 143
LTC1439, 5.2V–25V In, 5V/3A, 3.3V/3A, 2.9V/2.5A Out 5
LTC1473, 28V In, 5V/3A and 12V/250mA Out 4
LTC1504, 5V In, 3.3V/0.5A Out 24
LTC1504, Improved Transient Response 25
LTC1504, SCSI-2 Terminator 25
LTC1504, Supply Splitter 25
LTC1553, 5V In, 1.8V–3.5V/14A Out 14
LTC1558, Battery Backup with LTC1435 Main System
Regulator 147
LTC1622, 2.5V–8.5V In, 2.5V/1.5A Out 43
LTC1622, Improved Transient Response 44
LTC1624, 4.5V–25V In, 3.3V/2A Out 20
LTC1624, 4.8V–20V In, 1.3V–3.0V Out 32
LTC1624/LTC1706, 4.8V–20V to 1.3V–2.0V 143
LTC1625, 12V–28V In, 12V/2.2A Out 39
LTC1625, 5V–28V In, 2.5V/5A Out 38
LTC1627, 1.8V/0.3A/3.3V/100mA 37
LTC1627, 2 Li-Ion to 3.3V/0.5A 37
LTC1627, Single Li-Ion to 2.5V/0.5A 37
Buck-Boost 56–57
LT1371, 2.7V–20V In, 5V Out 56
LTC1515, 3- or 4-Cell to 3.3V or 5V/50mA 57
Charge Pump
LT1054, Generates –5V for LTC1419 ADC 59
LTC1430, Assisted by LTC1517 22
Current-Sharing
LT1506, 6V–15V In, 5V/12A Out 30
Efficiency 149–152
Flyback 65–69
LT1172, 10V–15V In, 24V/200mA Out Isolated Flyback 68
LT1316, –48V to 5V Flyback 63
LT1425, 5V to –9V/250mA Isolated LAN Supply 65
LT1425, Fully Isolated ±15V, ±600mA Supply 66
LTC1624, 4.75V–24V In, ±50V/75mA Out 69
Forward
LT1339, 15V–25V In, 5V/6A Out 27
Hybrid
12V to 3.3V/9A Switcher plus Linear 17
Inverting 57–64
Inverting, Negative-to-Positive
LT1316, –48V to 5V Flyback 63
LT1425, –36V to –72V In, 5V/2A Out Telecom Supply 67
LT1680, –48V to 5V/6A Telecom Supply 60
Inverting, Positive-to-Negative
LT1172, 12V to –48V/120mA Telecom Supply 61
LT1370, 2.7V–13V In, –5V/3A Out 54
LT1614, 5V In, –5V/200mA Out 62
LT1776, 10V-28V In, –5V/300mA Out 50
LTC1373, 5V to –5V for LTC1419 ADC 58
LTC1433, 3V–7.5V In, –5.0V Out 11
www.BDTIC.com/Linear
AN84-157
Application Note 84
Regulators—Switching (continued)
Isolated
LT1172, 10V–15V In, 24V/200mA Out Isolated Flyback 68
LT1339, 15V–25V In, 5V/6A Out 27
LT1425, 12V to 5V/1A Isolated Supply 67
LT1425, 5V to –9V/250mA Isolated LAN Supply 65
LT1425, Fully Isolated ±15V, ±600mA Supply 66
LCD Bias
LT1316, 20V/5mA/–10V/5mA LCD 80
Low Noise 70–76
LT1533, 24V to 5V/2A Converter 74
LT1533, 3.3V to 5V/350mA Boost Converter 72
LT1533, 5V to ±12V/80mA DC/DC Converter 71
LT1533, 5V to 12V/200mA Push-Pull Converter 70
LT1533, 5V to 12V/5A Converter 75
LTC1436-PLL, 5V/3A/3.3V/0.1A Supply 81
Microprocessor Supply
12V to 3.3V/9A Hybrid 17
2.9V Regulator for Portable Pentium Processor 9
LTC1435/LTC1706, Pentium II Processor Supply 142
Mobil Pentium II VID Power Converter 32
Multioutput 76–84
LT1316, 20V/5mA/–10V/5mA LCD Bias Supply 80
LT1377, 4V–10V In, ±12V/100mA Out 51, 52
LT1425, Fully Isolated ±15V, ±600mA Supply 66
LT1533, 5V to ±12V/80mA DC/DC Converter 71
LT1776, Dual-Output SEPIC (5V/–5V) 50
LTC1263/LTC1266, 3.3V/5A/12V/60mA Supply 79
LTC1435, 5V/0.1A, 3.3V/0.5A, –5V/0.5A Supply 77
LTC1436-PLL, 5V/3A/3.3V/0.1A Supply 81
LTC1439, 5.2V–25V In, 5V/3A, 3.3V/3A, 2.9V/2.5A Out 5
LTC1473, 28V In, 5V/3A and 12V/250mA Out 4
LTC1538-AUX, 3.3V/3.5A, 5V/3A, 12V/120mA, 5V/20mA 76
LTC1624, 4.75V–24V In, ±50V/75mA Out 69
LTC1627, 1.8V/0.3A/3.3V/100mA 37
No RSENSE
LTC1625, 12V–28V In, 12V/2.2A Out 39
LTC1625, 5V–28V In, 2.5V/5A Out 38
PolyPhase
LTC1430A, 2.5V/30A, 2-Phase Synchronous 40
SEPIC
100V/2A Bench Supply 146
LT1370, 2 Li-Ion Cells to 5V/2.9A 54
LT1776, Dual Output (5V/–5V) 50
LTC1624, 5V–15V In, 12V/0.5A Out 21
Step-Down. See Regulators—Switching: Buck
Step-Up. See Regulators—Switching: Boost; Regulators—
Switching: Flyback
AN84-158
Supply Splitter
LTC1504, 5V to 2.5V/±500mA 25
Switched Capacitor
LTC1515, 3- or 4-Cell to 3.3V or 5V/50mA 57
Synchronized
LTC1430, Dual Buck 17
LTC1436A-PLL/LTC1706, 4.5V–22V to 1.3V–2V/5A 143
Telecom
LT1172, 12V to –48V/120mA Telecom Supply 61
LT1425, –36V to –72V In, 5V/2A Out Telecom Supply 67
LT1680, –48V to 5V/6A Telecom Supply 60
LTC1504, SCSI-2 Terminator 25
VID Voltage Controlled
LTC1435A, 4.5V–22V In, 1.3V–2V/7A Out 32
LTC1436A-PLL, 4.5V–22V In, 1.3V–2V/5A Out 33
LTC1436A-PLL/LTC1706, 4.5V–22V to 1.3V–2V/5A 143
LTC1553, 5V In, 1.8V–3.5V/14A Out 14
LTC1624, 4.8V–20V In, 1.3V–3.0V Out 32
LTC1624/LTC1706 4.8V–20V to 1.3V–2.0V 143
Regulators—Switching (Micropower) 85
2-Cell Digital Camera Supply 101
Boost
LT1307, Single-Cell to 3.3V/75mA Converter 86
LT1307, Single-Cell to 3.3V/75mA Converter with Output RC
Filter 86
LT1308, Single-Cell Li-Ion to 5V/1A 100
LT1317, 2-Cell to 5V/200mA 102
LT1317B, 33V/10mA Varactor Bias Suppy 105
LT1610, 2-Cell to 5V/100mA 104
LT1610, Single Cell to 3V/30mA 103
Single-Cell NiCd to 3.3V/400mA 100
Buck
LTC1174, 9V to 5V Converter 85
LTC1474, 4V–18V In, 3.3V/200mA Out 92
LTC1475, with Push-Button On/Off Control 93
LTC1626, 2.7V–6V In, 2.5V/0.25A Out 97
LTC1626, 3- or 4-Cell NiCd/NiMH to 2.5V/0.25A 97
LTC1626, Single Li-Ion Cell to 2.5V/0.25A 96
Buck-Boost
LTC1626, Single Li-Ion Cell to 3.3V/500mA 98
Charge Pump
LTC1263, Flash Memory VPP Generator 90
LTC1263, Split-Supply Generator (12V/–7V) 91
LTC1516, 2-Cell to 5V/20mA 106
LTC1516, Ultralow Quiescent Current 5V Supply 89
Flyback
1.8V-6V to 9V, for Digital Panel Meters 94
www.BDTIC.com/Linear
Application Note 84
Isolated
1.8V-6V to 9V, for Digital Panel Meters 94
LED Driver
LT1307, 25mA LED Driver 89
Multioutput
LT1317, Single-Cell Li-Ion to ±4V 103
Negative Buck
LT1307B, 7V–25V In, 5V/400mA Out 99
SEPIC
LT1317, Single-Cell Li-Ion to ±4V 103
LT1610, Sigle-Cell Li-Ion to 3.3V/100mA 105
Switched Capacitor
LTC1514, 2.7V–10V In, 3.3V and 5V Out 96
LTC1515, 4-Cells to 5V/50mA or 3.3V/50mA 95
LTC1516, 2-Cell to 5V/20mA 106
VPP Generator
LTC1263, for 2 Flash Memory Chips 90
S
Switches
High-Side
LTC1177, Short-Circuit Protected 139
LTC1623, SMBus Controlled 140
P-Channel
LTC1623, SMBus Controlled 140
PC Card
LTC1623, 3.3V/5V Switch Matrix 140
Switching Regulators. See Regulators—Switching
T
Transformer
Details
of LT1339 5V/6A Forward Converter 28
www.BDTIC.com/Linear
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
AN84-159
Application Note 84
AN84-160
www.BDTIC.com/Linear
Linear Technology Corporation
an84f LT/TP 0400 4K • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
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