Download AN113 - Power Conversion, Measurement and Pulse Circuits

Application Note 113
August 2007
Power Conversion, Measurement and Pulse Circuits
Tales From the Laboratory Notebook, 2005-2007
Jim Williams
INTRODUCTION
This ink marks LTC’s eighth circuit collection publication.1
We are continually surprised, to the point of near mystification, by these circuit amalgams seemingly limitless
appeal. Reader requests ascend rapidly upon publication,
remaining high for years, even decades. All LTC circuit
collections, despite diverse content, share this popularity, although just why remains an open question. Why is
it? Perhaps the form; compact, complete, succinct and
insular. Perhaps the freedom of selection without commitment, akin to window shopping. Or, perhaps, simply
the pleasure of new recruits for the circuit aficionados
intellectual palate. Locally based electrosociolgists, spinning elegantly contrived theories, offer explanation, but
no convincing evidence is at hand. What is certain is that
readers are attracted to these compendiums and that calls
us to attention. As such, in accordance with our mission
to serve customer preferences, this latest collection is
presented. Enjoy.
all with outputs below 600mV, are typical power sources
for such a converter.
Figure 1, an N-channel JFET I-V plot, shows drain-source
conduction under zero bias (gate and source tied together)
conditions. This property can be exploited to produce a
self-starting DC/DC converter that runs from 0.3V to 1.6V
inputs.
Figure 2 shows the circuit. Q1 and T1 form an oscillator
with T1’s secondary providing regenerative feedback to
Q1’s gate. When power is applied, Q1’s gate is at zero
volts and its drain conducts current via T1’s primary. T1’s
phase inverting secondary responds by going negative
at Q1’s gate, turning it off. T1’s primary current ceases,
its secondary collapses and oscillation commences. T1’s
primary action causes positive going “flyback” events
at Q1’s drain, which are rectified and filtered. Q2’s ≈ 2V
Note 1. Previous efforts include References 4, 6, 7 and 23-26.
+
C1
LTC1440
JFET-Based DC/DC Converter Powered From
300mV Supply
A JFET’s self-biasing characteristic can be utilized to
construct a DC/DC converter powered from as little as
300mV. Solar cells, thermopiles and single-stage fuel cells,
100Ω
5
7
0.3VIN-1.6VIN
3
1.21M*
T1
1N5817
6
Q2
TP0610L
+
BAT-85
0.001µF
Q3
VN2222L
+
6.8µF
Q1
Figure 1. Zero Volt. Biased JFET I-V Curve Shows 10mA Conduction
at 100mV, Rising Above 40mA at 500mV. Characteristic Permits
DC/DC Converter Powered From 300mV Supply.
0.01µF
–
1.18V
REF. OUT
3.83M*
5VOUT,
2mA MAX
100µF
AN113 F02
* = 1% METAL FILM RESISTOR
T1 = COILTRONICS VP1-1400
TIE WINDINGS (1, 12) (2, 4) (8, 10) (9, 11)
Q1 = PHILIPS BF-862, 2 DEVICES IN PARALLEL
Figure 2. JFET-Based DC/DC Converter Runs From 300 Millivolt
Input. Q1-T1 Oscillator Output Is Rectified and Filtered. Load
Is Isolated Until Q2 Source Reaches ≈2V, Aiding Start-Up.
Comparator and Q3 Close Loop Around Oscillator, Controlling
Q1’s On-Time to Stabilize 5V Output.
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
www.BDTIC.com/Linear
an113f
AN113-1
Application Note 113
turn-on potential isolates the load, aiding start-up. When
Q2 turns on, circuit output heads towards 5V. C1, powered
from Q2’s source, enforces output regulation by comparing
a portion of the output with its internal voltage reference.
C1’s switched output controls Q1’s on-time via Q3, forming a control loop.
The 5V output can supply up to 2mA, sufficient to power
circuitry or furnish bias to a higher power switching
regulator when larger current is required. The circuit will
start into loads of 300µA at 300mV input; 2mA loading
requires a 475mV supply. Figure 4 plots minimum input
voltage versus output current over a range of loads.
Waveforms for the circuit include the AC coupled output
(Figure 3, trace A), C1’s output (trace B) and Q1’s drain
flyback events (trace C). When the output drops below
5V, C1 goes low, turning on Q1. Q1’s resultant flyback
events continue until the 5V output is restored. This pattern repeats, maintaining the output.
Q3’s shunt control of Q1 is simple and effective, but results
in 25mA quiescent current drain. Figure 5’s modifications reduce this figure to 1mA by series switching T1’s
secondary. Here, Q3 switches series connected Q4, more
efficiently controlling Q1’s gate drive. Negative turn-off
MINIMUM INPUT VOLTAGE TO MAINTAIN
VOUT = 5V
0.5
A = 0.1V/DIV
AC COUPLED
B = 5V/DIV
C = 5V/DIV
0.3
0.2
0.1
0.0
AN113 F03
HORIZ. = 5ms/DIV
START AND RUN
0.4
0
250 500 750 1000 1250 1500 1750 2000
OUTPUT CURRENT (µA)
AN113 F04
Figure 3. JFET-based Converter Waveforms. When Supply
Output (Trace A) Decays, C1 (Trace B) Switches, Allowing Q1
to Oscillate. Resultant Flyback Events at Q1 Drain (Trace C)
Restore Supply Output.
Figure 4. JFET-Based DC/DC Converter Starts and Runs into
100µA Load at VIN = 275mV. Regulation to 2mA Is Possible,
Although Required VIN Rises to 500mV.
1M
Q4
51Ω
5
0.3VIN-1.6VIN
3
T1
7
10M
6
+
6.8V
1N4692
5VOUT
Q2
TPO610L
1N5817
+
6.8µF
100µF
Q1
*
Q1, Q4
Q1
T1
=
=
=
=
=
1N4148
1% METAL FILM RESISTOR
PHILIPS BF-862
2 DEVICES IN PARALLEL
COILTRONICS VP1-1400
TIE WINDINGS (1, 12) (2, 4)
(8, 10) (9, 11)
+
1µF
3.83M*
+
≈ –12V
0.01µF
C2
LTC1440
Q3
TP0610L
1.21M*
–
470k
1.18V
REF. OUT
AN113 F08
Figure 5. Adding Q3, Q4 and Bootstrapped Negative Bias Generator Reduces Quiescent Current. Comparator Directed Q3
Switches Q4, More Efficiently Controlling Q1’s Gate Drive. Q2 and Zener Diode Isolate All Loading During Q1 Start-Up.
AN113-2
www.BDTIC.com/Linear
an113f
Application Note 113
bias for Q4 and Q1 is bootstrapped from T1’s secondary;
the 6.8V zener holds off bias supply loading during initial
power application, aiding start-up. Figure 6’s plot of minimum input voltage versus output current shows minimal
penalty (versus Figure 4’s data) imposed by the added
quiescent current control circuitry.
Bipolar Transistor-Based 550mV Input DC/DC Converter
Bipolar transistors may be used to obtain higher output
currents, although their VBE drop raises input supply re-
quirements to 550mV. Figure 7’s curve tracer plot shows
base-emitter conduction just beginning at 450mV (25°C)
with substantial current flow beyond 500mV. Figure 8’s
circuit operates similarly to FET-based Figure 2, although
the bipolar transistor’s normally off characteristic allows
more efficient operation. Figure 9’s operating waveforms
are similar to Figure 3, except the comparator’s output state
is reversed to accommodate the bipolar transistor. Figure
10’s start-run curves show 6mA output current at 550mV
input—3 times the FET circuit’s capacity. The “run” curve
MINIMUM INPUT VOLTAGE TO MAINTAIN
VOUT = 5V
0.5
START AND RUN
0.4
0.3
0.2
0.1
0.0
250 500 750 1000 1250 1500 1750 2000
OUTPUT CURRENT (µA)
0
AN113 F06
Figure 6. Start/Run Curve for Low Quiescent Current
JFET-Based DC/DC Converter. Quiescent Current Control
Circuitry Slightly Increases Input Voltage Required to
Support Load.
Figure 7. Bipolar Transistor Base Emitter Junction I-V Curve
Shows Conduction Beginning at 450 Millivolts (25°C).
Characteristic Forms Basis of DC/DC Converter Powered From
550 Millivolts.
–
BAT-85
C1
LTC1440
+
0.01µF
0.5VIN-1.6VIN
5
1.18V
REF. OUT
3
T1
7
6
Q2
TP0610L
1N5817
1k
Q1
ZTX-849
5.1V
1N751
BAT-85
A = 0.1V/DIV
AC COUPLED
+
1.21M*
B = 5V/DIV
3.83M*
C = 5V/DIV
5VOUT
+
6.8µF
100µF
HORIZ. = 2ms/DIV
AN113 F09
AN113 F08
* = 1% METAL FILM RESISTOR
T1 = COILTRONICS VP1-1400
TIE WINDINGS (1, 12) (2, 4) (8, 10) (9, 11)
Figure 8. Bipolar Transistor-Based DC/DC Converter Runs From
500 Millivolt (25°C) Input. Q1-T1 Oscillator Output is Rectified
and Filtered. Load is Isolated Until Q2’s Source Reaches ≈2V,
Aiding Start-Up. Comparator Closes Loop Around Oscillator,
Controlling Q1’s On-Time to Stabilize 5V Output.
Figure 9. Converter Waveforms. When Output (Trace A)
Decays, C1 (Trace B) Switches, Allowing Q1 to Oscillate.
Resultant Flyback Events at Q1 Collector (Trace C)
Restore Output.
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an113f
AN113-3
Application Note 113
indicates that, once started, the circuit will run at input
voltages as low as 300mV depending on loading.
When considering these circuits’ extremely low input voltages and output power limits it is worth noting that the
transformer specified is a standard product. A transformer
specifically optimized for these applications would likely
enhance performance.
MINIMUM INPUT VOLTAGE TO MAINTAIN
VOUT = 5V
0.6
START AND RUN
0.5
RUN
0.4
0.3
0.2
0.1
0.0
0
1
2
3
4
5
OUTPUT CURRENT (mA)
6
7
AN113 F10
Figure 10. Bipolar Transistor-Based DC/DC Converter Requires
≈ 550mV (25°C) Input to Start into 0mA to 6mA Loading. Once
Running, Converter Maintains Regulation Down to 300mV Inputs
for 100µA Load.
5V to 200V Converter for APD Bias
BEFORE PROCEEDING ANY FURTHER, THE READER
IS WARNED THAT CAUTION MUST BE USED IN THE
CONSTRUCTION, TESTING AND USE OF THIS CIRCUIT.
HIGH VOLTAGE, LETHAL POTENTIALS ARE PRESENT
IN THIS CIRCUIT. EXTREME CAUTION MUST BE USED
IN WORKING WITH, AND MAKING CONNECTIONS
TO, THIS CIRCUIT. REPEAT: THIS CIRCUIT CONTAINS
DANGEROUS, HIGH VOLTAGE POTENTIALS. USE
CAUTION.
Avalanche photodiodes (APD) require high voltage bias.
Figure 11’s design provides 200V from a 5V input. The
circuit is a basic inductor flyback boost regulator with a
major important deviation. Q1, a high voltage device, has
been interposed between the LT1172 switching regulator
and the inductor. This permits the regulator to control Q1’s
high voltage switching without undergoing high voltage
stress. Q1, operating as a “cascode” with the LT1172’s internal switch, withstands L1’s high voltage flyback events.2
Diodes associated with Q1’s source terminal clamp L1
originated spikes arriving via Q1’s junction capacitance. The
Note 2. See References 1 (page 8), 2 (Appendix D), and 3.
DANGER! Lethal Potentials Present — See Text
5V
+
33µF
1N5712
100k
L1
BAS521
Q1
IRF840
0.1µF
15V
1N4702
1N5256B
30V 5%
1N5819
300Ω
OUTPUT
200V
1µF = 2s–
0.47µF
250V
1M*
BAS521
SW
5V
LT1172
VIN
+
FB
1µF
VC GND
E2 E1
6.19k*
* = 1% METAL FILM RESISTOR
L1 = 33µH, COILTRONICS UP2B
0.47µF = PANASONIC ECW-U2474KCV
1k
+
AN113 F11
1µF
Figure 11. 5V to 200V Output Converter for APD Bias. Cascoded Q1 Switches High Voltage, Allowing Low Voltage Regulator
to Control Output. Diode Clamps Protect Regulator from Transient Events; 100k Path Bootstraps Q1’s Gate Drive from L1’s
Flyback Events. Output Connected 300Ω-Diode Combination Provides Short-Circuit Protection.
AN113-4
www.BDTIC.com/Linear
an113f
Application Note 113
high voltage is rectified and filtered, forming the circuit’s
output. Feedback to the regulator stabilizes the loop and
the RC at the VC pin provides frequency compensation. The
100k path from L1 bootstraps Q1’s gate drive to about 10V,
ensuring saturation.3 The output connected 300Ω-diode
combination provides short-circuit protection by shutting
down the LT1172 if the output is accidentally grounded.
Figure 12 shows operating waveforms. Traces A and C
are LT1172 switch current and voltage, respectively. Q1’s
drain is trace B. Current ramp termination results in a high
voltage flyback event at Q1’s drain. A safely attenuated
version of the flyback appears at the LT1172 switch. The
sinosoidal signature, due to inductor ring-off between
conduction cycles, is harmless.
Battery Internal Resistance Meter
It is often desirable to determine a battery’s internal
resistance to evaluate its condition or suitability for an
application. Accurate battery resistance determination is
complicated by inherent capacitive terms which corrupt
results taken with AC-based milliohmmeters operating in
the kHz range. Figure 13, a very simplistic battery model,
shows a resistive divider with a partial shunt capacitive
term. This capacitive term introduces error in AC-based
measurement. Additionally, the battery’s unloaded internal
resistance may significantly differ from its loaded value. As
such, a realistic determination of internal resistance must
be made under loaded conditions at or near DC.
Note 3. This circuit is not a fresh contribution but, rather, a belated mea
culpa. The original version suffered temperature dependent output error
due to its gate bias bootstrap scheme. See Reference 4.
A = 0.5A/DIV
INTERNAL
IMPEDENCE
TERMS
USER
TERMINALS
B = 100V/DIV
C = 20V/DIV
AN113 F13
2µs/DIV
AN113 F12
Figure 12. Waveforms for 5V to 200V Converter Include LT1172
Switch Current and Voltage (Traces A and C, Respectively) and
Q1’s Drain Voltage (Trace B). Current Ramp Termination Results
in High Voltage Flyback Event at Q1 Drain. Safely Attenuated
Version Appears at LT1172 Switch. Sinosoidal Signature,
Due to Inductor Ring-Off Between Current Conduction Cycles,
is Harmless. All Traces Intensified Near Center Screen for
Photographic Clarity.
Figure 13. Simplistic Model Shows Battery Impedance Terms
Including Resistive and Capacitive Elements. Capacitive
Component Corrupts AC-Based Measurement Attempts to
Determine Internal DC Resistance. More Realistic Results Occur
if Battery Voltage Drop Is Measured Under Known Load.
www.BDTIC.com/Linear
an113f
AN113-5
Application Note 113
Figure 14’s circuit meets these requirements, permitting
accurate internal resistance determination of batteries
up to 13V over a range of 0.001Ω to 1.000Ω. A1, Q1 and
associated components form a closed loop current sink
which loads the battery via Q1’s drain. The 1N5821 provides
reverse battery protection. The voltage across the 0.1Ω
resistor, and hence the battery load, is determined by A1’s
“+” input voltage. This potential is alternately switched,
via S1, between 0.110V and 0.010V derived from the
2.5V reference driven resistor string. S1’s 0.5Hz square
wave switching drive comes from the CD4040 frequency
9V
IN
LTC1798
2.5V
REFERENCE
2.5V
OUT
9V
237k
1%
10k
1%
1k
1%
SWITCHED
CURRENT SINK
divider. The result of this action is a 100mA biased 1A
0.5Hz square wave load applied to the battery. The battery’s
internal resistance causes a 0.5Hz amplitude modulated
square wave to appear at the Kelvin-sensed S2-S3-A2
synchronous demodulator. The demodulator DC output
is buffered by chopper stabilized A2 which provides the
circuit output. A2’s internal 1kHz clock, level shifted by Q2,
drives the CD4040 frequency divider. One divider output
supplies the 0.5Hz square wave; a second 500Hz output
activates a charge pump, providing a –7V potential to A2.
This arrangement allows A2 output swing to zero volts.
1N5821
FORCE 1µF
S1
0.110V
0.010V
7
9
A1
LT1077
6
–
MODULATOR
+
10k
+
2.7k
9V
SYNCHRONOUS
DEMODULATOR
1
Q1
IRLR-024
5
SENSE
S2
0.001µF
A2
LT1150 CLK
–1 V –
4
+
10k
1µF
BATTERY
UNDER TEST
≈1kHz
Square Wave
–
9V
0.1Ω
1%
3
SENSE
1µF
16
10k
14 LTC6943
2
≈ 0.5Hz SQUARE WAVE
NC
100k
CLK
+V
CD4040
÷ 2048
÷2
R
FREQUENCY DIVIDER
AND CHARGE PUMP
9V
13
9V
1µF = MYLAR
SINGLE POINT GROUND
AT 0.1Ω RESISTOR
# = LTC6943 PIN #
Q1 = HEAT SINK
200pF
S3
FORCE
15
OUTPUT
0V-1.000V =
0Ω-1.000Ω
≈ –7V
Q2
2N3904
200k
+
10µF
BAT-85
AN113 F14
BAT-85
+
10µF
Figure 14. Battery Internal Resistance Is Determined by Repetitively Stepping Calibrated Discharge Current and Reading Resultant
Voltage Drop. S1-Based Modulator, Clocked From Frequency Divider, Combines with A1-Q1 Switched Current Sink to Generate
Stepped, 1 Ampere Battery Discharge Cycles. S2-S3-A2 Synchronous Demodulator Extracts Modulated Voltage Drop Information,
Provides DC Output Calibrated in Ohms.
AN113-6
www.BDTIC.com/Linear
an113f
Application Note 113
wide range of battery voltage. The floating battery simulator
is substituted for a cell in the stack and any desired voltage
directly dialed out. Figure 15’s circuit is simply a battery
powered follower (A1) with current boosted (A2) output.
The LT1021 reference and high resolution potentiometric
divider specified permits accurate output setting within
1mV. The composite amplifier unloads the divider and
drives a 680µF capacitor to approximate a battery. Diodes
preclude reverse biasing the output capacitor during
supply sequencing and the 1µF -150k combination provides
stable loop compensation. Figure 16 depicts loop response
to an input step; no overshoot or untoward dynamics occur despite A2’s huge capacitive load. The battery monitor
The circuit pulls 230µA from its 9V battery power supply,
permitting about 3000 hours battery life. Other specifications include operation down to 4V with less than 1mV
(0.001Ω) output variation, 3% accuracy and batteryunder-test range of 0.9V to 13V. Finally, note that battery
discharge current and repetition rate are easily varied
from the values given, permitting observation of battery
resistance under a variety of conditions.
Floating Output, Variable Potential Battery Simulator
Battery stack voltage monitor development (Reference 5)
is aided by a floating, variable potential battery simulator.
This capability permits monitor accuracy verification over a
18V
2X9V
1X9V
75k*
–9V
18V
IN
50k
TRIM THIS VALUE
FOR THIS VOLTAGE
TRIM
LT1021 OUT
6
10.000V
BAT-85
+
100k**
A1
LT1012
–
18V
1k
1µF
* = 1% METAL FILM RESISTOR
** = ESI DP-1311 KELVIN-VARLEY DIVIDER
150k
A2
LT1010
OUTPUT
–9V
IN4001
+
680µF
AN113 F15
Figure 15. Battery Simulator Has Floating Output Settable within 1mV.
A1 Unloads Kelvin-Varley Divider; A2 Buffers Capacitive Load.
A = 200µA/DIV
VERT. = 0.5V/DIV
B = 50µV/DIV
NOISE
AVERAGED
AC COUPLED
HORIZ. = 20ms/DIV
AN113 F16
Figure 16. 150k-1µF Compensation Network Provides
Clean Response Despite 680µF Output Capacitor.
HORIZ. = 2µs/DIV
AN113 F17
Figure 17. Battery Simulator Output (Trace B) Responds to Trace A’s
Monitor Current Pulse. Closed Loop Control and 680µF Capacitor
Maintain Simulator Output within 30µV. Noise Averaged, 50µV/
Division Sensitivity Is Required to Resolve Response.
www.BDTIC.com/Linear
an113f
AN113-7
Application Note 113
determines battery voltage by injecting current into the
battery and measuring resultant clamp voltage (again, see
Reference 5). Figure 17 shows battery simulator response
(trace B) to trace A’s monitor current pulse into the output.
Closed loop control and the 680µF capacitor limit simulator
output excursion within 30µV. This error is so small that
noise averaging techniques and a high gain oscilloscope
preamplifier are required to resolve it.4
The LTC1799’s output is divided down to form a 2-phase
925Hz square wave clock. This frequency, harmonically
unrelated to 60Hz, provides excellent immunity to harmonic
beating or mixing effects which could cause instabilities. S1
and S2 receive complementary drive, causing the FET-A1
based stage to see a chopped version of the input voltage.
A1’s square wave output is synchronously demodulated
by S3 and S4. Because these switches are synchronously
driven with the input chopper, proper amplitude and polarity information is presented to DC output amplifier A2.
This stage integrates the square wave into a DC voltage,
providing the output. The output is divided down (R2 and
R1) and fed back to the input chopper where it serves as
a zero signal reference. Gain, in this case 1000, is set by
the R1-R2 ratio. The AC coupled input stage’s DC errors
do not affect overall circuit characteristics, resulting in the
extremely low offset and drift noted.
40nVP-P Noise, 0.05µV/°C Drift, Chopped FET Amplifier
Figure 18’s circuit combines the LTC6241’s rail-to-rail
performance with a pair of extremely low noise JFETs
configured in a chopper-based carrier modulation scheme
to achieve extraordinarily low noise and DC drift. This
circuit’s performance suits demanding transducer signal
conditioning situations such as high resolution scales and
magnetic search coils.
Note 4. This may be viewed as a historic event in some thinly populated
circles. Figure 17 marks the author’s first published use of a digital
oscilloscope (Tektronix 7603/7D20), updating him to the 1980s.
5V
TO LTC201 V+ PIN
–5V
TO LTC201 V– PIN
1µF
+
+
5V
18.5kHz
V+
74C90 ÷ 10
DIV LTC1799 OUT
RSET
5V
74C74 ÷ 2
Q
54.2k*
TO
Ø1
POINTS
8
909Ω**
300Ω
INPUT
6
7
S1
S2
0.47µF
TO
Ø2
POINTS
909Ω**
1µF
Ø2
1µF
1
–
LSK389
11
10
Q
925Hz
5V
Ø1
1µF
5V
A1
1/2 LTC6241HV
9
499Ω**
Ø2
100k
+
1µF
3
2
S3
S4
10M
100k
–5V
0.001µF
10k
240k
14
15
16
Ø1
1µF
* = 0.1% METAL FILM RESISTOR
** = 1% METAL FILM RESISTOR
= LTC201 QUAD
= LSK389
= LINEAR INTEGRATED SYSTEMS
FREMONT, CA
NOISE = 40nVP-P 0.1Hz TO 10Hz
OFFSET = 1µV
DRIFT = 0.05µV/°C
R2 + 1
GAIN =
R1
OPEN-LOOP GAIN = 10 9
I BIAS = 150pA
–
A1
1/2 LTC6241HV
OUTPUT
+
R2
10k
R1
10Ω
AN113 F18
Figure 18. Chopped FET Amplifier Has 40nVP-P Noise and 0.05µV/°C Drift. DC Input Is Carrier Modulated, Amplified by A1,
Demodulated to DC and Fed Back From A2. 925Hz Carrier Clock Prevents Interaction with 60Hz Line Originated Components.
AN113-8
www.BDTIC.com/Linear
an113f
Application Note 113
Figure 19, measured over a 50 second interval, shows
40nVP-P noise in a 0.1Hz to 10Hz bandwidth. This is spectacularly low noise for a JFET-based design and is directly
attributable to input pair area and current density.
Wideband, Chopper Stabilized FET Amplifier
The previous circuit’s bandwidth is limited because the
chopping occurs within the signal path. Figure 20’s circuit
circumvents this restriction by placing the stabilizing element in parallel with the signal path. This maintains DC
performance although noise triples to 125nV in a 0.1Hz
to 10Hz bandpass.
FET pair Q1 differentially feeds A2 to form a simple low
noise op amp. Feedback, provided by R1 and R2, sets
closed loop gain (in this case 1000) in the usual fashion.
Although Q1 has extraordinarily low noise characteristics,
its offset and drift are relatively high. A1, a chopper stabilized amplifier, corrects these deficiencies. It does this by
measuring the difference between the amplifier’s inputs and
adjusting Q1A’s channel current to minimize the difference.
Q1’s drain values ensure that A1 will be able to capture the
VERT. =
20nV/DIV
AN113 F19
HORIZ. = 5sec/DIV
Figure 19. Amplifier 0.1Hz to 10Hz Noise Measures
40nVP-P in 50 Second Sample Period.
0.02µF
499Ω*
–
* = 1% METAL FILM
RESISTOR
VOS = 3µV
TCVOS = 0.05µV°/C
NOISE = 125nV
0.1Hz -10Hz
Ib = 500pA
R2 + 1
IN
GAIN =
R1
6
A VOL = >10
BW = 29kHz
A1
LTC2050HV
100k
5V
+
2k
2k*
1k*
0.22µF
499Ω*
–
A2
LT1797
0.02µF
+
100k
A
R2
10k
B
Q1
LSK-389
R1
10Ω
499Ω*
AN113 F20
–5V
Figure 20. Placing Stabilizing Amplifier Outside Signal Path Permits Bandwidth Increase
over Previous Circuit. Noise Triples to 125nV in 0.1Hz to 10Hz Bandpass.
VERT. =
50nV/DIV
VERT. =
0.2V/DIV
HORIZ. = 20µs/DIV
AN113 F21
Figure 21. Figure 20 Responds to a 1mV Input. 12µs Rise
Time Indicates 29kHz Bandwidth at A = 1000.
HORIZ. = 1sec/DIV
AN113 F22
Figure 22. Chopper Stabilized FET Pair Noise
Measures 125nV in 0.1Hz to 10Hz Bandpass.
www.BDTIC.com/Linear
an113f
AN113-9
Application Note 113
offset. A1 supplies whatever current is required to Q1A’s
channel to force offset within 5µV. Additionally, A1’s low
bias current does not appreciably add to the overall 500pA
amplifier bias current. As shown, the amplifier is set up
for a noninverting gain of 1000, although other gains and
inverting operation are possible.
better than any monolithic chopper stabilized amplifier,
while retaining low offset and drift.
Submicroampere RMS Current Measurement for
Quartz Crystals
Quartz crystal RMS operating current is critical to longterm stability, temperature coefficient and reliability. Accurate determination of RMS crystal current, especially
in micropower types, is complicated by the necessity to
minimize introduced parasitics, particularly capacitance,
which corrupt crystal operation. Figure 23’s high gain, low
noise amplifier combines with a commercially available
Placing the offset correction in parallel with the signal path
permits high bandwidth. Figure 21 shows response to a
1mV input. The 12µs risetime indicates 29kHz bandwidth
at A = 1000.
Figure 22’s photo measures noise in a 0.1Hz to 10Hz
bandwidth. The performance obtained is almost 6 times
CRYSTAL OSCILLATOR
TEST CIRCUIT
TEKTRONIX CT-1
CURRENT PROBE
5mV/mA (5µV/µA)
32.768kHz
2V
2M
2V
+
1M
LTC1440
CI
–
1.2M
10pF
A = 224,000
A = 1120 (CT-1 GAIN ERROR AT
32.768 kHz ≈ 12%; SEE TEXT)
+
49.9Ω*
–
A = 200
5V
A1
LT1028
+
39pF
1.5k*
1µF
A2
LT1222
–
49.9Ω*
+
1740Ω*
–5V RCAL
39Ω
(SEE TEXT)
1k
A3
LT1222
+
–
825Ω*
A4
LT1222
–
825Ω*
10µF
61.9Ω*
49.9Ω*
43k
63.4Ω*
5V
32.7kHz
5V
BANDPASS FILTER
* = 1% METAL FILM RESISTOR
= EPSON C-100R
10µF, 1µF = WIMA MKS-2
LTC1563-2
LP
V+
SA
21k*
INVA
20k*
LPA
LPB
42.2k*
5.62k*
I1
0.01µF
+V OUT
LTC1968
I2
10µF
–
24.9k*
A5
LT1077
+
1µF
R
OUT
0V-1V =
0µA-1µA
RMS
84.5k*
E
INVB
SB
G
10k
GND
V–
RMS DC
CONVERTER
0.1µF
AN113 F23
EN
10k
–5V
5V
Figure 23. A1 to A4 Furnish Gain of >200,000 to Current Probe, Permitting Submicroamp Crystal Current
Measurement. LTC1563-2 Bandpass Filter Smooths Residual Noise While Providing Unity Gain at 32.768kHz.
LTC1968 Supplies RMS Calibrated Output.
AN113-10
www.BDTIC.com/Linear
an113f
Application Note 113
closed core current probe to permit the measurement. An
RMS-to-DC converter supplies the RMS value. The quartz
crystal test circuit shown in dashed lines exemplifies a
typical measurement situation. The Tektronix CT-1 current
probe monitors crystal current while introducing minimal
parasitic loading. The probe’s 50Ω terminated output feeds
A1. A1 and A2 take a closed loop gain of 1120; excess
gain over a nominal gain of 1000 corrects for the CT-1’s
12% low frequency gain error at 32.768kHz.5 A3 and A4
contribute a gain of 200, resulting in total amplifier gain
of 224,000. This figure results in a 1V/µA scale factor at
A4 referred to the gain corrected CT-1’s output. A4’s
LTC1563-2 bandpass filtered output feeds an LTC1968-A5
based RMS → DC converter which provides the circuits
output. The signal processing path constitutes an extremely
narrow band amplifier tuned to the crystal’s frequency.
Figure 24 shows typical circuit waveforms. Crystal drive,
taken at C1’s output (trace A), causes a 530nA RMS
crystal current which is represented at A4’s output
(trace B) and the RMS → DC converter input (trace C).
Peaking visible in trace B’s unfiltered presentation derives
from parasitic paths shunting the crystal.
Typical circuit accuracy is 5%. Uncertainty terms include
the transformer’s tolerances, its ≈1.5pF loading and resistor/RMS → DC converter error. Calibrating the circuit
A = 2V/DIV
B = 1µA/DIV
C = 1µA/DIV
HORIZ. = 10µs/DIV
AN113 F24
Figure 24. C1’s 32.768kHz Output (Trace A) and Crystal Current
Monitored at A4 Output (Trace B). RMS Converter Input Is
Trace C. Peaks in Trace B’s Unfiltered Waveform Derive From
Inherent and Parasitic Paths Shunting Crystal.
reduces error to less than 1%. Calibration involves driving
the transformer with 1µA at 32.7kHz. This is facilitated by
biasing a 100k, 0.1% resistor with an oscillator set at 0.100V
output. The output voltage should be verified with an RMS
voltmeter having appropriate accuracy (see Reference 8’s
Appendix B). Figure 23 is calibrated by padding A2’s gain
with a small resistive correction, typically 39Ω.
Direct Reading Quartz Crystal-Based Remote
Thermometer
Although quartz crystals have been used as temperature
sensors (see References 7 and 10) there has been almost
no adaptation of this technology. The primary impediment
has been lack of standard product quartz crystal temperature sensors. The advantages of quartz-based temperature
sensing include nearly purely digital signal conditioning,
good stability and a direct, noise immune digital output
ideally suited to remote sensing.
Figure 25 utilizes an economical, commercially available
(Reference 9) quartz temperature sensor in a direct reading thermometer scheme suited to remote data collection.
An LTC485 transceiver, set up in transmit mode, forms
a quartz-based, Pierce class oscillator. The transceiver’s
differential line driving outputs provide frequency coded
temperature data to a 1000 foot cable run. A second RS485
transceiver differentially receives the data, presenting a
single ended output to the PIC-16F73 processor. The processor converts the frequency coded temperature data to
its °C equivalent, which appears on the display. Figure 26
is a software listing of the processor’s program.6 Accuracy
over a sensed –40°C to 85°C range is about 2%.
Note 5. The validity of this gain error correction at one sinusoidal
frequency –32.768kHz—was investigated with a 7-sample group of
Tektronix CT-1s. Device outputs were collectively within 0.5% of 12%
down for a 1µA 32.768kHz sinusoidal input current. Although this tends
to support the measurement scheme, it is worth noting that these results
are as measured. Tektronix does not guarantee performance below the
specified –3dB 25kHz low frequency roll-off.
Note 6. Mark Thoren of LTC designed the processor-based circuitry and
authored Figure 26’s software.
www.BDTIC.com/Linear
an113f
AN113-11
Application Note 113
10M
1M
100pF
5V
6
4
2
5
6
100Ω
7
LTC485
3
7
8
100k
1000 FEET
TWISTED PAIR
8
+
–
5
1k
1
LTC485
5V
3
VCC
2
10k
5V
Y1
10pF
3.3M
10pF
15pF
D7
D6
D5
CONTRAST D4
EN
RW
GND
RS
2 × 16 CHARACTER
LCD DISPLAY
(OPTREX DMC-162498
OR SIMILAR)
SENSOR IS Y1 = EPSON
HTS-206
40kHz/25°C
–29.6ppm/°C
19
17
16
15
14
13
12
11
28
27
26
25
24
23
22
21
7
6
5
4
3
2
VDO
RC7
RC6
RC5
RC4
RC3
RC2
RC1
RC0
RB7
RB6
RB5
RB4
RB3
RB2
RB1
RB0
28
5V
0.1µF
OSC1
OSC2
MCLR
9
Y1
10MHz
10
1 R1, 10k
5V
PIC-16F73
RA5
RA4
RA3
RA2
RA1
RA0
D1
BAV74LT1
8
18
AN113 F25
Figure 25. Quartz Crystal-Based Remote Thermometer Has 2% Accuracy over –40°C to 85°C Sensed Range,
Drives 1000ft Cable. RS-485 Transceiver Oscillates at Y1 Quartz Sensor Determined Frequency and Drives
Cable. Second Transceiver Receives Data and Feeds Processor. Display Reads Directly in °C.
/*
Thermometer based on Epson HT206 temperature sensing crystal.
Output is to a standard Epson HD447980 based alphanumeric
LCD display. LCD driver functions are part of compiler library
Written for CSS compiler version 3.182
*/
#include <16F73.h>
#device adc=8
#fuses NOWDT, HS, PUT, NOPROTECT, NOBROWNOUT
#use delay (clock=10000000) // Tell compiler how fast we’re going
#use rs232(baud=9600, parity=N, xmit=PIN_C6, rcv=PIN_C7, bits=8)
#include “lcd.c” // LCD driver functions
void main( )
{
int16 temp;
unsigned int16 f;
setup_adc_ports(NO_ANALOGS);
setup_adc(ADC_OFF);
setup_spi (FALSE);
setup_counters (RTCC_INTERNAL, RTCC_DIV_1);
setup_timer_1 (T1_EXTERNAL|T1_DIV_BV_1);
setup_timer_2 (T2_DISABLED, 0, 1);
lcd_init( ); // Initialize LCD
while(1)
(
set_timer1(0);
setup_timer_1(T1_EXTERNAL|T1_DIV_BY_1);
delay_ms(845);
delay_us(412);
setup_timer_1(T1_DISABLED);
f = get_timer1();
temp = 33770 – f + 25;
//**
//**
//
//
//
//
//
//
//
Reset counter
Turn on counter
0.845412 is the magic time to count
-it gives 1 less plus per degree C
turn off counter
Read result
Convert to temperature
At this point ‘f’ is the temperature in degrees C
**//
For this experiment, dump to standard HD44780 type LCD display **//
lcd_putc(‘\f’);
lcd_gotoxy(1, 1);
printf(lcd_putc, “%ld”, temp);
}
// Clear screen
// Goto home position
// And display result
}
Figure 26. Software Listing for PIC Processor Program. Code Converts Frequency to °C Equivalent, Drives Display.
AN113-12
www.BDTIC.com/Linear
an113f
Application Note 113
5V
120Ω
CORE
OSCILLATOR
120Ω
12k
1µF
0.1µF
NC
Q1
2N3906
Q2
2N3904
5V
74HC4060 ÷ 16
D1 Q1
I1
7.5k
CLK1
I2
PI CLR1
CLK2
D2 Q2
C1
20pF
D1
FOUT
1Hz TO 100MHz
74AHC74 ÷ 4
01
Q4
CLR
P2 CLR2
5V
FEEDBACK
DIVIDER
≈–3VDC HERE
1Hz
2k
–V BIAS GENERATOR
510k
510k
5V
+
SERVO AMP.
10µF
10µF
33Ω
20Ω
–
CHARGE PUMP
7
+
CLK
A1
LTC1150
1µF
+
9
10k
10k
6
100pF**
10
12
∑
14
11
5
LTC6943
1
4
100Ω*
100pF**
D1 = JPAD-500; LINEAR SYSTEMS
= 74ACH14, GROUND UNUSED INPUTS
BAT-85
IRC-MAR-6, 1% RESISTOR
WIMA FKP-2
WIMA MKS-2
AIR WIRED, MINIMUM AREA, MINIMIZE STRAY
CAPACITANCE, RELIEVE GROUND PLANE
20pF = AG MICA
DASHED AREA = LEAKAGE GUARD TRACE
*
**
1µF, 0.22µF
CIRCLED AREA
=
=
=
=
=
6.19k*
0.22µF
2
LT1460
2.5V
5V
1µF
LINEARITY
(60MHz)
16
2k
100MHz
1k
13
1.5k*
AN113 F27
VIN – 0V TO 5V
Figure 27. 1Hz to 100MHz V→ F Converter Has 160dB Dynamic Range, Runs From 5V Supply. Input Biased Servo
Amplifier Controls Core Oscillator, Stabilizing Circuit’s Operating Point. Wide Range Operation Derives From Core
Oscillator Characteristics, Divider/Charge Pump-Based Feedback and A1’s Low DC Input Errors.
www.BDTIC.com/Linear
an113f
AN113-13
Application Note 113
1Hz-100MHz V→ F Converter
Figure 27’s circuit achieves a wider dynamic range and
higher output frequency than any commercially available
voltage to frequency (V→ F) converter. Its 100MHz fullscale output (10% overrange to 110MHz is provided) is
at least ten times faster than available units. The circuit’s
160dB dynamic range (8 decades) allows continuous
operation down to 1Hz. Additional specifications include
0.1% linearity, 250ppm/°C gain temperature coefficient,
1Hz/°C zero shift, 0.1% frequency shift for VSUPPLY = 5V
± 10% and a 0V to 5V input range. A single 5V supply
powers the circuit.7
A1, a chopper stabilized amplifier, servo biases a crude
but wide range core oscillator in Figure 27. The core
oscillator drives a charge pump via digital dividers. The
averaged difference between the charge pump’s output
and the circuit’s input appears at a summing node (∑) and
biases A1, closing a control loop around the wide range
core oscillator. The circuit’s extraordinary dynamic range
and high speed derive from core oscillator characteristics,
divider/charge pump-based feedback and A1’s low DC input
errors. A1 and the LTC6943-based charge pump stabilize
circuit operating point, contributing high linearity and low
drift. A1’s low offset drift allows the circuit’s 50nV/Hz gain
slope, permitting operation down to 1Hz at 25°C.
The positive input voltage causes A1 to swing negatively, biasing Q1. Q1’s resultant collector current ramps
C1 (trace A, Figure 28) until Schmitt trigger inverter I1’s
output (trace B) goes low, discharging C1 via Q2. C1’s
discharge resets I1’s output to its high state, Q2 goes off
and the ramp-and-reset action continues. D1’s leakage
dominates all parasitic currents in the core oscillator,
ensuring operation down to 1Hz. The ÷64 divider chain’s
output clocks the LTC6943-based charge pump. The charge
pump’s two sections operate out-of-phase, resulting in
charge transfer at each clock transition. Charge pump stability is primarily determined by the LT1460 2.5V reference,
the switches low charge injection and the 100pF capacitors. The 0.22µF capacitor averages the pumping action
to DC. The averaged difference between the input derived
current and the charge pump feedback signal is amplified
by A1, which biases Q1 to control circuit operating point.
Core oscillator nonlinearity and drift are compensated by
A1’s servo action, resulting in the high linearity and low
drift previously noted. A1’s 1µF capacitor provides stable
loop compensation. Figure 29 shows loop response
(trace B) to an input step (trace A) is well controlled.
Some special techniques enable this circuit to achieve its
specifications. D1’s leakage current dominates all parasitic
currents at I1’s input; hence Q1 must always source current
to sustain oscillation, assuring operation down to 1Hz. The
100MHz full-scale frequency sets stringent restrictions
on core oscillator cycle time. Only 10ns is available for a
Note 7. Reference 12 (1986) contains a circuit with comparable
specifications although considerably more complex than the one presented
here. The advent of high speed CMOS logic permitted replacing the earlier
designs ECL elements, facilitating a dramatic decrease in complexity.
Comparing the designs permits viewing the impact a technology shift can
have in realizing a circuit function. In this case, the effect is pervasive,
directly or indirectly influencing nearly every aspect of circuit operation.
While circuit architecture is consistent, this incarnation is substantially and
favorably altered.
1V/DIV
AC COUPLED
2V/DIV
(ON 0.05VDC)
2V/DIV
1V/DIV
HORIZ. = 5ns/DIV
AN113 F28
Figure 28. V→ F Operation at 40MHz. Core Oscillator Waveforms Viewed in 670MHz Real Time Bandwidth Include Q1
Collector (Trace A) and Q2 Emitter (Trace B). Ramp-and-Reset
Operating Characteristic Is Apparent; Reset Duration of 6ns
Permits 100MHz Repetition Rate.
AN113-14
HORIZ. = 10ms/DIV
AN113 F29
Figure 29. Response (Trace B) to an Input Step (Trace A)
Shows 30ms Settling Time at Summing Junction (Σ). A1’s
1µF Capacitor Shapes Response, Stabilizing Feedback Loop.
Clamped Response on Negative Going Input Step is Due to
Summing Junction Limiting.
www.BDTIC.com/Linear
an113f
Application Note 113
complete ramp-and-reset sequence. The ultimate speed
limitation is the reset interval. Figure 28, trace B, shows
a 6ns interval, comfortably within the 10ns limit.
A scaled resistive path from the input to the charge pump
corrects small nonlinearities due to residual charge injection. This input derived correction is effective because
the charge injections effect varies directly with input
determined frequency.
Prototype and small lot construction may proceed using
the schematic and its notes, but component selection
should be considered for volume production. Figure 30
lists applicable components and their selection targets.
To calibrate this circuit apply 5.000V and trim the 100MHz
adjustment for a 100.0MHz output. Next, ground the input
and adjust the 1Hz trim for 1Hz output. Allow for long settling time, as charge pump update rate at this frequency is
once every 32 seconds. Note that this trim accommodates
either offset polarity because of the –V bias derived from
A1’s clock output. Finally, set the 60MHz adjustment for
60.0MHz with 3.000VIN. Repeat these adjustments until
all three points are fixed.
Delayed Pulse Generator with Variable Time Phase,
Low Jitter Trigger Output
Fast circuitry often requires a pulse generator that also
supplies a variable time phase trigger output. It is desirable
that the main output pulse occurrence be continuously
settable from before to after the trigger output with low
time jitter. Figure 31’s circuit produces a 360ps risetime
output pulse with trigger output time phase variable from
–30ns to 100ns. Jitter is 40ps.
Q1 and Q2 form a current source that charges the 1000pF
capacitor. When the LTC1799 clock is high (trace A, Figure 32) both Q3 and Q4 are on. The current source is off
and A2’s output (trace B) is at ground. C1’s latch input
prevents it from responding and its output remains high.
When the clock goes low, C1’s latch input is disabled and
its output drops low. The Q3 and Q4 collectors lift and
Q2 comes on, delivering constant current to the 1000pF
capacitor. The resulting linear ramp at A2 (trace B) is applied to bounded current summing amplifiers A3 and A4.
Both amplifiers compare ramp induced current with fixed,
opposite polarity currents derived from A1 – Q6. A1-Q6,
in turn, is referred to the +5 supply rail which also sets
Q1-Q2 current and hence, ramp slope. This ratiometric
connection promotes supply rejection. When A4 and A3
(traces C and F, respectively) come out of diode bound
and cross zero, comparators C2 and C1 (traces D and G,
respectively) are heavily overdriven and switch rapidly.
C2’s output path includes components which form trace
E’s trigger output pulse. C1 triggers output pulse generator
Q5, operating in avalanche mode (trace H).8
The “delay adjust” control sets the ramp amplitude that
A3-C1 switches the main output at, providing the desired
variable time phase with respect to the A4-C2 controlled
trigger output. Time jitter between C1 and C2 outputs is
minimized because A3 and A4 effectively multiply ramp
transition rate as their outputs enter the active region,
provide gain and cross zero.
Note 8. Avalanche mode pulse generation is a subtle, arcane technique
requiring extensive discussion. The text’s cavalier treatment is deliberately
brief in order to maintain focus on this circuit’s low timing jitter
characteristics. More studious coverage can be found in References 13-22.
SELECTION PARAMETER
(25°C)
TYPICAL YIELD
(%)
ICER < 20pA at 3V
90
Q2
IEBO < 20pA at 3V
90
D1
IREV < 500pA, > 75pA at 3V
80
I1
IIN < 25pA
80
A1
IB < 5pA at VSUPPLY = 5V
90
74ACH74
Operate with 3.6nS Wide
(50% Point) Input Pulse
80
COMPONENT
Q1
Figure 30. Selection Criteria for Components Ensure V→ F Performance. First Five Entries
Enhance Operation Below 100Hz. Last Entry Assures Reliable Feedback Divider Operation.
www.BDTIC.com/Linear
an113f
AN113-15
+
221Ω
P
0.001µF
274Ω*
162Ω*
100Ω*
(100ns
CALIB.)
5V
Q2
www.BDTIC.com/Linear
M
P
PNP
NPN
*
**
A1
A2-A4
C1, C2
L1
L2
+
–5V
4.7k
–5V
4.7k
10µF
10k*
+
A1
330Ω
330Ω
47pF
+
–
+V
5V
5V
DIV
10k
25Ω
CLOCK
–5V
Q6
250Ω
200Ω*
+
22µF
2k*
(–30ns
CALIB.)
499Ω*
200Ω*
+
–
+
–
4.02k*
A4
4.02k*
A3
200Ω*
SLOPE AMPLIFIERS
100V
P6KE100A
≈60V
+L
–
C1
+
–
L
C2
5V
1k
330Ω
30pF
BAT-85
5V
100Ω
L2
13k
10k
10pF
74AHC14
470Ω
RINGING
1N5711
6 FERRITE
BEADS
5pF
(SEE NOTES)
SLOPE COMPARATORS
M
1µF
100V
1k
10nS
0.25V
TRIGGER
OUTPUT
200Ω**
200Ω**
200Ω**
2N2501/
2N2369
Q5
(SELECTED–
SEE TEXT)
3pF
AN113 F31
50Ω
200Ω**
PULSE OUTPUT
SMA CONNECTOR
MINIMIZE LEAD LENGTHS –
MOUNT Q5 EMITTER AND
ASSOCIATED 200Ω
RESISTORS DIRECTLY AT
OUTPUT CONNECTOR.
GROUND 200Ω RESISTORS
DIRECTLY AT OUTPUT
CONNECTOR.
0.5pF TO 3pF
EDGE TIME/PEAKING
40", 50Ω COILED HARD LINE
AVALANCHE PULSE OUTPUT STAGE
Figure 31. Pulse Generator Output Time Phase Varies –30ns to 100ns with Respect to Trigger Output; Jitter Is 40ps. Clocked Ramp
at A2 Produces Variable (A3-C1) and Fixed (A4-C2) Delays Driving Pulse and Trigger Outputs, Respectively. A3-A4 Provide Gain to
Comparators at Trip Point, Minimizing Time Jitter Between Outputs.
GND
S
A2
100Ω
6.19k*
M
4.7µF
100V
DELAY ADJUST
–30ns TO 100ns
TRIP POINT BIAS
AVALANCHE
BIAS
ADJUST
100Ω
LTC1799
OUT
1.2k*
49.9k*
39µF
10k
100V
1N4148
2N5087
2N2369
1% FILM RESISTOR
1% FILM, 1206 SIZE
LT1006
LT1809
LT1394
COILCRAFT D03316P-224
1TURN#22 WIRE 0.05”
DIAMETER AIR CORE
= FERRITE BEAD, FERRONICS#21-110J
= POLYPROPELENE
= MYLAR
=
=
=
=
=
=
=
=
=
=
Q4
Q3
1µF
FB
300Ω
VC
SW
RAMP GENERATOR
10µF
Q1
LT1082
GND
VIN
BAV-21
AVALANCHE HIGH VOLTAGE BIAS
L1
220µH
+
AN113-16
–
5V
Application Note 113
an113f
Application Note 113
The A3-A4 amplifier gain is the key to low jitter between
C1 and C2’s switching times. The amplifiers augment
the comparator’s relatively low gain, promoting decisive
switching despite the ramp input. Figure 33 shows A4
(trace A)-C2 (trace B) response to the ramp crossing the
trip point. As the ramp nears the trip point A4 comes out
of bound, providing an amplified version of the ramp’s
transition rate to C2. C2 responds by switching decisively
6ns after A4 crosses zero volts at center screen. A3-C1
waveforms are identical. Figure 34, Q5’s pulse output, taken
with the oscilloscope synchronized to the trigger output,
shows 40ps jitter in a 3.9GHz sampled bandpass.
Circuit calibration is accomplished by first adjusting the
“-30ns cal” so the main pulse output occurs 30ns before
the trigger output with the “Delay Adjust” set to minimum.
Next, with the “Delay Adjust” set to maximum, trim the
“100ns cal” so the main pulse output occurs 100ns after
the trigger output. Slight interaction between the 30ns and
100ns trims may require repeating their adjustments until
both points are calibrated. As mentioned, the avalanche
output stage is illustrative only and not detailed in this
discussion. Its optimization and calibration are covered
in Reference 13.
A = 5V/DIV
B = 5V/DIV
C = 2V/DIV
A = 0.5V/DIV
D = 5V/DIV
E = 5V/DIV
F = 2V/DIV
B = 1V/DIV
G = 5V/DIV
H = 20V/DIV
HORIZ. = 100ns/DIV
AN113 F32
HORIZ. = 10ns/DIV
Figure 32. Low Jitter Delayed Pulser Waveforms Include
Clock (Trace A), A2 Ramp (B), A4 (C), C2 (D), Trigger Output (E),
A3 (F), C1 (G) and Delayed Output Pulse (H). Trigger-to-Output
Pulse Delay Is Continuously Variable From –30ns to 100ns.
AN113 F33
Figure 33. A4 (Trace A) - C2 (Trace B) Response to A2’s Ramp
Crossing Trip Point. C2 Goes High 6ns after A4 Crosses Zero
(Center Screen). A3-C1 Waveforms Are Identical.
VERT. =
2V/DIV
HORIZ. = 100ps/DIV
AN113 F34
Figure 34. Main Pulse Output Synchronized to Trigger Output
Shows 40ps Jitter in 3.9GHz. Sampled Bandpass.
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Application Note 113
REFERENCES
1. Williams, Jim, “Bias Voltage and Current Sense
Circuits for Avalanche Photodiodes,” Linear Technology Corporation, Application Note 92, November
2002, p. 8.
2. Williams, Jim, “Switching Regulators for Poets,” Appendix D, Linear Technology Corporation, Application
Note 25, September 1987.
3. Hickman, R. W. and Hunt, F. V., “On Electronic Voltage
Stabilizers,” “Cascode,” Review of Scientific Instruments, January 1939, p. 6-21, 16.
4. Williams, Jim, “Signal Sources, Conditioners and
Power Circuitry,” Linear Technology Corporation,
Application Note 98, November 2004, p. 20-21.
5. Williams, Jim, and Thoren, Mark, “Developments in
Battery Stack Voltage Measurement,” Application Note
112, Linear Technology Corporation, March 2007.
6. Williams, Jim, “Measurement and Control Circuit
Collection,” Linear Technology Corporation, Application Note 45, June 1991. p. 1-3.
7. Williams, Jim, “Practical Circuitry for Measurement
and Control Problems,” Linear Technology Corporation, Application Note 61, August 1994. p. 13-15.
8. Williams, Jim, “Instrumentation Circuitry Using RMSto-DC Converters,” Linear Technology Corporation,
Application Note 106, February 2007. p. 8-9.
9. Seiko Epson Corp. Crystal Catalog. Models HTS-206
and C-100R. See also, p. 10-11, “Drive Level.”
10. Benjaminson, Albert, “The Linear Quartz Thermometer—A New Tool for Measuring Absolute and Difference Temperatures,” Hewlett-Packard Journal,
March 1965.
11. Williams, Jim, “Applications Considerations and
Circuits for a New Chopper Stabilized Op Amp,”
1Hz-30MHz V→ F, Linear Technology Corporation,
Application Note 9, p. 14-15.
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12. Williams, Jim, “Designs for High Performance Voltage-to-Frequency Converters,” “1Hz-100MHz V → F
Converter,” p. 1-3, Linear Technology Corporation,
Application Note 14, March 1986.
13. Williams, Jim, “Slew Rate Verification for Wideband
Amplifiers,” Linear Technology Corporation, Application Note 94, May 2003.
14. Braatz, Dennis, “Avalanche Pulse Generators,”
Private Communication, Tektronix, Inc. 2003.
15. Tektronix, Inc., Type 111 Pretrigger Pulse Generator
Operating and Service Manual, Tektronix, Inc. 1960.
16. Hass, Isy, “Millimicrosecond Avalanche Switching
Circuit Utilizing Double-Diffused Silicon Transistors,” Fairchild Semiconductor, Application Note 8/2,
December 1961.
17. Beeson, R. H., Haas, I., Grinich, V. H., “Thermal Response of Transistors in Avalanche Mode,” Fairchild
Semiconductor, Technical Paper 6, October 1959.
18. G. B. B. Chaplin, “A Method of Designing Transistor
Avalanche Circuits with Applications to a Sensitive
Transistor Oscilloscope,” paper presented at the 1958
IRE-AIEE Solid State Circuits Conference, Philadelphia,
PA., February 1958.
19. Motorola, Inc., “Avalanche Mode Switching,” Chapter 9,
p. 285-304. Motorola Transistor Handbook, 1963.
20. Williams, Jim, “A Seven-Nanosecond Comparator for Single Supply Operation,” “Programmable
Subnanosecond Delayed Pulse Generator,” p. 32-34,
Linear Technology Corporation, Application Note 72,
May 1998.
21. D. J. Hamilton, F. H. Shaver, P. G. Griffith, “Avalanche
Transistor Circuits for Generating Rectangular Pulses,”
Electronic Engineering, December 1962.
22. R. B. Seeds, “Triggering of Avalanche Transistor Pulse
Circuits,” Technical Report No. 1653-1, August 5,
1960, Solid-State Electronics Laboratory, Stanford
Electronics Laboratories, Stanford University, Stanford, California.
www.BDTIC.com/Linear
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Application Note 113
23. Markell, R. Editor, “Linear Technology Magazine Circuit
Collection, Volume I,” Linear Technology Corporation,
Application Note 52, January 1993.
26. Williams, Jim, “Circuitry for Signal Conditioning and
Power Conversion,” Linear Technology Corporation,
Application Note 75, March 1999.
24. Markell, R. Editor, “Linear Technology Magazine
Circuit Collection, Volume II,” Linear Technology
Corporation, Application Note 66, August 1996.
27. Williams, Jim, “Instrumentation Applications for a
Monolithic Oscillator,” Linear Technology Corporation, Application Note 93, February 2003. “Chopped
Amplifiers,” p. 9-10.
25. Markell, R. Editor, “Linear Technology Magazine
Circuit Collection, Volume III,” Linear Technology
Corporation, Application Note 67, September 1996.
www.BDTIC.com/Linear
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
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Application Note 113
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Linear Technology Corporation
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