Download Licentiate Thesis

On the design of hybrid DC-breakers consisting of a
mechanical switch and semiconductor devices
JESPER MAGNUSSON
Licenciate Thesis
Stockholm, Sweden 2015
TRITA-EE 2015:011
ISSN 1653-5146
ISBN 978-91-7595-481-3
KTH Skolan för Elektro- och Systemteknik
Avd. Elektroteknisk teori och konstruktion
SE-100 44 Stockholm
SWEDEN
Akademisk avhandling som med tillstånd av Kungl Tekniska högskolan framlägges
till offentlig granskning för avläggande av teknologie licenciatexamen i elektro- och
systemteknik fredagen den 8 maj 2015 klockan 10.00 i Hörsal E2, Lindstedsvägen
3, Kungl Tekniska högskolan, Stockholm.
© Jesper Magnusson, April 2015
Tryck: Universitetsservice US AB
iii
Abstract
The interest of using direct current in networks for both transmission and
distribution of power is increasing due to the higher efficiency compared to
the alternating current used today. As no natural zero crossings exist in direct current, the interruption of fault currents becomes a challenge. Several
circuit breaker topologies have been proposed to fulfil the requirements for
DC grids. One such topology is the hybrid DC-breaker consisting of three
parallel branches: a mechanical switch, a semiconductor branch, and a metal
oxide varistor.
The current interruption in the hybrid DC-breaker is made in three steps.
A mechanical switch carries the nominal current with low losses during normal
operation. When the breaker is tripped to interrupt the current, the mechanical switch is opened and commutates the current into the semiconductor
branch. This branch will then conduct the current as the mechanical switch
regains its voltage withstand. The semiconductors turn off and force the current into the varistor branch where the magnetic energy is absorbed and the
current is forced to zero.
This thesis is based on simulations and experiments to obtain design rules
for such a DC-breaker. It has been shows that several aspects needs to be
considered. Simulations are performed with several different models to obtain
the requirements of each of the components in the DC-breaker.
First of all, the choice of the semiconductor is important. There are a
number of components available in the market, but typically they are optimized for fast switching applications like inverters rather than circuit breaker
applications that only requires one single switching. Due to the high current
and voltage ratings and the easy control, the IGBT seems to be the best
choice among the commercially available components.
Simulations on the mechanical switch show that there is an optimal combination of opening time and arc voltage of the to obtain a successful commutation into the semiconductor branch. The actuator is a key component since
a relatively low increase in performance of the actuator drive circuit, significantly decreases the requirement of the other components in the DC-breaker.
A significant part of the work has been put on the voltage transient during
the turn-off of the semiconductor. As the current is forced into the varistor
branch, the stray inductance in that loop will result in an over-voltage due to
the high current derivative. A new type of snubber has been investigated using
another varistor mounted close to the semiconductor. It has been shown that
the function of the varistor snubber can be divided into two regions depending
on the ratio between the snubber and the main varistor. If the ratio is high
enough, the energy absorbed in the snubber varistor is only a few percent of
the total energy.
v
Sammanfattning
Intresset för att använda likström i både transmission och distribution av
elkraft har ökat tack vare en högre verkningsgrad och lägre förluster. En begränsande faktor för utbyggnad av likströmsnät är utvecklingen av brytare för
likström. Svårigheten att bryta en likström jämfört med växelström är att det
saknas en naturlig nollgenomgång där strömmen kan brytas. För att tvinga
fram en nollgenomgång kan man använda en hybrid brytare som består av
tre parallella grenar: en mekanisk switch, en gren med krafthalvledarkomponenter och en gren med en varistor.
Hybridbrytaren bryter en ström i tre steg. I normalfallet leder den mekaniska kontakten strömmen för att hålla nere de elektriska förlusterna. När
strömmen ska brytas öppnas den mekaniska kontakten och strömmen kommuteras över till krafthalvledargrenen. När krafthalvledaren stängs av, trycks
strömmen över till den tredje grenen där varistorn begränsar spänningen över
brytaren och absorberar den magnetiska energi som finns lagrad i nätet.
Strömmen avtar och när systemet är avmagnetiserat når strömmen noll.
Denna avhandling sammanfattar ett arbete att med hjälp av simuleringar
och experiment få fram designparametrar för en sådan brytare. Flera simuleringsmodeller har av använts för att få fram designkriterier för de olika
komponenterna i hybridbrytaren.
Ett första steg är att välja krafthalvledarkomponent. Dessa komponenter
är främst utvecklade för omvandling mellan lik- och växelström där de slås av
och på i korta intervall. En brytartillämpning skiljer sig markant från detta
eftersom strömmen bara behöver brytas enstaka gånger vilket ändrar kraven
på komponenten. Enkelheten i att kontrollera komponenten tillsammans med
dess höga ström- och spänningstålighet gör IGBTn till det bästa valet bland
de kommersiellt tillgängliga komponenterna.
Simuleringar visar att en optimal design av den mekaniska switchen är
en kombination av snabbhet och ljusbågsspänning. En nyckelkomponent är
den snabba aktuatorn som driver switchen. En relativt liten förbättring av
drivkretsen kan leda till minskade krav och därmed kostnader på övriga komponenter i brytaren.
En stor del av arbetet har lagts på att studera den transienta överspänningen då krafthalvledarkomponenten bryter strömmen. Den höga strömderivatan medför att även små ströinduktanser leder till relativt stora överspänningar som riskerar att skada brytaren. En ny typ av snubberkrets, bestående
av en mindre varistor som monteras nära krafthalvledaren, har föreslagits. På
detta sätt separeras överspänningsskyddet från absorptionen av den magnetiska energin. Experiment och simuleringar visar att om spänningsförhållandet
mellan de två varistorerna är tillräckligt stort absorberas endast ett par procent av energin i snubbern.
vii
Acknowledgements
Först och främst vill jag tacka min tidigare gruppchef på ABB, Mikael Dahlgren,
som var den första att överhuvudtaget få mig att fundera på att doktorera.
Vidare vill jag tacka min huvudhandledare Göran Engdahl för alla långsiktiga och
visionära diskussioner.
Jag vill tacka min bihandledare Lars Liljestrand som alltid tar sig tid att diskutera och dela med sig av sina kunskaper och alltid får mig att känna att jag
åstadkommer någonting vettigt. Jag är även glad för alla trevliga kollegor på ABB,
inte minst det härliga gänget jag ofta äter lunch med och Magnus Backman som
alltid får mig att känna mig välkommen på ABB. Ett speciellt tack går till Robert
Saers som inte bara hjälpt mig med ett par mycket bra forskingsuppslag, utan även
stöttat mig i att få undersöka dessa noggrannare.
Självklart vill jag tacka alla kollegor, och tidigare kollegor, på KTH. Det finns
mycket kunskap och välvilja att dela med sig, både bland doktorander och seniorer.
Ett extra tack till Carin Norberg som alltid hjälpt mig att hantera alla administrativa ärenden.
Ett jättestort tack går till min närmaste kollega Ara Bissal för allt samarbete de
senaste åren, för alla roliga och intensiva diskussioner och självklart också för de
tekniska kunskaperna. Cong-Toan Pham förtjänar även ett tack för att du står ut
med våra diskussioner på kontoret och för att du tagit över många av diskussionerna när Ara inte varit där.
Jag vill tacka min familj för att ni alltid stöttat mig och trott på mig. Jag vill
även passa på att tacka min ingifta familj för att ni alltid finns nära.
Sist men inte minst vill jag tacka min underbara fru Jessica och mina två döttrar Laura och Ellinor för att ni ger mig en bra anledning att inte jobba för mycket!
Contents
Contents
ix
1 Introduction
1.1 Motivation . . . .
1.2 Main contributions
1.3 Publications . . . .
1.4 Thesis outline . . .
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1
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2 Background
2.1 Power transmission
2.2 Applications of DC
2.3 DC grids . . . . . .
2.4 Faults in AC grids
2.5 Faults in DC grids
2.6 DC breakers . . . .
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3
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3 The Hybrid DC breaker
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3.1 Topology . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
3.2 Operation principle . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
4 The
4.1
4.2
4.3
4.4
mechanical switch
The actuator . . . . . . . . . .
Current commutation . . . . .
Voltage blocking . . . . . . . .
Design of the mechanical switch
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5 The
5.1
5.2
5.3
5.4
5.5
choice of semiconductor components
High power semiconductor switches . . . .
Available components . . . . . . . . . . .
Component requirements . . . . . . . . .
Component comparison . . . . . . . . . .
The optimal component . . . . . . . . . .
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x
CONTENTS
6 The
6.1
6.2
6.3
snubber circuit
43
Snubber circuits . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43
Possible solutions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
The parallel MOV snubber . . . . . . . . . . . . . . . . . . . . . . . 48
7 The
7.1
7.2
7.3
7.4
7.5
7.6
energy absorbing branch
Background . . . . . . . . . . . . . .
Basic working principle of the MOV
Dimensioning considerations . . . . .
The MOV simulation model . . . . .
Requirements on energy rating . . .
Choosing the MOV configuration . .
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67
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8 Conclusions
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9 Future Work
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List of Figures
79
Bibliography
83
Chapter 1
Introduction
1.1
Motivation
Recent developments in DC technologies and renewable energy increase the interest
in building large DC-grids. To have full control over possible faults and increase
the reliability of such a grid, DC-circuit breakers will be required. Due to the
low impedance of the DC-grid, the DC-circuit breaker needs to be faster than
conventional AC-circuit breakers. The hybrid solution, combining a mechanical
switch with power semiconductors, shows promising features to fulfil the demands
of such a circuit breaker.
The different components in the hybrid breaker are existing technologies, but are
used in a different way than they were initially developed for. This project aims at
investigating all of the building blocks of a hybrid breaker to determine demands on
each component. This will lead to better understanding of the interaction between
the components and clear design rules for the circuit breaker.
1.2
Main contributions
The main contributions of this thesis to the state of the art are:
• A proposed snubber circuit solution consisting of parallel varistors to limit
the over voltage across the semiconductor during switching. The solution
separates the energy absorption from the over-voltage protection and increases
the utilization of the semiconductor.
• Simulation models that show the trade-off between the different parameters
affecting the performance of the hybrid DC-breaker. The models consider the
whole breaker: the actuator of the mechanical switch including its energy storage, the current commutation between the different branches, the transient
over voltages during the semiconductor switching, and the non-linear voltagecurrent characteristics and energy absorption capabilities of the varistors.
1
2
CHAPTER 1. INTRODUCTION
1.3
Publications
The content of this thesis is mostly based on work published in scientific journals
and conferences. The thesis is based on the following papers:
I) A. Bissal, J. Magnusson, and G. Engdahl, “Comparison of Two UltraFast Actuator Concepts”, IEEE Transactions on Magnetics vol. 48, num
11, November 2012.
II) J. Magnusson, A. Bissal, G. Engdahl, R. Saers, Z. Zhang, and L. Liljestrand, “On the use of metal oxide varistors as a snubber circuit in solid-state
breakers”, IEEE ISGT Europe, October 2013.
III) J. Magnusson, R. Saers, L. Liljestrand, and G. Engdahl, “Separation of the
Energy Absorption and Over-voltage Protection in Solid-State Breakers by
the Use of Parallel Varistors”, IEEE Transactions on Power Electronics vol.
29, num. 6, June 2014.
IV) J. Magnusson, J. A. Martinez-Velasco, A. Bissal, G. Engdahl, and L. Liljestrand, “Optimal design of a medium voltage hybrid fault current limiter”,
IEEE EnergyCon, May 2014.
V) J. Magnusson, A. Bissal, G. Engdahl, and J. A. Martinez-Velasco, “Design Aspects of a Medium Voltage Hybrid DC Breaker”, IEEE ISGT Europe,
October 2014.
1.4
Thesis outline
The thesis is outlined as follows:
• Chapter 1 (this chapter), states the motivation and main contributions of
the thesis and papers the thesis is based on.
• Chapter 2 describes the background as the power distribution and transmission as well as the need of DC-breakers.
• Chapters 3-7 is the main work of the thesis. It describes the different parts
of a hybrid DC-breaker in detail and the design criteria to consider.
• Chapters 8 and 9 summarizes the main conclusions of the work and states
some future work.
Chapter 2
Background
2.1
Power transmission
In the late 19th century there was a discussion whether to use direct current (DC)
or alternating current (AC). Both systems existed in parallel with their own benefits
and drawbacks. At this time, the DC motor was dominant for the conversion of
electrical energy to mechanical energy in e.g. elevators. DC is also preferred for
lighting as the light should be constant. However, this problem can be easily solved
by choosing a high enough frequency of the AC so that the human eye cannot notice
the blinking.
When the use of electricity increased rapidly, there was a need to increase the
efficiency of power generation and distribution. One way to do this is to increase
the size of the generators, which in turn requires moving the generation away from
the consumer. This quickly resulted in too high losses in the transmission lines since
for a set amount of transferred power, the resistance of the lines will give resistive
losses in the form of heat. If the voltage of the line is increased, the required current
though the line is decreased. As the losses are proportional to the square of the
current, an increase in the voltage by a factor of 10 decreases the losses with a
factor of 100.
The desired voltage level at the load should be low enough to avoid injuries to
humans. Even though the standards classify voltages below 50 V AC as harmless,
voltages of 250 V are standard at the consumer side today [1]. If this voltage
level was to be used in power transmission, the distance would be limited to a few
kilometres due to the high losses.
Using a transformer, an AC-system can easily change the voltage level in different parts of the system and hence significantly decrease the losses and increase
the transmission distance. This enabled large power plants and larger generators
further away from the consumer resulting in cheaper production of electric power
and better environmental conditions in the cities. At that time there was no efficient way to change the voltage level in a DC-system, so AC-systems became the
3
4
CHAPTER 2. BACKGROUND
natural choice.
With the increase of renewable power, the power generation is becoming more
and more distributed in the form of small photovoltaic and combined heat and
power plants. However, when introducing large offshore wind parks and connecting
remote hydro power plants, there is still a large need of long transmission lines.
Further, ongoing discussions to utilize e.g the solar power in desert areas will even
further increase the need of long distance power transmission [2].
Development in power semiconductors [3] and switching technologies, has enabled efficient rectifiers and inverters to convert AC to DC and DC to AC respectively. Small changes in DC voltage levels, about a factor of three [4], can be made
directly by DC/DC converters, whereas larger changes use an intermediate step
with a high frequency AC transformer to change the voltage level.
2.2
Applications of DC
DC is used in transmission lines with high voltage (HV) DC and is discussed to
be used also for lower voltage levels due to the lower losses. To compare the losses
between AC and DC transmission in a fair way, one can consider to use the same
lines for either AC or DC. For DC, the current is only carried with two conductors
instead of three in the AC, resulting in higher losses for the same wire [5]. However,
the operating voltage of a DC line that would replace an existing AC line can be
higher than the AC
√ RMS-voltage since the AC system has to be insulated for the
peak voltage, i.e. 2VRM S . Calculation shows that for a three-phase double circuit
AC-line, the transmission capacity can be increased with 47% [6]. Hence, if the
systems are designed for equal losses, the percentage of losses is decreased with
68%. This means that when a system is designed with a specified power capacity,
thinner conductors can be used with DC, resulting in lower installation cost.
The other main benefit of DC compared with AC is the removal of all frequency
related effects. In a steady-state, the capacitance and inductance of the system will
have no effect on the current or voltage in a DC-system. Long AC overhead lines
are generally considered inductive and this inductance limits the power transfer
capacity of the line. With DC, this inductance has no effect on the voltage and is
not limiting for the power transfer. Similarly, the capacitance of a cable will draw
a charging current. For a 380 kV cable of 600 A, the charging current equals the
maximum current capacity at a length of 40 km. Hence, no useful power can be
transferred above this length. By series compensation with one inductor in each
end, the maximum length can be doubled [6]. The length can be further increased
by series compensation along the line, but this is not always possible, e.g. with subsea cables. Hence, in the case with sub-sea systems the use of DC can be motivated
even at shorter distances compared to on land.
Even though AC is dominant for power transmission and distribution today, DC
is available in several places. Some loads are DC-loads by nature and are possible
2.2. APPLICATIONS OF DC
5
to feed with a common DC supply to decrease cost. DC power also exists for a
wide range of voltage levels, even if just for some specific applications.
Low voltage DC
Low voltage (LV) DC is today used where there are natural DC loads. One example is data-centres since one larger rectifier is more cost-effective that several
smaller ones. Another reason is the easier implementation of battery-storage for
uninterruptible power systems that are naturally DC-systems.
Another common LVDC system is in traction, especially in metro systems since
the highly transient power flows are easier to handle in DC. Modern metro systems
also use regenerative breaking and produce electric power when the train is decelerating. The power is then fed back into the system and used by another train.
When using DC, there is no need to synchronize the generated power to the grid
voltage as if AC was used.
In Finland, there is ongoing research to use LVDC for the distribution network
[7]. As DC is considered low voltage up to 1.5 kV, compared to 1 kV for AC [1],
higher voltages can be used to comply with the current standards. The idea is to
replace the MVAC network and LVAC distribution of 20 kV and 0.4 kV respectively
with a 1.5 kV LVDC cable. The main benefit is to get rid of the MV over-head lines
that cause a significant part of the faults in rural areas. The lower losses compared
to the LVAC compensates for the increase in losses compared to the MVAC and
results in a system with the same losses as before and higher reliability.
Medium voltage DC
Wind power plants are generally producing AC power, but with a rectification
on the output. The DC rectification makes it easier to produce constant power
regardless of the wind speed. The collection grid, where the wind power plants are
connected together, is usually an MVAC grid, but research on MVDC collection
grids are ongoing [8, 9].
Another MVDC application is the electric systems aboard ships [10, 11]. As
more and more electrical equipment is integrated into ships, discussions on their
electric systems are ongoing. The US military has run several projects aiming
at integrating an MVDC grid on ships. The benefits compared to AC are easier
synchronisation of generators and to replace big, heavy 60 Hz transformers with
small lightweight DC/DC converters [12].
High voltage DC
Since the first commercial installation of HVDC in the 1950s [13], the number
of installations has been increasing. HVDC offers high power transmission for
very long distances both on land and under water. The possibility of transferring
bulk power is an important aspect for utilizing the available green energy that is
6
CHAPTER 2. BACKGROUND
located far from populated areas. One example is large scale hydro power in western
China being transferred over 2000 km to the eastern parts of the country [14] as
well as plans to use the enormous amounts of solar energy available in the Sahara
desert [15].
The other interesting feature of HVDC is the possibility to connect asynchronous
AC networks and AC networks of different frequency, i.e. in Japan and the US [15].
Since DC-links also are fully controllable regarding power flow, they can be used
to stabilize weak AC networks and increase reliability [16].
2.3
DC grids
HVDC lines today are point to point connections where power is transmitted mainly
from one end to the other. In the case of a fault in the DC line, the power is
interrupted on the AC side and the whole line is disconnected.
There are plans to build an HVDC grid across Europe as a backbone to strengthen
the existing AC transmission grid [17]. If there is a fault in such a grid, there has
to be a possibility to disconnect the faulted line and redirect the power flow rather
than disconnecting the entire grid. Such operations requires HVDC breakers that
can interrupt the fault current and isolate the faulted section. There are no such
installations to the present day, even if a lot of research and development is ongoing
in the area [18–20].
The same holds for a MVDC collection grids for offshore wind parks as the
system must be able to shut down one faulted wind mill and continue production
with the healthy equipment. The remote location makes maintenance and visits
time consuming and costly so automatic solutions will be required rather than single
shot devices like fuses.
For LVDC distribution grids the case is a bit different. If the grid is the last
step close to the loads, it is possible to disconnect the whole grid in case of a fault.
This is already done in the MVAC and LVAC systems that will be replaced by the
LVDC grid. Instead the most important feature is the ability to provide enough
fault current in the case of a fault on the AC side so that existing AC protection
devices function as desired [21].
2.4
Faults in AC grids
When a fault occurs in an AC-system, the behaviour of the fault current will depend
on the angle of the system voltage at the instant of the fault. Two distinct different
cases are often considered: the symmetric and asymmetric faults.
Figure 2.1 shows the current (dashed red) and voltage (solid blue) of a system
in the case of a symmetric fault. At the instant of the fault, the voltage is at
its maximum and the current is zero. Since the grid at fault conditions is mainly
inductive, the current will be lagging the voltage by 90◦ . As the current was zero
when the fault occurred, the fault current is symmetric around zero.
2.5. FAULTS IN DC GRIDS
7
Figure 2.1: Symmetric fault in an AC-system.
If the fault occurs when the voltage is zero, the fault current will not be symmetric as shown in Fig. 2.2. The initial rise of the current will be lower due to the
lower voltage, but the current will keep rising as long as the voltage is positive, i.e.
half a period. Hence the current will reach a peak that is up to 1.9 times the peak
current in the symmetrical fault. The asymmetric fault current can be seen as a
symmetric current with an additional DC component that will decay exponentially
with the system time constant τ = L/R.
For an AC circuit breaker, that interrupts the current at a natural zero crossing,
the asymmetric fault is worse than the symmetric as the peak current is higher and
puts more stress on all components in the system. Also, it is harder to interrupt a
current at a zero crossing after a larger current due to the higher temperature and
possibly higher ionization of the breaking media.
For a fault current limiter [22, 23], that has to limit the fault current before
the first peak, the symmetric fault is harder to handle [24]. Even though the first
current peak is higher in the asymmetric case, the slower rise of the current makes
it easier to limit the peak to a lower value in the asymmetric case than in the
symmetric case.
2.5
Faults in DC grids
As a fault occurs in a DC-system with a stiff DC source, the current will start to
rise from its initial value towards the maximum current determined by the system
8
CHAPTER 2. BACKGROUND
Figure 2.2: Asymmetric fault in an AC-system.
voltage and the remaining resistance in the system. The rate of rise of the current
will be limited either by the natural inductance of the system or by a current
limiting inductor [25]. The shape of such a rising fault current is shown in Fig. 2.3.
This inductive-resistive rise of the fault current is only valid if the DC-system is
fed from a constant DC-source or if the inductance in the system is high enough to
give a high time-constant. In the case of transmission or distribution systems, these
will always be fed by some kind of rectification of the voltage from a three phase
AC-system. Therefore, the short circuit current will be limited by the impedance
of the AC-system rather than the remaining resistance of the DC-system under DC
fault conditions.
Figure 2.4 shows the rising fault current on the DC side from a 6-pulse diode
rectification of the feeding AC-system. It can be seen that the fault current has a
behaviour highly influenced by the AC current and rises rapidly in the beginning.
The current oscillates with a 50 Hz frequency for several periods before settling at
the steady state value. The peak current is slightly less than 2 times the steady
state DC fault current.
Figure 2.5 shows the peak current, steady state current and the current after
1 and 2 ms depending on the voltage angle at the time of the fault. The voltage on
the DC-side is equal to the highest difference of the three AC voltages. Hence, the
DC fault will always be similar to a symmetric fault on the AC side. The steadystate fault current will be the same regardless of the fault angle, but the peak value
of the current will vary. Due to the 6-pulse rectifier, the DC fault will always result
2.5. FAULTS IN DC GRIDS
9
Figure 2.3: Fault in an inductive resistive grid with a stiff DC-source.
Figure 2.4: Fault current from the rectification of a three-phase AC system.
in a fault within ±30◦ of a symmetric fault in one of the phases. Hence, the initial
rise of the fault current will always be close to what can be considered the worst
case for an AC fault current limiter.
10
CHAPTER 2. BACKGROUND
Figure 2.5: Level of the fault current with a rectified AC system after different
times showing that the benefit of a fast opening of the mechanical switch is a lower
current to commutate.
In the description and simulations in this work, the modelling of the faults has
ignored the capacitors in the inverters and the impedance of the possible DC cables.
The capacitors in the converter can store significant energy [26] and including these
components would further increase the initial peak of the fault current [27, 28].
2.6
DC breakers
To interrupt a fault current in an inductive DC network, the current has to be forced
to zero, unlike in an AC-system where the current naturally crosses zero two times
each period, i.e. every 10 ms in a 50 Hz system. In the simplest possible model of a
DC-grid consisting of a source, an inductor and a breaker, the relationship between
current and voltage can be written
Usystem = L
di
+ Ubreaker .
dt
(2.1)
Hence, to decrease the current in the system, the voltage across the breaker has
to exceed the system voltage. The breaker also has to supply this higher voltage
until the current reaches zero. During this time, the magnetic energy stored in the
grid inductance is absorbed by the breaker. This is discussed further in Chapter 7.
It can be seen from Fig. 2.4 that to minimize the stress on the system, and avoid
the high current peak, an operation time in the millisecond range is desirable. A
2.6. DC BREAKERS
11
breaker with an operation time of 2 ms, limits the current to about 40% of the peak
current and with an operation time of 1 ms, the current can be limited to about 25%
of the peak current. This shows that for a DC-grid a very fast breaker is required
and the current technique to interrupt the current on the AC side with conventional
AC circuit breakers is not a feasible solution. However, other measures can be taken
to limit the impact of fault currents [29] as an alternative, or complement to the
DC-breakers.
Mechanical breaker
In a pure mechanical breaker, the current is interrupted by the counter voltage
of the electric arc between the contacts. Generally this is only possible if the
system voltage is low enough. These breakers are used for LVDC and the lower
region of MVDC in e.g. traction systems. Many other switches that does not
need to interrupt fault currents, e.g. disconnectors and load break switches are also
mechanical.
For high voltage systems, a very high arc voltage is required to force the current
down and absorb the magnetic energy stored in the grid. This results in very high
energy absorption in the arc, and a high wear on the contacts.
Solid state breaker
A solid state breaker consists of a fully controllable semiconductor in the current
path [30, 31]. The main benefit is that the switch is very fast and can switch
the current in some microseconds. This makes the solution attractive in sensitive
systems or systems with very low inductance, i.e. very fast rising fault currents.
As semiconductors are already in the current path in rectifiers and inverters, they
have already proven a high reliability.
The main drawback is the high on-state losses. A semiconductor with a voltage
rating of a few kV has an on-state voltage drop in the order of some volts. This
means constant losses in the kilowatt range when conducting some hundred ampere
which will require a large and complex cooling system. The power loss is maybe
not an economical problem, but rather rimes bad with current discussions on efficiency. Hence the solid state breaker is a good choice with high demands on short
interruption times or for low power systems.
Resonant breaker
The idea of a resonant breaker is to get the current through the main branch
of the breaker to oscillate and hence provide a natural zero crossing. This way a
conventional AC-breaker can be used as the main breaker in the current path. Such
a breaker is based on conventional technology, which makes it both less expensive
and easier for the market to accept.
12
CHAPTER 2. BACKGROUND
The oscillation can be created with a parallel LC circuit by discharging a precharged capacitor through the breaker in opposite direction of the main current [19].
Another option is to use a resonance circuit in parallel with the main breaker and
let the natural oscillations in the arc trigger a resonance to obtain the current zero
crossing. However, the current zero crossing is only local in the main breaker and
an energy absorbing element is required to absorb the magnetic energy from the
grid.
The hybrid DC-breaker
To combine the low losses of a mechanical switch in the main current branch with
the high switching performance of the semiconductors, the hybrid DC-breaker has
been proposed [32–35]. The basic topology consists of three parallel branches further described in the next chapter. The losses will be similar to the mechanical
breaker and the interruption time will be shorter than for the mechanical breaker
as the switch never has to interrupt current, and hence can be made lighter and
faster than a mechanical circuit breaker.
Chapter 3
The Hybrid DC breaker
3.1
Topology
The hybrid DC breaker consists of three parallel branches to handle different tasks
of the breaker. The first branch contains a mechanical switch that will carry the
nominal current with metallic contacts resulting in conduction losses similar to
conventional, mechanical circuit breakers. The second branch consists of semiconductors with a high switching performance. The third branch is metal oxide
varistors (MOV) to limit the transient voltages and absorb the magnetic energy
stored in the system.
Figure 3.1: Basic layout of the three parallel branches of a hybrid DC-breaker.
Figure 3.2 shows the current through the hybrid breaker and Fig. 3.3 shows the
voltage across the breaker when interrupting a rising fault current. The interruption of the current can be divided into five steps: fault detection, commutation,
semiconductor conduction, semiconductor turn-off, and current limitation. The
intervals are described in detail below.
13
14
3.2
CHAPTER 3. THE HYBRID DC BREAKER
Operation principle
Before the time t = 0, the system is in steady state with a constant DC current
of 1 per unit (p.u.) flowing through the breaker. At t = 0, the load impedance
is suddenly decreased to simulate a fault between the breaker and the load. This
causes the current to start to increase towards the peak current limited by the remaining resistance. The rate of rise of the current will be limited by the inductance
in the system and all the current will be conducted by the mechanical switch due
to the significantly lower resistance in this branch compared to the other parallel
branches.
Fault detection
At this point the breaker control has to identify that there is a fault in the system. Due to natural variations such as switching transients and over-loading the
breaker cannot be allowed to open only because the current becomes higher than
the nominal current. In complex power systems, this is done with sophisticated
algorithms. In this work, the fault is considered detected once the current reaches
2 p.u. as this current is too high to be considered a temporary over-loading. Hence,
as the current becomes 2 p.u., the fault is identified and the opening interruption
sequence is started by sending the opening command to the mechanical switch.
If the semiconductors are in blocking mode, they should also be switched in to
conduction mode.
Commutation
As the mechanical switch is trigged, there will be a delay before the mechanical
movement starts and the contacts separate. During this time, the current continues
to rise. At t = t1, the contacts of the mechanical switch separate and the arc voltage
appears across the breaker. The arc voltage is demonstrated in Fig. 3.3 with a
constant voltage in dotted red. This voltage will try to commutate the current over
to the semiconductor branch. As current is commutated into the semiconductors, a
counter voltage will be seen due to the forward voltage drop of the semiconductors,
shown in dash-dotted blue. In high voltage applications, the arc voltage will be
significantly lower than the system voltage and the commutation will not affect the
system current. Hence, by considering only the commutation loop in an ideal case,
the commutation can be described as
Uarc = Usemiconductor + Lstray
di
,
dt
(3.1)
where Lstray is the undesired inductance in the commutation loop.
Hence it can be seen that the duration of the commutation will be determined
by the voltage difference between the arc and the semiconductor as well as the stray
3.2. OPERATION PRINCIPLE
15
inductance in the loop. This equation holds only if the voltages are independent of
the current in the two branches, but shows the basic relationship.
In Fig. 3.2, the commutation is seen as the current in the mechanical switch
(dashed green) decreases and the current in the semiconductor (dotted red) increases. The main current (black) through the breaker remains constant due to the
inductive system.
Semiconductor conduction
At t = t2, the current in the mechanical switch reaches zero and is interrupted.
All the current is now conducted by the semiconductor branch and their forward
voltage drop is shown across the breaker. The semiconductors will be required to
conduct the current for a certain time, to allow the mechanical switch to reach a
fully open position and to let its plasma deionize. Since the forward voltage drop
of the semiconductors are low compared to the system voltage, the current will
continue to rise during this interval. The highest allowed conduction time will then
be limited by either the highest allowed peak current in the system or the thermal
limit of the semiconductors.
Semiconductor turn-off
When the semiconductors turn off at t = t3, the current is forced in to the MOV
branch and the voltage across the breaker rises rapidly up to the protection level
of the MOV. The semiconductor turn-off transient has to be handled with care to
avoid large transient over-voltages that might harm the components.
Current limitation
When the current is conducted by the MOV, the voltage across the breaker is higher
than the system voltage and hence the current will decrease. The magnetic energy
in the system will be absorbed as heat in the MOV. At t = t4, the system current
reaches zero and the voltage across the breaker will be equal to the system voltage.
16
CHAPTER 3. THE HYBRID DC BREAKER
Figure 3.2: Currents in the different branches of the hybrid DC-breaker during
interruption of a fault current.
Figure 3.3: Voltage across the hybrid DC-breaker during interruption of a fault
current.
Chapter 4
The mechanical switch
The mechanical switch is the part that will conduct the continuous current and
has to provide low conduction losses. It also has to be able to commutate the
current into the semiconductor when the current should be interrupted. Once the
semiconductor has interrupted the current, the mechanical switch has to block the
transient over-voltage as the current is decreasing.
Another major benefit of having a mechanical switch in the main current path
compared to pure semiconductors is the possibility to handle temporary overloading. Unlike semiconductors that have quite strict requirements in maximum
current to avoid over-heating, metallic contacts can handle currents above the rated
current for longer periods.
4.1
The actuator
To obtain the desired contact opening times a very fast actuator is required. In this
work, actuators based on electromagnetic repulsion has been used. Two different
concepts have been considered: The Thomson coil (TC) [36], and the Double sided
coil (DSC) [37].
Working principle
Both the TC and the DSC are based on flat spiral shaped coils as shown in Fig. 4.1.
To the left is the TC where an armature of a conductive material, e.g. copper or
aluminium, is placed in close vicinity of the coil and to the right is the DSC where
two identical coils are connected in series. The coils are connected so that the
direction of current in the two coils are opposite.
When a current pulse is sent through the coil, there is a fast increase in magnetic
flux. By Lentz law, currents are induced in the conducting armature to oppose the
change in flux. Hence the current in the armature will be in opposite direction of
the current in the coil and a repulsive force is obtained due to the interaction of
17
18
CHAPTER 4. THE MECHANICAL SWITCH
Figure 4.1: Sketch of the Thomson coil (left) and Double sided coil (right).
the inducted current and the magnetic field from the coil. This way a fast current
pulse is required to induce enough current in the armature to obtain the desired
repulsive force.
In the DCS, the induced current of the armature is replaced by a secondary
coil so that the same amount of current flows through both coils. Hence there is
no need to induce currents and the DCS would also give a repulsive force for DC
currents.
The force in the actuator is described by the Lorentz force as
f = J × B,
(4.1)
where J is the current density and B is the magnetic flux density. The total force
is then obtained by integrating the force density in the whole armature.
As the coils are wound like flat spirals, the system can be seen as close to
axisymmetric. The magnetic flux from the azimuthal current coil can then be
divided into two components: radial, and axial. The change in the axial field will
induce azimuthal currents in the armature, and the interaction with the radial field
will give an axial force density. Hence, to obtain a high repulsive force, one needs
to maximize both the radial magnetic field and the current density in the armature.
Simulation model
Figure 4.2 shows the energizing circuit of the TC and DSC. It consists of a precharged capacitor bank, controlled by a thyristor. Since the capacitors are electrolytic, a free-wheeling diode is used to carry current and avoid negative voltages
across the capacitor. When modelling the set-up some parameters have to be taken
into consideration. Those are: the resistance of the capacitors; resistance and induc-
4.1. THE ACTUATOR
19
tance of the wires connecting the capacitor bank to the actuator; and the resistance
and inductance of the actuator itself.
Figure 4.2: Equivalent circuit of the Thomson coil actuator system with its capacitive energy storage.
In a DSC, the inductance can be described as a function of the distance between
the two coils. As the coils are close to each other, the mutual coupling between them
will decrease the total inductance of the actuator. The resistance and inductance
are not much influenced by the current transients [38] and the inductance can be
modelled as only a function of the distance between the two coils. The inductance
can be calculated at several separation distances between the coils for an arbitrary
current and a continuous function can be obtained by interpolation of the acquired
values.
For the TC, the simulation model becomes much more complex. As the mutual
coupling between the two parts, coil and armature, will depend on the induced currents in the armature, the inductance will no longer be a pure geometrical property.
The induced currents will in turn depend on the rate of change of the magnetic flux
and the inductance and resistance will become functions of the frequency of the
applied current.
By coupling the circuit in Fig. 4.2 to a FEM model of the TC geometry, a
more accurate model is obtained. The inductance and resistance of the TC-part
is then calculated as function of the applied voltage and current as well as the
separation distance. The dimensions of the DSC is given in Table 4.1. For the TC,
the secondary coil is replaced by a copper ring of the same dimensions.
Simulation results
Simulations are run to compare the TC and the DSC for two different cases. In
the first case the actuated mass is only the mass of the moving part of the actuator
20
CHAPTER 4. THE MECHANICAL SWITCH
Table 4.1: Geometrical parameters of the double sided coil used in the simulations.
Number of turns
Inner radius
Outer radius
Conductor width
Conductor height
10
27.5 mm
47.5 mm
2 mm
4 mm
and in the second case, an extra mass of 1 kg is added to the moving part. The
circuit is energized from a capacitor bank of 16.5 mF charged to 200 V.
Figure 4.3 shows the current pulse in the four cases. The low resistance and
inductance of the circuit gives a fast rise of the current. Initially the movement
is small and the electric circuits for all four cases are similar resulting in a similar
rise of current in all four cases. Since the current in the TC only flows through one
coil compared to the two coils of the DCS, the TC will experience a higher current
peak due to the lower resistance. Just before the current peak, the voltage of the
capacitor reaches zero and the current is taken over by the free-wheeling diode.
The equivalent circuit then changes from an RLC circuit to an RL circuit and the
fall of the current is much slower than the rise.
It can be seen that the two cases with a 1 kg load experiences a higher current
peak than the unloaded cases. The reason is that the higher mass results in less
movement and hence the low separation distance keeps the inductance low for a
longer time allowing the current to rise higher.
Figure 4.4 shows the resulting force in the same four simulations. The loaded
cases have a higher force due to the higher current described in the previous section.
The DSC shows a higher force than the TC due to the higher current density in
the armature. This shows the benefit of the DCS compared to the TC: the series
connection of two coils results in a higher current in the moving part compared to
the TC that relies on induced currents.
The force in the TC and DSC both peak at 250 µs at 11.2 and 12.6 kN respectively. Hence both actuators show the desired performance of fast reaction to give
a short delay from trigger to contact separation.
Apart from a short time to contact separation, it is also desirable to separate
the contacts with a high velocity to rapidly obtain a sufficient contact separation
distance to handle the transient over-voltages after the commutation. Figure 4.5
shows the resulting velocity as function of time for the four simulated cases. It can
be noted that all four set-ups reach the final velocity within 1 ms from the time of
trigger.
Due to the higher force, the DSC will obtain a higher velocity than the TC both
with and without loading. However, even though the loaded set-ups shows a higher
force, their resulting velocity is lower than that of the unloaded cases. The reason
is that the increased force is not enough to compensate for the increased mass.
4.1. THE ACTUATOR
21
Figure 4.3: Current pulse through the Thomson coil and Double sided coil.
Figure 4.4: Obtained force in the Thomson coil and Double sided coil when discharging a capacitor bank through the coil.
22
CHAPTER 4. THE MECHANICAL SWITCH
Figure 4.5: Velocity as function of time for the discussed set-ups.
To compare the different cases, the efficiency can be defined as the ratio of the
final kinetic energy and the initial electric energy in the capacitor
2
mvfinal
Ek
=
.
(4.2)
Ei
CUi2
The efficiency of the four cases are summarized in Table 4.2. It can be noted
that the efficiency of the TC and DSC drops by 73% and 65% respectively as the
actuators are loaded with 1 kg. Further, it can be seen that the DSC shows a
superior efficiency to the TC and that the difference increases for the loaded case.
η=
Table 4.2: Efficiency of the four simulated cases of the TC and DSC with and
without 1 kg external load.
η
TC
18.5%
Loaded TC
5%
DSC
24.6%
Loaded DSC
8.5%
Another way to explain the difference between the TC and the DSC is by looking
at the inductance. Figure 4.6 shows the inductance of the actuator seen from the
connections from the capacitor bank.
For the DSC, the initial inductance is low since the magnetic flux from the second
coil effectively cancels the flux produced by the primary coil at low separation
distances. As the coils separate, the magnetic coupling between them decreases
4.1. THE ACTUATOR
23
and the inductance increases towards the inductance of the two coils connected in
series.
For the TC, the cancellation of magnetic flux relies on the induced currents in
the armature. In the beginning where the current rapidly rises, high currents are
induced in the armature, and the magnetic flux is cancelled almost as good as in the
case with the DSC. When the current has reached its peak, the induced currents
will decrease and the magnetic coupling will decrease. The inductance will then
increase towards the inductance of the primary coil alone, i.e. half the inductance
compared to the DSC.
Figure 4.6: Comparison of the inductance of the Thomson coil and the Double
sided coil as the armature separates from the primary coil.
Even though the efficiency is a measure of the performance of the actuator, it
is not the main key parameter of a switch. Unlike for example a rotating machine
that operates continuously, the breaker will only be operated at specific events when
there is a fault, or the line needs to be taken down for maintenance. Hence the
operational energy of the actuator will not have a significant effect on the running
costs of the breaker. However, an actuator with higher efficiency requires less input
energy and enables a smaller and possible cheaper drive system.
The performance of the actuator can be described as a requirement of moving a
certain distance in a specified time. Table 4.3 shows a comparison of the previously
discussed TC and DSC set-ups with two additional set-ups of the DSC where the
position after 6 ms is compared. The higher efficiency of the DSC results in a longer
travelled distance after the specified time. This enables to decrease the input energy
of the DSC to show the same performance. In DSC T1, the initial voltage of the
24
CHAPTER 4. THE MECHANICAL SWITCH
capacitor bank has been decreased so that the travelled distance matches that of
the TC. In DSC T2, the voltage is kept constant and the capacitance is decreased.
As seen in the table, the higher efficiency of the DSC enables a decrease in
input energy of 21% and 23% respectively if the voltage or capacitance is decreased.
Further, the peak current is decreased resulting in less losses and hence less heat
to be handled in the actuator system.
Table 4.3: Summary of the simulation results for the Thomson coil, Double sided
coil, and Double sided coil with adjusted input energy.
C [mF]
Vc [V]
Ei [J]
Iˆ [kA]
Fˆ [kN]
x @ 6 ms [mm]
η [%]
TC
16.5
200
330
6.21
11.15
154
18.5
DSC
16.5
200
330
5.47
12.57
178
24.6
DSC T1
16.5
178
261
4.97
10.39
153
23.3
DSC T2
12.7
200
254
5.21
11.56
154
23.6
The simulations have shown that the DSC has a better performance resulting
in the possibility of a lower input energy for the same performance. However, it
comes at the cost of mechanical performance. As the two coils in the DSC are
connected in series, a current carrying wire has to be connected also to the moving
coil. This connection is difficult to make mechanically durable due to the extreme
accelerations, in the order of 70,000 m/s2 involved. As the armature of the TC is
merely a disc of conductive material, its mechanical reliability is much higher.
4.2
Current commutation
As described in (3.1), the success of the commutation from the mechanical switch
to the semiconductor will depend highly on the arc voltage, but also the voltage
drop across the semiconductor. This means that to commutate a higher current, a
higher arc voltage is required. Figure 4.7 shows a failed commutation in a hybrid
breaker trying to commutate a rising fault current. The combination of switch
opening time and arc voltage is not enough to commutate the current into the
semiconductor branch.
The commutation starts at t1 and the current in the semiconductor continues to
increase. The voltage drop increases until it becomes too high and the current in the
mechanical switch starts to rise again around t2. At this point, the commutation
has failed and an interruption of the current will not be possible.
There are two possible ways to avoid a commutation failure. One is to decrease
Tsw , the time from trigger until the contacts separate and the arc forms. Figure 4.8
shows a simulation where the arc voltage is the same as in the failed commutation,
4.3. VOLTAGE BLOCKING
25
Figure 4.7: If the combination of switch opening speed and current during the
commutation is not sufficient, the commutation into the semiconductor branch will
fail and the breaker will not manage to interrupt the current.
but where the opening time has been decreased. This decrease causes the commutation to start at t1 with lower current and hence the commutation is finished at
a lower current at t2. This way the arc voltage is kept sufficiently higher than the
voltage drop of the semiconductor and the commutation is successful.
Another possibility to obtain a successful commutation is to increase the arc
voltage, Varc . Figure 4.9 shows a simulation where the commutation starts at
the same time as in the failed case, but with a slightly higher arc voltage. As
the commutation is faster due to the higher arc voltage, all current is successfully
commutated into the semiconductor branch at t2. The total time from the fault
(at 0) to successful commutation (at t2) is the same in both Fig. 4.8 and 4.9.
4.3
Voltage blocking
When the semiconductor turns off and forces the current into the MOV, the contact
system of the mechanical switch has to withstand the transient over-voltage. For
long distances in atmospheric pressure, the breakdown voltage can be assumed to
increase linearly with the separation distance. Hence the final velocity given by the
actuator will determine the time where it is possible to turn off the semiconductors.
If a sufficient separation distance is not reaches when the semiconductor switch is
turned off, a breakdown will occur between the contacts of the mechanical switch,
and the control of the switching sequence is lost.
26
CHAPTER 4. THE MECHANICAL SWITCH
Figure 4.8: By decreasing the switch opening time, the commutation will occur
earlier, i.e. with a lower current and succeed.
Figure 4.9: By increasing the arc voltage of the switch, the higher arc voltage will
force a faster and successful commutation.
4.4
Design of the mechanical switch
To obtain the possible design parameters for the mechanical switch, a large set
of simulations have been performed. A DSC has been used as the actuator, not
4.4. DESIGN OF THE MECHANICAL SWITCH
27
mainly because of the performance, but rather due to the simpler simulation model.
As described previously, the DSC is less sensitive to the frequency of the applied
current pulse and can be simulated using only an inductance model. By calculating
the inductance and resistance as function of the separation distance in a FEM
model, these parameters can be extracted and used in the simulation of a pure
electric circuit.
The arc voltage is modelled as a constant voltage and the voltage drop across
the semiconductor is modelled as a constant voltage plus a resistive component.
The hold-off time, where the semiconductor is allowed to conduct the full current
is set to 500 µs, as this will push the semiconductor towards its thermal limit. The
peak current is not allowed to exceed 16 kA.
If the problem is stated as an optimization problem, where the objective is
to obtain the lowest possible Varc and the highest possible Tsw , that result in a
successful interruption, an infinite number of solutions can be obtained. To find
the optimal parabola know as the Pareto front [39] the importance of the two
parameters can be swept using a third parameter. By introducing a weighting
parameter α according to
f = (1 − α)
Tsw
Varc
−α
,
V0
T0
(4.3)
one can instead minimize this function for different values of α. As the absolute
value of Tsw is much smaller than Varc , both parameters are normalized with an
initial value to be around 1. By sweeping α from 0 to 1, points along the Pareto
front can be obtained. The normalization will only affect the distribution of the
obtained solutions to be more evenly distributed.
Figure 4.10 shows the result of the parametric sweep. The area to the left of the
Pareto front fulfils the design criteria and the area where the interruption fails is
greyed out. The dashed black lines shows the strict limitations of the two parameters respectively. The switch opening time cannot exceed 620 µs as this together
with the delay due to fault detection and the semiconductor hold off time gives a
current higher than 16 kA. In the same way, with an instant opening of the switch
(Tsw = 0), the arc voltage has to exceed the voltage drop of the semiconductor
at the end of the commutation. The simulations show that the lowest possible arc
voltage is 42 V.
For a low α, the focus will be on minimizing the arc voltage and hence the
solutions will end up down in the left corner of the allowed space. As α increases,
the solutions will move up to the right along the front towards the vertical black
line showing the solution with highest possible Tsw .
The optimal solution depends on other parameters such as cost and utilization
of the components. Without any further relationship between the two parameters
regarding e.g. cost, all these solutions are an equally good design. However, generally the points close to the dashed black lines are not the best solutions as they
push one of the parameters to the extreme. Consider the points α = 0 and α = 0.2
28
CHAPTER 4. THE MECHANICAL SWITCH
where the first gives Varc = 42 and Tsw = 0. By increasing the arc voltage with
only 6 V (14%), the opening time of the switch is allowed to increase from 0 to
170 µs.
Figure 4.10: Possible combinations of switch opening time and arc voltage to fulfil
the commutation.
Figure 4.11 shows a different design approach, where two other parameters are
considered. One is the arc voltage but with an improved model of the arc. As
one of the main benefits of the DSC actuator is the high velocity, the arc model is
improved to contain both a constant voltage drop and a component proportional
to the length of the arc as
Varc = N V0 + El,
(4.4)
where N is the number of series contacts, V0 = 20 V is the cathode voltage drop
of each contact, and E = 1000 V/m is the electric field along the length of the arc
(l). The value of E is taken from [40], where it is experimentally obtained for free
burning arcs.
The second parameter is the charge voltage of the capacitor bank energizing the
actuator. With a constant capacitance of 10 mF, the final velocity of the contact
system will depend directly on the initial voltage of the capacitor bank.
Simulations are run to find the minimum combination of these two parameters,
where the current is interrupted below 12 and 16 kA respectively. Further, a condition is set that the switch should be able to handle the transient over-voltage when
the IGBT is turned off. Hence, it will require the switch to both provide a sufficient
4.4. DESIGN OF THE MECHANICAL SWITCH
29
arc voltage for the commutation and to reach a sufficient distance to handle the
MOV voltage during current limitation.
Figure 4.11: Different design options for the mechanical switch to obtain the required arc voltage.
Figure 4.11 shows the minimum combination of the two parameters for two
different peak current ratings. If the number of series contacts are increased, the
arc voltage originating from the cathode voltage drop will increase and the required
distance will decrease. However, as the number of contacts are increased to two, the
arc voltage is already dominated by the distance dependent term. Further increase
in number of contacts gives little decrease in required capacitor energy. All the way
to the right, when increasing from 8 to 9 series contacts, the capacitor voltage is not
decreased at all. This means the minimum required capacitor voltage for achieving
sufficient voltage withstand has been reached. If deceased below this value (100 V
for 12 kA and 70 V for 16 kV) the switch will not reach a sufficient distance to be
able to handle the over-voltage when the semiconductor is switched off.
The other thing to notice is that the performance of the breaker can be increased
with a slight increase in capacitor voltage. For example with two series contacts,
an increase in capacitor charge voltage from 140 to 180 V, the peak current in the
system is decreased from 16 kA to 12 kA, i.e. by 25%. Even though the cost of
the capacitor bank will increase with the higher voltage, a saving in 25% of the
semiconductors required to handle the peak current, should be a significant cost
saving.
Chapter 5
The choice of semiconductor
components
5.1
High power semiconductor switches
Traditionally, power semiconductor devices have been used in rectifier or inverter
applications [41], [42] where several switchings per fundamental power frequency
period are required. To reduce the harmonics in the sinusoidal output signal, the
switching frequency has been pushed to several hundreds of Hertz or even to the kHz
range [43]. To obtain reasonable losses, the losses in the semiconductor component
at each switching instant have to be very low. Since the component will be in the
main current path, also the forward voltage drop, i.e. the on state power loss has
to be kept low [44].
Due to the fast and controllable switching ability, the interest in using the
semiconductor components in circuit breaker topologies has increased. This use
changes the demands set on the components and this project presents a summary
of available components and their benefits and drawbacks for use in a DC breaker
topology.
There are several high power semiconductor components commercially available.
The definition of high power is not unique and components are available in different
power ranges.
5.2
Available components
Biplolar Junction Transistor (BJT)
The BJT [45] is one of the oldest semiconductor components. It consists of a pnp
or npn structure where the outer contacts are called collector and emitter, and the
middle contact is called base as seen in Fig. 5.1. The npn transistor is faster than
the pnp due to the higher mobility of electrons compared with holes.
31
32
CHAPTER 5. THE CHOICE OF SEMICONDUCTOR COMPONENTS
E
E
n
p
n
B
B
C
C
Figure 5.1: Symbol and schematic drawing of the BJT.
As a positive voltage is applied between the collector and emitter in an npn
transistor, the first pn-junction (collector-base) will be reverse biased and block the
flow of current. If a current is then injected through the base to the emitter, the
main part of the electrons injected from the emitter into the base region will be
swept through the depletion layer and end up in the collector region. Hence, the
injected drive current from base to emitter can be seen as magnified to a larger
main current between the collector and emitter.
Unlike most other components, the BJT is current controlled. Hence it requires
a rather high, continuous base-current to stay in conduction mode which results in
a rather poor efficiency.
Metal Oxide Semiconductor Field Effect Transistor (MOSFET)
An n-channel MOSFET [46] consists of two contacts: drain and source, with heavily
n-doped semiconductor as shown in Fig. 5.2. Those regions are embedded in a
lightly doped p-substrate and hence not directly connected electrically. The gate
contact is centred between the drain and source contacts and separated from the
semiconductor substrate by an insulating layer of metal oxide (shown in yellow).
D
D
G
S
G
n
p
n
S
Figure 5.2: Symbol and schematic drawing of the MOSFET.
5.2. AVAILABLE COMPONENTS
33
When a voltage is applied between the gate contact and the bottom of the device,
the capacitance will start to attract electrons. Since the metal oxide is insulating,
the negative charges will accumulate close to the surface and compensate for the
excess of holes due to the p-doping. As the voltage is further increased, the number
of electrons will increase until the region closest to the metal oxide is effectively
in excess of electrons and a channel of negative charges is formed between the
drain and source regions. The conduction of current can now be performed by the
electrons in the channel. If the gate voltage is removed, the excess electrons will
be removed and the region will return to its initial p-doped state. The electron
channel will vanish and the current is be interrupted.
The conduction losses of the MOSFET have resistive behaviour since it depends
only on the available concentration of charges and no effective pn-junctions are in
the conduction path. The control is performed with the gate voltage and only low
capacitive currents which results in low energies required in the control circuit. By
the physics of the MOSFET, the component should be bidirectional. However, due
to other factors in the component design, this is generally not the case.
The two main benefits of the MOSFET are: the simple control through the gate
voltage, and the very high switching speed.
Junction Field Effect Transistor (JFET)
The JFET [47] is also a field effect transistor but is based on another structure
than the MOSFET. In a p-channel JFET, shown in Fig. 5.3, the drain and source
contacts are connected by a p-doped semiconductor. On the surface between the
contacts, or surrounding the device, there is a layer of n-doped material covered by
a gate contact.
D
D
G
G
S
p
n
S
Figure 5.3: Symbol and schematic drawing of the JFET.
When a voltage is applied between the drain and source contacts, the connected
p-region will carry the current. If a voltage is then applied on the gate-contact, the
p-n junction will be reverse-biased and a depletion layer will form on both sides of
the junction, shown in grey colour in the figure. When enough voltage is applied,
34
CHAPTER 5. THE CHOICE OF SEMICONDUCTOR COMPONENTS
the whole p-channel is depleted and the current will be interrupted since there are
no charges available to carry the current.
The control of the component is made by voltage and only small currents are
required to remove the charges in the depletion area. The component is in a normally on-state meaning that it can conduct current without a gate-signal and the
gate signal is rather applied to interrupt the current. Like the MOSFET, the JFET
is normally not bidirectional even though the physics behind could allow this.
Thyristor
The thyristor [48] consists of a pnpn structure where the component is turned into
its on-state with a current pulse through the gate (p) and cathode (n). This causes
a turn-on of the npn transistor which in turn allows the second pnp structure to
turn on. Once this happens, the external control of the current is lost and the
thyristor latches in conduction mode. When the current reaches zero again, e.g. in
a AC circuit, the transistors will turn off and the component will block any forward
voltage until a new trigger signal is given on the gate.
A
A
G
G
C
p
n
p
n
C
Figure 5.4: Symbol and schematic drawing of the thyristor.
The thyristor is classified as a semi-controllable component as it can be turned
on but not turned off. A turn off of the thyristor can only be obtained by forcing
a zero crossing with external elements. The conduction is unidirectional and under
negative voltage, the thyristor will behave as a reverse biased diode and block the
voltage.
The main benefit of the thyristor is the high current and voltage ratings available and that the component has been developed a long time ago resulting in a well
proven concept and reliable components. Also the forward voltage drop is considerably lower than most other semiconductors. The main drawback is the lack of
turn-off control signal.
5.2. AVAILABLE COMPONENTS
35
Gate Turn-off Thyristor (GTO)
The GTO [49] is a fully controllable version of the thyristor where a reverse current
pulse can be sent through the cathode and gate to turn the switch off. The structure
of the component is slightly modified from the thyristor, by adding a region with
higher doping level around the gate contact as shown in Fig. 5.4.
A
A
G
p+
C
G
p
n
p
n
C
Figure 5.5: Symbol and schematic drawing of the GTO.
By drawing a current from the cathode, out via the gate contact, the flow of
charges between the n and p regions are stopped so that the device will limit the
current and return to a blocking state. The required amplitude of the turn-off
current is about 20-30% of the forward current, i.e. 70-80% lower than the required
reverse current to turn off a thyristor under the same conditions.
The main benefit of the GTO is the low forward voltage drop in combination
with a relatively simple turn-off operation compared to the thyristor. However,
since the GTO is current controlled, the switching energy is much higher than in
voltage controlled components making it more suitable for lower switching frequencies.
ABB have their own version of a GTO, known as the insulated gate commutated
thyristor (IGCT) [50]. The evolution is that the gate drive circuit is integrated into
the component to ensure a very low inductance and hence a high current peak when
discharging a capacitive energy storage though the gate and cathode. By turning
the component off with a higher reverse current, all of the minority carriers are
swept out and the turn-off time decreases. It is also possible to get the IGCT with
reverse blocking capability, but it comes at the cost of increased on-state voltage
drop due to the thicker lowly doped p-body-region.
Insulated Gate Bipolar Transistor (IGBT)
The IGBT [51] is a bipolar device that is easily controlled with a voltage signal via
a MOSFET structure. Figure 5.6 shows the schematic lay-out of the component.
Close to the collector contact, there is a highly doped p-layer. Above this, there
is an n-layer which takes care of the voltage when the component is in forward
36
CHAPTER 5. THE CHOICE OF SEMICONDUCTOR COMPONENTS
blocking mode. The top layer has a structure of both a gate and an emitter contact
with a larger p-region and a smaller n-region.
C
E
G
G
E
E
n
p
n
p
C
Figure 5.6: Symbol and schematic drawing of the IGBT.
The working principle is that when a voltage is applied to the gate contact, an
n-channel is created through the p-base in the same way as in a MOSFET. Once
the two n-regions are set in contact with each other, the large flow of electrons will
result in an injection of holes into the n-base. This bipolar conduction behaviour
leads to that the device can operate at current densities of 20 times that in a
MOSFET.
The switching speed of the IGBT is rather slow due to high modulation of charge
carriers. By removing, or even reversing, the gate voltage, the n-channel is removed
and the current starts to decay rapidly down to a threshold. However, due to the
high number of electrons in the n-base layer the current will decay exponentially
as the carriers recombine usually referred to a tail-current.
Looking at the basic structure of the IGBT, the component would also have
reverse blocking capability. However, this is usually not the case due to the small
scale structures of the p and n regions close to the emitter. Thus, the IGBT is
almost always used with an anti-parallel diode, separately or integrated into the
same package.
In the early days, the IGBTs had problems with latching, i.e. the inherent
thyristor pnpn structure got latched due to a too high voltage drop in the p-base
region. This results in that the control of the component is lost and it will behave
as a thyristor. This problem was solved already in the mid-80s and is no longer an
issue.
The main benefits of the IGBT are that it is easily controlled via the MOS
structure and that the high current density leads to components with high current
ratings [52].
Another version of the IGBT is the Bi-mode Insulated Gate Transistor [53],
(BiGT), which is a reverse conducting IGBT. The main benefit is that the antiparallel diode is integrated into the same chip as the IGBT. This means the chip
will experience a more homogeneous and constant heating regardless of the current
direction. It also enables a higher current in components consisting of several sub-
5.3. COMPONENT REQUIREMENTS
37
modules. As all modules in the component are BiGTs, compared to conventional
IGBT packs where every third module is an anti-parallel diode, the current rating
of the component increases with about 30%.
Other novel components
Several other evolutions of existing components are proposed in literature but not
commercially available.
One example is the HUBFET [54], which is a combination of the IGBT and
Power MOSFET. It combines the benefits of the two components as it is unipolar
at low drain-source voltages and bipolar at higher voltages. Thus the on-state
voltage drop will be low for both low and high currents.
Another example is the MOS Controlled Thyristor (MCT) [55]. Even though
the main idea of the component is not new, it has not fully found its way to
commercialisation. The reason is the difficulty to precisely manufacture the narrow
regions, which makes the structure inherently sensitive to fast transients in both
current and voltage.
5.3
Component requirements
The semiconductor components are generally optimized for fast switching and used
in high-frequency applications. The requirements if the component is used in a
circuit breaker application will be completely different.
In converter applications, the efficiency is calculated based on the on-state and
switching losses in relation to the power put through [56]. When the current is
interrupted in a circuit breaker, the energy stored in the grid is already considered
lost and just needs to be dissipated as heat. Hence, the switching losses of the
semiconductor are not relevant in this application.
Solid-state breakers
In a solid state circuit breaker [57], the semiconductor component will be in the
main current path, i.e. it will constantly conduct current when used in on-state.
According to this, the conduction losses will be high. In the previously suggested
application, with a load current of 2 kA, each volt of forward voltage drop will
result in a continuous power dissipation of 2 kW. An optimal component would
have a very low on-state voltage drop to minimize these losses, but it is impossible
to avoid the use of a water cooling system.
Further, in a circuit breaker application, there is no correlation between the
terms switching frequency and turn-off speed. Although a fast switching operation
is generally held as one of the main benefits of a solid-state breaker, the time-scale is
totally different. A conventional mechanical circuit breaker will interrupt a current
with in some 10 milliseconds. In this sense, any breaker that can react and operate
in less than one millisecond can be considered fast. Instead, due to the high current
38
CHAPTER 5. THE CHOICE OF SEMICONDUCTOR COMPONENTS
and voltages involved, as well as the high rate of rise of a fault current, the need for
a fast semiconductor switch is to limit the stresses on the semiconductor itself [43].
Since the solid-state breaker is fully controllable, there is no need for large
over-current capabilities. As the fault is identified, the current can be interrupted
almost instantaneously. However, any over-rating of current capabilities provides
extra time for better analysis and decision making.
Hybrid breakers
The idea of the hybrid breaker is to reduce the losses in the conduction mode but
still to be able to benefit from the fast switching of the solid state switch. By letting
a mechanical contact carry the nominal current and commutate the current to the
semiconductor switch only for a short time just before interruption, the demands
on the semiconductor reduces and changes.
Since there are no conduction losses in the semiconductor during nominal current, the on-state voltage drop becomes irrelevant. However the voltage drop during
higher currents becomes critical for two reasons. First, this voltage drop will determine the required arc voltage of the mechanical switch to ensure a safe and fast
enough commutation of the fault current [24]. Secondly, the power dissipation will
determine the heating of the semiconductor itself and limit how long it will be
allowed to carry the current before it is interrupted. Compared to the solid state
breaker where the power dissipation is merely a question of cost and complexity of
the cooling system, in this case the voltage drop is fundamental as it determines
the specifications of the mechanical switch.
5.4
Component comparison
Comparing different semiconductors for a purpose like a breaker application is not
straight forward. Most of the components are optimized for their most common
applications. Also the data-sheets contain different information depending on the
type of semiconductor, most likely because they are more application specific.
Small components
Most of the components are commercially available from several manufacturers in
slightly different versions. Table 5.1 shows the highest available component ratings
from an online electronics store [58] and the price for one component. Two examples
are given for the IGBT as the highest voltage rating is only available for relatively
low currents.
Since the ratings, and hence the prices, of the components are very different,
a relative price is calculated to make a more fair comparison. From this it can
be seen that the IGBT is several times cheaper than the other components. The
very low price of the thyristor is not relevant as a fully controllable component will
be required to force a current zero. The relatively low price of the BJT might be
5.4. COMPONENT COMPARISON
39
Table 5.1: Highest available component ratings. For the IGBT, two different values
are given since the highest voltage rating is only available for relatively low current
levels.
Voltage [kV]
Continuous current [A]
Price [SEK]
Relative price [SEK/A]
BJT
1.5
2.5
13.7
5.48
MOSFET
1.7
180
5091
28.3
JFET
1.7
4
471
118
Thyristor
1.7
1600
2156
1.35
IGBT
1.2/1.7
270/44
1021/36
3.78/0.8
Table 5.2: On-state forward voltage drop for ABB high power semiconductor components. The voltage drops are given for a junction temperature of 25◦ and 125◦ C
respectively. Values in parentheses are interpolated values as they are exceed the
component ratings.
Rated voltage
Rated current
Max turn-off current
Voltage drop at 2 kA
Voltage drop at 10 kA
Thyristor [59]
4200
2192
1.65/1.55
2.8/3.15
GTO [60]
4500
2000
2000
-/3.5
(10.3)
IGCT [61]
4500
2670
4000
1.75/1.85
(3.25)/3.75
IGBT [62]
4500
2000
2000
2.7/3.4
(10.1)
misleading as it is calculated from a component with very low rating; however, the
price of a MOSFET with similar ratings still gives a very high reference price.
The IGCT and GTO are components specifically developed for high voltage
applications and are generally not available to the public but rather directly from
the manufacturers to the power industry.
High voltage components
For a real high power application such a DC circuit breaker, the current rating
requires slightly different components. There are only a few manufacturers of these
components, and one of the major players is ABB. Table 5.2 shows some of the
components that are available from ABB in the higher power range. The components are chosen to be as close as possible to 4500 V blocking voltage and 2000 A
continuous current. The reason for this is the assumed continuous current of 2 kA
discusses earlier. Even though some of the components are available with higher
ratings, this level is chosen to give a fair comparison.
The forward voltage drop of each component is compared at the rated current
as this is the on-state losses of the solid state circuit breaker. The voltages are
also compared for 10 kA since this is the system current when the commutation is
finished in the hybrid circuit breaker and hence the voltage the arc in the mechanical
switch has to overcome. The numbers correspond to the voltage drop in volts at 25
40
CHAPTER 5. THE CHOICE OF SEMICONDUCTOR COMPONENTS
and 125 degrees junction temperature respectively. As seen in the table, the IGCT
fulfils its purpose to have a low voltage drop. Even if the voltage drop is adjusted
for the higher current capability of the IGCT, it is almost 30% lower than for the
IGBT.
Even though all the components can handle to conduct higher currents than
2 kA, the IGCT is the only one that can interrupt this over-current which makes
it very interesting. The voltage drops at 10 kA are in parentheses since they are
estimated or interpolated from the data-sheets. All components should handle
surge currents of this amplitude for a limited period of time, but it is not clear
from the data-sheet how the voltage drop will change. Since several components,
or at least chips, will be required in parallel to fulfil the current requirement, a
positive temperature coefficient is desired to enhance equal current sharing. As
seen in Table 5.2, this is generally fulfilled for high power devices at rated current.
However, it can be seen in the data-sheets that it is not the case for low currents.
Often it is a desired feature that is enabled by smart design of the chip. For the
IGBT this is necessary as several chips are already used in parallel in the same
component.
5.5
The optimal component
It is clear that the requirements of the semiconductor in a DC circuit breaker differ
very much from those in a converter application. Also the optimal component seems
to be different between a pure solid-state topology and a hybrid.
For the solid-state breaker, due to the on-state conduction losses the IGCT
seems to be the best choice. The low forward voltage drop compared to other
components decreases the losses and the demand on the cooling system. Also
no continuous energy feed is required due to the thyristor structure, where the
component locks in conduction mode until actively turned off. Since the IGCT
also can handle transient currents of about double the continuous current, there
should be margin for both over-load and time-delays for intelligent fault detection
schemes.
The choice of optimal component for the hybrid seems to be harder. Also here,
the IGCT with its low voltage drop is beneficial to reduce the demands on the
mechanical switch. However, since the semiconductor branch in a hybrid circuit
breaker is only required to conduct current for about one millisecond, components
requiring high input energies during conduction can be used. Due to this, it could
be possible to design a specific component for this application. For example a BJT
of MOSFET could be used if the component ratings could be increased for a limited
conduction time. The benefit would be a less complicated component that provides
more robustness and also a lower price due to easier manufacturing.
New more efficient power semiconductors are developed in novel materials as
silicon carbide [63]. Even though this development might reduce the on-state losses
in the components with a factor of 100 [64], the high currents in a DC grid will
5.5. THE OPTIMAL COMPONENT
41
still generate high power losses that will be hard to handle without large cooling
systems. Hence this development will be beneficial not only for the pure solid-state
circuit breakers, but also enhance the use of hybrid circuit breakers in the future.
Chapter 6
The snubber circuit
6.1
Snubber circuits
Switching times of power semiconductors are often in the µs range. Considering
high voltages and currents, this might cause undesired transient phenomena or
trigger oscillations between capacitive and inductive elements in the circuits. To
avoid these undesired transients, external components are often added in series and
parallel to semiconductors to slow down the switching. These external capacitors
or inductors are generally referred to as snubber circuits.
In most cases, the snubber circuits are added to protect the semiconductor
component itself or to reduce the losses in the semiconductor during the switching
sequence. Snubber circuits may be required during turn-on, turn-off, or at both
switching operations.
Turn-on snubber
Thyristors and GTOs will be turned on by a current pulse on the gate that injects
excess carriers into the component. As the component is forward biased with a
voltage, the component will start to conduct current. Large carrier densities will
build up close to the gate region, and starts to spread into the rest of the component.
As it takes time to spread the charge carriers into the whole conduction area of the
component, there is a highest allowed rise of the current that is specified in the
component data-sheet.
If the external circuit would cause a rise in the current higher than the allowed
limit, the current becomes too high before the charge plasma has fully formed.
The internal heating of the component will be very high and local and might cause
the component to be damaged or even fail. The solution is to limit the rise of the
current with a snubber inductor in series with the component. The turn-on snubber
is undesired as the inductor has to withstand the full system current and voltage.
43
44
CHAPTER 6. THE SNUBBER CIRCUIT
Turn-off snubber
The turn-off characteristics of the semiconductor will depend highly on the choice
of component. However, all active turn-off high power semiconductors will force
the current to zero by removing the charge carriers inside of the component. Due
to the internal structure, some components have restrictions in the highest allowed
voltage derivative during turn-off. One example is the GTO where a too fast rise
in anode-cathode voltage might cause a re-triggering of the GTO into conduction
mode. This would result in a failed interruption, and depending on the external
circuit, a destroyed component. By connecting a snubber capacitor in parallel to
the component, the rise of the voltage is limited and the component can be kept
within its ratings.
Even with components that don’t require a turn-off snubber for a successful
interruption, a snubber circuit might help to increase the safe operating area of
the component or to decrease the switching losses. The former might be desired as
it can reduce the number of components to use in parallel and series to fulfil the
system requirements. The latter is desired in high speed switching where the losses
from the switching constitute a significant part of the total losses in component.
In many applications the semiconductor is turned off with a parallel branch.
One example is converters where the current is commutated into a free-wheeling
diode. In this case, the only voltage appearing across the semiconductor during
switching is the inductive voltage drop in the loop between the semiconductor and
the free-wheeling diode. The voltage stress on the semiconductor is then small and
a snubber circuit might not be required to keep the component within the safe
operating area.
To minimize the cost of the semiconductor branch, it is desirable to switch the
components as close to their voltage rating as possible. In the DC-breaker application, the parallel branch is a metal oxide varistor that should give a clamping
voltage between one and two times the system voltage. As this will set the voltage requirements for the semiconductor branch, there is no margin for transient
over-voltages during the switching. This chapter describes the possible actions to
limit the voltage during switching and maximize the utilization of the semiconductors. The working principle of the MOV in the energy absorbing branch is further
discussed in 7.
Switching over-voltages in DC-breakers
In a DC-breaker, the semiconductor switch will only force the current into the
parallel energy absorbing branch. A simplified schematic drawing of the grid and
breaker is shown in Fig. 6.1. The energy absorbing element in the form of an MOV
will provide a voltage higher than the system voltage to force a current zero.
By considering the commutation loop between the semiconductor and the MOV,
and the undesirable stray inductance in that loop, the voltage across the semiconductor switch can be written
6.1. SNUBBER CIRCUITS
DC Grid
Rseries Lseries
+
Usource
45
Breaker
Lst MOV
Fault
PE switch
Load
Figure 6.1: Schematic diagram of a DC grid in the case of a fault between the
breaker and the load impedance.
Usemiconductor = UMOV + Lstray
diMOV
.
dt
(6.1)
Even though the stray inductance might be small the over-voltage can still
be significant. In the case of a high power semiconductor, a current of several
kiloampere is switched in a few microseconds, resulting in an over-voltage of 1
kilovolt per each microhenry stray inductance. An example from a low voltage
experiment is shown in Fig 6.2. It is desired that the IGBT voltage (green) should
closely follow the MOV voltage (magenta) to minimize the voltage stress on the
IGBT. In this set-up the stray inductance is exaggerated to compensate for the low
currents involved in the experiment.
The main reason behind the problem is the powerful switching performance
of the IGBT. Fig. 6.3 shows the typical waveform of an IGBT during turn-off.
The figure contains the three relevant parameters: the gate voltage (blue), current
(dotted red), and the voltage across the component (dashed green). As the control
of an IGBT is based on a MOSFET structure, the turn-off process is similar to that
of a MOSFET [65].
When the turn-off signal is sent at t = 0, the component is conducting nominal current with a sufficient voltage applied on the gate. The voltage across the
component is equal to the forward voltage drop. After a short dead time, the component will start the turn-off process at t1 as the gate voltage starts to decrease.
As the gate voltage drops, it will reach an intermediate level known as the Miller
plateau at t2. This voltage is the minimum gate voltage required to keep the IGBT
in conduction mode for the conducted current. The voltage across the component
starts to rise while the gate voltage remains constant. At t3, the voltage across
the component has reached its rated value and the charge-carriers will start to be
removed through the gate. The current through the component will now rapidly
decrease linearly down to a few percent of the nominal current. At t4, the gate
voltage reaches the threshold value where the component is fully turned off. From
46
CHAPTER 6. THE SNUBBER CIRCUIT
Figure 6.2: Transient over-voltage across the IGBT due to undesired stray inductance during switching in a low-voltage experimental set-up.
here, the last of the current decreases exponentially as the last charge carriers are
removed by internal recombination.
The problems with over-voltages during the switching comes mainly during the
linear decrease of the current. During this decrease, the current is forced down with
a constant derivative almost regardless of the surrounding circuit. The result is an
induced over-voltage across any inductance in the circuit as previously seen in Fig
6.2.
6.2
Possible solutions
The obvious solution to this problem is to decrease the stray inductance. This
should be done as much as possible as there is no benefit of having an inductance
in the loop. However, there will be a limit how small this inductance can be. First
of all, there will always be an inductance in the leads and a self inductance in the
components, so zero inductance can not be reached. Further, depending on the voltage level of the breaker, some distance will be required between the different nodes
to ensure a low electric field in the air to obtain the desired insulation strength.
Finally, the distance between the semiconductor and MOV will also depend on the
thermal design. As all energy absorbed by the MOV will be dissipated as heat and
unnecessary heating of the semiconductor should be avoided, some clearance will
be required.
6.2. POSSIBLE SOLUTIONS
47
Figure 6.3: Typical turn-off waveform of an IGBT.
Gate resistance
Another possibility for decreasing the over-voltage is by decreasing the switching
speed of the semiconductor [66]. For an IGBT, the rate of decrease of charge carriers
can be controlled by changing the gate resistance. With a higher gate resistance,
the draining of the component will be slower and hence the current decrease will
be less steep. Since the full voltage is already across the IGBT when the current
starts to decrease, there will be a massive power dissipation inside the IGBT as the
current decreases. This solution will then be limited by the amount of energy that
can be absorbed by the IGBT without a destructive over-heating [67].
C-snubber
In cases where a snubber is required during turn-off of a semiconductor switch, the
most common solution is the C-snubber [68]. It basically consists of a capacitor
that is connected in parallel to the semiconductor. As the semiconductor is turned
off, the voltage rise across the component will be slowed down. When the current
starts to decrease, it will partially be commutated into the snubber and the voltage
rise will be determined by the current and the capacitance. If the capacitance
is high enough, the current commutation to the MOV will be slowed down and
the over-voltage will be limited. Another benefit of this snubber is that since the
voltage rise across the semiconductor is delayed, the power dissipation will decrease
which may increase the operation limits of the semiconductor.
48
CHAPTER 6. THE SNUBBER CIRCUIT
RCD-snubber
The problem with a simple capacitor in parallel with the semiconductor is that the
capacitor will hold the full charge after turn-off. When the component is turned on
again, this would lead to a very fast and high current pulse that might harm the
semiconductor. In fast switching applications it would also increase the turn-on
losses which decreases the system efficiency. The solution is to limit the inrush
current by connecting a resistor in series with the capacitor [69]. However, during
the turn-off the resistor would destroy the effect of the capacitive snubber. Hence, a
diode is connected in parallel to the resistor to form the Resistor-Capacitor-Diode
(RCD) snubber shown in Fig. 6.4. During turn-off the semiconductor will see a
capacitor in series with a diode, but during turn-on the inrush current will be
determined by the capacitor in series with the resistor. Another benefit of this
configuration is that the resistance will help to damp any oscillations between the
snubber capacitor and the stray inductance in the MOV-loop. The drawback is the
complexity as three components are required to obtain the desired limitation of the
switching over-voltage.
R
C
D
IGBT
Figure 6.4: Basic circuit layout of an IGBT with an RCD snubber.
6.3
The parallel MOV snubber
The newly proposed solution is to use another MOV as a snubber [70] by installing
it much closer to the IGBT than the energy absorbing MOV. This results in a much
lower stray inductance and hence shows no problems with over-voltages due to the
fast change in current. The main idea is to separate the two tasks of the MOV:
limitation of over-voltage and absorption of the inductive energy. The voltage across
the semiconductor would be determined by the snubber MOV and the energy should
be absorbed by the main MOV. As the snubber MOV is not supposed to absorb
more than a fraction of the energy, a smaller component can be used. This in turn
enables the possibility to connect the snubber MOV close to the semiconductor
with a very low stray inductance. The basic circuit diagram of the snubber MOV
is shown in Fig. 6.5 where the snubber is added with dashed blue lines.
6.3. THE PARALLEL MOV SNUBBER
49
Lst
MOVE
′
Rseries
Usource
Lseries
Lst MOVov
IGBT
Figure 6.5: Circuit diagram of the breaker set-up with the proposed MOV snubber
in dashed blue.
Experimental set-up
This new concept of snubbers have been evaluated with a low power experiment.
The test circuit was built up according to the system in Fig. 6.1 with a 12 V battery
as the voltage source. The interruption tests were performed with a constant load
current, i.e. no fault was introduced in the circuit. The components were soldered
to a circuit board and a photograph of the set-up can be seen in Fig. 6.6. The load
resistance was chosen as 3.3 W and the current limiting inductance is an air wound
inductor of 2.3 mH. The resistance of the inductor is a few milliohm and hence
negligible compared to the load resistance.
For this low power set-up there are a number of possible semiconductor components that could have been used as the switching element. However, as the
concept is to be generalized for different power levels, an IGBT was chosen since
it is available in all ranges from a few amperes to several kiloamperes rating. The
IGBT used has ratings well over the requirements to make sure it can handle the
occurring over-voltages without destruction.
In order to properly see the inductive over-voltage even with the low current
and current derivatives in the experiment, a large inductance was added in the loop
between the IGBT and the MOV. The inductance is a single square turn in air,
mounted on a wooden board. The loop was 710 by 550 mm and the inductance
was estimated to just above 3 µH.
The previously discussed graph in Fig. 6.2 of an IGBT over-voltage during
switching was obtained using this set-up.
In this set-up, varistors from EPCOS [71] have been used. The type used
is called S05KXX which is the lowest power rating available of these varistors.
They are here referred to as KXX, where XX should be replaced with a number
corresponding to the highest AC voltage level the component can be continuously
exposed to without risk of over-heating. Each type also have a corresponding DC
50
CHAPTER 6. THE SNUBBER CIRCUIT
level which is typically 25-30% higher than the AC value. e.g. the S05K11 can
handle 11 V AC and 14 V DC and is referred to as K11.
These are varistors for electronics applications that can handle a transient peak
current of maximum 100 A. All experiments are run with an energy absorption
below 1% of each component rating to ensure there is no degradation [72] in the
components between the tests. According to the specification, the components
should handle more than 100 operations at this energy level. The component with
the lowest voltage rating used is the K11, hence all components are suitable for use
in the 12 V experimental set-up.
Figure 6.6: Typical turn-off waveform of an IGBT with a parallel capacitor as a
snubber.
Experimental Results
When interrupting a current in the solid-state breaker considered in the set-up the
current will be commutated into the MOV where the magnetic energy is absorbed.
Figure 6.7 shows the normal operation where a constant current is interrupted by
the breaker. The IGBT turns off and the current is commutated in to the MOV
that absorbs the magnetic energy. The conduction voltage of the MOV is almost
constant which causes the current to decrease almost linearly. The fast transition
6.3. THE PARALLEL MOV SNUBBER
51
from conducting to blocking of the MOV causes an oscillation when the current
reaches zero. The frequency shows that the oscillation occurs between the series
inductance of the circuit and the capacitance in the MOV. This transient is not
further discusses here.
Figure 6.7: Typical current interruption in the solid-state breaker considered in the
LV test set-up. The IGBT is turned off and the current is rapidly pushed into the
MOV where the magnetic energy is absorbed.
Figure 6.8 shows a zoom of Fig. 6.7. The stray inductance between the IGBT
and MOV has been minimized here by mounting them both closely on the same
circuit board. It can be seen that the IGBT and MOV voltages follows each other
closely and there is no transient over-voltage during the IGBT switching. The small
over-voltage on both voltages compared to the constant conduction voltage is due to
the self-inductance in the MOV. It can also be noted that the switching of the IGBT
is done in just fractions of a microsecond showing that there will be a significant
current derivative even with low currents. If the IGBT voltage is compared with
the IGBT voltage in Fig. 6.2, the large effect of the stray inductance can be seen.
The main idea of the concept is clearly shown by Fig. 6.9. Here a K11 is used as
the outer, energy absorbing MOV, and a K20 is used as a snubber. When the IGBT
is turned off, the current is rapidly interrupted in the IGBT and taken over by the
two MOVs. The snubber MOV takes the current initially as the stray inductance
in this branch is negligible. Hence the current in the snubber MOV rises and the
voltage across it and the IGBT rises according to the current-voltage characteristics
of the K20. The voltage across the IGBT will reach it peak as the current in the
IGBT becomes zero. Since this voltage is higher than the corresponding voltage of
52
CHAPTER 6. THE SNUBBER CIRCUIT
Figure 6.8: Typical current interruption in the solid-state breaker considered in the
LV test set-up. The IGBT is turned off and the current is rapidly pushed into the
MOV where the magnetic energy is absorbed.
the K11 at the same current, the current will also start to rise in the outer MOV.
The current is now commutated from the snubber MOV into the outer MOV and the
voltage across the IGBT decreases as the snubber current decreases. At t = 0.6 µs,
the snubber current reaches zero and all current is conducted by the outer MOV.
The energy is now absorbed in the MOV and the current is forced to zero as in a
normal operation. This is not seen in the graph, as the time scale for the current
decrease is 300 µs.
This over-voltage can be compared with Fig. 6.2 to see the limiting effect of the
snubber. The voltage peak is decreased from 120 V to 50 V and the oscillation is
eliminated. Since the snubber MOV only conducts current for a very limited time,
the amount of absorbed energy will be very small and hence the possibility to mount
the snubber varistor close to the IGBT with low stray inductance is motivated.
Even though the peak voltage across the IGBT is significantly decreased using
the K20 as a snubber, the peak voltage is still higher than the steady-state voltage
of the outer MOV. This requires a higher voltage rating of the IGBT and hence
it is desirable to further decrease this voltage. Figure 6.10 shows an experiment
where a K14 has been used as a snubber, still with a K11 to absorb the energy as
the outer MOV.
It can be seen that the voltage peak is further limited due to the lower voltage of
the K14 compared to the K20 in Fig. 6.9. The peak current in the MOV is mostly
controlled by the inductance and hence is not much affected by the voltage level
6.3. THE PARALLEL MOV SNUBBER
53
Figure 6.9: Successful test with a K20 as snubber and a K11 as outer MOV.
of the snubber. However, since the voltage difference between the K14 and K11 is
not too big, the commutation of the current to the outer MOV will be slow. This
means the snubber MOV will conduct current for a longer time and the absorbed
energy will be larger. As seen in the experiments, the voltage peak is limited to
35 V, but the current commutation from the snubber MOV to the outer MOV
is too slow. The difference in voltage rating between the two components is not
enough to rapidly commutate the current and even after 1.5 µs the snubber MOV
is conducting about one third of the current. This means the snubber MOV will
absorb a significant part of the energy and hence the motivation of using a smaller
component that can be mounted with low stray inductance falls. Therefore, the
test is considered to fail even though the voltage characteristics are desirable.
It is evident from the previous tests that a component with a voltage rating
between the K20 a K14 would be desired as the K20 provides a too high voltage
peak while the K14 absorbs too much energy. Unfortunately no such component
is available from the manufacturer. Another way to slightly alter the currentvoltage characteristics is by parallel connection of several components. This way
the current is shared between several components so that each component carries
less current and the effective voltage across the components for a specified current
will be slightly lower.
In this case, additional components have been added in parallel to the outer
MOV. This is not a reasonable design procedure as the requirements of the outer
MOV are set by the system voltage and the magnetic energy to be absorbed in the
MOV. However, connecting several components in parallel as the outer MOV could
54
CHAPTER 6. THE SNUBBER CIRCUIT
Figure 6.10: Failed test with K14 as snubber and K11 as outer MOV. The snubber
absorbs too much energy for the concept to be successful.
correspond to using an outer MOV with higher energy rating than the snubber
MOV. This is in line with the idea of the snubber MOV, but one has to be careful
when comparing the results with the previous tests. The main purpose of the test
is to change the ratio between the voltage levels of the outer MOV and the snubber
to obtain a better performance.
Figure 6.11 shows a test where a K14 is used as snubber with 10 parallel K11 as
the outer MOV [73]. Since the effective voltage level of the outer MOV is decreased,
this test is somewhere between the previous tests with a K20 and a K14 as snubber.
The graph also shows the expected behaviour where the voltage peak is limited by
the K14 to about 35 V as in the previous test. However, the parallel connection
decreases the conduction level of the outer MOV from about 27 V to 23 V. This
decrease gives enough difference between the K14 snubber and the outer MOV to
commutate the current in around 1 µs. Hence the test is successful in both limiting
the over-voltage by the snubber MOV and absorbing the energy in the outer MOV.
Figure 6.12 shows the peak voltage and the absorbed energy in the K14 used
as snubber MOV when the number of parallel K11 are changed as the outer MOV.
It shows that the IGBT peak voltage is almost entirely determined by the snubber
MOV and that the absorbed energy decreases when the characteristics of the outer
MOV are changed. With a good choice of components, it is possible to limit the
absorbed energy to just a few percent enabling the choice of a smaller component
as snubber and hence justifying the concept of an MOV as snubber.
The concept of using MOVs as a snubber still relies on the different voltage levels
6.3. THE PARALLEL MOV SNUBBER
55
40
10*ITOT
10*IIGBT
35
10*IMOV_E
Current [A], Voltage [V]
30
10*IMOV_ov
U
IGBT
25
UMOV_E
20
15
10
5
0
0
0.2
0.4
Time [µs]
0.6
0.8
1
Figure 6.11: Successful test with a K14 as snubber and 10 parallel K11 as outer
MOV.
50
IGBT peak voltage
Absorbed energy in MOVov
45
Voltage [V], Part of energy [%]
40
35
30
25
20
15
10
5
0
0
2
4
6
8
Number of parallel K11 as MOVE
10
Figure 6.12: The peak-voltage and energy absorption for 10 different configurations
with a K14 as snubber and 1 to 10 parallel K11 as outer MOV.
of the two MOVs used. Figure 6.13 shows the IGBT peak voltage and the absorbed
56
CHAPTER 6. THE SNUBBER CIRCUIT
energy in the snubber MOV when a K11 is used as snubber and the number of
parallel K11 as outer snubber is varied. It would be expected that the absorbed
energy starts at 50% with only one K11 as outer MOV as the two MOVs would
share the current equally after the first initial switching transient. However, the
curve starts lower than this due to the variation of the exact voltage level of the
MOV. In this case it seems that the outer MOV has slightly lower voltage than the
snubber MOV resulting in that most of the energy is absorbed in the outer MOV.
When more components are connected in parallel to the outer MOV, the amount
of energy absorbed in the snubber decreases further. It would be expected that the
absorbed energy should as 1/x, where x is the number of parallel components, i.e.
from 50% with one component to 10% with 9 parallel components. Due to the one
component with lower voltage characteristics, the absorbed energy will always be
lower than this as a large share of the current and hence the energy is taken by this
component.
The experiment shows nothing of how the energy absorbed in the outer MOV
is divided between the parallel components, but this is also not too interesting. In
a real application one would not use parallel components if there are other components with higher current and energy ratings available. Due to the sharp currentvoltage characteristics of the MOVs one has to be careful when connecting them
in parallel as just a small difference in voltage level, even within the specification,
might severely affect the distribution of absorbed energy during transients [74].
50
IGBT peak voltage
Absorbed energy in MOVov
45
Voltage [V], Part of energy [%]
40
35
30
25
20
15
10
5
0
0
2
4
6
8
Number of parallel K11 as MOVE
10
Figure 6.13: The peak-voltage and energy absorption for 10 different configurations
with a K11 as snubber and 1 to 10 parallel K11 as outer MOV.
6.3. THE PARALLEL MOV SNUBBER
57
Modelling
The experiments can only be performed with a limited number of components due
to the available components from the manufacturer. The experimental data is also
subject to some variations in the components within the component specifications.
To generalize the study, a circuit simulation model was built in EMTDC/PSCAD.
The circuit diagram of the model is shown in Fig. 6.14. As the switches used in
PSCAD are ideal switches, some modelling work is necessary to obtain the observed
over-voltages during switching. The model is built using basic components to mimic
the different intervals of the turn-off and some of the component values have no
physical interpretation.
The model consists of a voltage source and a series connection of a resistor and
an inductor limiting the current and providing the necessary magnetic energy for
making the current harder to interrupt. The main IGBT in the experimental set-up
is modelled by the IGBT called IGBT1. To mimic the turn-off delay of the IGBT,
another IGBT is connected in parallel and turned off with a slight time delay. In
series with this IGBT is the capacitor C1 that causes the voltage across the IGBT
to increase during the time-delay. In parallel with this there is a network of resistors
and capacitors to mimic the IGBT turn-off and the oscillations that might occur.
The model also contains the two MOVs, the snubber MOV closest to the IGBT
and the main energy absorbing outer MOV in the top of the diagram. Each MOV
is modelled with the built in, non-linear resistor model with custom current-voltage
characteristics data. In series with the non-linear resistor model is a resistor and
an inductor representing the resistance of the leads and the self inductance of the
component. In parallel to this is a capacitor, whose value is taken from the MOV
data-sheet as well as a resistor to damp oscillations.
In each branch, there is also a stray inductance that slows down the current
commutation.
For the model to have a good precision, accurate current-voltage characteristics
of the MOVs have to be used. PSCAD offers a non-linear resistor component in
the standard library where the current-voltage characteristics can be provided in
an external file with more than 100 points. Figure 6.15 shows measured currentvoltage characteristics for three different MOVs used in the experiments. All three
components have the same diameter and differ only by the voltage level. As the
characteristics should be a pure material property, the curves for each component
can be scaled to obtain a current-voltage characteristic that is valid for all voltage
levels. By normalizing each of the curves in Fig. 6.15 to the rated voltage level of
the respective component, this universal characteristics curve was obtained. After
the normalization, the mean value of the three curves was used in the model.
This way, different MOV configurations can be simulated by just changing the
rated voltage of the non-linear resistor model and the capacitance according to the
components to be simulated. To increase the number of possible components in the
simulations, the capacitance can be interpolated from the existing components for
the non-existing voltage levels.
58
CHAPTER 6. THE SNUBBER CIRCUIT
Figure 6.14: The circuit diagram of the simulation model used in PSCAD to generalize the investigation of the parallel MOV snubber concept.
Model verification
To calibrate the model, one of the previous experiments with a K11 as outer MOV
was used. The component values were varied to get a good fit of simulation results
compared to the measured currents and voltages. Figure 6.16 shows the agreement
of the simulation model and the experiment.
6.3. THE PARALLEL MOV SNUBBER
59
Figure 6.15: Measured current-voltage characteristics for three of the different MOV
models used in the experiments. The spread between the data points is due to the
noise in current and voltage measurements.
To simulate different configurations, only three parameters are changed: the
rated voltage of the MOV; the capacitance of the MOV; and the inductance in the
loop between the IGBT and the outer MOV. They are all real physical parameters
from the circuit or the data-sheet which makes the simulation model useful.
To validate that the model is reflecting the actual switching properties for a
wider range of components, three different inductances were used as shown in table
6.1. Figure 6.17 shows the difference in the peak IGBT voltage from the simulations compared to the experimental data. The plot contains 9 experiments and
simulations, where the first one was used to calibrate the model parameters. The
dashed line shows that the mean error in the nine cases are slightly above 3%. It
should also be noted that none of the configurations show a significantly higher
error than the other meaning that there is no clear systematic error in the model.
In the simulations, only loop A is considered.
Simulation results
The simulations represent a generalization of the experimental results where the
components are ideal in the sense that there are no variations between the components. To compare different set-ups, the parameter α is defined as the voltage ratio
of the snubber MOV and the outer MOV. The voltage of each MOV is measured
for a DC-current of 10 µA and is used to calculate α. In the simulations α is calcu-
60
CHAPTER 6. THE SNUBBER CIRCUIT
Figure 6.16: The model was calibrated to a good fit using an experiment with
K11 as outer MOV and the inductance according to loop A. The graph shows
the experimental curves in solid lines and the corresponding simulation curves in
dashed lines.
Table 6.1: Three different loops have been used to represent the stray inductance,
Lst , between the IGBT and the MOV in validation of the model.
Height [mm]
Width [mm]
Estimated Lst [µH] [75]
Loop A
710
550
3.03
Loop B
340
550
2.00
Loop C
0
lated from the rated voltage of the components. Both methods are based on that
the current-voltage characteristics of the MOVs are pure material parameters and
hence scalable with the voltage level.
Figure 6.18 shows a set of experiments and simulations where the absorbed
energy is plotted as function of α. The simulations are shown as stars and the
experiments as circles. Each colour represents a specific combination of MOVs and
can be deducted from the legend in Fig 6.19.
The dashed orange lines represents two exponential functions, seen as straight
lines in the log-scale, fitted to the simulation data in two intervals. The left line
is fitted to the five leftmost points and the right to the three rightmost. The left
line should also be compared to the black square that represents the case α = 1,
6.3. THE PARALLEL MOV SNUBBER
61
Figure 6.17: The calibrated model was validated by comparing the difference in
IGBT peak voltage of the model to the measured value in 9 different configurations.
which should render 50% energy absorbed in each MOV. It can be seen that the
line slightly underestimates this level.
The same kind of fit is performed for the experimental points in the same intervals and shown with the dotted pink lines. It can be seen that the two left lines
have the same slope, but with a vertical offset. The line from the experiments is
obviously wrong as the interpolated value for α = 1 ends up above 100%. For the
right interval, the experiments get a lower slope due to the rightmost point close to
α = 2. The dash-dotted cyan line is a vertical fit of the data points with the same
slope as the fit from the simulations.
It seems that the uncertainty in the calculation of the energy in the experiments
is too large to draw any conclusions. However, the fits from the simulation curves
clearly splits the graph into two intervals where the absorbed energy can be considered low or high. For the concept to make sense, the absorbed energy should be
kept low and hence an α in the right part should be chosen, i.e. α > 1.3.
Figure 6.19 shows the result of 8 experiments with different combinations of
MOVs. The legend shows the two MOVs used where the first (with higher rating)
is the snubber MOV and the latter is the outer MOV.
The uncertainty in the measurements of the MOV voltage at 10 µA is large.
Since this is used to calculate α, the spread in horizontal position of the experiments
might be significant. Figure 6.19 shows the same experiments as Fig 6.18 together
with the theoretical α calculated from the rated voltage of the components. The
arrows shows the uncertainty [76] propagation of errors in α due to the specified
62
CHAPTER 6. THE SNUBBER CIRCUIT
Figure 6.18: The absorbed energy in the snubber MOV as function of the voltageratio between the voltage levels of the snubber MOV and the outer MOV. Simulations as stars and the experimental data as circles. The plots can be divided into
two distinct intervals shown with the lines.
variations (±10%) in voltage level of the MOVs. It can be seen that the measured
values of α are within the expected intervals, even if the spread is large.
It can be seen that the variation within the components specifications can have
a disastrous effect on the result. Consider for example the experiment of the K35K25 set-up shown in orange. From the component ratings one is expecting α =
35/25 = 1.4 which is in the low energy absorption region according to the fit from
the simulations (dashed lines). The energy absorption would end just above 2%.
However, due to the variations within the component specifications, the arrows
show that the real value of the combination will be within 1.2 < α < 1.6. An α
as low as 1.2, ends up in the high energy absorption area, resulting in an absorbed
energy of above 6%, i.e. 3 times higher than the desired value.
Figure 6.20 shows the results of a set of simulations where alpha has been
swept from 1 to 4 with a K11 as outer MOV. The graph shows the design tradeoff one has to consider when choosing the voltage level of the snubber MOV. As
α is increased, the absorbed energy in the component decreases at the cost of
increasing peak voltage across the IGBT. Hence one should chose an alpha that is
as low as possible while still keeping the absorbed energy in the snubber within the
component specification.
It should also be noted here, that for all α, the peak IGBT voltage is lower than
without the MOV snubber.
6.3. THE PARALLEL MOV SNUBBER
63
Figure 6.19: Absorbed energy according to the experiments shown in circles. The
squares shows the expected α according to the rated voltage of the
In this study, the diameter of the MOV is the same for both the outer MOV
and the snubber which enables the comparison of voltage levels and the use of
the parameter α. In a real case, the energy rating of the snubber and hence the
diameter of the MOVs will be lower than for the outer MOV resulting in a shift
of the energy absorption curve to the left resulting in less energy absorbed by the
snubber MOV.
Benefits
The use of a snubber increases the maximum current that can be switched within
the voltage safe operating area (SOA) of the IGBT since the peak voltage is limited. This increases the utilization of the IGBT which is necessary to get the
semiconductor part of the hybrid DC-breaker competitive in price.
The idea of the parallel MOV snubber is to provide a more robust snubber with
high reliability. The three components in the RCD snubber are replaced by one
passive component. As an MOV is already used for absorbing the energy, no new
technologies are introduced in the circuit. The small MOV used as snubber is also
a cheap component compared to the RCD snubber.
By introducing the snubber MOV, the current commutation during the IGBT
turn-off the changed from current driven to voltage driven. With the current driven
commutation where the IGBT forces a negative current derivative, any inductance
will cause an over-voltage. With the snubber MOV, the voltage will be clamped
64
CHAPTER 6. THE SNUBBER CIRCUIT
Figure 6.20: There will be a trade-off between the peak voltage the IGBT is exposed
to and the energy that has to be absorbed by the snubber MOV.
across the IGBT and the voltage difference between the snubber MOV and the
outer MOV will commutate the current. The effect of the stray inductance is then
limited to slowing down the commutation and the peak voltage is determined by
the snubber MOV.
Limitations
Since the current is supposed to commutate from the snubber MOV to the outer
MOV, the voltage level of the snubber MOV has to be higher than the voltage level
of the outer MOV. This results in that the peak voltage across the IGBT will be
higher than the desired value set by the outer MOV.
Further, one must have a good idea about the magnetic energy stored in the
grid not to exceed the energy absorption capability of the MOVs. However, this is
also the case for dimensioning the outer MOV, so it should not cause any further
limitations.
The variations of current-voltage characteristics within the component specifications can be disastrous for the set-up. To use the concept with good reliability,
the voltage levels of the MOVs have to be specified with tight tolerances. This
is already done today for high power surge arresters in the power grid to closely
match components that are to be connected in parallel to ensure equal current and
energy sharing.
The snubber MOV is only possible to use in breaker applications where the
6.3. THE PARALLEL MOV SNUBBER
65
magnetic energy stored in the grid inductance is considered as waste. This way the
switching losses of the breaker are not relevant as the energy should be dissipated
as heat. In continuous switching applications like inverters or rectifiers, the snubber MOV would lead to very high switching losses and hence imply a decrease in
efficiency.
Chapter 7
The energy absorbing branch
7.1
Background
To protect the components in the power system from over-voltages due to lightning
strikes and switching transients, different kinds of surge arresters [77] have been
used. One solution is to use spark gaps in series with non-linear resistors, typically
made of silicon carbide. However, the breakdown of the spark gap typically results
in a transient over-voltage higher than the expected protection level for very fast
rising pulses [78].
After being introduced as over-voltage protection for electronic circuits, the
research on zinc oxide (ZnO) for high power surge arresters started in the 1970s [79].
From the beginning the higher cost of the ZnO made high voltage systems the
main applications even though the component is well suited also for a broad set
of applications also for lower voltage levels [80]. Today there are many different
versions of ZnO varistors for various voltage levels with both DC and AC [81].
7.2
Basic working principle of the MOV
The ZnO is a wide band-gap semiconductor that in combination with additives
shows excellent non-linear resistive properties. The relationship between the current
density and electric field is a pure material parameter [82]. The MOV consists of
small grains of ZnO separated by a mixture of other additives. The powder is heated
and pressed into discs with proper thickness and diameter for different applications.
The result is a ceramic like material with highly conductive grains of ZnO separated
by insulating boundaries.
In addition to the non-linear current-voltage characteristics, the MOV will give
a higher voltage during the transient where is starts to conduct current. In [80],
the voltage level of an MOV is studied for current pulses with rise times of 0.1 to
10 µs. It is seen that for a 1 µs pulse, the peak voltage is about 10% higher than
for the 10 µs pulse. For the 0.1 µs pulse, the corresponding value is slightly below
67
68
CHAPTER 7. THE ENERGY ABSORBING BRANCH
1.3. For a switching applications, this over-voltage is much undesired as it leads to
increased requirements on the semiconductor voltage rating.
7.3
Dimensioning considerations
For applications as surge arresters in power systems, the MOV is continuously
exposed to the system voltage. Hence the rated voltage of the MOV has to be chosen
above this level to limit the continuous power dissipation. As a semiconductor
material the Metal oxide will have a negative temperature coefficient. This means
that the current flowing in the component will increase the conductivity of the
material and hence allow more current to flow. At a certain voltage, the increase
in conductivity is self sustained by the power and the MOV will go into a thermal
runaway and be destroyed. Using the rated voltage as design parameter means that
the protection level will be determined by the current-voltage characteristics of the
component.
In a switching application, the dimensioning of the MOV will be based on the
amount of energy that needs to be dissipated in one single operation. The time
interval for this is short, in the order of some milliseconds, and there is not much
time to dissipate this energy as heat to the ambient. This means that the energy has
to be absorbed in the ZnO material by increasing the temperature of the component.
Hence the dimensioning parameter of the MOV energy capability in a switching
application is the total heat capacity, i.e. the mass or the volume of the ZnO block.
If the MOV is exposed to higher energy than its rated energy, the material will
degrade so that the leakage current at a certain voltage will increase [83]. If the
component is exposed to large enough energy pulses, the leakage current at the
system voltage will become too large and the MOV will fail from over-heating [72].
7.4
The MOV simulation model
The non-linear characteristics can be modelled as [84]
i=
v n
,
VN
(7.1)
where VN is the characteristic voltage of the MOV and n is the non-linearity coefficient. With higher n, the MOV is more non-linear and shows better current-voltage
characteristics. Typically n is in the order of 30 to 50.
However, in this work that model is not used. This simulation model uses the
built in MOV model and sets the current-voltage characteristics according to the
data supplied in an external data-file. This data is obtained from the data-sheet of
a commercially available component, B72280B0112K001, with a rated DC voltage
of 1465 V and an energy absorption capability of 6 kJ [85]. For the simulations,
the data of this component is scaled to mimic components with different voltage
and energy ratings.
7.5. REQUIREMENTS ON ENERGY RATING
7.5
69
Requirements on energy rating
To decrease the current in the inductive grid, the magnetic energy stored in the
grid inductance at the current peak has to be absorbed by the MOV. This magnetic
energy can be written as
Wmag =
1
L
I2
.
2 system peak
(7.2)
If the MOV is seen as an ideal component it will block the current fully if the
voltage is below the rated voltage and conduct current with a constant voltage
UM OV . If the resistance in the grid is neglected, the system consists of only the
feeding grid voltage, the grid inductance and the MOV. Hence the current through
the grid can be expresses as
1
di
=
(U
− UMOV ).
dt
Lsystem system
(7.3)
When the current is conducted by the MOV and decreasing towards zero, the
grid is still feeding energy into the system trying to maintain the current. Hence
the energy absorbed by the MOV will be larger than the energy calculated in (7.2).
By setting up the power in the MOV as
PMOV = UMOV Isystem ,
(7.4)
and integrating this from Isystem = Ipeak to Isystem = 0, the total absorbed energy
can be written as
Wtot =
Usystem
1
Lsystem I02 (1 +
).
2
Uvaristor − Usystem
(7.5)
As seen by the equation, a higher voltage rating of the MOV will lead to a
lower energy that needs to be absorbed. The minimal energy to be absorbed is
the initial magnetic energy calculated in (7.2) which would be for a case with an
infinite MOV voltage that would immediately force the current to zero. Hence a
high MOV voltage will minimize the energy absorbed in the MOV, but will give
a higher peak voltage leading to higher requirements on the rated voltage of the
semiconductors and hence more expensive components.
If the MOV voltage is chosen lower than the system voltage, the current will
not decrease, and the energy absorbed in the MOV will increase until the MOV is
destroyed due to over-heating.
70
7.6
CHAPTER 7. THE ENERGY ABSORBING BRANCH
Choosing the MOV configuration
This section is based on simulations of a hybrid DC breaker in a 12 kV DC system
[86]. The semiconductor branch is considered to consist of IGBTs with a voltage
rating of 4.5 kV and limit the current peak at 12 kA.
Choosing the MOV configuration is a trade-off between the number of parallel
MOVs (or the diameter of the MOV) and the voltage rating (thickness) of the
MOV. As the two desired design parameters are ideally only geometry dependent,
any voltage and energy ratings are possible for the MOV. In reality, the choice is
limited to existing products, but in this work arbitrary ratings are considered.
Figure 7.1 shows simulated values of the peak voltage across the IGBT when
limiting a fault current at 12 kA. As the peak voltage sets the voltage rating requirements of the IGBT during the turn-off, the voltages are divided into discrete
levels corresponding to different number of IGBTs connected in series. The lowest
level (dark blue) is below 9 kV corresponding to only 2 IGBTs in series. The levels
then increase in steps of 4.5 kV corresponding to adding another IGBT in series.
For the highest level (dark red) the voltage is up to 27 kV which would require at
least 6 IGBTs in series.
Figure 7.1: The colour shows the peak voltage across the semiconductors depending
on the rated voltage level of the MOV. The graph is split into discrete voltage levels
corresponding to the voltage rating of 2-6 IGBTs in series.
The y-axis shows the voltage rating of the component. An increase in this value
will make the peak voltage higher and put more stress on the system. Along the
x-axis is the number of components connected in parallel. When the number of
7.6. CHOOSING THE MOV CONFIGURATION
71
parallel components are increased, the current is shared between the parallel paths
and hence decreases the peak voltage. It is desirable to choose a combination in the
top of one of the intervals as this maximizes the utilization of the semiconductors.
As described in (7.5) the absorbed energy in the MOVs will depend both on the
current and the voltage level of the MOV. Figure 7.2 shows the absorbed energy
in the MOVs for the same set of simulations as in 7.1. The absorbed energy is
normalized to the total energy absorption capability of the MOVs connected in
series and parallel. Hence, to guarantee a safe operation, the absorbed energy must
be kept below 1.
When the voltage rating of the MOV is increased, the amount of absorbed
energy will decrease due to the faster decrease of the current. If the number of
parallel components is kept constant, i.e. moving vertically in the graph, the energy
rating of the set-up will also increase since the volume of the MOV will increase.
When the voltage rating is kept constant and the number of parallel components
are increased, the absorbed energy will increase as the voltage across the MOVs
is slightly decreased. However, the energy rating of the set-up will increase and
hence the normalized value will decrease when moving horizontally to the right in
the graph.
Figure 7.2: The colour shows the absorbed energy in the MOVs as function of
the MOV voltage rating and number of components in parallel. The energy is
normalized to the energy rating and hence should be below 1, i.e. in the dark blue
area.
Figure 7.3 shows the combination of the two previous graphs where data has
been reduced to only the lines separating the different levels. As discussed before,
72
CHAPTER 7. THE ENERGY ABSORBING BRANCH
the absorbed energy should be kept below 1, and preferably close to 1 to utilize the
energy rating of the MOV. To utilize the semiconductor, the peak voltage should
be close to the voltage rating. The intersection of these lines gives three optimal
designs marked in the graph. From a system perspective, the absorbed energy is
very small and the cost of the energy itself is negligible.
The red asterisk shows a 9 kV MOV where 5 components are required in parallel.
The peak voltage is 22.5 kV requiring 5 IGBTs in series.
The blue star shows an MOV of slightly below 8 kV. Due to the lower voltage,
7 components are required in parallel, but only 4 IGBTs are required in series.
The lilac square shows an MOV with voltage rating 6.7 kV and requires 17
components in parallel. Here, only 3 IGBTs are required in series.
In can be noted that the set-up with only 2 IGBTs in series does not offer a
valid solution. The reason is that the peak voltage has to be kept below 9 kV, i.e.
lower than the system voltage of 12 kV. Hence it will not be possible to force the
current to zero and a finite absorbed energy in the MOV is not possible to obtain.
Which one of these designs that is optimal has to be determined from other
parameters such as cost and reliability analysis as well as evaluation of other system
parameters. Some examples are:
• The cost of the semiconductors are generally higher than the cost of the
MOVs. The total cost can be minimized with the right combination.
• The reliability generally decreases with an increased number of components
as the risk of failure increases. This way some combinations incorporating
e.g. many components in series or parallel might be excluded.
• In a hybrid breaker, the number of semiconductors in series will put higher
demands on the arc voltage of the mechanical switch. The same applies in
a solid-state breaker where more semiconductors in series will increase the
losses.
• The system and adjacent components should be considered. If the system
contains extra sensitive components, it may be worth to decrease e.g. the
transient over-voltage even with an increased cost as it might decrease the
overall system cost.
From a cost point of view, one would generally like to limit the over-voltage to
just above the system level, i.e. the choice of 3 IGBTs in series discussed above. This
would require a high energy absorption capability of the MOVs, but would save a
lot of cost due to the decreased number of semiconductors. However, when choosing
the voltage rating of the MOVs, one also has to consider the post operational current
of the system, i.e. the leakage current through the MOV when the breaker is open.
A disconnecting switch will always be required in series with the hybrid breaker
to provide galvanic isolation after an opening operation. If the voltage rating of the
MOV is chosen below the system voltage, it will conduct some current even after
7.6. CHOOSING THE MOV CONFIGURATION
73
Figure 7.3: The combination of the two previous figures show the three possible
designs to optimize the utilization.
the breaker has opened. This will lead to two demands: the disconnector has to
be able to interrupt this current; and it has to be fast enough to open before the
MOV overheats.
Chapter 8
Conclusions
The design of a hybrid DC-breaker is not straight forward. All different aspects
have to be considered either by several iterations or by the means of an optimization. Even when optimal designs are found for the technical performance, external
parameters as cost and reliability has to be considered for the final design and those
parameters are not always easy to quantify.
Some general design conclusions can be drawn from the studies in this work:
• The hybrid breaker is a good choice of topology if the system voltage exceeds
a few kilovolt so that the arc voltage of a mechanical breaker is not sufficient
to limit the current. With a mechanical switch in the main current path, the
topology has losses as low as in mechanical breakers and can switch fast rising
fault currents. Even though the hybrid breaker contains both semiconductors
and a mechanical part it is not necessarily more expensive than the pure
solid-state breaker as the need of a cooling system is decreased or removed.
Also since the semiconductors only conduct current for a very limited time,
components with lower current rating can be used.
• When switching DC-currents with high power semiconductors one has to be
careful in the design of the circuit not to exceed the voltage ratings of the
components. Due to the high current derivative, also a small stray inductance may result in harmful over-voltages. The use of parallel varistors as
snubber has been proposed and shown to decrease the over-voltages. This
increases utilization of the semiconductor and also reduces the demand of a
low inductive connection between the semiconductor and the varistor branch.
• The most critical part of the hybrid breaker is the mechanical switch. The
reaction time and commutation to the semiconductor will determine the interruption time of the breaker and hence the peak current in the system during
a fault. The other parts of the breaker has to be designed to handle this
peak current and this will determine the cost of the set-up. Both the semiconductors and the varistors are components commonly connected in series
75
76
CHAPTER 8. CONCLUSIONS
and parallel to increase current and voltage ratings respectively. Hence the
challenge is merely a question of obtaining the correct component ratings.
• The requirements of the mechanical switch depends highly on the system
configuration and the requirements on current limitation. It has been shown
that for a 12 kV DC system, the allowed opening time of the mechanical
switch has to be a few 100 microseconds. Further a fast contact movement
is desired as it results in a longer arc and hence a higher arc voltage. A
switch with this performance is definitely achievable using a electromagnetic
repulsion actuator in the form of a Thomson coil or a Double sided coil.
Chapter 9
Future Work
The hybrid DC-breaker shows promising results and should be further investigated.
Among the possible things to investigate in the future are:
• The parallel varistor concept should be studied in detail and verified with
high power.
• Further experiments should be performed to validate the models, especially
the parts for arc voltages and current commutations.
• A final design of the mechanical switch with both high switching performance
and high number of operations should be obtained. A prototype of the switch
should be manufactured and tested to verify the design.
77
List of Figures
2.1
2.2
2.3
2.4
2.5
3.1
3.2
3.3
4.1
4.2
4.3
4.4
4.5
4.6
4.7
4.8
4.9
Symmetric fault in an AC-system. . . . . . . . . . . . . . . . . . . . . .
Asymmetric fault in an AC-system. . . . . . . . . . . . . . . . . . . . . .
Fault in an inductive resistive grid with a stiff DC-source. . . . . . . . .
Fault current from the rectification of a three-phase AC system. . . . .
Level of the fault current with a rectified AC system after different times
showing that the benefit of a fast opening of the mechanical switch is a
lower current to commutate. . . . . . . . . . . . . . . . . . . . . . . . . .
Basic layout of the three parallel branches of a hybrid DC-breaker. . . .
Currents in the different branches of the hybrid DC-breaker during interruption of a fault current. . . . . . . . . . . . . . . . . . . . . . . . .
Voltage across the hybrid DC-breaker during interruption of a fault current. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Sketch of the Thomson coil (left) and Double sided coil (right). . . . . .
Equivalent circuit of the Thomson coil actuator system with its capacitive energy storage. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Current pulse through the Thomson coil and Double sided coil. . . . . .
Obtained force in the Thomson coil and Double sided coil when discharging a capacitor bank through the coil. . . . . . . . . . . . . . . . .
Velocity as function of time for the discussed set-ups. . . . . . . . . . .
Comparison of the inductance of the Thomson coil and the Double sided
coil as the armature separates from the primary coil. . . . . . . . . . . .
If the combination of switch opening speed and current during the
commutation is not sufficient, the commutation into the semiconductor branch will fail and the breaker will not manage to interrupt the
current. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
By decreasing the switch opening time, the commutation will occur earlier, i.e. with a lower current and succeed. . . . . . . . . . . . . . . . . .
By increasing the arc voltage of the switch, the higher arc voltage will
force a faster and successful commutation. . . . . . . . . . . . . . . . . .
79
7
8
9
9
10
13
16
16
18
19
21
21
22
23
25
26
26
80
List of Figures
4.10 Possible combinations of switch opening time and arc voltage to fulfil
the commutation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
4.11 Different design options for the mechanical switch to obtain the required
arc voltage. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
29
5.1
5.2
5.3
5.4
5.5
5.6
Symbol
Symbol
Symbol
Symbol
Symbol
Symbol
32
32
33
34
35
36
6.1
Schematic diagram of a DC grid in the case of a fault between the
breaker and the load impedance. . . . . . . . . . . . . . . . . . . . . . .
Transient over-voltage across the IGBT due to undesired stray inductance during switching in a low-voltage experimental set-up. . . . . . . .
Typical turn-off waveform of an IGBT. . . . . . . . . . . . . . . . . . . .
Basic circuit layout of an IGBT with an RCD snubber. . . . . . . . . .
Circuit diagram of the breaker set-up with the proposed MOV snubber
in dashed blue. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Typical turn-off waveform of an IGBT with a parallel capacitor as a
snubber. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Typical current interruption in the solid-state breaker considered in the
LV test set-up. The IGBT is turned off and the current is rapidly pushed
into the MOV where the magnetic energy is absorbed. . . . . . . . . . .
Typical current interruption in the solid-state breaker considered in the
LV test set-up. The IGBT is turned off and the current is rapidly pushed
into the MOV where the magnetic energy is absorbed. . . . . . . . . . .
Successful test with a K20 as snubber and a K11 as outer MOV. . . . .
Failed test with K14 as snubber and K11 as outer MOV. The snubber
absorbs too much energy for the concept to be successful. . . . . . . . .
Successful test with a K14 as snubber and 10 parallel K11 as outer MOV.
The peak-voltage and energy absorption for 10 different configurations
with a K14 as snubber and 1 to 10 parallel K11 as outer MOV. . . . .
The peak-voltage and energy absorption for 10 different configurations
with a K11 as snubber and 1 to 10 parallel K11 as outer MOV. . . . .
The circuit diagram of the simulation model used in PSCAD to generalize the investigation of the parallel MOV snubber concept. . . . . . . .
Measured current-voltage characteristics for three of the different MOV
models used in the experiments. The spread between the data points is
due to the noise in current and voltage measurements. . . . . . . . . . .
6.2
6.3
6.4
6.5
6.6
6.7
6.8
6.9
6.10
6.11
6.12
6.13
6.14
6.15
and
and
and
and
and
and
schematic
schematic
schematic
schematic
schematic
schematic
drawing
drawing
drawing
drawing
drawing
drawing
of
of
of
of
of
of
the
the
the
the
the
the
BJT. . . .
MOSFET.
JFET. . .
thyristor. .
GTO. . . .
IGBT. . .
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
28
45
46
47
48
49
50
51
52
53
54
55
55
56
58
59
List of Figures
6.16 The model was calibrated to a good fit using an experiment with K11 as
outer MOV and the inductance according to loop A. The graph shows
the experimental curves in solid lines and the corresponding simulation
curves in dashed lines. . . . . . . . . . . . . . . . . . . . . . . . . . . . .
6.17 The calibrated model was validated by comparing the difference in IGBT
peak voltage of the model to the measured value in 9 different configurations. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
6.18 The absorbed energy in the snubber MOV as function of the voltageratio between the voltage levels of the snubber MOV and the outer
MOV. Simulations as stars and the experimental data as circles. The
plots can be divided into two distinct intervals shown with the lines. . .
6.19 Absorbed energy according to the experiments shown in circles. The
squares shows the expected α according to the rated voltage of the . . .
6.20 There will be a trade-off between the peak voltage the IGBT is exposed
to and the energy that has to be absorbed by the snubber MOV. . . . .
7.1
7.2
7.3
The colour shows the peak voltage across the semiconductors depending
on the rated voltage level of the MOV. The graph is split into discrete
voltage levels corresponding to the voltage rating of 2-6 IGBTs in series.
The colour shows the absorbed energy in the MOVs as function of the
MOV voltage rating and number of components in parallel. The energy
is normalized to the energy rating and hence should be below 1, i.e. in
the dark blue area. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
The combination of the two previous figures show the three possible
designs to optimize the utilization. . . . . . . . . . . . . . . . . . . . . .
81
60
61
62
63
64
70
71
73
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