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Transcript
2.
The MOSFET
In the first chapter we adopted a simplified picture of the Metal-OxideSemiconductor Field Effect Transistor (MOSFET) as a switch – a switch that
could be turned ON and OFF by means of an electrical control voltage.
Furthermore, we tried to make feasible that such a simple model is sufficient
for understanding how to design CMOS logical gates. However, the switch
model is not detailed enough for evaluating circuit performance such as
switching speed of a CMOS logic gate. Therefore, we shall look more into
details of MOSFET models for integrated circuit design. This chapter will be
far from complete when it comes to MOSFET modeling, but it will provide a
useful platform for the beginner circuit designer.
Since the MOSFET has two ports, the main port and the control port, a
two-port representation as shown in Fig. 2.1 is useful. Here, the gate-tosource voltage VGS is the input control voltage not only used for turning the
device ON or OFF, but also for controlling to what level the device is ON.
Mathematically, both port currents IGS and IDS are nonlinear functions of the
two port voltages, VGS and VDS, the output port voltage1.
+ IDS Gate VDS + VGS ‐ ‐ Source
Drain + G VGS ‐ S IDS
IGS
The MOSFET as a two‐port
D + VDS ‐ S
Fig. 2.1.The MOSFET as a two-port.
A first-order two-port representation of the MOSFET as a switch is shown in
Fig. 2.2. The input voltage on the controlling gate appears across a capacitor
1
Mathematically this can be formulated as follows using an admittance matrix  I GS   y11 y12   VGS 


. 
 I DS   y21 y22   VDS 
25 CG
D + VDS ‐ RON
S
S OFF REGION + G VGS ‐ IDS
OUTPUT VOLTAGE CG, the gate capacitor. The output voltage appears across a nonlinear
resistor. In the ON state, it is denoted RON, while in the OFF state it is
denoted ROFF. In theory ROFF is infinite. The input capacitor is a consequence
of the MOSFET being a field-effect device. In Fig. 2.2, the MOSFET field
of operation is divided into the OFF region and the ON region defined by a
certain threshold voltage, VT.
ON REGION RON ROFF CONTROL VOLTAGE
Fig. 2.2. Simplified MOSFET two-port model and its regions of operation.
A closer look at the output characteristic of the MOSFET reveals that RON is
both voltage-controlled and nonlinear. The MOSFET output characteristic is
plotted in Fig. 2.3 for two different gate voltages to illustrate this behavior.
saturation region VDS OFF REGION IDSAT linear region OUTPUT VOLTAGE IDS SATURATION REGION LINEAR REGION
CONTROL VOLTAGE Fig. 2.3. Nonlinear MOSFET behavior and the two regions of operation.
26 The graph reveals that the device behaves as resistor with resistance
RON(VGS) for small drain voltages, but that the current through the device
saturates for large drain voltages. Therefore the ON region of operation can
be divided into two subregions, the linear region and the saturation region.
The border line between these two regions of operation is given by the
straight line VDS=VDSAT=VGS-VT.
Piecewise linear (PWL) models approximating the real device behavior
are often used, particularly for calculations by hand using pencil and paper.
A piecewise linear MOSFET model uses a conductor GON to approximate the
behavior in the linear region, and a constant-current source IDSAT to
approximate the voltage-independent behavior in the saturation region. This
model is illustrated in Fig. 2.4. In the context of a piecewise linear model,
the more accurate nonlinear model can be seen as a “smoothing curve”
between the linear region and the saturation region. While the nonlinear
model saturates for VDS=VDSAT, the simplified piecewise linear model
saturates for VDS=VDSAT/2.
linear region IDSAT VDS OFF REGION saturation region OUTPUT VOLTAGE IDS SATURATION REGION IDS=IDSAT IDS=GON VDS
LINEAR REGION CONTROL VOLTAGE Fig. 2.4. Piece-wise linear MOSFET model and its regions of operation.
In the piecewise linear model, the drain current is given by the following
simplified relationships
G V
I DS   ON DS
 I DSAT
VDS  VDSAT / 2
VDS  VDSAT / 2
(2.1)
27 In a piecewise linear model we use one two-port model to represent the
MOSFET in the linear region, and another two-port model to represent the
MOSFET in the saturation region. These two-port models are shown in Fig.
2.5.
MOSFET saturation region, VDS>VDSAT/2 MOSFET linear region, VDS<VDSAT/2 drain
gate +
vGS
CG
GON
-
drain
gate
+ vDS ‐
source
+ vGS ‐ CG + vDS ‐
IDSAT source Fig. 2.5. MOSFET two-port models for the linear and saturation regions.
Example 3.1. Calculate the ON resistance RON of a fully ON MOSFET if it
saturates at VDSAT=0.8 V and the saturation current is 640 A?
Solution: The ON resistance is given by RON = 0.8/2/640 = 625 .
According to the classical long-channel MOSFET model developed by
William Shockley, the ON conductance and the saturation current is given
by the following equations
 GON  k VGS  VT 


k
2
 I DSAT  VGS  VT 
2

,
(2.2)
where two new parameters k and VT have been introduced. Here, VT is the
threshold voltage, i.e. the minimum gate voltage required to turn the
MOSFET ON. The ON conductance is proportional to the gate overdrive
voltage VGS-VT, often conveniently denoted VGT or VGST, while the saturation
current obeys a square-law dependence.
The parameter k contains both geometry and technology-dependent
parameters according to the following model
28 k
W
W
Cox  k´, where "k prime"=Cox .
L
L
(2.3)
The geometry dependency is given by the MOSFET channel aspect ratio
between the width, W, and the length, L. The aspect ratio can be defined by
the circuit designer, while the technology parameter “k prime” is given by
the manufacturer since it contains technology parameters such as the carrier
mobility,  and the oxide capacitance, Cox, per unit area.
Example 3.2. Calculate the ON conductance GON of a fully ON MOSFET
and the maximum current it can deliver or sink, given the following
parameters: W/L=4, VDD=1,2 V, VT=0,3 V, Cox= 20 fF/m2, =100 cm2/Vs?
Solution: The process parameter ‘k-prime’ is given by k’=Cox=200 A/V2.
The device aspect ratio is W/L=4. The gate voltage overdrive for a fully ON
device is given by VGST=VDD-VT=0.9 V. Hence,
GON  kVGST  4  200  0.9  720  A/V

.
k 2

2
 I DSAT  2 VGST  4  100  0.9  324  A
For VGS=VDD, the PWL breakpoint between the linear and saturation regions
appears at VDS=324/720=0.45 V, which is half the gate voltage overdrive.
The nonlinear saturation voltage is equal to the gate voltage overdrive, i.e.
VDSAT=0.9 V.
The complete long-channel MOSFET model derived by Shockley2 is given
by
I DS
 
VDS 
k  VGST  2 VDS

 
k V 2
 2 GST
VDS  VGST
linear_region
(2.4)
VDS  VGST
saturation_region
2
W. Shockley, A Unipolar ‘Field‐Effect’ Transistor, Proc. IRE, 40, 1365‐1376, (1952) 29 2.1
Second-order effects
For modern sub-micron short channel length MOSFETs there are a number
of second-order effects that makes device behavior deviate from the ideal
long-channel model. The three most important of these short-channel effects
(SCE) are the mobility degradation, the velocity saturation, and the output
conductance, an effect that is due to a combination of drain-induced barrierlowering (DIBL) and channel length modulation (CLM).
2.1.1
The charge carrier mobility in a MOSFET is limited by an effect called
scattering. The carriers are scattered by the silicon lattice, and by the silicon
surface. The vertical electric field between the gate and the channel attracts
the charge carriers to the surface thereby increasing the surface collision
frequency and reducing the mobility. The higher the scattering rate the lower
the mobility; it is like it is more difficult to walk fast on a crowded side-walk
if you bump into the house wall. Chen et al. have found the following
empirical models for the effective mobility of electrons and holes in a
MOSFET channel
eff ,n 
540
1.85
 V  VTN 
1   GS

 0.54  tox 
and eff , p 
185
 V V
1   SG TP
 0.34  tox



, respectively, (2.5)
where tox is the oxide thickness of the MOSFET gate capacitor.
Example 3.3. Calculate the effective mobilities for 65 nm n-channel and pchannel MOSFETs for the two cases of VGS=0.5 and VGS=1.2 V. The threshold
voltages are ±0.3 V. Assume tox=1.7 nm.
Solution:
eff ,n 
eff ,n 
30 540
1.85
=300 cm 2 /Vs and eff , p 
185
=80 cm 2 /Vs
 0.8 
1 

 0.34  1.7 
1.85
=150 cm 2 /Vs and eff , p 
185
= 50 cm2 /Vs
 1.5 
1 

 0.34  1.7 
 0.5  0.3 
1 

 0.54  1.7 
540
 1.2  0.3 
1 

 0.54 1.7 
These examples clearly shows how the effective mobility is reduced to about
half or even less when the gate voltage VGS is increased from the low bias
voltage of 0.2 V used by the analog designer to the full-rail input voltage used
by the digital designer in rail-to-rail logic designs.
2.1.2
Velocity saturation
Like Ohm’s law, the square-law Shockley MOSFET model relies on a linear
relationship between the charge carrier drift velocity and the electrical field
along the channel. The slope of this relationship defines the mobility
discussed in the previous section. However, beyond a certain critical field Ec,
the velocity saturates at a maximum velocity vsat, approximately 107 cm/s.
In a piecewise linear velocity model,
 E
v   eff
 vsat
E  Ec
,
E  Ec
(2.6)
the critical field is given by EC=vsat/.
The MOSFET saturation voltage in the case of velocity saturation can be
derived by the following simplified reasoning. The current in a MOSFET
channel is given by the mobile charge induced in the channel by the gate
overdrive voltage across the MOSFET input capacitor, multiplied by the
charge drift velocity. In the linear region of the piecewise linear MOSFET
model, the drain current is given by the source charge multiplied by the
carrier drift velocity,
V 

I DS  WCoxVGST    eff DS  .
L 

(2.7)
In the case of velocity saturation at the drain end of the channel, the
saturation current is given by the drain charge multiplied by the carrier
saturation velocity,
I DSAT  WCox VGS T  VDSAT   vsat .
(2.8)
In a piecewise linear model, the current in the linear region reaches the
saturation level at the model breakpoint VDS=VDSAT/2. Equating the two
expressions for the drain current at this breakpoint yields the following
saturation voltage
31 VDSAT 
VGST VC
,
VGST  VC
(2.9)
where VC=2LEC=2Lvsat/eff, and the following saturation current
VC
k
k
.
I DS  VGST VDSAT  VGST 2
2
2
VGST  VC
(2.10)
For negligible velocity saturation, i.e. for VB >> VGST, we can see that VDSAT
approaches VGST, but for velocity saturation becoming a dominating
phenomenon, i.e. for VB << VGST, VDSAT approaches VC.
Example 3.4. Find the critical voltage VC for fully ON nMOS and pMOS
transistors using the effective mobilities from example 3.3.
Solution: Using the equation VC=2LEC=2Lvsat/ we obtain for low gate
voltages,
cm
cm
2  50nm  107 2  50nm  107 s =0.33 V and V 
s =1.25 V
VC  n 
C p
cm2
cm2
300
80
Vs
Vs
and for high gate voltages (VGST=0.9 V),
cm
cm
2  50nm  107 2  50nm 107 s =0.65 V and V 
s =2 V
VC  n 
C p
cm2
cm2
150
50
Vs
Vs
These latter values for VGST=0.9 V correspond reasonably well with the
values given by Weste & Harris in their example 2.4.
The following square-law model will act as a smooth replacement for
IDS=kVGSTVDS in the linear region

V
I DS  kVGST VDS 1  DS
 2VDSAT

.

(2.11)
A typical output characteristic using the saturation voltages calculated in
example 3.5 is shown in Fig. 2.6. From the graph it is obvious that the device
is more of a linear device than a square-law device due to a combination of
the velocity saturation and mobility roll-off effects.
32 Fig. 2.6. MOSFET output characteristics.
Example 3.5. Find the nMOS and pMOS transistor saturation voltages for
VGST=0.2 V and VGST=0.9 V using the critical voltages from example 3.4.
V V
Solution: Using the equation VDSAT  GST C , we obtain
VGST  VC
0.2  0.33
0.2  1.25
VGST  0.2 V: VDSAT , n 
 0,125 V, and VDSAT , p 
 0,17 V
0.2  0.33
0.2  1.25
VGST  0.9 V: VDSAT , n 
0.9  0.65
0.9  2
 0,38 V, and VDSAT , p 
 0,62 V
0.9  0.65
0.9  2
We can see that for VGST=0.2 V the saturation voltage VDSAT is quite close to
VGST (at least for the p-channel device, while for VGST=0.9 V it is not.
It is also clear that values extracted for the parameter k at low gate voltages,
for instance by using the method illustrated in Fig. 2.7, cannot be used for
calculating the maximum current driving capability of the MOSFET using
the long-channel square-law model. This is obvious from Fig. 2.8.
Nevertheless, it can be concluded from the figures that the square-law
Shockley model is quite useful for transconductance and voltage
amplification estimates in analog designs where MOSFETs are biased in this
low-bias region.
33 Fig. 2.7. MOSFET transfer characteristics.
Fig. 2.8. MOSFET transfer characteristics.
2.1.3
Discussion and practical approach
So, what is the main conclusion of this model discussion? Where has this
discussion taken us? The analog designer, on the one hand, is often quite
satisfied using the long channel square-law MOSFET model based on model
parameters k and VT extracted at low values of VGS. The digital designer, on
the other hand, is most interested in the maximum drain saturation current
34 ION that the MOSFET can deliver or sink for gate voltages equal to the
supply voltage. In this region of operation, both mobility roll-off and
velocity saturation are dominating effects in today’s short channel devices.
Obviously, it is too complicated to handle both mobility roll-off and
velocity saturation in first-order approximations. To our advantage is the fact
that the mobility*VDSAT product varies quite slowly with VGS. This means that
we can use the same low VGS values for model parameters k and VT as the
analog designer in combination with an ‘effective’ critical voltage VC
determined to yield the correct ION. It is a quite simple approach, the main
disadvantage of which is that it will somewhat exaggerate the saturation
region as shown in Fig. 2.9.
Fig. 2.9. Comparison between MOSFET models.
2.1.4
DIBL and channel-length modulation (CLM)
On top of the problem with these two second-order effects there is also the
non-negligible problem of non-saturating currents in the saturation region.
This short-channel effect is due to two physical phenomena: drain-induced
35 barrier-lowering (DIBL) and channel-length modulation (CLM). For hand
calculations, the digital designer solves this problem by fitting his model to
the average current for delay calculations determined at VDS=3VDD/4. This
approach is illustrated in Fig. 2.10.
For the analog designer, the problem is not that severe for bias
calculations, but for small-signal analysis a non-zero output conductance,
gDS, corresponding to the slope of the IDS vs. VDS curve must be added to the
small-signal MOSFET model, as shown in Fig. 2.11.
Fig. 2.10. Average model fitted to data for non-saturating drain currents.
MOSFET saturation region, VDS>VDSAT drain gate
+
vGS, vgs
CG
IDSAT or gmvgs
‐
gds
+ vDS, vds ‐ source
Fig. 2.11. MOSFET two-port model for non-saturating drain currents.
36 2.1.5
Subthreshold leakage
As can be seen in Fig. 2.7, the device is not fully turned off below the
threshold voltage, but there seems to be a subthreshold current flowing. This
current is quite visible when the transfer characteristic is plotted in a
semilogarithmic graph, as shown in Fig. 2.12. As indicated by this
semilogarithmic graph, where the subthreshold current fits a straight line, the
subthreshold turn-off behavior is exponential.
Fig. 2.12. Semilogarithmic graph of the MOSFET transfer characteristic.
The exponential behavior indicates to the device physicist that the subthreshold current is a diffusion current and not a drift current. The inverse of
the slope of the straight line is called the subthreshold swing, S. Typical
values for the subthreshold swing is around 100 mV per decade.
Another performance parameter that becomes more and more important
as technology and supply voltages scale is the ION/IOFF ratio3. On the one
hand, the designer wants as high effective gate voltage overdrive, VGST=VDDVT, as possible. On the other hand, the threshold voltage should be as high as
possible to ensure a low IOFF. For a typical subthreshold swing of 100
mV/decade, a 100 mV increase of the threshold voltage means a factor of ten
reduction of the OFF leakage current. For a 65 nm device delivering an ON
current of 1 mA, the OFF current is typically 50 nA. These numbers result in
an ION/IOFF ratio of 20,000; a factor of ten less than shown in Fig. 2.12.
3
ION=IDS(VGS=VDS=VDD), IOFF=IDS(VGS=0, VDS=VDD). 37 2.2
MOSFET capacitances
In this section we shall complete our simple MOSFET two-port model by
adding the input and output capacitances.
2.2.1 Intrinsic gate capacitance
Let us start by calculating the intrinsic gate capacitance. This is easily done
knowing the MOSFET channel width W and length L, yielding
CG  WLC ox .
(2.12)
For simplicity, we placed this gate capacitance between the gate and source
in the beginning of the chapter when we introduced the two-port model.
These models are repeated in Fig. 2.13 for our convenience.
MOSFET saturation region, VDS>VDSAT MOSFET
linear region, VDS<VDSAT drain
gate + vGS ‐ CG
GON
source drain
gate
+ vDS ‐
+ vGS ‐ CG IDSAT + vDS ‐
source Fig. 2.13. MOSFET two-port models for the linear and saturation regions.
In reality, the situation is much more complicated since the distributed gateto-channel capacitance should be split between the source and the drain in a
voltage-dependent way. In essence, the gate capacitance should be split
equally between source and drain in the linear region, while the capacitance
from gate to drain can be neglected in the saturation region. On the other
hand, the capacitance to the source is reduced to 2/3CG in the saturation
region due to the charge distribution in the channel, see Fig. 2.14.
2.2.2
Gate overlap and fringing field capacitances
To make modeling even more complicated, we must also consider the
parasitic capacitances due to the gate-to-source/drain overlap and the
38 MOSFET
saturation region, VDS>VDSAT MOSFET linear region, VDS<VDSAT drain gate + vGS ‐ + vDS ‐ CG/2 CG/2 GON drain
gate
+ vGS ‐ + vDS ‐
IDSAT
2CG/3
source source Fig. 2.14. More correct two-port models for the intrinsic gate capacitance.
fringing field along the edges of the gate. These capacitances are illustrated
in Fig. 2.15.
CGSO Metal gate
source STI CGDO W
drain L STI Fig. 2.15. Side view of the gate overlap and fringing field capacitances.
The gate overlap and fringing field parasitic capacitances are given by
CGSO  W  C GSOL , CGDO  W  CGDOL ,
(2.13)
where typically CGSOL=CGDOL= 0.2-0.4 fF/m. These parasitic capacitances
should be added to the intrinsic gate capacitances. However, for hand
calculations the existence of capacitances between the input and the output
introduces unwanted complications. Therefore, in circuits where the output
voltage is an amplified version of the input signal, vout=-Avin, the Miller
theorem can be used to arrive at an equivalent circuit with capacitances only
to ground. This is illustrated in Fig. 2.16.
39 In digital circuit applications, where the equivalent voltage amplification is
equal to one, the gate to drain capacitance CGD results in an equivalent
parasitic capacitance of 3CGDO at the input and an equivalent load of 2CGDO
at the output. Since the intrinsic gate capacitance LCox per unit width as
about 1 fF/m for the 65 nm node, the parasitic capacitance 3CGDOL per unit
width is actually more than the missing third of the gate-to-source
capacitance in the saturation region. Nevertheless, not to complicate matters
more than necessary, let us agree on the simplified two-port model in Fig.
2.13. Of course, circuit simulators like SPICE contain more detailed capacitance models that also take the voltage dependence of the gate capacitance
into account.
MOSFET
parasitic gate‐to‐drain capacitance
drain
gate + vin ‐ drain
gate
+
CGDO vout=‐Avvin
‐
source MOSFET equivalent Miller capacitances + vin ‐ (1+A)CGDO + vout=‐Avin (1+1/A)CGDO
‐
source Fig. 2.16. The parasitic gate-to-drain capacitance and its equivalent Miller
capacitances.
Example 3.6. Calculate the gate capacitance per unit width of a 65 nm
CMOS n-channel MOSFET device with an effective gate length of 50 nm
given the following parameters:
Cox=20fF/m2, CGSOL=CGDOL=0.2fF/m?
Solution: The gate capacitance per unit width is given by CG’=WCox, yields
CG’=1.0 fF/m. This is the gate capacitance per micron transistor width. The
parasitic overlap and fringing field capacitances from the two edges of the
MOSFET gate as seen from the input is equal to COVL=CGSOL+2CGDOL=0.6
fF/m, which is a non-negligible part of the total gate capacitance. Also, the
parasitic input capacitance is about twice the missing third of the gate
capacitance in the saturation region.
40 However, we will not change our MOSFET two-port model because of this
since the capacitance values are rather rough estimates anyway as are the
exact device geometries.
2.2.3
Source and drain parasitic junction capacitances
In addition to the gate, the source and drain are also associated with
capacitances. Like the gate fringing field and overlap capacitances, these
capacitances are regarded as parasitic since they are not fundamental to
MOSFET operation. However, they do impact circuit performance. The
source and drain capacitances arise from the pn junctions between the source
or drain and the body substrate. Hence, they are called CSB and CDB. They are
illustrated in Fig. 2.17 and depend both on the junction bottom plate area and
the junction perimeter. Hence, these capacitances are given by
 C SB  WLS  C j  (W  2 LS )C jsw  WC jswg
.

C DB  WLD  C j  (W  2 LD )C jsw  WC jswg
MOSFET switch model (2.14)
Channel width W D MOSFET symbol CDB
GON
LD
channel length, L G CG
S CSB
LS
Fig. 2.17. MOSFET switch model and layout.
As indicated by the capacitance model, the source and drain junction
capacitances consist of a bottom plate capacitance proportional to the
source/drain area,
WLS  C j
Cbottom _ plate  
WLD  C j
source
,
drain
(2.15)
41 where Cj is the bottom plate junction capacitance per unit area and A=WLX is
the bottom plate area. Also indicated by the capacitance model in (2.14) is
the sidewall capacitance of the source/drain regions,
 W  2 LS   C jsw  WC jswg
C sidewall  
W  2 LD   C jsw  WC jswg
source
,
drain
(2.16)
where Cjsw is the sidewall capacitance due to the shallow trench isolation and
Cjswg is the sidewall capacitance to the channel. In deep submicron processes
below 0.35 m, where shallow trench isolation is used, the sidewall
capacitance along the nonconductive trench tends to be much smaller than
the sidewall capacitance facing the channel. A side view of the MOSFET
and its parasitic junction capacitances is shown in Fig. 2.18.
To further complicate modeling, the junction capacitances are voltage
dependent as the decrease with increasing source/drain reverse bias. For the
source it is not so much of a problem since it is usually grounded. However,
for the drain it is more complicated since the drain voltage changes during
charging and discharging of the MOSFET that occurs as a response to input
gate voltage variations.
LD LS Csidewall source Csidewall Metal gate drain W STI Cbottom Cbottom STI Fig. 2.18. Side view of the source and drain parasitic capacitances.
Example 3.7. Calculate the source/drain capacitance of a 65 nm CMOS nchannel MOSFET device given the following parameters: W=1 m,
LS=LD=0.125 m, Cj=1fF/m2, Cjswg= 0.36fF/m and Cjsw= 0.1 fF/m?
42 Solution: The source/drain bottom plate capacitance is C’bottom=0.125 fF/m
channel width. The sidewall capacitance consists of a nonscalable part
2.LX.Cjsw= 0.025 fF, and a scalable part Cjswg+ Cjsw= 0.46fF/m.
The total junction capacitance is then CSB=CDB=0.025+W.(0.125+0.46) fF,
which for a one micron wide MOSFET accounts to a total capacitance of
0.61 fF. For a 100 nm wide MOSFET the junction capacitances reduce to
0.084 fF, i.e. 84 aF.
The junction capacitances are voltage dependent according to the following
model,
C j (0)


M
 1  VSB / Vbi 
C j 
C j (0)

 1  V / V  M
DB
bi

source
,
(2.16)
drain
where Vbi is the built-in potential of the junction. On top of this, modeling is
even more complicated due to the fact that the capacitance parameters are
different for n-channel and p-channel devices, and they are different for each
of the bottom plate and sidewall capacitances. All this must be carefully
considered when extracting model parameters for Spice simulations.
However, for hand calculations we will use the zero-bias capacitance values.
This is of course a rather conservative approach since we are then basing our
estimates on capacitance values larger than the actual values, for instance
during switching when the drain voltage is switching from one rail to the
other. On the other hand, to some extent this conservative approach makes
up for the output Miller capacitance from the gate overlap and fringing field.
2.3
Why do we need dynamic models?
Digital designers need a dynamic MOSFET model for estimating the
switching speed and propagation delay of the logic gates they are designing.
The simplified piecewise linear current model we have presented in this
chapter is good enough for preliminary and approximate hand calculations. It
provides a rough estimation that usually less than a factor of two from the
real value. In fact, accuracy can be quite good, particularly if the driving
43 capabilities and capacitances of the MOSFETs are calibrated to Spice
simulations of some simple logic gates like two and three input NAND and
NOR gates. For more accurate delay estimations circuit simulations must be
performed using the Spice simulator where detailed MOSFET models are
used (like the BSIM model developed at University of California at Berkeley
with almost one hundred model parameters).
A typical switching situation where we will have use of our dynamic
MOSFET two-port model is shown in Fig. 2.19 where one MOSFET is seen
charging and discharging the gate of another MOSFET. There will be more
about switching of MOSFETs in chapter 4.
VOUT
M1
VIN
M2 M1 two‐port model
drain
gate
+ VIN ‐ M2 two‐port model
gate + CG
IDSAT
CDB
driver
VOUT ‐
CG load Fig. 2.19. Equivalent two-port model for one MOSFET charging/ discharging the gate capacitance of another MOSFET.
44
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