Download Variable Frequency Drive - UCF EECS

Variable Frequency Drive
Merritt Robbins, Justin Barwick, Will Santos,
and Chris Guido
Dept. of Electrical Engineering and Computer
Science, University of Central Florida,
Orlando, Florida, 32816-2450
Abstract — A first of a kind at UCF; the Variable
Frequency Drive designed and implemented by Group F is
a simplified approach to the problem of vector motor
control. Using open source software and original hardware
design, the drive allows precisely variable speed and torque
of a three phase AC induction motor of up to 1/2hp. The
variable frequency drive designed by group F implements
field oriented vector control to achieve the goal of variable
speed and torque operation. The design focuses on
efficiency and safety for the end user and his or her
equipment being driven.
Index Terms — Variable speed drives, Induction motors,
Sensorless control, Machine vector control, Space vector
pulse width modulation.
I. OVERVIEW OF INDUCTION MACHINES
Since Nikola Tesla patented the alternating current
induction motor in 1896 [1], the simple action of a
rotating magnetic field inducing an opposing field has
been the driving force for the majority of heavy industry.
The induction motor provides many distinct advantages
over a DC motor of the same power; the induction motor
is more efficient, more reliable, easier to manufacture,
smaller, less costly to the environment, quieter, and can
be more easily sealed for harsh environments. Due to
these advantages, the induction motor has dominated
industry and individual life for constant speed
applications.
The catch with induction motors is their lack of
variable speed operation without varying both the current
(or voltage) and frequency of the power signal sent to the
motor. Varying the voltage or current of a signal is a
trivial challenge, but varying frequency is not. The
Variable Frequency Drive (VFD) designed by group F
generates a 325V DC bus and then implements flux
oriented vector control to synthesize an output waveform
with variable voltage from 2.3V to 230V and power rated
up to 500W input.
II. VARIABLE FREQUENCY DRIVE APPLICATIONS
Solving the problem of variable speed control for
induction motors is massively useful. Without the
limitation of constant speed, applications where an
inefficient and heavy DC motor was used simply due to
a requirement of variable speed can now implement AC
motors and take advantage of their significant benefits.
The primary target industry of the project is
transportation; an innately variable speed drive system.
Modern electric vehicle manufacturers use high
performance VFDs which cost tens of thousands of
dollars to drive hundreds of Kilowatts to their motors.
Other key applications include machine tools, where the
addition of a VFD to an existing tool allows for a massive
performance improvement for the machine.
Variable speed can be extremely useful on near 100%
duty cycle motors in HVAC systems as well. Rarely does
a building need the full capacity of its air conditioning
system compressors, but the compressors can only be
turned on or off because they are driven by induction
motors. The problem can be partly solved by running
fewer compressors, but controlling the speed of the
compressor blades dynamically (thus avoiding excessive
power cycling) to match the current load can have
massive energy savings for the building owner.
Beyond the current applications where variable speed
is the only justifiable reason to use a VFD, if
implemented on a large scale on large motors (over about
50hp), has the potential to increase grid efficiency by
moving the load power factor to near unity. This can save
money for the owner (if they get billed by the VA) and
can save power for everyone but reducing reactive
loading.
III. MAJOR DESIGN GOALS AND SPECIFICATIONS
The VFD design is focused on four major areas;
operator and system safety, drive dynamic performance,
simplicity of design, and efficiency.
A. Operator and System Safety
Safety is the number one priority of Group F for the
VFD. Cartridge fuses are placed in every power path in
the system, furthermore the motor has a fuse for each
phase. The system is based around a differential
positive and negative 162.5V DC link. The choice of a
differential rail was made to reduce the high voltage
potentials found on the board by a factor of two relative
to ground. All microcontrollers are chassis grounded.
The chassis and high voltage power board are both earth
grounded. The DC link is not ground referred directly, it
is indirectly ground referenced by earth grounding the
center tap of the power transformer secondary.
An array of nine NTC thermistors is implemented to
monitor real time temperatures of all power switching
and drive devices in the high voltage power system as
well as the housing of the motor under test, In the case
of over – temperature, the MSP430 that oversees the
system will prevent catastrophic component failure by
disconnecting the power to the power transformer.
Further contributing to overall system safety is the
large derating of all ceramic capacitors, integrated
circuits, and power switches. All ceramic capacitors are
voltage derated by a factor of at least 1.75. All drivers
have carefully chosen current limiting resistors and are
massively over – specified for drive current levels based
on our switching frequency of 20kHz.
B. Drive Dynamic Performance
Due to the main target application for the VFD being
the transportation sector, having an extremely wide
dynamic range of control is essential. The VFD
designed by group F achieves 1% to 110% operating
speed range, ramping motor start up, 0.1% accurate
speed and torque control, as well as the ability to shift
the speed of the motor at a real time pace.
C. Simplicity of Design
Simplicity of design results in greater reliability, as
well as easier troubleshooting and repair. To keep the
design simple, group F elected to limit all resistor
packages to 0603 package size minimum, disallow any
integrated circuit packages without exposed leads, and
use highly common packages for all permitting devices.
One important step taken during prototyping phase was
to leave all board vias un-tented thus allowing many
convenient oscilloscope probe locations built into the
signal path.
C. Efficiency
Efficiency is a major design goal of nearly all modern
power systems. Group F’s VFD is no exception. The
system achieves overall power efficiency of greater than
85% under full load. This efficiency is achieved by the
selection of a highly efficiency rectifier design as well as
a space vector modulation algorithm which minimizes
switching cycles, thus minimizing switching losses. The
switches themselves are selected for optimal conduction
characteristics while maintaining reasonably low gate
capacitance to reduce switching losses. The drive
integrated circuits are selected specifically to optimize
switching speed and minimize conduction losses due to
long rise and fall times.
III. POWER SYSTEM DESIGN
The VFD power system will be broken into two
sections for discussion; high voltage and low voltage.
The high voltage section is the power path which
supplies the motor itself. The low voltage section
provides regulated DC busses to supply the various
integrated circuits, sensors, and microcontrollers in
contained in the system.
A. High Voltage Power System
The high voltage power system incorporates three
main power blocks; The synchronous rectifier, the DC
link, and the power inverter. The system takes mains
power at 115VAC and steps up the voltage to 230VAC
with a power transformer. The center tap of the power
transformer secondary is grounded to chassis ground.
The 230VAC is fed into a high efficiency MOSFET
synchronous rectifier. The MOSFETs are configured
such that their body diodes act as rectifier diodes when
no gate drive is applied. A smart gate drive is used to
switch the MOSFET on when the body diode starts
conducting and switch it off just before the body diode
stops conducting. This method results in a close
approximation of the ideal diode rectifier, resulting in
only a few hundred millivolts drop across the rectifier,
and very low losses in the switching devices.
The output of the synchronous rectifier is a nominal
325VDC, but because the center tap of the secondary on
the power transformer was grounded, the output appears
as positive and negative 162.5V relative to ground. The
choice to use dual polarity rail was made for safety. This
configuration allows the VFD to maintain half of the
potentials to ground in its power system versus a single
polarity, thus lowering the chance of electrocution or
equipment damage. The choice has no impact on the
drive system as the motor only responds to the
differential voltage on the windings, they are not
grounded.
The output of the synchronous rectifier is fed into a
capacitor bank which decouples the inductance of the
power transformer from the inverter, providing the
inverter with a large amount of reactive power, and
smoothing switching noise on the DC link to maintain a
predictable output signal.
The DC link outputs power to the power inverter
block. The power inverter consists of three identical
phases which all consist of a pair of IGBTs driven by a
gate driver PMIC and associated passive components.
The gate drivers have built in isolation circuitry to allow
one drive IC to manage one high-low pair of IGBT
switches.
The gate drivers receive their signals relative to the DC
link negative rail. This requires that the drive signal be
level shifted down to be a known PWM with respect to
the DC link negative rail. To accomplish this there are
six opt isolating bridges which level shift the signal from
earth ground reference down to DC link negative
reference while isolating the microcontroller from any
high voltages.
B. Low Voltage Power System
The Low Voltage Power System provides proper
supply and digital I/O voltages for the LCD Display,
rotary encoder, and MCU interfaces. Mains 115VAC
power is fed to another dual output toroidal transformer
that provides two 15VAC isolated outputs: each as inputs
to a Schottky Bridge Rectifier block.
The Bridge Rectifier blocks have identical designs
and follow the conventional diode bridge rectifier
topology.
The requirement for two bridge rectifier blocks comes
from a ground reference consideration for certain
integrated circuits in the system. The gate driver PMICs
and opt isolating bridges require that their signals be
referenced to the DC link negative reference, therefore
the switch mode power supply for the ICs in these
systems must also be referenced to the negative terminal
of the DC Link. The Buck Regulators each receive their
input voltages from the Schottky bridge rectifiers, so any
power delivered to High Voltage Power System ICs must
also be referenced to the negative terminal of the DC
Link.
On the low voltage power/control board all MCU
interfaces and any LDO voltage regulators providing
analog/digital supply voltages are referenced to chassis
ground, so the bridge rectifier providing the input rail to
the buck regulator engaged in the switch mode power
supply for these blocks is also referenced to chassis
ground.
IV. SENSOR DATA ACQUISITION AND PROCESSING
The term ‘sensorless’ in the context of a motor control
system is somewhat of a misnomer. A sensorless motor
controller is one which does not require any internal
measurements of direct motor parameters such as flux
density and rotor current. The critical measurements
which the VFD uses to model the behavior of the motor
are simply voltage and current delivered to the stator. We
provide an optional third feedback to our controller in the
form of a three channel rotary encoder which gives the
controller a value for actual rotor speed, which can be
estimated via mathematical modeling, but removes a
significant computational load by measuring it directly.
Further sensors include DC link voltage monitoring and
an array of NTC thermistors which monitor real time
system temperatures to protect from over – heating.
A. Voltage Measurement Acquisition
Probing of high voltages at the motor phase
connections, as well as the DC link voltage, is
accomplished with a resistor divider. The divider has one
special feature, however, in that it includes a constant DC
offset applied to the measurement. This is done so that
none of the active filters to follow require negative power
rails, and guarantees that the measurement voltage fed to
the ADC will never be negative. This offset is
accomplished by using a precision shunt voltage
reference effectively raising the ground of the voltage
divider up by a constant value. This value is fed to the
ADC as well to reduce cost by allowing for lower
precision shunt references since the actual offset can be
measured and can therefore accept a higher tolerance
shunt.
B. Current Measurement Acquisition
Measurement of phase current is accomplished via a
current transformer which produces an output which has
a highly linear relationship to the current flowing through
the measured wire. The current transformers were
specifically selected for appropriate bandwidth for the
VFD application at hand as well as a high turns ratio to
minimize power losses.
The secondary of the current transformer is passed
through a precision resistive shunt which is set at a
constant offset to ground potential in the same manner as
the voltage measurements described above.
C. Current And Voltage Signal Processing
Both current and voltage sensor signals are filtered
through a second order Butterworth filter with its cutoff
frequency set at approximately 200Hz. This gives a
nominal -80dB attenuation to our 20kHz switching
frequency and attenuates a 60Hz signal by less than
0.1%. The output of these filters will be the time average
signals sent to the motor. This discrete analog filtering
removes the need for filtering of the signals in software,
reducing complexity of the code as well as computational
load.
D. Temperature Measurement Acquisition.
The method of thermal measurement is achieved by
using negative temperature coefficient resistor networks
to provide a suitable voltage range for processor use. The
effective temperature range for measurement is 20 to 100
degrees Celsius and, from a reference of 1.2V at 20
degrees Celsius, a 10 degree rise in temperature
corresponds to a .1V increase in the output voltage of the
thermistor network. The network receives power from a
3V regulated DC rail. A set of fixed resistors in series
with the thermistor serve to limit current applied to the
thermistor and set the top resistor value for a potential
divider that sets the effective output voltage range of the
network. The thermistors are thermally coupled to each
gate drive IC, the motor housing, and the 15V regulator
IC on the high voltage side of the power system.
voltmeter and ammeter connections made to achieve the
DC resistance. The Reliance Electric P56H5069G stator
windings are configured in a 1Y topology in order to use
the low voltage setting to drive the motor.
Figure 1 -Wiring Diagram for determination of 𝑅𝐷𝐶 in
the following equations
E. Rotary Encoder Data Acquisition
The rotary encoder chosen utilizes a novel technology
reliant on magnetics for motion control and sensing,
thereby eliminating any mechanical contact and
consequently, wear and tear. Operating at a single supply
voltage of 5V taken from the low voltage switch mode
power supply, it has a three channel output: two that
indicate direction of rotation. When channel A of the
encoder leads channel B by 90 electrical degrees, the
shaft of the motor is spinning in the clockwise direction,
the opposite holding true for counterclockwise rotation.
The third channel provides a pulse to indicate every full
rotation of the shaft: thereby providing a suitable counter
for use in software to determine the rotation speed of the
shaft. The encoder is mounted such that the base will be
attached to a quarter-inch shaft with a piece of 10-32
threaded bolt coupled to the motor shaft via matching
threads. This connection is purely mechanical.
F. Determination of Induction Motor Parameters
A set of electrical tests determine the parameters used
to characterize the motor and develop torque-speed
characteristic curves. Among important considerations in
the steady-state analysis of a three-phase induction motor
are the variation of current, power, losses and torque
associated with various operating conditions. Through
completion of these tests, the equivalent circuit of the
stator per phase can be modeled for development of
torque-speed characteristics with multiple slip
parameters to be generated. A set of simple DC tests
determine the stator resistance of the motor. First,
connection of any two stator leads to a variable voltage
DC power supply, then adjustment of the power supply
to provide the rated stator current for the motor. The
series DC resistance is determined as the ratio of voltage
to current from voltmeter and ammeter readings. In the
case of this system, the motor will be utilizing a wyeconnected stator, so then the wye-connected stator
resistance is determined as half the series DC resistance.
The following figures illustrate this idea in addition to the
𝑅𝐷𝐶 = 𝑉𝐷𝐶 ⁄𝐼𝐷𝐶 (1)
Figure 2-The wye connection configuration and
illustration of stator resistances per phase
𝑹𝑫𝑪 = 𝟐𝑹𝟏,𝒘𝒚𝒆
(2)
Consequently the stator resistance per phase is
determined as
𝑅1,𝑤𝑦𝑒 = 𝑅𝐷𝐶 ⁄2
(3)
i) The blocked rotor test
The blocked rotor test determines each of the rotor and
stator reactances referred to the line and load sides of
the motor equivalent circuit. In addition, it determines
the rotor resistance when combined with data from the
DC test. The rotor is blocked so that it will not turn and
a variable AC supply is connected to each phase of the
motor to provide rated current. The following figures
provides the wiring diagram for the blocked rotor and
no load tests, and an illustration of the blocked rotor
equivalent circuit [1].
Figure 3- Wiring diagram for no load and blocked rotor
tests, V and A represent voltmeters and ammeters
respectively [1]
The blocked rotor impedance is then corrected to rated
frequency of the motor via the scalar factor of 60/15
coupled to the blocked rotor reactance value calculated
[1]. The blocked rotor reactance at rated frequency
relates X1 and X2 as follows
𝑋1 + 𝑋2 = 𝑋𝐵𝑅 (9)
Figure 4 - Simplified motor equivalent circuit for the
blocked rotor test. R1 and R2 are the stator and rotor
resistances of while X1 and X2 represent the stator and
rotor reactances, respectively. [1]
Per IEEE test standard 112-984 the blocked rotor test is
performed using a quarter of the rated frequency with
the test voltage adjusted The magnetizing inductance
for the stator and to obtain approximately rated current.
The 60 Hz Reliance Electric motor would use a 15 Hz
test voltage. However, because the motor only provides
a 0.5hp, rated frequency suits the needs of the test. The
following presents the determination of R1 and R2. The
blocked rotor resistance is calculated as [1]:
𝑅𝐵𝑅 =
𝑃𝐵𝑅𝑧
2
𝐼𝐵𝑅𝑧
(4)
The numerator is the true power applied to the motor
and can be obtained from the product of voltmeter and
ammeter readings, and the blocked rotor current is
determined as the ammeter reading [1]. Then it is
known that R1 and R2 in series create the blocked rotor
resistance
The determination of X1 and X2 is achieved by
dividing the blocked rotor reactance evenly among them
per the NEMA design class of the motor.
ii) The no load test
The no load test allows for the determination of the
magnetizing impedance, inductance and core, friction,
and winding losses. The rotor is allowed to rotate freely
and run unloaded at rated voltage and frequency,
230V@60Hz. In the no-load case, the speed of the rotor
is very close to synchronous speed, equivalent to the
speed of the magnetic fields, and hence the slip is very
close to zero, causing the current in R2/s to be very
small, and may be ignored in calculations [1]. The
equivalent circuit becomes very simple as the core
resistance and rotor resistance may be ignored.
𝑅1 + 𝑅2 = 𝑅𝐵𝑅 (5)
R2 can now be determined as the difference between
the blocked rotor resistance and R1 obtained from the
DC test data [1]. The blocked rotor impedance is
calculated as
𝑍𝐵𝑅 =
𝑉𝐵𝑅𝑧
𝐼𝐵𝑅𝑧
(6)
The blocked rotor impedance relates the blocked rotor
reactance and resistance as follows
2
2
𝑍𝐵𝑅 = √𝑅𝐵𝑅
+ 𝑋𝐵𝑅
(7)
The blocked rotor reactance can then be expressed as
2
2
𝑋𝐵𝑅 = √𝑍𝐵𝑅
− 𝑅𝐵𝑅
(8)
Figure 5 - Simplified motor equivalent circuit for the no
load test. [1]
The no-load true power is determined as the product of
no-load voltmeter and ammeter readings. The complex
power is the product of the no-load voltage and current
phasors. Its magnitude is related to the magnitudes of
real and reactive power supplied to the motor as
2
2
𝑆𝑁𝐿 = √𝑃𝑁𝐿
+ 𝑄𝑁𝐿
(10)
Therefore the magnitude of no load reactive power
becomes
2
2
𝑄𝑁𝐿 = √𝑆𝑁𝐿
− 𝑃𝑁𝐿
(11)
The magnitude of no load reactive power is then related
to the no load reactance and current as
2
𝑄𝑁𝐿 = 𝐼𝑁𝐿
𝑋𝑁𝐿 (12)
Dividing the magnitude of the no load reactive power
by the no load current yields the no load reactance and
may be related to X1 obtained from the blocked rotor
test and the magnetizing reactance XM. The
magnetizing reactance is then calculated as the
following [1].
𝑋𝑀 = 𝑋𝑁𝐿 − 𝑋1 (13)
rotor is equal and is calculated as [2]
𝐿𝑀 =
𝑋𝑀
2𝜋𝑓
(14)
The combined friction, winding and core loss can be
expressed as a function of no-load data
2
𝑃𝑁𝐿 − 𝐼𝑁𝐿
𝑅1,𝑤𝑦𝑒 = 𝑃𝑐𝑜𝑟𝑒 + 𝑃𝑓,𝑤 (15)
V. VECTOR CONTROL ALGORITHM
A. Clark and Park Transformation
The Clarke transform projects the three abc-phases onto
a new set of axes where one of the axes is the zerosequence component and the other two axes are the
alpha and beta components of the new three phase
system. The zero-sequence component is equidistant
from the three abc-phase axes resulting in a phase that
produces no effect onto the rotor. The alpha axis is
aligned with the a-phase axis and the beta axis is
aligned so that it is made up of only the b-phase and cphase components. This allows for the manipulation of
two sinusoidal signals (alpha and beta) to control the
three abc-phase signals sent to the stator The equations
governing the Clarke transformation can be seen below.
1
𝛼
2
[𝛽 ] = √ [ 0
3
0
1/√2
−1/2
√3/2
1/√2
−1/2
𝑎
−√3/2] ∗ [𝑏 ]
𝑐
1/√2
signals. The zero component of the alpha-beta-zero
phase system is equal to the zero component of the d-qzero phase system. The frequency and reference signal
from the encoder allows for the Park transform to take
place. The equations can be seen below where the dcomponent controls the flux and the q-component
controls the torque.
[d q 0 ]=[cos (θ) sin (θ) 0 -sin (θ) cos (θ) 0 0 0 1 ]*[α β
0]
(17)
This sampled signal is compared to the d- and qcomponents of a reference signal and the difference is
sent through a proportional-integral (PI) controller. The
two d-component values of the sampled and reference
signals should have the same value but the q-component
values will differ. This is due to the constant flux motor
operating condition. Constant flux is accomplished
using a volts-over-hertz (V/Hz) look-up table defined
by the parameters of the motor. To maintain constant
flux, the voltage of the stator signals must vary with
respect to the frequency of signals sent to the stator (as
defined by Faraday’s Law). As the frequency of the
signal changes, the q-component of the reference signal
will also change maintaining constant flux, also
manipulating the output torque.
The output signal from the PI controller passes through
an inverse Park transform block converting the d and q
components back into alpha and beta components.
These waveforms then produce the new PWM signals
using space vector modulation.
B. Space Vector Pulse Width Modulation
Space vector modulation projects the alpha and beta
signals onto each of the inverter state vectors. These
projections determine the duty cycle for each inverter
state.
(16)
After the Clarke transformation, the alpha-beta-zero
phases are passed through the Park transformation. This
transformation rotates the three-dimensional plane of
the alpha-beta-zero phases about the zero-sequence axis
at the same frequency of the sampled signal, projecting
the two alpha and beta phases onto the new set of
rotating axes resulting in two dc quantities (the d- and
q-components) that now control the three abc-phase
For each of the sectors between the inverter states the
calculations for the duty cycles of each of the abc-phase
signals are shown below. The first set of duty cycle
equations are for the a-phase duty cycles per sector.
1
2
𝑡𝑎
𝑡𝑚
[1 +
1
=
2
3𝑓𝑐
2𝑣𝑑𝑐
[1 +
1
𝑣𝑏𝑒𝑡𝑎
(−𝑣𝑎𝑙𝑝ℎ𝑎 −
3𝑓𝑐
2𝑣𝑑𝑐
(−2
√3
𝑣𝑎𝑙𝑝ℎ𝑎
√3
3𝑓𝑐
)]; 𝑠 = 1,4
)]; 𝑠 = 2,5
(18)
𝑣𝑏𝑒𝑡𝑎
{2 [1 + 2𝑣𝑑𝑐 (−𝑣𝑎𝑙𝑝ℎ𝑎 + √3 )]; 𝑠 = 3,6
The second set of calculations are for the b-phase duty
cycles per sector.
1
2
𝑡𝑏
𝑡𝑚
[1 +
1
=
2
3𝑓𝑐
2𝑣𝑑𝑐
(𝑣𝑎𝑙𝑝ℎ𝑎 − √3𝑣𝑏𝑒𝑡𝑎 )]; 𝑠 = 1,4
3𝑓𝑐
[1 +
1
2𝑣𝑑𝑐
(−2
𝑣𝑏𝑒𝑡𝑎
√3
3𝑓𝑐
)]; 𝑠 = 2,5
(19)
𝑣𝑏𝑒𝑡𝑎
{ 2 [1 + 2𝑣𝑑𝑐 (𝑣𝑎𝑙𝑝ℎ𝑎 − √3 )]; 𝑠 = 3,6
The third set of calculations are for the c-phase duty
cycles per sector.
1
2
𝑡𝑐
𝑡𝑚
[1 +
1
=
2
1
3𝑓𝑐
2𝑣𝑑𝑐
[1 +
(𝑣𝑎𝑙𝑝ℎ𝑎 +
𝑣𝑏𝑒𝑡𝑎
3𝑓𝑐
)]; 𝑠 = 2,5
2𝑣𝑑𝑐
(2
𝑣𝑏𝑒𝑡𝑎
√3
√3
)]; 𝑠 = 1,4
(20)
3𝑓
𝑐
{2 [1 + 2𝑣𝑑𝑐 (𝑣𝑎𝑙𝑝ℎ𝑎 + √3𝑣𝑏𝑒𝑡𝑎 )]; 𝑠 = 3,6
The variable fc is a compensation factor for when the
amplitude of the signal approaches the maximum
voltage. The variable tm is the period of each pulse. The
duty cycles are then used to generate the PWM signals
control the IGBTs.
VI. EMBEDDED PROCESSING
A. Microcontroller Selection
In order to implement the control algorithm, we will
need to have a microcontroller with high speed and
precision when it comes to heavy computations. Based
on these requirements, we have selected the
TMS320f28027f microcontroller to drive the inverters. It
has multiple 12-bit ADC channels for high precision in
signal processing, a 60MHz clock for incredibly fast
performance, and four enhanced PWM modules for
generating the output signal to the inverter gates.
We also wish to have an LCD display that gives the
user information on the operations of the motor, such as
speed the motor is outputting or the torque being
produced. In order to drive this display, we have selected
the MSP430f5529 microcontroller. It features 128KB of
flash memory for storing a large amount of information,
a 25MHz clock for the performance necessary to drive
the display, and a large number of general purpose inputoutput pins for sending a large amounts of data to the
LCD. While this is not as powerful as the
TMS320f28027F, it is more than sufficient for the task at
hand.
B. Control Algorithm Implementation
In order to implement the Space Vector PWM control
algorithm, the TMS320f28027f will first receive various
feedback signals as inputs. This feedback will provide us
with information about the current state of the motor. We
will also receive an input signal that will tell us at what
frequency we want the motor to run at. Based off of that
desired input, we can generate a reference signal to
model what we want our output to be. All of these values
will be generated through the various ADC pins on the
microcontroller.
Once we have all of our inputs collected, we generate
the reference signal that we will compare our feedback
with. We run these signals through the alpha-beta
transformation to translate these signal values to their
alpha-beta equivalent coordinates. We can then run these
coordinates
through
the
direct-quadrature-zero
transformation for the equivalent values in the rotating
reference frame. These values are run through a PI
controller in order to properly match up the feedback
signal to the reference signal, and then translated back
from the direct-quadrature-zero system to the alpha-beta
system using the inverse Park transformation.
The next step is to map the alpha and beta values that
we have onto the inverter-state map. This will determine
the inverter-state signals that will be sent out, as well as
the duration of the signals. Once we know the angle of
the vector, as well as the magnitude, we will be able to
calculate the inverter states and switching frequencies.
The PWM is generated and sent as an output.
The MSP430f5529 microcontroller will simply be
receiving various analog signals as inputs and send those
values to the LCD display. This will be done by reading
in analog inputs and translating them into the appropriate
values and units we wish to have on display. Those
values will then be translated through a lookup table
created so that each character will be able to be displayed
on the 128x64 resolution display.
The microcontrollers that we are using for this system
will also be implementing SPI communication to send
information to each other. Specifically, the
TMS320f28027f will be gathering ADC inputs for the
MSP430f5529, and will need to send that data to it so that
it can be used on the LCD display. The MSP430f5529
will act as the Master, generating clock signals indicating
the TMS320f28027f, acting as the Slave, to send the
appropriate information.
VII. CONCLUSION
The variable frequency drive developed by Group F
serves to provide an exploratory opportunity to develop
a motor control algorithm to suit general purpose needs.
With a battery management system and regenerative
braking system the VFD can be incorporated into an
inverter driven AC motor in a drive train to suit electric
vehicle needs. The system incorporates robust power
system, sensor interface, and signal processing blocks to
achieve the proper signal conditioning required for the
MCUs performing the many computations required to
achieve vector control. With appropriate closed loop
control, torque and speed is controlled effectively.
ACKNOWLEDGEMENT
The authors wish to acknowledge the assistance and
support of Douglas Maukonen and Bobby Wong.
REFERENCES
[1] “Determination of Induction-Motor Parameters”.
University of Massachusetts Dartmouth. ECE 441.
www.faculty.umassd.edu/xtras/catls/resources/binarydoc/
3581.ppt
[2] Sinisa Jurkovic.
“Induction Motor Parameters
Extraction.” Massachusetts Institute of Technology.
http://web.mit.edu/kirtley/binlustuff/literature/electric%2
0machine/motor-parameters.pdf
[3] R. H. Park “Two-Reaction Theory of Synchronous
Machines” NAPS, University of Waterloo, Canada, pp.
81-95, October 2000