Download A Fresh Approach to Switching Regulator Topologies and

A Fresh Approach to Switching
Regulator Topologies and
Implementations
Global Power Seminar 2006
1
Agenda
• Switching Topologies Overview
• Off-Line Regulators
• FPS
• FPS Quasi-resonant Mode
• FPS Buck & SEPIC Modes
• LED Lighting
• Off-line
• DC-DC
2
1
Switching Topologies Overview
3
Switched Conversion Techniques Advantages and Disadvantages
Charge Pumps
Inductor Based
• Simple & moderate cost
• Can be step up/down or invert
• Require only resistors and
capacitors
• Noise performance is
architecture dependant
• Can be step up/down or
invert
• Can handle a wide range of
input and output voltages
• Generally higher cost
• Can obtain high current levels
• Issues include:
• Issues include:
• Limited choices for Vout vs. Vin
• Can be noisy
• Limitations on output power
• Need for magnetics
• Board layout can be critical.
• Require more components
than charge pumps
4
2
Simple Single Output, Single Inductor
Topologies
Buck
M(D) = D
Boost
Buck
Buck Boost
1
4
0
0.8
3
-1
M(D) 2
M(D) -2
1
-3
0
0.6
M(D)
Buck Boost
M(D) = -D/(1-D)
Boost
M(D) = 1/(1-D)
0.2
0.4
0.6
0.8
1
0.4
0.2
0
0
0
0.2
0.4
0.6
0.8
-4
0
1
0.2
0.4
0.6
0.8
1
D
D
D
M(D) vs. D relationship assumes Continuous Conduction Mode
D = Duty Cycle on horizontal axis, M(D) = voltage conversion ratio, on vertical axis
5
Simple Single Output, Dual
Inductor Topologies
SEPIC
M(D) = D/(1-D)
CUK
M(D) = -1/(1-D)
CUK
SEPIC
0
0
0.2
0.4
0.6
-1
0.8
1
4
3
M(D) 2
M(D) -2
1
-3
0
0
-4
D
0.2
0.4
0.6
0.8
1
D
M(D) vs. D relationship assumes Continuous Conduction Mode
D = Duty Cycle on horizontal axis, M(D) = voltage conversion ratio, on vertical axis
6
3
Simple Transformer-Based Topologies
Forward
M(D) = D / N
In the forward, the transformer is a
true transformer and (hopefully)
stores no energy
Flyback
M(D) = 1 / (N *(1-D))
In the flyback, the transformer
is really coupled inductors
and stores energy
M(D) vs. D relationship assumes Continuous Conduction Mode
D = Duty Cycle, M(D) = voltage conversion ratio, N is the transformer turns ratio, always >1
7
Conduction Modes and Control Modes
• Conduction Modes
• Discontinuous – current in the inductor remains at zero at some time in the
cycle
• Continuous – current in the inductor NEVER remains at zero at some time
in the cycle
• Systems can operate in both continuous or discontinuous modes depending
on load and input voltage
• Control Modes
• Take the output voltage, compare with a reference & feed the difference
back to control the duty cycle
• Voltage Mode – error voltage directly controls the duty cycle
• Current Mode – error voltage controls the current through the switch
8
4
Flyback Converter – Discontinuous
Conduction Mode (DCM)
n
IR
Switch
Control
ON
VO
C1
Vin
ISW
Vs
T1
Vin+nVo
VSW
RL
OFF
ON
Vin
ILm
ISW
SW1
IR
VSW
9
Flyback topology - circuit and waveforms,
continuous conduction mode (CCM)
D1
n
IR
Switch
ON
OFF
ON
VO
C1
Vin
ISW
SW1
RL
Vs
T1
VSW
Primary Inductance
value is larger than a
comparable DCM
system.
VSW
Vin+nVo
ILm
ISW
IR
10
5
Discontinuous vs. Continuous Mode
• Continuous – lower peak currents
Discontinuous
Inductor Current
2
Continuous
9 Lower conduction losses
9 Smaller input filter required to
reduce EMI
8 Bigger transformer inductance
1
8 Higher cost fast recovery diodes
required on the secondary side
0
40W, UK voltage
SMPS, DCM
40W, UK voltage
SMPS, CCM
• Lp = 350uH
• Lp= 1mH
• Ipk = 1.7A
• Ipk= 1.1A,
Imin=0.6A
• Switch conduction
loss = 1.3W
• Switch conduction
loss = 1.1W
11
• Discontinuous – higher peak
currents
8 Higher I2R losses, Bigger
core/Skin Effect losses
8 Bigger Input Filter to reduce EMI
9 Smaller transformer inductance
9 Lower cost secondary side
rectifier diodes
Control Modes: Voltage or Current
Mode
Voltage mode
• The PWM clock turns the switch on. An
internally generated ramp is compared to
Vfb to determine when to turn the switch off
• In voltage mode, the output voltage is
compared to a reference and controls the
PWM duty cycle directly
• If the output is too low, the PWM duty cycle
is increased, which, in turn, causes the
output voltage to increase
• Hence, the feedback voltage directly
controls the duty cycle
PWM
Vfb
Ref &
Feedback
Load
Vfb
Isw
D
12
6
Control Modes: Voltage or Current
Mode
Current Mode
• Similarly to voltage mode, the output voltage is
compared to a reference and the difference fed
back as Vfb.
• In current mode, the clock turns the switch on.
• When the switch current crosses a threshold, the
switch is turned off. The threshold is set by VFB.
• In this way, the maximum current in the switch is
limited for each PWM pulse: “pulse-by-pulse current
limit”
• Due to the interaction of the voltage and current
feedback loops, the operating threshold is variable:
• If the output is too low, the current threshold is
increased, which increases duty cycle
• Since di/dt = V/L, the system will automatically
compensate for changes in V with no change required
in VFB
PWM
Ref &
Feedback
VISW
VFB
Load
VFB
Isw
Ipk
13
Voltage Mode vs. Current Mode
• Current mode has better line regulation.
• The control loop does not have to respond to a change in line voltage. The drain current
slope is defined by Vin/L. Hence if Vin increases, di/dt also increases and the duty cycle
is automatically changed with no change in Vfb.
• Voltage mode systems often have greatly differing dynamic loop parameters between CCM
and DCM resulting in different stability and different transient response.
• A CCM systems will enter DCM at low loads and/or high input voltages
• Current mode automatically limits the current in the primary winding
• This reduces the maximum current specifications for the transformer, the line filter and
the rectifier, saving costs
• Easy to implement many protection functions: over-load, secondary or transformer
shorts etc.
• Voltage Mode allows duty cycles greater than 0.5
• Current mode requires slope compensation to do this
• Voltage mode has improved load regulation
• The current loop can appear to initially to go the wrong way
• Voltage mode does not require current sensing
• No power loss or expense due to current sensing (assumes that lossless sensing is not
used).
• Voltage mode requires a higher order of loop compensation
14
7
Fairchild Power Switch
15
Fairchild Power Switch (FPSTM) – What
is it?
Snubber
Circuit
+
MOSFET
PWM IC
Fairchild Power Switch
Control IC
WIRE BONDING
HIGH VOLTAGE
INSULATOR
+
SENSE
FET
External Devices
PWM
IC
LEAD FRAME
Additional Functions
16
8
Green FPS™ Portfolio
Green FPS™ Part
Number
Output Power (W)
85265Vac
Burst
230Vac ± Mode
15%
7
7
8
11
11
13.6
18.4
18.4
18.4
18.4
28
28
28
28
46
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Function
Static
Switching Frequency
DrainOperating
Frequency Modulatio
Soft Start
Mode
Source on
n
(kHz)
Resistanc
Peak
Current
Limit (A)
Drain
Voltage
MAX (V)
0.3
0.3
0.55
0.7
0.7
1.2
1.5
1.5
1.5
1.5
2.2
2.2
2.2
2.2
1.8
700
700
650
650
650
650
700
650
650
650
650
650
650
650
650
FSD210Bx
FSD210
FSDM311x
FSDH321x
FSDL321x
FSDL0165Rx
FSDM0270RNB
FSDM0265Rx
FSDM0265RNB
FSDH0265RN
FSDM0365Rx
FSDL0365Rx
FSDM0365RNB
FSDL0365RNB
FSDM0465RB
4.8
4.8
5.6
8
8
12
16
16
16
16
24
24
24
24
38
32
32
19
19
19
10
7.3
6
6
6
4.5
4.5
4.5
4.5
2.6
FSDM0565RB
48
56
Yes
2.3
650
2.2
FSCM0565R
FSCQ0565RT
FSCQ0765RT
FSCM0765R
FSDM07652RB
FSES0765RG
FSCQ0965RT
FSCQ1265RT
FSDM1265RB
FSCQ1465RT
FSCQ1565RT
56
56
85
68
60
70
110
140
90
160
170
68
68
100
76
68
90
130
170
110
190
210
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
2.5
3.5
5
3
2.5
4
6
7
3.4
8
8
650
650
650
650
650
650
650
650
650
650
650
2.2
2.2
1.6
1.6
1.6
1.6
1.2
0.9
0.9
0.8
0.65
134
134
67
100
50
50
67
67
67
100
67
50
67
50
67
Protection
OLP
TSD
OCP
OVP
Yes
Yes
No
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
No
PWM
PWM
PWM
PWM
PWM
PWM
PWM
PWM
PWM
PWM
PWM
PWM
PWM
PWM
PWM
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
No
No
No
Restart
Restart
Restart
No
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
No
No
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
67
No
PWM
Yes
Restart
Restart
Restart
Restart
67
Variable
Variable
67
67
Variable
Variable
Variable
67
Variable
Variable
Yes
NA
NA
Yes
No
NA
NA
NA
No
NA
NA
PWM
QRC
QRC
PWM
PWM
Sync
QRC
QRC
PWM
QRC
QRC
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Latch
Latch
Latch
Latch
Restart
Restart
Latch
Latch
Restart
Latch
Latch
Restart
Latch
Latch
Restart
Restart
Restart
Latch
Latch
Restart
Latch
Latch
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Restart
Package
DIP, SOP
DIP
DIP, SOP
DIP, SOP
DIP, SOP
DIP, SOP
DIP
DIP, SOP
DIP
DIP, SOP
DIP, SOP
DIP, SOP
DIP
DIP
TO-220F-6
TO-220F-6,
I2PAK-6
TO-220-5
TO-220F-5
TO-220F-5
TO-220-5
TO-220F-6
TO-220-6
TO-220F-5
TO-220F-5
TO-220F-6
TO-220F-5
TO-220F-5
Devices with power ranges from 7 to 210W (230VAC +/- 15%)
17
Features of Fairchild Power Switch
• Current mode operation (FSD2XX are voltage mode)
• Directly limits current flowing through the transformer
• Fully avalanche-rated and 100% tested SenseFET
• Lossless current sensing
• Cheaper snubber network
• More robust against transients
• Protection
• Over Voltage Protection (OVP) - protects against output over voltage due
to open circuit opto for example.
• Over Temperature Protection (OTP) – self explanatory
• Over Load Protection (OLP) – shuts down in the event of over-load
• Abnormal Over-current (AOCP) – protects against short circuit secondary
diodes and transformer shorts
Together, these benefits lead to a lower cost, more reliable
18 system solution
9
Block Diagram of FSDM0365RN
#2
#5
VSTR
VCC
Internal
BIAS
L
VREF(5V)
H
Burst Block
EMI reduction
Block
0.3V/0.5V
#6,7,8
12/8V
DRAIN
Built-in Start up
With High voltage J-FET
FREQUENCY
MIDULATION
0.5V
VCC
5uA
5V
OSC
(70kHz )
5V
BURST
LIMIT
1mA
Soft Start
15ms
#3
Built-in soft
start
VFB
S
Q
Tr=34ns
Tf=32ns
R
LEB
#1
2.5R
R
GND
#4
Offset
Ilimit
UVLO
6.0V
R
Over Voltage
Protection(19V)
Current Limit
Block
Q
Rsense
S
Thermal Shutdown
(140)
2V
OVER CURRENT LIMIT
Protection
block
19
Typical Single Output FPS Schematic
Transformer &
Secondary side
RCD Snubber
T1
TRNSFMR EF20 AUXWIN
1
R102
15K
2
3
L101
8
7
C103
10nF
1000V
6
D201
SB540
22uH
+ C201
1000uF
16V
C202
470uF
16V
+
1
2
J1
CONN RECT 2
4
D102
1N4148
D101
DF10M
D106
1N4148
FS101
230V/T1A
2
VStr
VCC
IC101
FSDM0365RN
1
GND
5
R104
10R
6
-
C101
22uF
400V
7
~
1
Drain
3
+
8
2
~
Drain
+
4
Drain
LF101
2 x 10mH
ILim
C100
150nF
1
VFB
2
+
3
C105
1uF
50V
R201
1K
IC103
FOD2741BTV
4
CONN101
B2P3-VH
R103
22K
C107
47nF
50V
R202
1K
0.25W
R208
3.9K
R205
3.9K
C207
22nF
25V
R204
1K
Input EMI filtering
& bridge rectifier
20
Power switch
Reference &
Feedback
10
Features: Built-in Start-Up with Soft
Start
Peak Drain Current, ID
15 steps
1.6mS
t
• The built-in start-up provides trickle current to charge
Vcc until the FPS supply voltage reaches a preset
voltage threshold called Vstart
Output voltages
• Switching then starts and the peak current allowed is
increased in fixed increments over time.
• Soft start advantages
• Helps prevent high currents during start-up from
damaging the components in the SMPS
• Limits output voltage overshoots
Drain current &
voltage
21
Features: Burst Mode Operation
• In standby mode, outputs draw very
little to no load current.
• Vfb is proportional to load.
Burst Operation
VFB
Normal Operation
VBURSTH
• When the feedback voltage goes
V
below VBURSTH threshold, switching
Current
continues at a fixed current driving VFB waveform
down further.
BURSTL
• When the feedback voltage goes
below VBURSTL threshold, switching
stops in the FPS device, saving
considerable power.
• Depending on load current, the output
voltages will eventually fall causing the
feedback voltage to increase.
• Once the feedback voltage crosses
VBURSTH threshold, switching starts
again and the process repeats.
Switching OFF
Switching OFF
Standby
Power on
Burst operation
@ 85Vac
Standby
Power on
Burst operation
@ 265Vac
22
11
Features: Frequency Modulation for EMI
Internal
Oscillator
138kHz
Peak Level Limit
Peak Level Limit
Average Level Limit
Drain to
Source
voltage
Average Level Limit
Peak waveform
Average waveform
Drain to
Source
current
Vds
Waveform
8kHz
130kHz
138kHz
Turn-on
With Frequency Modulation, freq =134kHz
Without Frequency Modulation,
freq =100kHz
134kHz
Turn-off
point
• Fixed frequency oscillators generate EMI (Electromagnetic Interference) in a narrow band of
the frequency spectrum
• Fairchild’s latest FPS devices modulate its frequency over a ±4kHz frequency range
• EMI is thus spread over a wider range of frequencies
• Allows simple EMI filters to be employed for meeting worldwide EMI requirements
• Refer to Application Note AN-4145: Electromagnetic Compatibility for Power Converters for
additional information
23
Over-Load Protection
• The switching current is defined by VFBI
which quickly increases with output power,
as CFB is charged by IFB.
Internal OLP
block. CFB is
external.
IFB = 900µA
IDELAY = 5µA
• Once VFB passes 3v at T1, the device is
switching at maximum current & VFBI cannot
rise any further
• VFB continues to rise slowly, charged by
IDELAY since D1 is now reverse biased
• Once VFB passes 6v at T2, the device shuts
down
• VCC will now fall and once it passes the
lower UVLO threshold, the device will
attempt auto-restart
• OLP is therefore timed allowing short term
transient over-loads without shut-down.
Delay times can be set by CFB value. Extra
delay can be introduced with a zener &
capacitor in parallel with CFB
24
6v
VFB
3v
T1
T2
t
All switching stops
Now at max switching
current
12
Normal Operating Waveforms
Traces from top to bottom:
• Drain voltage
• Drain current 0.5A / div
• Voltage across secondary diode of 24V output
Input Voltage: 195VAC
Input Voltage: 265VAC
Note: The switch current is the same in both cases. Therefore, Vfb will be the
same.
25
Switching Waveform at a Minimal Load
Input Voltage: 230VAC
Traces from top to bottom:
• Drain voltage
• Supply voltage
• Feedback voltage
• The power loading is very much
reduced and so the duty cycle is
small
• The system is clearly running in
DCM
• The ripple- once the secondary
side has finished conducting- is
normal and is caused by
resonance of stray capacitance in
the switch and the primary
inductance
26
13
Quasi-Resonant Operation
quasi
adj : having some resemblance; "a quasi success";
"a quasi contract“ – “somewhat resonant”
In our case, by strategically adding an L-C tank circuit
to a standard PWM power supply, we can utilize the
resonant effects of the tank to “soften” the transitions of
the switching device. This will help reduce the
switching losses and EMI associated with hard
switching converters.
The fact that we are utilizing the resonant circuit only
during the switching transitions to an otherwise
standard square wave converter gives rise to the name
Quasi-Resonant!
27
Flyback Parasitics
• The MOSFET switch has parasitic
capacitances
• COSS = CDS + CDG It is critical to switching
losses in hard switching converters
• CISS = CGS + CDG
• The transformer has a leakage
inductance. This can be modelled as a
(hopefully) small inductance in series with
the primary
• The transformer also has a winding
capacitance but this is often ignored.
• CTOT = total parasitic capacitance
• LL = leakage inductance
28
14
A Closer Look at the Drain Resonances
Secondary
Current
• There are two resonances
of interest
1. CTOT resonating with LL at
primary switch off
2. CTOT resonating with LP at
secondary switch off
Drain Voltage
1
2
Note: this is not present in
CCM
• Note the effect of the
Snubber at switch-off
29
Now Let’s Turn This into a QRC System
1. Remove the Snubber
Now we have a real problem
since we risk causing the switch
to avalanche. The CTOT, LL
resonance voltage is huge
V2 = IPK2 x LL / CTOT
T1
TRNSFMR EF20 AUXWIN
Note: without the snubber the
switching device would
avalanche and be destroyed.
1
R102
15K
C103
10nF
1000V
8
7
2
3
6
4
D102
1N4148
FPS
Q1
30
15
Now Let’s Turn This into a QRC System
2. Add extra capacitance across the
switch
• CTOT is increased so V has
decreased
• The frequency of both resonances
has also decreased
• The technique is to increase the
capacitance until the peak drain
voltage is acceptable. The lower
the capacitance the lower the
switching loss but the higher the
drain voltage
T1
TRNSFMR EF20 AUXWIN
1
8
7
2
3
6
4
FPS
Q1
C104
2.2nF
1000V
31
More on Turning This Into a QRC System
3. Adjust the switching frequency & duty
cycle by adjusting the point at which the
device switches on
• Now the switch turns on at the bottom of
the CTOTLP resonance
• Since the switch off edge is slowed, we
have soft switching reducing EMI.
• Since VDS is reduced or zero at switch on,
we have lower COSS switching losses and
reduced EMI.
• A QRC system has variable switching
frequency defined by load and input voltage Note: In reality, the Snubber spike
would probably be smaller
• Since the DC link will always have some
but this is shown for clarity.
32
ripple, even with constant load, the
switching frequency will be modulated
by the ripple helping to spread EMI
energy.
16
Detecting the Valley with an “FSCQ”
Device
• To detect the drain voltage valley with
a Fairchild FSCQ device, the VCC bias
winding is monitored
• First, RSY1 and RSY2 should be chosen
so that the divided signal is less than
the OVP trip point of 12 V, typically 34V less
• Next, CSY needs to be selected such
that the decay of the sync waveform
ensures it passes the sync threshold
when the drain voltage is at the bottom
of its resonance.
Synchronization circuit
33
Detecting the Valley
• Sync pin components
selected correctly
• The sync pin voltage passes
through the threshold at the
minimum of the CTOT, LP
resonance
• Switching occurs at minimum
or zero VDS
34
17
A Limitation of QRC Mode
• The limitation
• As the load decreases or input voltage
increases, the switching frequency
increases
• When the system reaches its maximum
switching frequency, it starts to cycle skip
leading to reduced regulation
• Alternatively we could end up with a
minimum frequency within the audio band.
Switching
Frequency
90kHz
Extended QR
Operation
Normal QR
Operation
45kHz
• The solution
• As the system approaches an upper limit for
the switching frequency, switch on the
second minimum
Output Power
• Switching frequency will drop allowing the
load to further decrease without cycle
skipping
35
Extended Quasi-Resonant Switching
• Fixing the QRC limit
• At low power levels, the device will
operate in extended QRC
• As the output power falls, the switching
frequency increases
• Once the frequency reaches the upper
limit of 90kHz, extended QRC operation
takes over
• The device now switches on the second
minimum and the switching frequency
drops
• The same effect can be seen as input
voltage rises
• When the device goes into extended
QRC mode, the switching frequency
decreases, allowing more range for it to
increase before cycle skipping occurs.
36
18
Non-Isolated Topologies
37
Non-Isolated Topologies: Buck vs
SEPIC
• Buck – can only step down
• High Side Switch
+12V output, 1 inductor
• SEPIC – can step up or down
• Low Side Switch
+12V output, 2 inductors, 1 high voltage
capacitor
• The current ripple on the SEPIC can be
minimized by having large inductor.
38
19
FSDL0165: Buck Mode Implementation
39
Buck Mode Operation – the On Time
375V
Note: the FPS is operating as a high side switch
375+15V
375V
Diode Blocked
15V
0V
40
20
Buck Mode Operation – the Off Time
375V
Diode Conducts
15V-Vf
Vf
15V
15V
-Vf
Vf
0V
41
Test Results for FSDL0165 Buck Board
• Low load input power
• Highest value 370mW at 3oC, 265V
• Minimum load was set to 10mA
• Efficiency
• Exceeds 78% over temperature and voltage for loads of 100mA, 200mA and 300mA
• Efficiency changes
• Best at high temperatures.
•Core losses which decrease with temperature dominate over conduction losses which increase.
• Best at mid voltages
•Worse at low voltages. (Rds(on) losses increase).
•Worst at high voltages (switching losses dominate)
• Accuracy
• 14.8V – 15.5V, -1.3% to +3.3%
• (0-300mA load, full temperature range)
• Device variations not included
• The FSDL0265 and FSDL0365 devices can be used for higher power levels
42
21
Single Ended Primary Inductance
Converter (SEPIC)
• The boost converter limitation is that VOUT must always be greater than VIN since
there is a DC path between input and output
• The SEPIC removes this limitation by inserting a capacitor between the inductor
and the rectifier to block the dc path
• However, the rectifier anode must be connected to a known potential by L2
Boost
SEPIC
• So what’s the advantage of SEPIC since it contains an additional inductor
and capacitor?
43
•
•
•
Vout can be above OR below Vin
Better EMC – more of this later
Potentially lower cost – more of this later
FSDL0165 SEPIC Implementation
Advantages:
• Input current may be continuous => choose large L1
• Switching losses are smaller compared to buck topology
• Wide input and output voltage range is possible
L1
2200uH/120mA
C3
+
+12V
D5
UF4005
3.3uF
400V
D6
1N4007
D2
R5
1N4148
22R
0.25W
6
Drain
8
1
FS2
230V/250mA
Drain
Drain
VStr
GND
5
VCC
ILim
C1 +
22uF
400V
VFB
3
2
1
B3P-VH
CONN2
2
3
+
C4
100nF
50V
C2
1uF
50V
L3
470uH
?A
D4
13V
0.5W
R6
Q1
BC337-25
470R
0.25W
R3
100R
0.25W
2
1
R4
xxxR
0.25W
C5
470uF
25V
IC1
FSDM0165RN
4
D3
1N4007
7
+
Note: this application is available as a completed evaluation board
CONN1
B2P3-VH
44
22
SEPIC Converter: On Time Current Flow
L1
2200uH/120mA
C3
+
+12V
D5
UF4005
3.3uF
400V
D6
1N4007
D2
R5
1N4148
22R
0.25W
8
6
Drain
Drain
Drain
1
FS2
230V/250mA
VCC
ILim
VStr
GND
5
VFB
2
3
+
L3
470uH
?A
C2
1uF
50V
B3P-VH
CONN2
D4
13V
0.5W
R6
C4
100nF
50V
Q1
BC337-25
470R
0.25W
R3
100R
0.25W
2
R4
xxxR
0.25W
1
3
2
1
C5
470uF
25V
IC1
FSDM0165RN
4
C1 +
22uF
400V
7
+
D3
1N4007
CONN1
B2P3-VH
Idmax = Iin + Iout + (ΔIL1 + ΔIL2) / 2
45
SEPIC Converter: Off Time Current Flow
L1
2200uH/120mA
C3
+
+12V
D5
UF4005
3.3uF
400V
D6
1N4007
D2
R5
1N4148
22R
0.25W
6
Drain
8
1
FS2
230V/250mA
Drain
Drain
VStr
GND
5
VCC
ILim
C1 +
22uF
400V
VFB
3
2
1
B3P-VH
CONN2
2
3
+
C4
100nF
50V
C2
1uF
50V
L3
470uH
?A
D4
13V
0.5W
R6
Q1
BC337-25
470R
0.25W
R3
100R
0.25W
2
1
R4
xxxR
0.25W
C5
470uF
25V
IC1
FSDM0165RN
4
D3
1N4007
7
+
CONN1
B2P3-VH
46
23
Buck vs. SEPIC Comparison
Input Voltage / VAC
Min
185
Output Voltage / VDC
Output Current / A
15
0.3
Efficiency at Vin = 230VAC
Max
265
Pout / W
Buck
4.59
SEPIC
4.56
Pin / W
Efficiency
5.78
79.4%
5.7
80.0%
68
63
FPS Device Temperature / C
For similar specified converters, the performance is similar
47
Input Switching Current Harmonics
• In all converters, the input switching current is provided by the DC link capacitor. It’s not
perfect, so this generates a voltage:
•
V = Isw * ESR
• The buck mode switching current is a large impulse.
• It is always discontinuous and contains a high harmonic content.
• The SEPIC mode switching current is a small saw-tooth.
• It is continuous and contains a low harmonic content.
• Hence, the EMI is reduced
• It may be possible to meet EMI requirements without the use of an input filter and hence
offset the cost of the extra capacitor and inductor in the SEPIC
• Hence, cost is reduced since an EMI filter often costs more than the extra inductor and
capacitor required by the SEPIC.
48
24
Buck vs SEPIC EMI Harmonics in
Practice
• The two plots show the EMI spectra of the SEPIC (left) and the buck board
(right). No extra EMI filtering was used. Peak power was measured.
• The SEPIC is very close to meeting the EMI levels shown.
• The buck is well above the limits and needs extra LC filtering
• These results are to be expected from the large differences in the peak currents
49
LED Lighting Solutions
50
25
Off-line Isolated Current Source with FPS
Basic Control Circuit
51
Isolated Current Source with FPS
• Standard voltage
regulator with KA431
• Low cost current regulator
with bipolar transistor
52
26
Isolated Current Source with FPS
• Application: LED Driver
• Application Note: AN-4138
53
SEPIC Converter for LED lighting
54
27
Fairchild’s LED Drivers & DC-DC
Converters for Low Voltage Applications
• Fairchild’s extensive state-of-the-art LED driver family provides designers
with a multitude of LED driver choices—including the most popular options
and features for various lighting applications. These low-dropout drivers
provide a complete backlighting solution for portable devices
Parallel
No Boost/Bias
Drivers
Serial
Switched Cap Boost
Matched Current
Constant & Matched
Current
FAN5611, FAN5612
FAN5613, FAN5614
FAN5610
FAN5609, FAN5607
FAN5602 (Flash LED)
Inductor Boost
FAN5606, FAN5608
FAN5331, FAN5333
55
LED Drivers Product Portfolio
Fairchild’s LED drivers save space, reduce part count, and provide uniform
brightness. These LED drivers can regulate the LEDs either directly from the battery
or through a DC/DC converter, and come in small (SC-70, MLP, and TSOT)
packages to fulfill even the smallest PCB area requirements
Part Number
Max
current
Number of LEDs per (mA per
Channels channel channel)
FAN5602 4.5v backlight
N/A
1-6
200
FAN5602 5.0v flash
Brightness
control
method
Input Voltage
Boost / Method
A, P
Unregulated / battery
Yes / charge pump
Shutdown
current
Max
(max uA) Efficiency
1
85%
Packages
MLP 3x3
N/A
1-6
200
A, P
Unregulated / battery
Yes / charge pump
1
85%
MLP 3x3
FAN5606
1
6
20
D,A,P
Unregulated / battery
Yes, inductor built in diode
1
85%
MLP 3x3
FAN5331
1
5
50
A,P
Unregulated / battery
Yes, inductor
1
80%
SOT-23
FAN5332
1
6
75
A,P
Unregulated / battery
Yes, inductor
1
80%
SOT-23
FAN5333
1
6
75
A,P
Unregulated / battery
Yes, inductor
1
80%
SOT-23
FAN5608
2
6
20
D,A,P
Unregulated / battery
Yes, inductor built in diode
1
85%
MLP 3x3, 4x4
FAN5607
4
1
20
A,P
Unregulated / battery
Yes, adaptive charge pump
2
92%
MLP 4x4
FAN5609
4
1
20
D,P
Unregulated / battery
Yes, adaptive charge pump
3
86%
MLP 4x4
FAN5610
4
1
21
D,P
Unregulated / battery
Not required
1
91%
MLP 3x3
FAN5611
4
1
40
A,P
Regulated
Not required
1
90%
SC70-6, TSOT-6
FAN5612
3
1
40
A,P
Regulated
Not required
1
90%
SC70-6, TSOT-6
FAN5613
4
1
40
A,P
Regulated
Not required
1
90%
MLP 2x2
FAN5614
2
1
80
A,P
Regulated
Not required
1
90%
SC70-6, TSOT-6
* Brightness Control Method: D=Digital, A=Analog, P=PWM
56
28
FAN5606: Serial LED Driver
Features
• Drives up to six LEDs at 20mA constant
current
• Digital, analog or PWM brightness control
• 2.7V to 5V input voltage range
• Adaptive output voltage for high efficiency
• Built-in current-regulated DC/DC boost
• Soft-start
• Built-in Schottky diode
• Single resistor current set in analog mode
• LED short circuit protection
• Shutdown current <1µA
• Small footprint 8-lead 3x3 MLP
Applications
• Portable video devices
• Cell phones
• Digital still cameras (DSCs)
• PDAs
57
FAN5608: Dual Serial LED Driver
Features
• Drives 2x6 LEDs at 20mA constant current
• Digital, analog or PWM brightness control
• 2.7V to 5V input voltage range
• Adaptive output voltage for high efficiency
• Built-in current-regulated DC/DC boost
• Accurate current matching between LED
strings
• Soft-start
• Built-in Schottky diode
• Single resistor current set in analog mode
• LED short circuit protection
• Shutdown current <1µA
• Small footprint 8-lead 3x3 MLP
Applications
• Portable video devices
• Cell phones
• Digital Still Cameras
• PDAs
58
29
FAN5331 - 20V, 1A Boost OLED
Driver / DC-DC Converter
•Features
•VIN = 2.7V to 5.5V
•1A peak switch current
•Adjustable output voltage
•50mA @ 15V
•1.6MHz operating frequency
•Low noise PWM operation
•Cycle-by-cycle current limit
•Over-voltage protection
•5-lead SOT-23 package
•Applications
•Portable video devices
•Cell phones
•Hand Held computers
•Portable electronic equipment
•Digital Cameras
59
FAN5332 - 30V, 1.5A Boost OLED
Driver / DC-DC Converter
•Features
•VIN = 2.7V to 5.5V
•1A peak switch current
•Adjustable output voltage
•75mA @ 20V easily achieved
•1.6MHz operating frequency
•Low noise PWM operation
•Cycle-by-cycle current limit
•Over-voltage protection
•5-lead SOT-23 package
•Applications
•Cell phones
•Hand Held computers
•Portable electronic equipment
•Digital Cameras
60
30
FAN5333A/B
20V @ 75mA Boost LED Driver
Features
• VIN = 1.8V to 5.5V
• 1A peak switch current
• Adjustable output voltage
• 75mA @ 20V
• 0.1V feedback voltage
• High Frequency PWM Dimming
• 1.6MHz operating frequency
• Low noise PWM operation
• Cycle-by-cycle current limit
• Over-voltage protection
• 5-lead SOT-23 package
Applications
• Torches
• Bicycle Front Light
• Portable electronic equipment
• Digital cameras
61
Related Links
The pdf version of the Power Seminar presentations are available on the our external
website. To access or download the pdfs, please visit
www.fairchildsemi.com/power/pwrsem2006.html
For product datasheets, please visit www.fairchildsemi.com
For application notes, please visit www.fairchildsemi.com/apnotes
For application block diagrams, please visit www.fairchildsemi.com/markets
For design tools, please visit the design center at www.fairchildsemi.com
For more information on Fairchild Power Switch, please visit
http://www.fairchildsemi.com/whats_new/fps.html
For more information on LED Drivers, please visit
http://www.fairchildsemi.com/whats_new/led_drivers.html
62
31
Appendix 1 – Additional FPS operating
waveforms
63
Short Circuit Protection
Input voltage: 230V AC
Traces from top to bottom:
• Drain voltage
• Supply voltage
• 24V output
• When the device is switching,
the current increases rapidly
since the magnetic core is
unable to reset each cycle
• The switch current is limited by
the over-current protection and
therefore the duty cycle is small
• Vfb rises and the device shuts
down when Vfb passes the OLP
threshold
64
32
Secondary Diode Short Circuit
Input voltage: 230V AC
Traces from top to bottom:
• Drain voltage
• Supply voltage
• When the switch turns on, the current
through the inductor ramps up very fast.
This is because the inductance is
effectively reduced to the leakage
inductance only
• The on time is limited by the switch
current limit which results in a very short
duty cycle
• The magnetic core is unable to reset
each cycle
• After a few cycles the device stops
switching
• A long time-base measurement would
show that Vcc drops to the lower UVLO
threshold and the device auto-restarts
65
Shorted Optocoupler LED
Input voltage: 230V AC
Traces from top to bottom:
•
•
•
•
Drain voltage
Supply voltage
Feedback voltage
Supply voltage
• Since the opto LED is shorted, no
current can flow through the optotransistor and Vfb rises quickly to
3.5V
• At this point, the device is switching
at maximum duty, Vcc rises quickly
and passes the over-voltage
protection (OVP) threshold at 24V
• The device then stops switching
and Vcc starts to decay
• Vfb continues to rise but this has no
effect. Once Vcc decays to the lower
ULVO threshold the device autorestarts and the cycle repeats
66
• Start-up is normal
33
Feedback Shorted to Ground
Input voltage: 230V AC
Traces from top to bottom:
•
•
•
•
Drain voltage
Supply voltage
Feedback voltage
Supply voltage
• The device tries to start as normal:
Vcc rises and passes the UVLO
threshold
• The device then draws more
current and tries to switch but
cannot as Vfb is grounded
• The Vcc capacitor discharges
because of the increased Vcc
current and falls below the lower
UVLO threshold where the device
tries to start again
67
34