Download A Design Study of a Future 10 kW Converter Sebastian Fant

A Design Study of a Future 10 kW Converter
Examensarbete utfört i elektroniksystem av
Sebastian Fant
LiTH-ISY-EX--08/4093--SE
Linköping 2008
A Design Study of a Future 10 kW Converter
Examensarbete utfört i elektroniksystem
vid Linköpings tekniska högskola
av
Sebastian Fant
LiTH-ISY-EX--08/4093--SE
Examinator: Kent Palmkvist
Handledare: Jonny Lindgren
Institution och avdelning
Institutionen för systemteknik
Presentationsdatum
2008-03-18
Publiceringsdatum (elektronisk version)
Department of Electrical Engineering
Språk
Typ av publikation
Svenska
x Annat (ange nedan)
Licentiatavhandling
x Examensarbete
C-uppsats
D-uppsats
Rapport
Annat (ange nedan)
Engelska
Antal sidor
93
ISBN (licentiatavhandling)
ISRN LiTH-ISY-EX--08/4093--SE
Serietitel (licentiatavhandling)
Serienummer/ISSN (licentiatavhandling)
URL för elektronisk version
http://www.ep.liu.se
Publikationens title
A Design Study of a Future 10 kW Converter
Författare
Sebastian Fant
Sammanfattning
This master thesis aim to design and evaluate a high power 3-phase DC/AC and AC/AC converter. The
purpose is to use it for an electric motor in an aircraft possibly driving electric actuators or a propeller
in an UAV or a small vehicle. Factors such as power loss and weight are of importance and will be
estimated using known models supplied by various manufacturers of components. Different topologies
of semiconductors suitable for this purpose are examined and presented. Extensive resources have been
put to properly select the most suitable switching device according to their power loss and weight.
The need for filters and protective circuits will be estimated according to regulations of common
military avionic standards and will be included in the resulting estimation along with simulations to
evaluate their need and importance.
Snubber circuits will be presented and their specific ability to reduce voltage transients and
switching losses will be examined along with some simulations to illustrate their performance.
In the final part an estimation of efficiency and weight of higher and lower power models of the
same inverter has been made using the same procedure as presented in this paper. Engineering rules
have been formed from these estimations to simply be able to calculate the proportions of a future
inverter of arbitrary rated power.
Antal sidor: 93
Nyckelord
Inverter design, Power electronics, Snubber circuits, filter, IGBT
Abstract
This master thesis aim to design and evaluate a high power 3-phase DC/AC and AC/AC
converter. The purpose is to use it for an electric motor in an aircraft possibly driving
electric actuators, a propeller in an UAV or a small vehicle. Factors such as power loss
and weight are of importance and will be estimated using known models supplied by
various manufacturers of components. Different topologies of semiconductors suitable
for this purpose are examined and presented. Extensive resources have been put to
properly select the most suitable switching device according to their power loss and
weight.
The need for filters and protective circuits will be estimated according to regulations of
common military avionic standards and will be included in the resulting estimation along
with simulations to evaluate their need and importance. Snubber circuits will be
presented and their specific ability to reduce voltage transients and switching losses will
be examined along with some simulations to illustrate their performance. In the final part
an estimation of efficiency and weight of higher and lower power models of the same
inverter has been made using the same procedure as presented in this paper. Engineering
rules have been formed from these estimations to simply be able to calculate the
proportions of a future converter of arbitrary rated power.
Keywords: Inverter design, Power electronics, Snubber circuits, filter, IGBT
I
Sammanfattning
Det här examensarbetet syftar till att designa och utvärdera en högeffektskonverter
kapabel att konvertera lik eller trefas växelspänning till en trefas växelspänning med
högre frekvens. Syftet är att driva en elektrisk motor för möjligtvis en domkraft eller en
propeller för en obemannad eller småskalig flygfarkost. Faktorer såsom förlusteffekt och
vikt är viktigt och kommer att uppskattas med hjälp av välkända modeller framtagna av
flertalet tillverkare av komponenter. Olika topologier av halvledarkomponenter passande
ändamålet är undersökta och presenterade. Mycket tid läggs på att hitta de perfekta
halvledarkomponenterna för ändamålet med tyngdpunkt på dess förlusteffekt och vikt.
Behovet av filter och skyddskretsars omfattning är uppskattade enligt standardiserade
bestämmelser för flygfarkoster och kommer i slutskedet att bidra med både vikt och
effektförluster. Simuleringar utförs i Matlab och Simulink för att visa dess behov och
prestanda. Olika typer av snubberkretsar presenteras med dess unika egenskaper som
syftar till att undertrycka spänningstransienter over de känsliga halvledarkomponenterna.
Simuleringar utförs i Pspice för att illustrera varje snubberkrets respektive för och
nackdelar samt prestanda. I slutskedet av rapport har det genomförts en uppskattning av
verkningsgraden och vikten gällande för konverters av högre och lägre märkeffekt. Det
har gjorts analogt med det tillvägagångssätt som använts i denna rapport. Ingenjörsregler
har blivit approximerade för att enkelt kunna uppskatta vikt och förlusteffekt för
godtycklig märkeffekt.
Nyckelord: Inverter design, kraft elektronik, snubberkretsar, högeffektsfilter, IGBT
II
Acknowledgements
This report is a result of my master-thesis project carried out at SAAB Aerosystems in
Linköping under supervision of the department of Electrical Engineering at the
University of Linköping. As a student at the programme for applied physics and electro
engineering I chose the alignment which felt the most natural and rewarding to me,
electronics. This master-thesis was chosen among several available projects due to my
interest in analog and high power electronics although the amount of the courses taken in
the field was few. As I expect the need for this competence will grow significantly in the
near I made the tactical choice by accepting the proposal offered to me.
Looking back over the time spent at SAAB Aerosystems it has been very interesting,
rewarded and has completely fulfilled my expectations. I have been given opportunities
to reflect the daily life of an avionic electrical engineer and the problems they face.
Great insight into the avionic industry has been given by my brilliant supervisor Lars
Austrin and I would like to thank him dearly for that. I would also like to thank Eduardo
Figueroa for giving me many discussions about the future life as an engineer and where it
might lead. I would like to thank Andreas Johansson for his revising and the many
practical hints he gave me. I would like to thank my co-worker Daniel Eidborn for a job
well done and I wish you all the luck in the future.
Thank you SAAB and especially the TDG department for providing the finances and
office necessary for completing this master-thesis.
Last but not the least I would like to thank my family and my friends for supporting me
and my studies throughout the years.
Linköping
March 2008
Sebastian Fant
III
Table of contents
LIST OF FIGURES.................................................................................................................................... VI
LIST OF TABLES...................................................................................................................................... VI
1.
INTRODUCTION............................................................................................................................... 1
1.1.
1.2.
1.3.
1.4.
1.5.
1.6.
2.
OBJECTIVES ................................................................................................................................. 1
BACKGROUND .............................................................................................................................. 1
CHALLENGES ............................................................................................................................... 1
RESEARCH METHOD ..................................................................................................................... 2
DELIMITATIONS ........................................................................................................................... 2
STRUCTURE OF REPORT ................................................................................................................ 2
THEORY ............................................................................................................................................. 5
2.1.
CONVERTER PRINCIPLES............................................................................................................... 5
2.2.
PRINCIPLES FOR CONTROL ........................................................................................................... 7
2.2.1. Switching frequency................................................................................................................ 7
2.3.
POWER COMPONENTS ................................................................................................................... 8
2.3.1. Switching transistors .............................................................................................................. 8
2.3.1.1.
2.3.1.2.
2.3.1.3.
2.3.2.
IGBT ............................................................................................................................................ 8
MOSFET.................................................................................................................................... 12
BJT............................................................................................................................................. 13
Free-wheeling-diodes ............................................................................................................13
2.3.2.1.
2.3.2.2.
Silicon Schottky ......................................................................................................................... 14
Silicon Carbide Schottky............................................................................................................ 14
2.3.3. Rectifier .................................................................................................................................14
2.4.
FILTERS .......................................................................................................................................16
2.4.1. Input filters ............................................................................................................................16
2.4.2. DC-bus filter ..........................................................................................................................19
2.4.2.1.
2.4.2.2.
Capacitor bank ........................................................................................................................... 19
Choke inductor........................................................................................................................... 25
2.4.2.3.
Output filters .....................................................................................................................26
2.5.
SNUBBER CIRCUITS .....................................................................................................................30
2.5.1. Increased SOA .......................................................................................................................30
2.5.2. Reducing losses......................................................................................................................35
2.6.
PRINCIPLES OF COOLING..............................................................................................................36
2.6.1. Thermal resistance ................................................................................................................36
2.6.2. Heat transfer..........................................................................................................................36
2.7.
CHASSIS DESIGN ..........................................................................................................................37
2.7.1. Cooling scenarios ..................................................................................................................39
2.7.1.1.
2.7.1.2.
2.7.1.3.
2.8.
2.9.
2.10.
3.
Exterior heatsink ........................................................................................................................ 39
Interior heat sink ........................................................................................................................ 42
Internal fluid cooling.................................................................................................................. 43
TRANSISTOR DRIVES ...................................................................................................................43
POWER SUPPLY............................................................................................................................45
CONTROLLER ..............................................................................................................................46
LOSSES ..............................................................................................................................................47
3.1.
SWITCHING LOSSES .....................................................................................................................47
3.1.1. Transistor ..............................................................................................................................47
3.1.2. Diode .....................................................................................................................................48
3.2.
CONDUCTION LOSSES ..................................................................................................................49
3.2.1. Transistor ..............................................................................................................................49
3.2.2. Diode .....................................................................................................................................49
3.3.
LOWEST LOSS ESTIMATION OF IGBT AND FWD .........................................................................50
3.4.
RECTIFIER ...................................................................................................................................51
IV
3.5.
FILTERS .......................................................................................................................................52
3.5.1. Input filter ..............................................................................................................................52
3.5.2. DC-bus filter ..........................................................................................................................52
3.5.3. Output filter ...........................................................................................................................53
3.6.
CONTROLLER ..............................................................................................................................53
3.7.
TRANSISTOR DRIVES ...................................................................................................................53
3.8.
INTERNAL POWER SUPPLY ...........................................................................................................54
4.
RESULTS ...........................................................................................................................................55
4.1.
4.2.
4.3.
4.4.
4.5.
POWER DENSITY ..........................................................................................................................55
SCALABILITY ..............................................................................................................................56
EFFICIENCY .................................................................................................................................56
RELIABILITY ...............................................................................................................................57
SIMULATIONS ..............................................................................................................................58
5.
CONCLUSIONS ................................................................................................................................59
6.
FUTURE WORK ...............................................................................................................................60
BIBLIOGRAPHY .......................................................................................................................................61
APPENDIX A1 – OUTPUT CHARACTERISTICS USING SURGE CURRENT LIMITER .........................................63
APPENDIX A2 – OUTPUT CHARACTERISTICS USING NO SURGE CURRENT LIMITER ..................................64
APPENDIX A3 – RECTIFIER/IGBT CURRENTS USING SURGE CURRENT LIMITER .......................................65
APPENDIX A4 – RECTIFIER/IGBT CURRENTS USING NO SURGE CURRENT LIMITER .................................66
APPENDIX B – SYSTEM SIMULATION MODEL ............................................................................................67
APPENDIX C1 – MAIN GRID VOLTAGE HARMONICS WITHOUT FILTER ......................................................68
APPENDIX C2 – MAIN GRID CURRENT HARMONICS WITHOUT FILTER ......................................................69
APPENDIX C3 – MAIN GRID VOLTAGE HARMONICS WITH FILTER .............................................................70
APPENDIX C4 – MAIN GRID VOLTAGE HARMONICS WITH 12-PULSE RECTIFIER .......................................71
APPENDIX C5 – MAIN GRID CURRENT HARMONICS USING 12-PULSE RECTIFIER ......................................72
APPENDIX D1 – IGBT CHART 1 .................................................................................................................73
APPENDIX D2 – IGBT CHART 2 .................................................................................................................74
APPENDIX D3 – MOSFET/DIODE CHART ....................................................................................................75
APPENDIX E1 – DECOUPLING CAPACITOR SNUBBER CIRCUIT SCHEMATICS..............................................76
APPENDIX E2 – RESTRICTED DECOUPLING CAPACITOR SNUBBER CIRCUIT SCHEMATICS .........................76
APPENDIX E3 – RCD CHARGE/DISCHARGE SNUBBER CIRCUIT SCHEMATICS ...........................................77
APPENDIX E4 – RCD CLAMP-SNUBBER CIRCUIT SCHEMATICS .................................................................77
APPENDIX F – LAYOUT OF COMPONENTS WITHIN ENCLOSURE...................................................................78
V
List of figures
FIGURE 1 : INVERTER OVERVIEW ..................................................................................................................... 5
FIGURE 2 : RECTIFIER STAGE ........................................................................................................................... 5
FIGURE 3 : SMOOTHING STAGE ........................................................................................................................ 5
FIGURE 4 : SWITCHING STAGE ......................................................................................................................... 6
FIGURE 5 : FILTERING OF PULSE TRAIN IN OUTPUT STAGE ............................................................................... 7
FIGURE 6 : SIMPLIFIED MODEL ........................................................................................................................ 8
FIGURE 7 : SYMBOL LAYOUT ........................................................................................................................... 9
FIGURE 8: TURN-OFF BEHAVIOR ...................................................................................................................... 9
FIGURE 9: INTERNAL STRAY CAPACITANCES IN AN IGBT...............................................................................10
FIGURE 10 : TYPCIAL IGBT TURN-ON ............................................................................................................11
FIGURE 11 : TYPICAL IGBT TURN-OFF ..........................................................................................................12
FIGURE 12: IGBT WITH FWD ........................................................................................................................13
FIGURE 13: OVERVEIW OF THE RECTIFIER ......................................................................................................15
FIGURE 14 : INPUT FILTER ..............................................................................................................................17
FIGURE 15 : TYPICAL IMPEDANCE FOR INPUT FILTER VS. HARMONICS OF GRID FREQUENCY ...........................17
FIGURE 16 : PROTECTIVE CIRCUITS: SIMPLE, SERIES AND PARALLEL CONNECTION .........................................21
FIGURE 17 : CHARGE AND DISCHARGE TIME OF CAPACITOR BANK .................................................................22
FIGURE 18 : ELECTRICAL LAYOUT OF CAPACITOR BANK ................................................................................24
FIGURE 20 : OUTPUT FILTER ...........................................................................................................................26
FIGURE 21 : DECOUPLING AND RESTRICTED DECOUPLING CAPACITOR ...........................................................30
FIGURE 22 : SWITCING DEVICE VOLTAGE WITHOUT SNUBBER CIRCUIT ...........................................................31
FIGURE 23 : DEVICE VOLTAGE WITH DECOUPLING CAPACITOR.......................................................................32
FIGURE 24 : DEVICE VOLTAGE WITH DISCHARGE RESTRICTED DECOUPLING CAPACITOR................................33
FIGURE 25 : RCD CHARGE-DISCHARGE SNUBBER AND RCD CLAMP-SNUBBER ............................................34
FIGURE 26 : RCD CHARGE-DISCHARGE SNUBBER .........................................................................................34
FIGURE 27 : RCD CLAMP-SNUBBER...............................................................................................................35
FIGURE 28 : SCETCH OF CHASSIS ....................................................................................................................40
FIGURE 29 : THERMAL RESISTIVITY ESTIMATION ...........................................................................................40
FIGURE 30 : HEATSINK LAYOUT .....................................................................................................................41
FIGURE 31: THERMAL RESISTANCE FACTOR VS. AIRFLOW ..............................................................................42
FIGURE 32 : DESCRIPTIVE DRIVER INTERNAL CIRCUIT (IGBT WITHIN DASHED LINE) .....................................43
FIGURE 33 : APPROXIMATE OUTPUT CHARACTERISTICS .................................................................................49
List of tables
TABLE 1 : INPUT CURRENT HARMONICS..........................................................................................................16
TABLE 2 : VOLTAGE HARMONICS ON GRID .....................................................................................................18
TABLE 3 : INPUT FILTER COMPONENTS ...........................................................................................................19
TABLE 4 : DC-BUS FILTER COMPONENTS ........................................................................................................26
TABLE 5 : OUTPUT FILTER COMPONENTS ........................................................................................................29
TABLE 6 : SNUBBER CHARACTERISTICS ..........................................................................................................36
TABLE 7 : SWITCHING CHARACTERISTICS VS. DRIVER DIMENSIONS ................................................................44
TABLE 8 : DRIVER COMPONENTS ....................................................................................................................45
TABLE 9 : POWER SUPPLY SPECIFICATIONS AND DIMENSIONS ........................................................................46
TABLE 10 : CONTROLLER COMPONENTS .........................................................................................................46
TABLE 11 : SUITABLE DISCRETE IGBTS .........................................................................................................50
TABLE 12: SUITABLE IGBT MODULES ...........................................................................................................51
TABLE 13 : SUITABLE RECTIFIERS ..................................................................................................................52
TABLE 14 : DC-BUS FILTER LOSSES WHEN USING 230/400 VAC INPUT ..........................................................52
TABLE 15 : DC-BUS FILTER LOSSES WHEN USING 115/200 VAC INPUT ..........................................................53
TABLE 16 : APPROXIMATE SEPARATED WEIGHT FOR 230/400VAC APPLICATIONS.........................................55
TABLE 17 : APPROXIMATE SEPARATED WEIGHT FOR 115/200VAC APPLICATIONS.........................................55
TABLE 18 : APPROXIMATE SEPARATED LOSSES @ 6 KHZ SWITCHING FREQUENCY.........................................57
VI
TABLE 19 : APPROXIMATE SEPARATED LOSSES @ 10 KHZ SWITCHING FREQUENCY.......................................57
TABLE 20 : APPROXIMATE SEPARATED LOSSES @14 KHZ SWITCHING FREQUENCY .......................................57
VII
List of symbols
VDC The voltage achieved on the DC-bus after the following smoothing
VAC Input/Output voltage in alternating mode
Collector current
IC
VCD Collector-Emitter voltage
VGE Gate-Emitter voltage
RDS Drain-Source resistance
VDS Drain-Source voltage
ID
Drain current
Abbreviations
AC
Alternating Current
DC
Direct Current
PWM Pulse Width Modulation
PM
Permanent Magnet
FET Field Effect Transistor
IGBT Insulated Gate Bipolar Transistor
SiC Silicon Carbide
PLD Programmable Logic Device
µC
Micro Controller
EMI Electro Magnetic Interference
PMSM Permanent Magnet Synchronous Motor
SOA Safe Operating Area
FWD Free Wheeling Diode
LC
Inductive-Capacitive
RLC Resistive-Inductive-Capacitive
RFI Radio Frequency Interference
MTBF Minimum Time Before Failure
ESR Equivalent Series Resistance
HFE small signal Forward Current Gain
MASL Meters Above Sea Level
LOP Lifetime Of Product
VIII
1. Introduction
As we all get more conscious about the damage the combustion of fossil fuels inflicts on
the environment more and more alternative solutions come to light. Hybrid drive,
biological petrol and power cells are terms most people are familiar with. Electric
propulsion in automotives is being promoted throughout the world as the most
environmental friendly solution on the market. Once electricity is produced the electric
propulsion is completely clean and it is easier to produce clean electricity in a controlled
environment such as a power plant. However, the environment issue is not the only
benefit. The increased control, performance and efficiency achieved are in some
applications the most wanted benefit, especially in avionics.
1.1.
Objectives
This master thesis aims to design and evaluate a 10kW three-phase converter handling
both AC/DC and DC/DC. The function of the converter is to drive a permanent
magnetized synchronized motor possibly working as a starter motor or driving an electric
actuator possibly in an aircraft. A main focus will be to pinpoint the parts responsible for
the largest power loss, how to reduce this magnitude and estimate the weight of the
product. As an extension the characteristics for a few imaginary converters with different
power ratings will be estimated. This to in the end form up approximate equations giving
the power loss and weight based of a few input parameters.
1.2.
Background
Today it is not uncommon to use highly efficient petrol driven generators which supplies
electronic equipment and motors with power and in the end reach a higher efficiency
compared driving the equipment directly with combustion engines. The middle step
between the engine and the power source is the converter, a device not seldom
responsible for a large loss due to inefficiency, demanding cooling and thereby
introducing further weight.
A new concept in aviation is the “More Electric Aircraft” which aims to replace all
hydraulics with an electric correspondence. This is to reach higher efficiency which will
reduce weight but also increase control and improved behavior.
With numerous motors demanding an individual converter its losses and weight becomes
prominent and has to be optimized. The long term goal is to remove the compressor and
improving the generator driven by the combustion engine to supply the electric power
needed.
1.3.
Challenges
The environments and conditions in an aircraft such as JAS 39 GRIPEN are very
different from sea-level. This complicates the objective as electronic devices behave
differently or possibly not at all when exposed to high temperature, cosmic rays and
mechanical stress.
1
Cooling gets more difficult when air density decreases as it does on higher altitudes. A
higher altitude also increases the failure rate due to cosmic radiation further extending the
need for effective cooling. Since weight is an emphasized problem and cooling is
responsible for most of it a well considered tradeoff is needed.
How can we maintain a low weight while still having a high reliability? How much will
new technology reduce power dissipation? Is it possible to use the aircrafts built-in
cooling system and will it improve overall efficiency?
1.4.
Research method
•
Literature study
•
Examine which switching topology and which components to select for the
optimum solution.
•
Estimate the amounts of power dissipated, simulate if possible to confirm
•
Estimate cooling needs and its size/weight
•
Design and simulate filters to follow standard regulations
•
Evaluate the operating area of the components and which counter measures to
take to guarantee reliability and functionality.
•
Form engineering rules to model scalability, power density and efficiency
1.5.
Delimitations
This master thesis will not cover the physical construction or commissioning of the
converter. The programming required in the controller will not be carried out. Some very
modern applications will be neglected because it is too time consuming to estimate the
gain over conventional solutions. Modeling of reliability will not be carried out very
detailed
1.6.
Structure of report
This report will be divided into seven chapters and below is a short description of each
chapter.
1. Introduction –
2. Theory – A general introduction to power inverters is given, how they operate,
difficulties etc. Also a light discussion and explanation of the concepts and behavior
of the different components and parts building up the converter is carried out as well
as some general calculations in aspect of cooling and weight of an enclosure.
3. Losses – A more specified estimation of the losses of the different components is
carried out accompanied by simulations to hopefully confirm these estimations and
2
illustrate their effectiveness. Loss in filter will be estimated and suggestions for
improvement will be carried out with respect to setbacks. Different snubber circuits
will be evaluated with their pros and cons.
4. Results – Results with aspect to losses and its respective cooling are presented for a
few suitable solutions along with their weight, reliability and cost.
A way of scaling is presented with their respective parameters.
5. Conclusions – Presents the conclusions of the work as well as gives answers to the
challenges mentioned concerning the converter. General thoughts on outlooks of
these applications are presented with regards to ongoing research in this field.
Also an evaluation of how the work has proceeded is included, which
problems have risen and how they were handled.
6. Future work – Discusses improvements and work to be carried out to further
optimize the converter. Which parts have been neglected etc?
7. Appendix – Various plots, spreadsheets and graphs from simulations as well as
schematics from simulations.
3
4
2. Theory
2.1.
•
Converter principles
Overview
Figure 1 : Inverter overview
1. Rectifier
In the first step of the inverter the rectifier rectifies the tri-phase figure 2:
Figure 2 : Rectifier stage
2. Smoothing stage
The oscillating rectified DC-bus voltage will cause problems further on and has do be
dealt with, this is done with a LC-filter (inductor-capacitor) which is low pass filtering
the voltage but also providing an extra current buffer. As explained in 2.3.3. the
oscillations can be as much as 9% of the DC-bus voltage and has to be decreased
otherwise the noise would spread further and cause loss and misbehavior. A change in
frequency or controlling the average DC-bus voltage with thyristors would alter the need
for a filter. A higher frequency would make the ripple more frequent requiring a smaller
capacitor but if the voltage is controlled a larger ripple would occur requiring larger
capacitors, more about this in 2.4.1. After the DC-bus filters the voltage will look like it
does in figure 3, with less ripple:
Figure 3 : Smoothing stage
5
If the DC-input would be used it would connected after the bypassed rectifier but still use
the DC-filter to relieve the source in the same manner as the VAC input filter covered in
chapter 2.4.1.
3. Switching stage
Now as the transistor bridge has a much more stable voltage to process it can start
chopping the voltage and forming the pulse width modulated sinus. The way of chopping
and switching is decided by a controller which gives signals to specified drives for the
switching transistors forming the pulse width modulated sinus wave. The transistor
bridge output wave can look as described in figure 4 where the voltage is measured
between one phase and another:
Figure 4 : Switching stage
4. Filtering and output stage
This rough sinus wave formed by the transistor bridge would inflict noise and EMI both
in wires and in the load due to its step nature and has to be low pass filtered at least just
before the load but rather directly after the transistor bridge. Having long wires with large
voltage spikes would create high magnitude radio frequency disturbance which is very
unwanted, especially in an aircraft. This filtration can be done using a three phase LCfilter but also in some cases the inductive coils of the motor can be used as a low pass
filter if the converter is close enough to the load.
In the picture 5 the pulse train has been filtered with a LC-filter. As can be seen it
resembles a sinus wave but with higher frequent noise. To remove this relative small
noise additional high power coils and capacitors has to be included further increasing
weight and size of the unit, again a tradeoff has to be made.
6
Figure 5 : Filtering of pulse train in output stage
2.2.
Principles for Control
2.2.1. Switching frequency
Calculating and choosing a proper switching frequency of a PWM inverter is not always
trivial, there as various advantages and disadvantages of choosing a too high or a too low
switching frequency.
f
(2.2.1.1)
Low mf ( m f = sw ≤ 21 ) [12].
f out
Where fsw is the switching frequency at which the transistors operate and fout is the
intended maximum frequency at which the load operates, fout is thereby a low-pass
filtered version of fsw. The major setback of a too low mf and too low switching
frequency is that the voltage-ripple on the sine-wave otherwise becomes prominent which
causes inefficient ripple within the load, as seen in figure 5. A synchronous PWM is
strongly advised since synchronizing the PWM means that the present output voltage is
fed back to the controller via a voltage-divider and then errors are compensated for,
greatly reducing the voltage-ripple and thus improving the efficiency. Synchronizing the
circuit requires mf to be an odd integer as well as more competent circuitry, including an
A/D-converter and perhaps hardware-support for arithmetic operations, however most
modern circuitry includes this. These voltage ripples or harmonics mainly cause loss in
effect but also can become a burden to people and mechanical devices since equipment
and especially motors can start to vibrate significantly but also generate other non
desirable side-effects. These vibrations can cause reduced lifetime in equipment due to
stress but also emit harmful and annoying sound in often heard frequencies. However if a
highly competent filter on the output is implemented many of the setbacks of a low mf
can be avoided, the current ripple in the load can be reduced heavily as well as reducing
the amount of emitted electromagnetic waves. The setback here is the additional weight
and possibly significant additional loss
f
High mf ( m f = sw ≥ 21 ) [12].
(2.2.1.2)
f out
When the switching frequency increases the need for a synchronized PWM decreases
since the amplitudes of the harmonics are very small and high frequent and thus
somewhat insignificant. Although when controlling an AC-motor with variable frequency
7
the currents at very low output frequencies can become prominent and undesired, despite
the very low amplitudes. As a result, synchronous PWM is almost always advised.
However, even if the overall negative effect of harmonics decreases with high switching
frequency the switching loss in the switching device increases in proportion to the
increased frequency which always is unwanted.
2.3.
Power components
In this part mostly the semi conductive components responsible for the main part of the
loss are considered, those are the main supply rectifier and the transistor-bridge.
Mainly two types of transistors have been considered, the IGBT and the MOSFET and
their advantages and disadvantages are explained further on. A third option exists, the
bipolar junction transistor (BJT) but as this one is current controlled it would require a
driver capable of providing a high current while switching very fast. A combination
which is very difficult to realize.
2.3.1. Switching transistors
2.3.1.1. IGBT
The Insulated Gated Bipolar Transistor is a device that combines the low forward
conduction loss, especially at high voltages, of a Bipolar Junction Transistor (BJT) and
the short switching times of the Metal Oxide Semiconductor Field Effect Transistor
(MOSFET). MOSFET on its own has very high conduction loss at high voltages while
BJT turns off and on much slower [12]. The two figures 6 and 7 illustrate a simplified
model of an IGBT and the symbol layout:
Figure 6 : Simplified model
8
Figure 7 : Symbol layout
The upper figure consist of a Darlington coupled pnp-doped BJT and an n-channel
MOSFET with a resistor that corresponds to the drain drift region in the IGBT.
It is also common to see an npn-doped transistor between the base if the BJT and the
emitter, this is to reduce the turn of tail illustrated in figure 8:
Figure 8: Turn-off behavior
As visible in figure 8, apart from this turn off tail the switching characteristics of an
IGBT resembles the ones of a MOSFET expect for the MOSFET usually being a lot
faster.
To design and dimension the drive stage knowledge of the internal stray capacitances are
required to reach a sufficient turn on and turn off time without a too large over-shoot and
to maintain within the safe operating area (SOA). Although the stray capacitances vary
with the voltage supplied over them making it a bit harder but there are usually provided
some graphs displaying this in the datasheet. To understand the switching characteristics
a more detailed explanation has to be done, however the IGBT-symbol can be simplified
for switching evaluation as described in figure 9:
9
Figure 9: Internal stray capacitances in an IGBT
• IGBT Turn-On
When a positive voltage is applied on the gate (VGE) a current (IG) will start to flow into
the gate through the gate resistor RG charging the Cge capacitor and the voltage rises
exponentially over the capacitor until reaching VGE(th). The Miller effect capacitance
(Cgc)at this point does not contribute much. Beyond this point the collector current (IC)
starts to increase quickly and linearly to an over-shoot level depending on the
semiconductor structure and the external circuit. The gate current decreases to a level
where it stabilizes as the VGE reaches the Miller plateau since the Cgc now gets charged
instead of Cge due to the low voltage at the collector. Since the voltage on the collector is
decreasing the voltage on the gate remains rather constant when charging the Cgc but
increases again after the VCE reaches the VCE(sat). To finally stop at the maximum VGE
when both the gate capacitors are fully charged. The speed of the whole Turn-On process
is directly linked to the gate resistor Rg, a smaller resistor speeds up the process while
causing excessive oscillations or voltage spikes in the circuit. If a snubber circuit is used
it can help to resize the components through filtering out unwanted parts of the signal and
then be able to reducing the gate resistor and make the switching process faster. Although
minimizing stray inductances in wiring and coils is the most effective way of reducing
noise without particular setbacks except practical. On the other hand a larger resistor
slows down the circuit but causes much less noise and voltage transients [9].
10
Figure 10 : Typcial IGBT Turn-on
The dissipated energy during each Turn-On can be calculated from the triangle infolded
by the collector current IC and the collector-emitter voltage VCE times the time period.
• IGBT Turn-Off
During Turn-Off the gate voltage turn to zero and current start to flow from the gate
through the gate resistor, discharging both the gate capacitances, Cge and Cce until the
Miller plateau is reached. Changing the gate resistor does not change the time of the
process like it did for the Turn-On except in a pure MOSFET where it is possible to
decrease Turn-Off time by reducing this resistor [9]. Then the collector-emitter voltage
(VCE) starts to increase until reaching the DC-bus voltage. The gate-emitter voltage (VGE)
continues to decrease until passing the threshold voltage (VGE(th)) and turning the IGBT
off. Due to the bipolar part of the IGBT a current tail will arise as shown in the figure
below, inflicting additional power loss. The current tail is highly unwanted but is very
hard (impossible today), to eliminate completely.
11
Figure 11 : Typical IGBT Turn-Off
The losses are calculated in the same manner as during Turn-On, the triangular area
infolded by the collector current and the collector-emitter voltage. In addition there is the
current-tail area multiplied by the collector-emitter voltage. In datasheets the dissipated
energy due to the current-tail is often included in the total Turn-Off energy.
2.3.1.2. MOSFET
A power Metal Oxide Semiconductor Field Effect Transistor (MOSFET) is just like a
regular small signal MOSFET but larger in every sense. Larger current and higher
voltages causes the internal capacitances and other critical parameters to suffer increases
both switching and conduction losses. Even though slower than a signal MOSFET it is
definitively faster than any IGBT. The appearance of the Turn-On and Turn-Off graphs
for the MOSFET is very similar to the ones of the IGBT except for a much faster process
and there are no current tails during Turn-Off. In addition, altering the gate resistor can
reduce both Turn-On and Turn-Off time unlike the case with the IGBT where the latter
was somewhat unchangeable. Even if the MOSFET is much faster it suffers large losses
during forward conduction, at least when operating in high voltage applications, this
partly due to the internal resistance growing exponentially with the rated VDS as
described by 2.3.1.1.
α
R DS ( on ) = R0V DS
(2.3.1.1)
Where α ≈ 1.6 , VDS is the maximum rated voltage and R0 the initial resistance [9].
This resistance along with the current forms up the voltage drop over the junction as in
2.3.1.2.
α
V DS ( sat ) = I D R DS ( on ) = I D R0V DS
12
(2.3.1.2)
As seen the forward voltage drop increases very quickly with increasing current and
especially the VDS voltage. The consequence of this is the inability to reach high
efficiency while operating under high voltages and large currents since the resistance
does not become small enough to compensate for the vast I D2 term. This loss is the
dissipated power as shown as in equation 2.3.1.3.
Pcond = I D2 RDS ( on )
(2.3.1.3)
2.3.1.3. BJT
The power Bipolar Junction Transistor (BJT) is one of the components forming the
IGBT. The benefits of using a BJT are its capability of handling high currents and high
forward voltages even if the reverse voltage capabilities are limited. The forward
saturation voltage is almost independent on the current which keeps the conduction loss
at a low level. In opposite to the MOSFET and the IGBT the BJT is a current controlled
device and high power devices usually have a very low HFE1, usually a value around 10
for a 10kW application.
This demands very high currents from the driver to saturate the BJT as an unsaturated
device will result in an unwanted high power dissipation most likely to cause failure in
the device. The high base current along with a switching speed near the one of the IGBT
makes the BJT an unsuitable device for this application.
2.3.2. Free-wheeling-diodes
A Free Wheeling Diode is an electronic component used to avoid damage to switching
transistors by reversing load current induction. When switching off an inductive load, the
current cannot go to zero in zero time since there is some energy stored in the magnetic
field. The coil produces a high voltage large enough to let the current continue to flow
over the contact gap, possibly causing permanent damage to the transistor as well as
radiating radio waves. The free wheeling diode is connected anti-parallel with the
transistor and by doing so it doesn't conduct normally as illustrated in figure 12:
Figure 12: IGBT with FWD
If the coil is switched off, the voltage across the coil reverses to maintain the direction of
the current. Now the diode carries the current until the energy is consumed by the inner
resistance of the coil and the forward voltage drop of the diode. This dissipated energy in
the diode is depending on the forward voltage as well as the switching characteristics of
the diode. Because of this, low forward voltage and small stray capacitances are wanted
1
Forward current gain
13
as well as low reverse recovery time which are characteristics that usually contradict each
other. The reverse recovery time is the time taken from forward conduction to blocking in
the reverse direction, this time directly causes loss on the circuit.
2.3.2.1. Silicon Schottky
Usually Schottky diodes are used which have very low reverse recovery time, slightly
lower forward voltage drop and being much faster (much lower stray capacitance)
compared to conventional diodes, although they have low maximum reverse voltage and
a relatively high reverse leakage current that also increases with increasing temperature
which makes them a bad choice in high voltage and high temperature applications.
2.3.2.2. Silicon Carbide Schottky
Since some ten years back other interesting materials are being researched. Diodes made
of Silicon Carbide have proven to have excellent characteristics for high voltage, high
frequency and high temperature. The reverse leaking current is up to 40 times less than
for a regular Shottky, directly reducing losses, reverse voltage up to 1200 V and
extremely low reverse charge as a result of junction capacitance, not stored charge. The
setback is high price and a relatively high saturation voltage, introducing increased loss
when conducting. By having high thermal conductivity and nearly no thermal runaway
also makes the Silicon Carbide the best choice in applications with high temperature.
With special packing junction operating temperatures as high as 500 °K (227 °C) is made
possible which opens up for a wide range of applications. The reverse recovery loss is
usually a significant part of the total switching loss in a hard switched2 IGBT and by
almost reducing it to zero great reductions in dissipated effect and heat can be made.
2.3.3. Rectifier
The rectifier forms a direct current from an alternating current, in this application from
three phase shifted sources of alternating current. In this case a 6-pulse rectifier model
has been chosen due to its simplicity. However, when handling disturbances on the main
grid a 12-pulse rectifier bridge is to prefer since it heavily reduces harmonics which
otherwise will require large filters. A simulation has been made using Matlab with
Simulink where a 6-pulse rectifier is used and the resulting voltage frequency spectrum
on the grid is measured, see Appendix C1 and C2. Analogous the same measures are
done using a 12-pulse rectifier where it is observed that the noise due to harmonics is
significantly lower, see Appendix C4 and C5. The setback of the 12-pulse bridge is the
needed high power transformer and an additional 6- pulse bridge, together largely
contributing to additional weight. The mean voltage archived on the DC side is calculated
as a combination of all the input voltages as seen in equation 2.3.3.1 [12]:
V DC _ AVE =
2
3 × 2 × VL− L
π
=
3 × 2 × 400
π
≈540 V
Switching with no snubber circuits or filters and an inductive load
14
(2.3.3.1)
This is an average voltage and it will actually oscillate between 490 and 566 volts at the
input frequency multiplied by six demanding a filter to provide a fixed voltage. For a
10kW application the maximum DC-bus current will be as large as 18.5A with
reservation for the result of the DC-bus filter temporarily capable of supplying more than
18.5A The choice of a rectifier bridge is mainly focused on low energy loss and weight
where low energy loss needs less cooling and therefore less weight although in some
applications disturbances and harmonics will cause problems and has to be given higher
priority. However the weight and size of the rectifier itself has in some cases proven to
differ a lot. The ability to withstand heat and to remove this heat should also be taken into
consideration, represented by the thermal resistance of the package. The layout of the
rectifier diodes can be found in figure 13 (encircled):
Figure 13: Overveiw of the rectifier
15
2.4.
Filters
2.4.1. Input filters
The non-linear nature of the converter with especially its rectifier and the inductive load
will form a load on the main grid that is far from ideal3. Noise generated back on the grid
is a problem for all the other connected equipment but will also emit RMI if not protected
either by shielded wires or an input filter that compensating for this behavior. Usually it
exist a lot of regulations concerning how inductive a load can be and how much noise a
load can inflict, especially in scenarios where the main supply is weak4 and where it
drives sensitive electronic equipment, such as in an aircraft or other vehicles.
In the scenario covered by this master-thesis a 400 Hz supply is given which implies that
it should not contain high magnitude components at other frequencies. Because of this a
band-pass filter has to be implemented at the input to reduce the magnitude of unwanted
components reflected back to the supply. The components are the fundamental frequency
component and its harmonics of order 6 k±15. This periodic order due to the switched
operation of the line commutated rectifier, in this case a three-phase diode bridge. The
currents harmonics can further be resolved into sequences according to the following
table:
Sequence
Positive
Negative
Harmonics
1,7,13,19,…
5,11,17,23,…
Table 1 : Input current harmonics
Usually it is trivial to filter out these harmonics but the high power application along
with the low weight goal makes it difficult to remain within the SOA of all the
components.
In figure 14 is the schematic of a simple input filter that attenuates the first two harmonic
components of the current as well as a wide range of high frequency noise. More filters
can be added to filter out additional harmonics but will add weight and volume to the
device. How much attenuation of respective harmonics is demanded is usually set by the
environment. In an aircraft there is usually very sensitive equipment on a weak power
source increasing the needs for good filtering. Most of the regulations used and followed
by SAAB AB are found in MIL-STD-704 produced by the US department of defense
[19]. Too some extent high frequency components emitting radio waves can be contained
using shielded cables.
3
Ideal is a purely sinusoidal current
The source is considered weak if the voltage is reduced significantly when loaded
5
K is any positive integer, which means the orders can be as in table 1
4
16
Figure 14 : Input filter
As it is hard to analytically calculate the behavior of currents and voltages in the filter
components an iterative approach using extensive simulations is usually taken. This is
due to the analytical result being valid only during the steady state and not during start up
or shutdown where surge currents will occur. These surge currents can cause failure or
reduce lifetime in the filter components and especially in the electrolyte capacitor if not
dimensioned properly.
Figure 15 : Typical impedance for input filter vs. harmonics of grid frequency
The amounts of harmonics reflected back from the inverter to the grid is also dependant
on the LC-filter between the rectifier and the switching bridge. This because currents
through the diodes is dependant on the charging currents of the capacitor which itself is
dependant on the choke inductor and the surge limiter. These harmonics are also hard to
analytically estimate and a system level simulation is favored. To estimate the harmonics
in this converter Matlab with Simulink will be used where the entire converter is modeled
and the frequency spectrum on the input is measured. When having the high amplitude
harmonics determined a band-stop filter for each undesired harmonic has to be
implemented with appropriate attenuations. This frequency spectrum measurement can be
seen in Appendix C1 where the harmonics can easily be identified. The maximum
allowed disturbance can be found in the MIL-STD-704D military standard document
17
[19]. In table 2 the larger harmonics in comparison with the maximum allowed levels and
the required attenuation can be viewed:
Harmonic #
fundamental
5
7
11
13
17
Frequency(hz)
400
2000
2800
4400
5200
7800
Amplitude peak/RMS V
550/388
48/34
16/11.3
20/14
14/10
8/5.7
Allowed level RMS
20
20
20
16
12
Attenuation dB
2.3
-
Table 2 : Voltage harmonics on grid
As in table 2 the given voltages arise due to harmonic currents through the source usually
represented by a generator the respective current components can be filtered instead. This
is carried out as explained in figure 14 with serial RLC circuit connected to next phase or
virtual ground. The impedance of the RLC circuit is usually calculated as in 2.4.1.1:
1 ⎞
⎛
RLC Z (ω ) = R 2 + ⎜ ωL −
⎟
ωC ⎠
⎝
2
(2.4.1.1)
The RLC impedance is at minimum at the resonance frequency ω = 1 LC where it
assumes the value R. This value R limits the current at the filtered frequency preventing it
to become unnecessary large. The L and C values can be chosen somewhat arbitrary as
long as the resonance frequency is the correct one. The current harmonics can be found in
appendix C along with the voltage harmonics.
As seen in table 2 the voltage at the 5th harmonic caused by its respective harmonic
current has to be reduced to 59% relying on the filter to consume this excessive current.
It is also the only component needed to be individually reduced, the higher frequency
components are attenuated with a high-pass filter as illustrated in figure 14. Given the
RMS voltage 34 V and the RMS current 2.52 A at the frequency 2000 Hz of which the
filter will consume 41 % gives the value of R according to 2.4.1.2:
R=
Vharmonics _ rms
I harmonic _ rms × 0.41
=
34
= 32.9Ω
2.52 × 0.41
(2.4.1.2)
Assuming the value 100µH for the inductor L gives along the resonance frequency 2000
Hz the capacitor C a value of 63 µF. The estimated voltage and current give a power loss
of 34W. Although this is per phase and should be multiplied by three when added to the
complete system. This has to be done with the components as well, adding weight and
space. The combined weight of the 5th harmonic filter per phase is estimated to 120g
which gives a total of 360g. The dimension can be seen in the layout sketch, in Appendix
F
18
The high-pass filters are calculated in the same manner but when calculating loss a few
more harmonic components are added, although smaller in magnitude. A 10Ω with a
100nF capacitor in series nearly does not interfere with the 5th harmonic filter but filter
higher order harmonics rather effectively. The accumulated power loss is estimated to be
about 3 watts per phase which will all be dissipated in the resistor. The estimated weight
per phase is 11 g, giving a total of 33g.
Component
Filter
R
R
5th
harmonic
5th
harmonic
5th
harmonic
High-pass
C
High-pass
L
C
Ratings Manufacturer
and Model
33Ω
ARCOL HS
Mass
(g)
3*31
Power
Volume
3
loss (W)
(mm )
3*(29x70x15) 3*34
100μH
3*19
3*(15x25x24)
3*70
3*(35x51)
3*5
3*(8x8x22)
3*7
3*(11x22x26)
70μF
10Ω
100nF
Pulse
L100
EVOX RIFA
PEH200
VITROHM
KH
EVOX RIFA
PHE845
3*3
Table 3 : Input filter components
2.4.2. DC-bus filter
2.4.2.1. Capacitor bank
Because of the nature of the rectifier the voltage on the DC-bus will vary with time, but
not only because of this but from the pulse shape of the output current as well. Due to this
a LC-filter is needed to stabilize the voltage as well as providing a current buffer. The
performance of the filter is limited by the weight and cost always wanted to be kept low.
As electrolyte capacitors are used the lifetime is limited and estimated using a simple
model to assure lifetime long enough according to regulations.
The dimensioning of the components requires some extensive calculations which in detail
will be explained in this chapter starting with reviewing the provided parameters.
•
•
•
•
•
Main power grid: 400VAC, 400Hz three-phase, 560VDC maximum
Capacity: 10kW with 10kHz switching frequency
Full wave bridge rectifier: 400*6 Hz ripple frequency
Maximum allowed ripple voltage: 3% of average DC-bus voltage i.e. 16 V.
Common MTBF: 5000 hours of flight/operation
Assuming all the energy is stored in the capacitor bank a calculation gives according to
the well known formula of the potential energy in a capacitor
1
Ecap = Ctot × V 2
2
(2.4.2.1.1)
19
and its extension to fit the actual case.
(
1
2
Pout × tripple = Ctot × VDC _ MAX − (VDC _ MAX − Vripple ) 2
2
)
(2.4.2.1.2)
Where Pout is the rated power of the inverter, VDC_MAX is the maximum voltage of the
rectified output, Vripple is the maximum ripple allowed and tripple is the period time of the
ripple. The ripple frequency for a full wave bridge is the main grid frequency multiplied
by six. Solving for Ctot in (2.4.2.1.3) with insertion of the proper values gives:
10000 ×
(
)
1
1
= C tot × 566 2 − (566 − 16) 2 → C tot = 4,67 × 10 − 4 = 467 μF (2.4.2.1.3)
2400 2
As very few capacitors exist capable of handling this high voltage and still maintain a
large capacitance multiple capacitors has to be connected in series. Although when serial
connected a resistor has to be in parallel with each capacitor as explained more in detail
later in this chapter. A common way is to use two legs in parallel with two identical serial
connected capacitors maintaining the same capacitance as a single one while doubling the
maximum voltage. This precaution due to the voltage peaks during start-up when the
voltage over the capacitor can rise to a level between 1.4 and 1.8 times normal depending
on the size of the choke inductor and the protective circuits. The surge current appear
because during start-up the capacitor bank is virtually short circuited and the choke
inductor will try to maintain this current, thus inflicting a high voltage transient. The
surge current and voltage over-shoot can be reduced or almost eliminated with these
protective circuits as illustrated in figure 16. The switch in this case is a high power
transistor, most likely an IGBT with a low saturation voltage. As the resistor will initially
limit the charge time of the capacitor it will also decrease the over-shoot. The surge
limiter can be expanded into several levels reducing the resistance sequentially although
many copies of this circuit in series will accumulate a large on-resistance in steady state
and therefore add a non negligible loss. However, if connected in parallel they can
replace each other instead and will end up with only one transistor in series causing a
smaller on-resistance and a lower power loss. When this rather fast switching behavior
has occurred it can be relieved by a relay, these are usually much slower but have a much
smaller on-resistance than any semiconductor almost eliminating the added loss. A bonus
of this circuit is that it limits the converter inrush current significantly from a level that
would probably destroy the rectifier to a reasonable current. Simulations of this behavior
with and without the surge limiter can be seen in appendix A which is based on a system
simulation done in Matlab and Simulink, the circuitry can be found in appendix B.
20
Figure 16 : Protective circuits: simple, series and parallel connection
Not only does the capacitor require a wide margin for voltage but also for the capacitance
where tolerance (±20%) and wear-out (-10% can reduce the capacitance with up to 28%
(1-0,8*0,9).
Taking these precautions suggests the Evox Rifa PEH200 electrolyte capacitor with the
ratings 400V and 680 µF with an individual weight of 180g. The four combined
capacitors get a rating of 800V maximum and still has the same capacitance as a single
one but of course weights four times as much i.e. 720g.
A doubling in the rated power of the inverter would result in at least doubling the
capacitance required suggesting the 1500 µF version which would weight 430g each
increasing the weight of the capacitor bank by at least 140%.
However, the next step would be to calculate the ripple currents from the AC-line and
from to the load by first calculating the capacitor charging time with 2.4.2.1.4 [15]:
⎛ V DC _ MAX − Vripple
arccos⎜
⎜
V DC _ MAX
⎝
TC =
2 × π × f grid
⎞
⎟
⎟
⎠
⎛ 566 − 16 ⎞
arccos⎜
⎟
⎝ 566 ⎠ =94.8 μs
=
2 × π × 400
The capacitor charge time can be illustrated and defined as in figure 17:
21
(2.4.2.1.4)
Figure 17 : Charge and Discharge time of capacitor bank
With the charge time and the period time of the ripple voltage it is easy to derive the
capacitor discharge time as in equation 2.4.2.1.5.
TDC = Tripple − TC =
1
− 0,0000948 =321.8 μs
2400
(2.4.2.1.5)
Based on the change in voltage vs. time (dV/dt), the peak and RMS charge current (the
rectifier output current) through each capacitor leg can now be calculated as in equation
2.4.2.1.6 and 2.4.2.1.7.
I Cpeak = C ×
dV
= 39.4 A
dTC
(2.4.2.1.6)
Where C is the capacitance in each leg, which is
Ctot
due to the serial connection.
2
2
I Crms = I Cpeak × TC × f ripple = 18.8 A
(2.4.2.1.7)
In the same manner the peak and RMS discharge currents can be calculated using the
discharge time instead as in equation 2.4.2.1.8 and 2.4.2.1.9:
dV
I DCpeak = C ×
= 11.6 A
(2.4.2.1.8)
dTDC
2
I DCrms = I DCpeak × TDC × f ripple = 10.2 A
(2.4.2.1.9)
With calculations done in equation 14 and 15 the ripple current resulting from the
rectification of the grid can now be calculated for each branch of the capacitor bank as in
equation 2.4.2.1.10.
2
2
I rms = I Crms
+ I DCrms
= 21.4 A
(2.4.2.1.10)
Thus, through the whole capacitor bank it flows twice as much since it has two legs
adding up to 42.8 A.
22
The load ripple current for the whole capacitor bank is calculated as in equation
2.4.2.1.11.
Prated
10000
I load _ rms =
= 17.92 A (2.4.2.1.11)
=
⎛ VDX _ MAX + (V DC _ MAX − Vripple ) ⎞ ⎛ 566 + 550 ⎞
⎟
⎟⎟ ⎜
⎜⎜
2
⎠
2
⎠ ⎝
⎝
Which gives a load current of 8.96 A per leg. The total makeup of the current through the
capacitor bank is 17.92 A @ 10 kHz and 42.8 A @ 2400 Hz.
The next thing is to determine the power loss achieved in the capacitor bank and whether
it is to large enough to require extensive cooling or a large choke inductor to maintain
sufficient lifetime. The power loss can be calculated by somewhat altering the well
known formula P = U x I as done in equation 2.4.2.1.12.
2
Ploss = U × I = R × I 2 = I rms × ERS
(2.4.2.1.12)
Where the ERS is the equivalent series resistance which can be found in diagrams within
the datasheet describing the capacitor. This ESR is strongly dependant on the frequency
and the hotspot6 temperature (Tht) which is typically 20º C warmer than ambient
temperature(Ta) of 70º C [15].
These expected typical values are found to be:
ESR(2400Hz) = 13.5 mΩ
ESR(10kHz) = 12.5 mΩ
Using the estimated ESR values and the calculated current at each frequency two
components of power loss can be estimated for each capacitor, trivially four times this
loss would give the total loss of the capacitor bank excluding the choke inductor.
Ploss (2400 Hz ) = 21.4 2 × 0.0135 = 6.18 W
Ploss (10000 Hz ) = 8.96 2 × 0.0125 = 1 W
(2.4.2.1.13)
This gives each capacitor a loss of about 7 W which is dissipated through heat. Whether
the capacitor can withstand this produced heat is dependant on the thermal resistance
between hotspot and the ambient air as well as the temperature of this ambient air.
A too high temperature at the hotspot will result in a reduced lifetime and this
temperature can be calculated with equation 2.4.2.1.14.
Ploss =
Tht − Ta
↔ Tht = Ploss × Rht _ a + Ta = 7.18 × 3.4 + 70 = 94.4°C
Rht _ a
(2.4.2.1.14)
The thermal resistance 3.4ºC/W between the hotspot and the ambience (Rht_a) can be
found in the datasheets under a variety of conditions, this one is without any heatsink or
6
The center of the capacitor where the highest temperature is likely to arise.
23
airflow. As it can be seen the temperature at the hotspot is very close to where the ESR
was specified negating the need for further more detailed calculations concerning the
loss. However, if this increase in temperature would affect the lifetime too much the
choke inductor can be designed to extensively filter the high frequency components.
Usually it is just meant for suppressing the high frequent ripple.
Suppose the temperature would be too high and the power loss needed to be reduced.
To keep the hotspot temperature at 90 º C 1.3 W is needed to be transferred to the choke
inductor as explained in chapter 2.4.2.2.
However, for the intended voltage sharing of the two serial capacitors to work correctly a
resistor has to be connected in parallel with each capacitor. The resistance of this resistor
is dependent on the capacitor value according to equation 2.4.2.1.15 [15]:
Rvsr =
1000
=143 kΩ
0.015 × C [μF ]
(2.4.2.1.15)
As a precaution the power rating of the resistor should be at least 50% higher than
estimated and the tolerance should be kept below 5% to prevent failure. The electrical
layout is illustrated as in figure 18, the physical layout on the other hand should not be
taken to easily as the inductance has to be kept low.
Figure 18 : Electrical layout of capacitor bank
According to the equation for lifetime the capacitor will live for at least 30 kH which is
far beyond the limit of 5 kH. The uncertainty is due to few values of expected rate of
failure where an average had to be taken. Although in the worst case the lifetime is still
enough by far.
LOP = A × 2
(85−Tht )
C
= 6 × 10 × 2
4
(85−94 )
12
≈36000 h
(2.4.2.1.16)
Where A is the lifetime at reference temperature and C is the rise in degrees required for
cutting the lifetime in half, both of these available from the data sheet.
24
However, this lifetime can still prove uncertain in avionic applications as the thermal
resistance due to convection will increase with altitude resulting in a higher hotspot
temperature. The air pressure will also affect the lifetime of the capacitor as the
electrolyte degrade faster.
2.4.2.2. Choke inductor
The choke inductor is mainly for reducing the transients capable of causing fatal failure
in the capacitor bank. However it can be designed to filter some of the high frequency
components of the charge current resulting in transferring the loss from the capacitor to
the inductor
Suppose the 94.4 ºC is too high and need to be reduced to 90 ºC, based on previous
calculations a reduction of 1.29 W is needed in each capacitor. This combined loss
reduction achieved in the four capacitors will be withdrawn from the inbound charge
current. A rough estimation for the needed reduction in voltage can be done by using the
maximum ripple voltage over the capacitors and the reduction in power as explained by
equation 2.4.2.2.1.
P
1.29
Vchoke = Vripple reduction = 16 ×
= 3.34V pk − pk
(2.4.2.2.1)
Pinitial
6.18
Vchoke _ rms =
V pk − pk
2.11
= 1.58Vrms
(2.4.2.2.2)
The 2.11 factor in (is different than the normal 2 × 2 since the voltage is not pure
sinusoidal to its form. The remaining fundamental ripple current can be estimated using
the new power loss and the ESR of the capacitors as in equation 2.4.2.2.3.
I ripple =
Ploss _ branch
ESRbranch
× nbranches =
2 × (6.18 − 1.29 )
× 2 =38.1 A
2 × 0.0135
(2.4.2.2.3)
This ripple current is a small reduction from the former value of 39.4 A and now the
proper inductance can be calculated as in equation 2.4.2.2.4:
X L = 2 × π × f ripple × L =
U
U
1.58
→L=
=
=2.75 μH
I
2 × π × f ripple × I 2 × π × 2400× 38.1
(2.4.2.2.4)
The power capabilities of the choke inductor are usually done with some empirically
established rules actually meant for transformers. However the figures given above are
enough to choose an inductor from an arbitrary manufacturer. Consider the DC3-56G
from WILCO, it weights approximately 30g each and measures 28 x 21.3 mm, to cover
the current two are used in parallel, hence the inductance of 5.6 µH.
Component
Ratings
Manufacturer
and Model
25
Mass
(g)
Volume
(mm3)
Power
loss (W)
L
5.6μH
C
680μF
WILCO
DC3-56G
EVOX RIFA
PEH200
2*30
3*(28x22)
4*1.3
4*180 3*(50x75)
4* 5.9
Table 4 : DC-bus filter components
2.4.2.3. Output filters
If there are long wires between the output of the inverter and the inductive load the stray
impedance (usually mostly capacitive) can induce annoying behavior both in the motor
and the driving transistors. The long wires would also serve as a good antennae emitting
EMI caused by the transistor transients when switching fast. Although, the problem with
emitted EMI can be treated with proper shielding of the conducting wire. In the following
chapter advantages/disadvantages and guidelines for an approximate design of an output
filter is carried out under the assumption that the cable is no longer than 30 meters. Near
and above this limit several phenomenons occur such as standing waves due to
impedance mismatch between the motor and the cable. Such standing waves can cause
serious over voltages and current oscillations too strong to be managed by a LC-filter.
Many motor manufacturers have recently published maximum dV/dt ratings for their
product, usually around 5 V/ns while the commutation of state of the art transistor can be
far above this rating, up to 10 V/ns or even more. Here a mean to reduce dV/dt at the
motor terminals are required and such a mean is often a simple LC-filter. It has also been
proven that placing the filter near the motor terminals can further improve this
misbehavior compared to placing it on the inverters board. However, as the task at hand
is to evaluate a complete inverter solution the choice is to have the filter within the
inverter unit, this filter will also be designed without any snubber circuits implemented.
In figure 20 is an illustration presented where a LC-filter has been added between the
output of the transistor-bridge and the motor, although only one phase is represented. The
motor has been modeled as a large inductor L2 corresponding to the phase/neutral
winding.
Figure 19 : Output filter
26
Where V1 represents the DC-bus voltage and the two capacitors represents the CollectorEmitter-capacitor (Cce) which also plays an important role during commutation.
Obviously the design of the filter is very simple and by assuming Ipeak is the peak motor
current, C1 can be chosen as in equation 2.4.2.3.1.
C1 =
I peak
(2.4.2.3.1)
dV / dt max
C1 softens the switching from the IGBTs so that the maximum dV/dt at the motor
terminals is kept within limits supplied by the manufacturer or the EMI specifications are
met. However, C1 cannot be left alone as there would run very large current peaks
through the IGBT leg during turn most likely to trigger the over-current protection
handled by the controlling circuit. This is where the inductor L1 comes in to block this
high charge current and the dimensioning of the inductor will be such that current peaks
in the IGBT (or C1) does not trigger the over-current protection, at least. The over-current
protection is dimensioned in advance with respect to the switching devices and their
current handling capabilities. As so far this is a simple LC circuit and it is driven by very
high dV/dt it will soon start go generate oscillations higher than the DC-bus voltage
which is far from what the filter components and the motor are designed for. A damping
resistor in series with the capacitor would suppress this behavior and in addition
removing un-useful power dissipation otherwise dissipated in the capacitor. When this L1
is involved the expression in equation 23 is obsolete due to the resonance between L1 and
C1. Thus the actual dV/dt will also be determined by L1 giving a new expression as
follows by equation 2.4.2.3.2 for the initial dV/dt:
dV
=
dt
V DC
(2.4.2.3.2)
L1 × C1
The value of L1 will be designed taking into account that each IGBT sees the phase
current of the motor, the recovery current of the opposite recovery diode and the peak
current of the filter. Although a fourth component could exist if the commutation causes
cross talk but it is not very common due to very potent controlling circuits and drivers.
After considering over-current limits and the peak recovery current the allowed peak
current contribution of the output filter can be determined. Excluding the damping
resistor R2 the current through the filter can be calculated as in equation 2.4.2.3.3 where
the ZC is the characteristic impedance of the filter as determined in equation 2.4.2.3.4.
I filter _ max =
ZC =
V DC
ZC
(2.4.2.3.3)
L1
C1
(2.4.2.3.4)
27
But as explained earlier the damping cannot be zero and thus the effect of R2 has to be
considered. Critically damping is achieved when R2 is equal to ZC but a lower value
could be chosen due to C8 and C9. Suppose R2 is chosen to ZC, then the current peak in
the IGBTs due to the output filter would be as determined in equation 2.4.2.3.5:
I filter _ max =
V DC
2 × ZC
(2.4.2.3.5)
As current limiting is not the only constraint for L1 it also has to be designed to ensure
enough build-up voltage in the motor’s coils during a very low duty cycle [19]. The
minimum duty cycle time should at least be twice as long as the resonance cycle caused
by the LC circuit. Neglecting the damping caused by R2 and adding some margins instead
gives the proper value of L1 as in equation 2.4.2.3.6 assuming the minimum turn-on time
is known.
π × L1 × C1 ≤ Ton _ min
(2.4.2.3.6)
Combining the 2.4.2.3.2 and 2.4.2.3.6 an expression for the maximum dV/dt related to
VDC and the minimum turn-on time is achieved by equation 2.4.2.3.7.
dV VDC × π
=
dt
Ton _ min
(2.4.2.3.7)
The result achieved in 2.4.2.3.7 seems highly logical as it would require a shorter turn-on
time to get a higher dV/dt. This overall gain in less EMI and enough slow dV/dt for the
motor does not come without loss, both in more weight and in more direct power loss.
The loss is dissipated in the damping resistor and cannot be neglected, in the case of R2
being equal to ZC the power loss can be estimated as in equation 2.4.2.3.8 [19]:
2
Poutput _ filter _ loss =
V DC
× Ton _ min × f sw
4 × R2
(2.4.2.3.8)
Applying the above given theory to the specific case examined in this paper gives some
guidelines for values and physical size of the filter components. As they should be
applied to each phase conductor pf course three of each has to be included.
Assuming as before the maximum dV/dt for the motor is the common 5V/ns and in the
data sheet for the IRG4PSH71UD it can be found that the typical dV/dt is 30 V/ns which
is much too fast. As the peak voltage through the IGBT is 20.5A ( 2 × 14.5 ) the value of
C1 can be calculated using 2.4.2.3.1, which gives the value 4.1nF. According to 2.4.2.3.7
the Ton_min is 350 ns which along with equation 2.4.2.3.6 gives the value for L1, 3μH. As
the characteristic impedance ZC is calculated as in equation 2.4.2.3.4 it receives the value
27Ω which is also the minimum value of R2, actually R2 can be chosen a bit larger to
suppress some overshoot but it should remain between ZC and 2 × Z C . Suppose the
28
maximum current due to opposite diode reverse recovery is 20.5 A and the maximum
current through the filter being according to 2.4.2.3.4 (if R2 = ZC) 10 A. This along with
the well known maximum current through and IGBT gives a maximum current of 51 A,
this is what the over-current protection level should be set to. The power loss mainly
dissipated in the resistor is calculated using equation 33 and is estimated to be 10 W. Yet
again, this is for only one phase and the loss and weight of components has to be tripled
to make up the actual figures. When of-the-shelf products are considered the weight
estimated to be 60g for L1, 6g for C1 and 7g for R2 adding up to73g per phase.
Filter
component
R
L
C
Value
27 Ω
3 μH
4.1 nF
Manufacturer
and model
ARCOL HS
Pulse L100
EVOX RIFA
PHE845
Mass(g)
Dimensions
3*7
3*60
3*5.2
3*(30x10x10) 3*10
3*(50x40x36) 3*(6x14.5x26) -
Table 5 : Output filter components
29
Power loss
2.5.
Snubber circuits
The main cause of implementing snubber circuits is to make sure the switching power
components remain within their safe-operating-area (SOA) both at turn-on and turn-off.
This is very important to ensure the longevity of the devices as well as reducing the
amounts of EMI emitted from the high power components. When introducing a snubber it
is not uncommon to be able to reduce switching losses in the switching devices, in
particular the turn-on switching loss. However, in general the losses removed from the
transistor are transferred to the snubber circuits instead while these components are
usually less sensitive to voltage spikes and are able to perform satisfactory in a wider
temperature span. As there are numerous variants of snubber circuits for different
purposes only the ones estimated to prove suitable will be presented below along with
their corresponding advantages and disadvantages. The simulations carried out are done
using Cadence Pspice Student Edition, the circuits can be viewed in Appendix E.
Although the simulations will not be used very much for estimating good snubber circuits
to obtain reduced power loss as much as ensuring longevity to the switching devices.
2.5.1. Increased SOA
Due to the non avoidable stray inductance in the DC-bus loop in addition to the internal
stray inductances in most of the components causes along with the very fast current
switching of modern semiconductors a large voltage over-shoot when switched off. To
some extent there is an over-shoot during turn-on as well but not nearly as large. The
faster switching and larger stray inductance the larger voltage over-shoot. The easiest and
least complex method to manage this over-shoot is to compensate for the stray inductance
in the DC-bus loop by connecting a small, fast and low inductance capacitor as close to
the IGBT terminals as possible as shown in figure 20 (left). Actually increasing the gate
resistor to slow down the switching process would work as well but switching losses
would rise according to it.
Figure 20 : Decoupling and restricted decoupling capacitor
There are capacitors manufactured for this purpose with very low self inductance and
designated connections to properly fit the concerned IGBT module. However when using
discrete components this gets harder as the components are separated which also causes
higher stray inductances. As the separation is one of the causes of choosing discrete
30
components the inductance is a large trade-off. The difference when using only the
decoupling capacitor can be seen in the figure 23.
Figure 21 : Switching device voltage without snubber circuit
As seen in figure 22 the transient at turn-off is easily large enough to cause failure in the
switching device, it is nearly twice as large as the DC-bus voltage. By attaching the
decoupling capacitor the over-shoot decreases to about 123% of DC-bus voltage but
instead an oscillating voltage will occur over the transistor as seen in figure 23. This
oscillating voltage will inflict a large current through the capacitor which often is
designed for small currents, with low self inductance as a benefit. The capacitor value can
be estimated with equation 2.5.1.1 [20].
2
Ls × I 0
Csn =
(Vpk − VDC )2
(2.5.1.1)
Where I0 is the maximum switched current, Ls being the DC-loop inductance and Vpk the
maximum allowed peak voltage over the switching device. This oscillating behavior is
seen in 23.
31
Figure 22 : Device voltage with decoupling capacitor
This oscillating behavior causing high RMS current through the capacitor is the major
disadvantage if the decoupling capacitor. The more inductive DC-loop and higher output
current the higher RMS current will run through the decoupling capacitor. This makes
this type of snubber circuit suitable for low current applications. The estimated power
loss due to in the capacitor can be estimated with 2.5.1.2 [20].
2
Ploss _ decoupling = ESR × I RMS = f sw × LS × I 0
2
(2.5.1.2)
This would not result in a very high power loss but still it can be too high for these kinds
of capacitors as they often are very delicate. The decoupling capacitor is a good solution
when having a switching frequency of only a few kHz. The loss and oscillating voltage
can be reduced by adding a diode and a resistor to restrict the de-charging and reduce
oscillations. The circuit can be seen in figure 21 (right). Although to get a small enough
over-shoot the capacitance needs to higher as the diode and the resistor introduce further
inductance to the loop. The diode should be of fast and soft recovery type to avoid severe
oscillations following Vpk at turn off. The resistor can be calculated using equation
2.5.1.3.
32
Rsn =
1
(2.5.1.3)
(6 × C sn × f sw )
Figure 23 : Device voltage with discharge restricted decoupling capacitor
However, as this circuit adds more components and therefore more stray inductance a
larger capacitor will be needed, although the power lost will mainly be dissipated in the
resistor and the diode as explained in equation 2.5.1.4.
Ploss _ restriced _ decoupling =
(
)
1
2
2
× C sn × V pk − V DC × f sw
2
(2.5.1.4)
The RCD Clamp-Snubber seen in figure 24 (right) does have a very favorable effect on
both the Turn-On and Turn-Off transients but have limited possibilities to reduce the
switching loss in the transistor. However, the loss dissipated in the snubber circuits is
much less than in the RCD Charge-Discharge Snubber (seen in figure 24, left) which is
calculated as in equation 2.5.1.5 [20].
Ploss _ RCD _ Ch arg e _ Disch arg e =
1
2
× C sn × V pk × f sw
2
(2.5.1.5)
Although the RCD Charge-Discharge Snubber inflicts some power loss it can help to
speed up both the Turn-On and in particular the Turn-Off processes directly reducing
power lost in the IGBT and diode. But it can be difficult finding component capable of
sustaining the internal loss while maintaining a low self inductance [20].
33
Figure 24 : RCD Charge-Discharge Snubber and RCD Clamp-Snubber
In figure 26 and 27 a switch has been simulated and plotted illustrating the difference
between the RCD Charge-Discharge Snubber and the RCD Clamp Snubber. As it can bee
seen they Turn-ON rather similar and cause about the same amount of over-shoot but the
RCD Charge-Discharge Snubber has a much quicker Turn-Off. As the Turn-Off with its
tail is the main part of the switching loss the total loss can be vastly reduced using a RCD
Charge-Discharge Snubber even without reducing the gate resistor. Reducing the resistor
would also reduce the losses but would do so by causing the over-shoot to increase once
again.
Figure 25 : RCD Charge-Discharge Snubber
34
Figure 26 : RCD Clamp-Snubber
2.5.2. Reducing losses
As the snubbers circuits help to attenuate the voltage transients the switching process can
be faster causing reduced switching losses by reducing the gate resistor. However,
precaution has to be taken to ensure the driver can withstand this higher peak current.
Although not actually reducing the losses as much as moving them is done with some
snubber as they consume then unwanted energy and dissipates it within its own
components. If this is a diode or a resistor they can withstand a lot of power loss at a high
temperature implying little need for excessive cooling.
Type
Decoupling
Capacitor
Advantages
Low snubber losses
Directly and favorably effects TurnOff and Turn-On on voltage stress
Special solutions of casings for
modules
Only a good choice in low current regions
Discharge
Low snubber loss
restricted
Directly reduces the Turn-Off voltage
decoupling over-shoots but also has as a favorable
Capacitor
effect on Turn-On voltage transients.
Much quieter switching as diode
blocks of oscillations.
Good in medium current ranges
RCD
Disadvantages
Produces voltage and current
oscillations in the DC-bus,
forcing usage of capacitor with
high RMS current limit
Additional circuitry increase
snubber inductance, making
protection less effective.
Snappy diode could produce high
recovery voltage spikes and
dv/dts across IGBT pair.
Very high snubber losses
35
ChargeReduces Turn-Off voltage over-shoots Requires more components
Discharge
Could substantially reduce Turn-Off
snubber
loss in switching device
circuit
NO oscillations on the DC-bus
Difficult components selection
Good for high current and low DC-bus voltage applications
RCD
Low snubber losses
Clamp
Directly reduces the Turn-Off voltage
Snubber
over-shoots but also has as a favorable
effect on Turn-On voltage transients.
NO oscillations on the DC-bus
Requires more components
Good for medium to high current applications
Table 6 : Snubber characteristics
A quick glance gives the impression that the RCD Charge-Discharge Snubber would be a
good choice for this inverter as the chosen transistor already has a very small gate resistor
as default. In this way loss could be reduced without adventuring a fatal over-shoot.
2.6.
Principles of cooling
2.6.1. Thermal resistance
A common design problem is to remove heat enough to guarantee satisfactory operation.
Then the thermal conductance between to adjacent mediums should be as high as possible
to in the end dump as much heat as possible in a large buffer, usually the surrounding air.
The thermal resistance is as usual the inverse of the conductance and is as a normal
resistance additive when connected in series respectively. An equation describing the
temperature where the power is lost in comparison to the ambient temperature in respect
to thermal resistance can be derived from Newton’s law of cooling and known relations.
ΔQ
= Pheat
Δt
T j − Ta
k×A
ΔQ k × A × ΔT
1
=
→
=
→ Pheat =
Rθja
Rθja
Δt
Δx
Δx
ΔT = T j − Ta
(2.6.1.1)
The different components can be fragmented into many if there are several layers of
different mediums between the source and the buffer. This makes it rather simple to
estimate heat sink needs when most manufacturers state the estimated thermal resistance
of the device. However, when having different cooling topologies a slight different
approach has to be taken.
2.6.2. Heat transfer
When using a cooling bed with circulating air or fuel as cooling medium the case is
somewhat different, a lower limit of coolant weight at a certain temperature has to be
derived. The allowed rise in temperature is dependant on the following devices in the
36
chain and how high temperature they can withstand. A higher flow of coolant will cause
it to warm up less in each device while a higher pressure would suppress this behavior as
more coolant can absorb more heat. The energy delivered to the cooling bed can be
calculated using equation 23 with the respective thermal resistance. This energy is to be
removed by the coolant and depending on the specific heat capacity and the throughput of
the coolant the cooling bed can be held at a reasonable temperature. Assuming it is
known how many degrees the whole chain devices can raise the temperature and how
many devices there are; an average per device of allowed temperature raise can be
derived. This along with the total power loss needed to be removed and the nature of the
coolant the needed throughput can be calculated as in equation 2.6.1.2:
ΔQ
P
ΔQ
(2.6.2.1)
= P → m × Δt =
→
ΔQ = m × C P × ΔT → m =
Δt
C p × ΔT
C p × ΔT
Where Cp is the specific heat capacity, ΔQ is the energy differential and ΔT is the
temperature differential. Suppose a 2°C raise in temperature is allowed and 200 watts
needs to be removed from the cooling bed by air. Using the formula above gives a
minimum flow of 100g/s. From consulting with the engineer Lars Austrin at SAAB AB it
was given that the onboard cooler is slightly less efficient than a heat sink with forced
convection at sea level.
2.7.
Chassis design
The main purpose of the chassis usually is to provide a safe and protected environment
for the usually sensitive equipment. The challenges are many and not always obvious, for
instance the most common challenges are:
•
•
•
•
•
•
•
Mechanical shock
Electrical hazard both to components and to environment
Dust and other small particles such as sand
Humidity and liquids
Electromagnetic environment
Cosmic radiation
Temperature
The mechanical shock is everything from a bump to the slightest vibration, both possible
to cause stress resulting in either short-circuit or interrupted conduction. As it can be
assumed that everything in an aircraft is very fixed the main issue is still vibration, the
remedy can be to mount everything elastically or just the chassis it self. Easily understood
it is more practical to only mount the chassis on suspensions but as it usually is much
heavier than smaller components these can suffer from a vibration of higher frequency
than the one filtered by the chassis. It is also of highly importance that the chassis does
not submit to fatigue especially when the temperature can vary in a wide span and
chemicals introduced in the environment possibly causing some material to decrease in
tensile strength.
Different materials can be used with their own advantages and disadvantages such as
polymers and metals, especially alloys of either aluminum or steel.
37
Polymers are usually very light and can be constructed in any imaginable form and shape,
however, it is very likely to submit to fatigue either from unfavorable temperature, strong
light or chemicals such as acids or different hydrocarbons. Metals and usually aluminum
alloys are heavier and less moldable but are much more resistant to external influences
and are much stronger. Within a great temperature span it is not affected, strong light can
also be neglected but some chemicals can cause corrosion, at least for untreated nonprecious metals. A big advantage with a metallic chassis is it being thermal conductive all
over. More characteristics for both metals and polymers will be explained further on.
Electrical hazard to components can be avoided by both protecting it from physical touch
with an enclosure and providing a fixed support for the internal conducting parts making
them firmly separated from each other. A chassis made of polymer is usually a very good
insulator which makes it very unlikely to be exposed to electric hazard by touching the
chassis in the case of an internal malfunction. In a fully conducting chassis a fatal
accident to devices or humans can occur in the case of an error making the chassis
electrifying. It also has some practical advantages due to its insulating nature when it
comes to mounting conduction parts on the chassis making individual insulators
unnecessary.
In the case of forced convection using a fan, dust and other particles can collect in the fan
or in a filter reducing its capacity and even eventually causing a failure, not to mention
the documented limited lifetime of a fan further increasing risk of failure.
Even in an external heat sink massive amount of dust can prevent natural airflow and
radiation also reducing the capacity, even if the airflow is high enough this problem is
unlikely to arise in either solution; however it is worth a thought. If a fully sealed
enclosure with no external heat sink can be used this problem is eliminated. This sealed
enclosure would further help reducing the amount of water and moist inside the chassis
which in some cases can prove fatal to circuitry and even make metals to corrode
possibly inflicting problems such as connection faults and reduced tensile strength.
A very common problem is when high power circuitry is mixed with logics. Due to
inductive and capacitive load high energy electromagnetic waves at radiofrequency is
radiated causing disturbances and noise in logic. In some extreme cases it can even
destroy sensitive devices. The noise can be generated by the device itself or generated by
other devices in the environment, in either way the susceptibility and emitted noise
should be kept to a minimal. Except from considering the cabling a shielding cage can be
made out of the chassis, it will work as the renowned Faraday cage and will keep noise
generated inside in and noise generated on the outside out. A metallic chassis
automatically inherits this benefit but a chassis made out of polymer has to have a film or
fine-meshed grating on either the inside or outside.
At high altitudes the failure rate in power-devices increases rapidly due to cosmic
radiation. Except from reducing the voltage over the device or the junction temperature
not much can be made to reducing the influence of cosmic radiation except forming an
38
enclosure out of a thick heavy material such as lead or concrete. However, as this task is
to optimize the power/weight-ratio this is a highly unsuitable solution.
2.7.1. Cooling scenarios
Besides the issues stated above the volume is of course critical but the main factor for
choice is how to implement cooling depending on which environment and which cooling
types that can be inherited from the surroundings.
In the case of an avionic application such as a fighter aircraft there are mainly three
possible solutions for cooling and each have to be evaluated individually:
•
•
•
Exterior heat sink integrated in chassis relying on a sufficient natural airflow
around the device
Interior heat sink relying on either a chassis mounted fan or airflow from external
system
Internal fluid cooling using coolant from external system
2.7.1.1. Exterior heatsink
To use a passively cooled heat sink to remove the heat is by far the heaviest and most
space consuming solution but it is also completely independent of an external system, has
no mechanical parts that wear out and cannot suffer from leakage or condense which is
possible in fluid cooling. In an aircraft such as JAS 39 GRIPEN the temperature within
the hull can easily reach 70°C and considering operating at 10000-15000 MASL7 the
convection contribution becomes less as the air becomes less dense. For instance from a
properly designed heat sink relying on natural convection at sea-level about 70% of the
heat is transferred by natural convection and 30% by radiation but at about 15000 MASL
the ratio is about the opposite. This further increase the size required for the heat sink
since the radiation part is not increasing, it is only the convection part that is decreasing.
This inverted ratio increases the overall thermal resistance to about 233% of the original
since the convection part of the heat transfer reduces to about 18% of the original while
the radiation part is constant [15].
In power equipment it is very common to make heat sinks form up the long side walls
with high loss components internally mounted on them. Then the short side is made of
sheet-metal making a suitable place for mounting indicators, connectors, switches or even
a fan. In the case of a fan an inlet or an outlet has to be made on the opposite side to
permit airflow, usually done by perforating the sheet.
The power-components should be spread evenly over the heat sink to reach the best
performance in respect to cooling. As the heat sinks usually are very sturdy, the other
sides can be made out of thinner material, reducing the weight and this principle is
illustrated in the sketch given in figure 28. A more detailed sketch with components fitted
can be viewed in appendix F.
7
Meters Above Sea Level
39
Figure 27 : Scetch of chassis
The weight and volume for an enclosure of this solution is highly dependant on the
actual heat sink whose figures are estimated from the dissipated energy and the different
thermal resistance of the different components as illustrated in figure 26. However this is
only for the top-side transistors since the two sides will be mounted on separate heat sinks
along with one out of two rectifiers. The transistors used in the calculations for thermal
resistivity are the IRG4PSH71UD with a built-in diode, the rectifier is from Semikron
and the thermal resistance values are given in their datasheets.
Figure 28 : Thermal resistivity estimation
RθTOT = Rθsa1 // Rθsa 3 // Rθsa 5 // Rθsarect ≈ 0.39 8
8
“//” is equal to parallel coupling
40
(2.7.1.1.1)
The internal thermal resistances can be found in the datasheets while the thermal
resistance of the heat sink has to be calculated for each one of the components and then
joined using a parallel coupling as seen in 2.7.1.1.1. The temperatures given where the
ambient temperature Ta = 70°C and the maximum junction temperature Tj = 125°C. This
way of calculating is done because some of the components have different thermal
resistance and dissipated energy and then requires different cooling than the next one.
The amount of cooling and space on the heat sink for each component will be determined
the easiest way with their specific thermal conductivity9 required from the heat sink, for
instance, the rectifier requires a thermal conductivity of about 0.59 which is about 23% of
the total 2.63 Rth-1. The layout of the heat sink used in the sketch of the chassis, placing of
the components and the actual distribution of the cooling area on the heat sink can be
viewed in figure 27.
Figure 29 : Heatsink layout
The heat sinks used in the model illustrated in figure 27 is from H S Marston, it measures
150x200x40mm and is optimized for natural convection having a thermal resistance of
0.4°C/W which is sufficient. However, this is at sea level only; an increase in altitude will
increase the thermal resistance significantly as explained before. The weight of 1.44 kg
for each side gives a rough estimation of the weight of the chassis since the steel sheets
forming the rest of the enclosure will weight much less.
Using the heat sinks as two sturdy walls and steel sheets of 250x150x1mm10 as the other
walls an internal volume of 7.5dm3 is reached, estimated to fit all the devices, circuitry,
filters and the capacitor-bank. The bottom and top plate can be made up from similar
steel sheets but the bottom should be slightly thicker to provide better support, thus
doubling the thickness. A 250x200x2mm and a 250x200x1mm along with the other walls
9
The conductivity is the inverse of the resistivity
Width x height x thickness,
10
41
made of aluminum weights about 610g which when added with the heat sink the whole
enclosure weights 3.5kg.
2.7.1.2. Interior heat sink
When mounting the heat sink inside the chassis the air circulation would be very limited
if not increased via an external source, the ambient temperature in the chassis along with
the junction temperature would probably rise to a critical level and the system would
eventually shut-down or burn. However even with a small fan airflow can be produced
through the interior raising the performance of the heat sink, often many times over as
illustrated in the figure below.
Figure 30: Thermal resistance factor vs. airflow
Where F is a factor of which the thermal resistance of a heat sink is reduced depending on
the airflow across the heat sink. The curve resulting in a horizontal line is mostly due to
the fact that airflow does not increase the radiated heat.
This would allow a reduction in weight due to removal of excess heat sink at the cost of a
few watts of lost effect for the fan and the possibility of fan-breakdown, causing a fatal
temperature build-up. The fan would at best push in air at the temperature up to 70°C and
the slightly warmer air coming out would blend with the ambient air and if the
compartment is not too small it would not raise the environment temperature
significantly. The increased internal temperature has to be included in the calculations
where size of the heat sink is determined. This solution would also suffer in cooling
capacity due to a decreasing air density but it can at least be reduced by serial-coupling
many fans which can create a strong pressure in the chassis by having a smaller outlet.
However it would not be nearly enough to compensate for the loss in pressure at higher
altitudes since even if the convection could be doubled or tripled it initially represents
such a small share of the total the reduction in thermal resistivity would be insignificant.
Most aircrafts and particularly JAS 39 GRIPEN have equipment for providing
pressurized air at a few degrees above 0°C and this can strongly reduce the need for a
large heat sink at the cost of a less stand-alone device. Since the air can contain moist and
particles the air cannot be sprayed directly on a heat sink or the component but is meant
to be ran through a cooling block similar to the one of conventional fluid cooling;
although not nearly as effective due to the difference in specific heat capacity. Here a
failure in the cooling system would inevitable lead to a failure in the inverter as well as
all other equipment depending on the same cooling system. It has to be considered here
42
whether a loss or a gain in weight is achieved depending on the efficiency of the cooler, if
it requires more weight and power to produce this very cooled pressurized air than
required by a passive heat sink a total gain in weight is achieved which was not the final
task.
2.7.1.3. Internal fluid cooling
A third way of cooling in an aircraft such as the JAS 39 GRIPEN is to use its fuel as a
coolant before it is combusted. The fuel is pumped around and reaches a temperature of
maximum 100°C keeping a cooling bed at the same temperature which is equivalent to a
very large heat sink with the same temperature.
This system is also very dependant on the external cooling system and will fail or become
able to only supply as much smaller load in case of a failure in the external cooling
system. The main advantage here is the reduction in weight but also no mechanical parts.
Designing the chassis also require less consideration since no wind-tunnels has to be
made. In the case of using this system each heat sink in the sketch is replaced with a
cooling bed of similar size although very likely to significantly remove weight of the unit.
2.8.
Transistor drives
The design of the transistor drive circuit is another subsystem which deserves attention as
a thoroughly designed driver along with snubber circuits can reduce switching losses
extensively. However, not only the losses but all switching behavior can be altered and
this is wanted as there is no universal perfect switching. All applications require
individual design, for example a highly inductive load such as a motor will cause a large
voltage over-shoot if the switching is too fast. This over-shoot is very unwanted as it may
bring the IGBT outside its safe-operating-area (SOA).
Figure 31 : Descriptive driver internal circuit (IGBT within dashed line)
The main factions to alter for a satisfying behavior are the positive bias voltage, the
negative bias voltage, or the gate resistor. However, the gate resistor can be in parallel
with a diode and a resistor in series as this will change the turn-on and turn-off behavior
independently.
The +VGE is often set on a level as close to the maximum gate voltage as possible with
regard to tolerance which can be as much as 10% in some cases. This because a high gate
voltage during the ON-state minimizes the ON-resistance (or ON-state VCE saturation
43
voltage) while a too high gate voltage may permanently damage the device. Respectively
a low -VGE will increase the OFF-resistance causing less loss during the OFF-state but
will increase chance of shoot through currents if too small [7].
Setting +VGE voltages high (and -VGE low) while keeping the gate resistor value constant
will result in a low switching time and thus low switching loss. However, a faster
behavior will as stated before cause unwanted surge currents and voltage spikes over the
devices possibly destroying the device.
A rule of thumb is to set the +VGE and –VGE values to 15 and -5 respectively while
mainly altering the gate resistor to achieve a satisfactory switching behavior. Below is a
table describing common transistor behavior when changing driver parameters.
Main characteristics
VCE(sat)
ton
Eon
toff
Eon
Turn-ON surge voltage
Turn-OFF surge voltage
dV/dT malfunction
Current limit value
Short circuit withstand
capability
Radiational EMI noise
+VGE rise
Fall
Fall
-VGE rise
-
RG rise
Rise
-
Fall
Rise
Rise
Rise
Rise
Fall
Rise
Fall
-
Fall
Fall
Fall
Fall
Rise
Rise
-
Fall
Table 7 : switching characteristics vs. driver dimensions
Using the values recommended by the rule of thumb the maximum current through the
gate resistor can be estimated and then the proper driver circuit can be chosen as well as a
sufficient power supply. The gate resistor value has been chosen as the default value
given in the datasheet for the transistor irg4psh71u from International Rectifier, with this
value the switching loss has also been estimated.
I G _ peak =
+ VGE + − VGE
RG + R g
=
15 + − 5
5
=4A
(2.8.1)
The Rg is an internal gate resistor common in larger transistor modules but for the models
represented in this thesis this resistance is negligible.
As well as the peak current a sufficient continuous current will be required from the
driver and the power supply which can be calculated as in equation 2.8.2.
I G = f sw × (Q g + C ies × − VGE ) = 10000 × (570nC + 6200 pF × 5) = 0.006 A
44
(2.8.2)
Where fsw is the carrier frequency, Qg is the gate charge from 0V to +VGE and Cies is the
input capacitance. As these currents are per driver and each transistor has its own driver
the average current supplied by the power supply has to be at least six times the above
stated value. The high peak current can be managed by a buffer capacitor but then the
power supply has to be slightly bigger. Without the buffer capacitor the power supply has
to be able to provide the peak current which will give a vast loss and a heavy device.
If all the power losses are completely consumed by the gate resistor then the power
required from the driver is shown in equation 2.8.3.
1
2⎞
⎛1
⎛ 570nC×15 6200pF× 25⎞ =0.0435 W (2.8.3)
+
Pg _ loss(on) = f sw ×⎜ Qg +VGE + Cies × −VGE ⎟ =10000×⎜
⎟=
2
2
2
⎝2
⎠
⎝
⎠
Since the gate turn-ON turn is as large as the turn-OFF loss the total loss is simply the
double turn-ON loss as in equation 2.8.4.
(
Pg _ loss = f sw × Qg + VGE + Cies × − VGE
2
) = 10000× (570nC ×15+ 6200pF × 25) = 0.087 W (2.8.4)
Selecting the Half Bridge Gate Driver IR2114SSPbF from International Rectifier easily
match the driving needs for each half bridge with the least amount of external
components.
The internal maximum loss of this driver is about 1.5 W each which adds up to the
dynamic loss as explained earlier. To provide the best performance both positive and
negative power supply is recommended, although it’s not crucial.
Driver
component
Driver
Buffer
capacitor
Resistor
Value
43 nF
5
Manufacturer and Mass(g)
model
IR2114SSPbF
3*4
Jamicon
3*2
Dimensions
(mm)
3*(8x9x2)
3*(5x11)
3*1,5
-
Phoenix
6*(3.2x2)
6*0.087
6*0.1
Loss(w)
Table 8 : Driver components
2.9.
Power supply
To have a versatile solution being able to provide the controlling system with power
regardless of whether the AC-source or DC-source is used a high voltage step-down
converter is necessary. It has to be connected to the DC-bus and output a double voltage
capable of supplying both the controller and the driver. A fly-back solution from
Fairchild has proven to be an excellent choice since with a high voltage IGBT transistor
instead of the common MOSFET it is capable of handling the voltage spikes common in
fly-back circuits [17]. This along with a transformer with double secondary windings two
different voltages with different polarity is achieved.
45
Input voltage:
Efficiency:
Output voltage:
Maximum output power:
Weight:
Size:
< 707V
~80%
-5V,+15V±5%
<25W
~70g
~90x50x30mm
Table 9 : Power supply specifications and dimensions
The physical figures are estimated from the application note [17] using the same
transformer core and the same transistor as used in the example in the same application
note. However a few components such as rectifier and smoothing capacitor can be
removed since it is fed with a direct current.
2.10.
Controller
The controller is logic device responsible for handling all signals and take appropriate
action to ensure operation. It handles all fault signals and causes a halt if a serious
problem arises. It also observes the actual voltage levels and steers it towards the proper
level. The controller is of the less crucial components when it comes to power losses or
weight and thus it is not examined in detail. Still, a suitable off-the-shelf component has
to chosen to have some guidelines. The IRMCK203 from International Rectifier is a fully
capable controller requiring minimal external circuits. However since it requires a 3.3V
power supply a fixed regulator has to be used to decrease the voltage from the power
supply.
Since the lost power due to the fixed regulator is dependant on the voltage drop and the
current through it the lowest voltage from the power supply should be used.
The above chosen controller dissipates 1.2 W @3.3V typically according to the data sheet
which gives a current of 400 mA. The combined loss of the controller and the regular are
thus 2 W (400mA × (1.7V + 3.3V )) . The space and weight consumed by the controller is
not of vast proportions and could easily be shared with the power supply and the driver
on the same circuit board.
Controller
compoenents
Controller
Value
-
Voltage
regulator
3.3 V
Buffer
capacitor
25 V
10 μF
Manufacturer
and model
International
Rectifier
IRMCK203
National
Semiconductor
LP3852
Jamicon
Mass
(g)
7
Dimensions
(mm)
15x15x5
Loss
(w)
1.2
9
10x21x5
0.8
2x2
5x3
-
Table 10 : Controller components
46
3. Losses
In this part mostly the components responsible for the main part of the loss are
considered, those are the main supply rectifier and the transistor-bridge. To some extent
the input filter and the capacitor bank will also add up to the losses. Mainly two types of
transistors have been considered, the IGBT and the MOSFET. The switching loss in the
rectifier is in this case ignored due to the low frequency and the low switching energies in
the diodes, which leave only the forward conduction loss remaining.
However in the IGBT-bridge the largest part of the loss is due to the high switching
frequency in the slow IGBT-transistors, a minor loss also arise due to forward voltage
drop. In the transistor-bridge the free wheeling diode is usually included further
contributing to power loss but in the cases of missing a built-in diode the IGBT-bridge
and a suitable diode will be evaluated individually. There will also be some loss in the
filter capacitances and the snubber circuit depending on which type chosen and the
severity of the compensation. Some minor power losses will occur in a switched voltage
supply and the control circuit but it is almost insignificant in amount and hard to
optimize.
In appendix D a spreadsheet has been formed over suitable transistors to ease the
choosing of the most suitable transistor, different technologies such as IGBT and
MOSFET has been evaluated as have different types of semi conducting materials. In the
spreadsheet there is a majority of IGBT before MOSFET, this is due to the difficulty
finding a MOSFET with a VDS rating of about 1200V, this is the recommended voltage
when having a line voltage of 400V due to large voltage peaks. We were able to find one
MOSFET with a VDS of 800V, however voltage peaks are likely to destroy this device
and the fact that is has a very large RDSon that generates a vast amount of dissipated
energy makes this device not suitable.
3.1.
Switching losses
3.1.1. Transistor
The switching losses can be derived from the switching energies found in datasheets but
usually requires some conversion since the rated currents and bus voltages might differ
from the application specific currents and voltages. The conversion can be done for the
current using equation 3.1.1.1 and 3.1.1.2 [7]:
E on = E on ' ( I ave / ratedI ave ) α
(3.1.1.1)
E off = E off ' ( I ave / ratedI ave ) β
(3.1.1.2)
47
Where α and β are multipliers depending on the device and Eon’ and Eoff’ are the energies
at the rated average current which is calculated according to equation 3.1.1.3.
2
I ave =
2 × I0
(3.1.1.3)
π
The same should be done analogous with the voltage since it usually differs as well.
•
Turn-On loss (Pon)
Pon = f c E on ( I ave )
(3.1.1.4)
Where fc is the switching frequency and Eon(Iave) is the Turn-On energy at the average
current Iave [7].
•
Turn-Off loss (Poff)
Poff = f c E off ( I ave )
(3.1.1.5)
Where fc is the switching frequency and Eoff(Iave) is the Turn-Off energy at the
average current Iave [7].
3.1.2. Diode
FWDrr = E rr ( I ave ) ×
fc
2
(3.1.2.1)
E rr = E rr ' ( I ave / ratedI ave ) χ
I ave =
2
π
(3.1.2.2)
2 × I0
(3.1.2.3)
Err is the reverse recovery energy per switch which with the switching frequency fc gives
the switching loss of the diode. Usually all characteristic values for the diode can be
found in the datasheet; however it may require some interpolation between values in
graphs to determine the appropriate value for the specific application, such as estimated
junction temperature.
It is very common for an IGBT or a power MOSFET to be co-packed with a silicon FWD
which gives some benefits such as shared cooling and less wiring reducing capacitive and
inductive noise but it also makes it slightly more difficult to experiment with other types
of diodes. A few IGBTs suitable for this application are available with and without a
FWD making interesting experiments possible, especially to use a SiC FWD to optimize
switching loss.
48
3.2.
Conduction losses
3.2.1. Transistor
The losses in the IGBT can be approximated using the information supplied in the
datasheet and some knowledge of the application specifics such as switching frequency
and the power factor of the load [13].
•
The ON-state power loss in an IGBT transistor can be calculated as in equation
54:
⎧⎪ 2 I 0 ⎫⎪⎧
⎛⎛
⎞⎫
8R ⎞
⎛π ⎞
⎛π ⎞
Psat = ⎨
⎟ × 2 × I 0 ⎟⎟⎬
⎬⎨V0 + ⎜ ⎟ × R × 2 × I 0 + ⎜ ⎟ × cos φ × ⎜⎜ ⎜V0 +
3π ⎠
⎪⎩ 2π ⎪⎭⎩
⎝4⎠
⎝4⎠
⎝⎝
⎠⎭
(3.2.1.1)
Where cosφ is the power factor of the load. I0 is the root mean square value (RMS) of the
phase output current and both V0 and R can be derived from the following picture
illustrating the output characteristics of an IGBT [13].
Figure 32 : Approximate output characteristics
• Total power loss (Plosstot)
The total power dissipation can be calculated as in equation 3.2.1.2 [13]:
Plosstot = Pon + Poff + Psat
(3.2.1.2)
3.2.2. Diode
The total loss and dissipated heat in the diode is dependent on many factors such as
switching frequency, characteristics of the diode and how inductive the load is. An easy
approximate model for power loss calculations is described in equation 3.2.2.1 [13]:
⎧⎪ 2I 0 ⎫⎪⎧
⎞⎫
⎛⎛
8R ⎞
⎛π ⎞
⎛π ⎞
FWDsat= ⎨
⎬⎨V0 + ⎜ ⎟ × R × 2 × I 0 − ⎜ ⎟ × cosφ × ⎜⎜ ⎜V0 + ⎟ × 2 × I 0 ⎟⎟⎬
3π ⎠
⎪⎩ 2π ⎪⎭⎩
⎝4⎠
⎝4⎠
⎠⎭
⎝⎝
(3.2.2.1)
Where cosφ is the power factor, I0 is the forward RMS current, R and V0 form up the
voltage drop of the diode in the same manner as for the IGBT but probably with slightly
different values. Usually all characteristic values for the diode can be found in the
49
datasheet; however it may require some interpolation between values in graphs to
determine the appropriate value for the specific application, such as estimated junction
temperature.
• Total power loss in the FWD (Plosstot)
The total power dissipation can be calculated as in equation 3.2.2.2 [13]:
Plosstot = Pon + Poff + Psat
3.3.
(3.2.2.2)
Lowest loss estimation of IGBT and FWD
In appendix D it can be found a MOSFET in comparison with various IGBT with almost
the same voltages and currents and as predicted the power loss in the MOSFET is greatly
exceeding any IGBT regardless of model. This is easily explained by the high voltage
ratings.
On the other hand, in other applications in example as an inverter with a DC-bus supplied
from a regular car battery (12V) which is consuming 50A an off-the-shelf IGBT with
VCEsat = 2.5V has the efficiency of 79%. An of-the-shelf MOSFET with RDS(on) = 0.007 Ω
has an efficiency of 97% and if we add switching losses the benefit of the MOSFET is
increased further more. The conclusion looks like choosing the IGBT with the lowest
power losses are the best choice without a doubt but there are other concerns as well.
By viewing the chart of suitable transistors/diodes again and removing all those which
may have a problem with either too high peak currents or too high peak voltages 11a few
good transistor solutions remain, in this case with respect to power loss per IGBT with
FWD. Calculations are done using the equation 55 and 57 with the parameters cosφ=1,
I0=15 A and the model specific parameters can be found in appendix D where all the
parameters have been normalized to fit the actual values. The switching frequency is
assumed to be 10kHz giving it a mf = 15.
PLOSSTOT12
Manufacturer
Model
IRF
IRF/CREE
IRG4PSH71UD 26,3W
IRGPSH71U/
22,8W
C2D05120
2 x CID100512 40W
CREE
SI IGBT, SI FWD
SI IGBT, 2 x SiC FWD
RθJC + RθCS
0.36+0.24
0.36+0.24,1.1+0.24
2 x SI IGBT, 2 x SiC FWD
1.15+0.24
Technology
Table 11 : Suitable discrete IGBTs
Although even if the discrete transistors CO-packed with free-wheeling-diodes above
seems to outperform the modules below in respect to loss the cabling has to be considered
since it will contribute with some capacitance and possibly some inductance. In some
cases the effects of this is large enough to be ineligible in favor of the big modules.
The discrete transistors also have the possibility to be mounted on a separate heatsink or
to displace the dissipated heat over a larger area making cooling easier in some
11
12
The rated voltage and current should be at least twice the nominal to handle overshoots [7].
Conduction loss and switching loss for one IGBT with FWD
50
applications. However, the modules are much easier to handle, one big bulky brick with
optimized internal wiring prepared for mounting a PCB on top along with built-in brake
IGBT and a high performing rectifier-bridge results in a lot less wiring and complexity.
On the other hand a module requires a larger heatsink or equal device capable of
removing a lot of heat from a relatively small area.
PLOSSTOT4
Manufacturer
Model
Infineon
FUJI
Infineon
FS25R12W1T4
29,6W
SI IGBT, SI FWD
7MBR35UB12013 33W
SI IGBT, SI FWD
FP25R12KT35
30,7W
SI IGBT, SI FWD
Table 12: Suitable IGBT modules
Technology
RθJC + RθCS
0.66 + 0.88
(0.76,1.07)+0.05
(0.8,1.35) + 0.3
In the end a more detailed approach than only the power loss has to be considered for
choosing the most suitable solution, mostly depending on the choice of cooling but also
on how much the transistor-performance can be increased with the help of snubber
circuits.
3.4.
Rectifier
The total conduction loss can be derived from the average current running through a leg
times the forward voltage drop of two diodes since the current always runs through two.
This voltage drop is somewhat depending in some grade on the current and the
temperature of the junction and the proper value can be found in the datasheet. The
average current is simply the output current divided by three, notice that this value will
not set the voltage drop since there is momentarily a higher current running through the
diode when actually conducting, they are just not conducting more than a half period at a
time.
PLOSSCOND = 2 × VSAT × I avg × 3
(3.4.1)
The “2” since the current is running through two diodes and the “3” since there are three
legs in the rectifier bridge as illustrated in figure 13. Mainly three rectifiers from three
different suppliers have been considered, FUJI, International Rectifier and Semikron.
They have all been chosen from a wide range of models and only the best suiting rectifier
for this application has been considered. Their main properties are described in the table
below.
13
The module is equipped with a built-in rectifier and brake function
51
Manufacturer
Model
Fuji
IRF
Semikron
6RI30E
36MT80
SKD 33
VSAT @14,5A
125° C
0,87
0,9
1,21
PLOSSTOT
Weight
Size(packet)
RθJC + RθCS
35,7 W
36,9 W
49,6 W
100g
20g
30g
52 x 22 x 26
29 x 26 x 21
64 x 30 x 21
0.80+ 0.10
1.16 + 0.2
0.417
Table 13 : Suitable rectifiers
As seen in the table above a tradeoff has to be made, the dissipated energy is lower in
some of the cases but it is also harder to remove the heat due to the larger thermal
resistance. For instance the dissipated energy in the Semikron diode bridge is
approximately 39% higher compared to the Fuji, however the Semikron has only about
46% of the thermal resistance making it a lot easier to cool. If an external efficient
cooling system is available then the Fuji would be the best choice to maximize efficiency
but with worse conditions for cooling the Semikron would be the best choice neglecting
the slightly lower efficiency.
3.5.
Filters
3.5.1. Input filter
As the current and voltage harmonics will not change in frequency with increasing rated
power of the inverter the components values can be scaled proportional as long as the
main voltage is considered fixed. Along with this proportional increase comes the power
loss. The factor that would have largest effect on the input filter components and loss
would be to change rectifier complexity. Stepping up from the present 6-pulse rectifier
bridge to at least a 12-pulse bridge would reduce initial harmonics significantly and
therefore almost completely removing the need for additional filter except for EMI.
However the EMI filter does barely represent any significant loss except in some
additional weight.
However if the 10kW input to a 6-pulse rectifier needs to filtered to an acceptable level
including a simple EMI filter it requires filters worth of 131g and 37 W of loss per phase,
which is not negligible. It adds up to a total of 393g and 111W.
3.5.2. DC-bus filter
As the losses already have been derived and explained some estimated values for
different output capacities and switching frequencies has been collected. A 3% ripple
voltage of the maximum DC-bus voltage is allowed in all scenarios.
Output power/switching
frequency @ 230/400VAC
2kW
10kw
20kW
6kHz 10kHz
14kHz
2.6
27
54
3.8
41
81
3.2
30
67
Table 14 : DC-bus filter losses when using 230/400 VAC input
52
Output power/switching
frequency @ 115/200VAC
2kW
10kw
20kW
6kHz
10kHz
14kHz
5
6.3
8.2
42
52
68
80
100
120
Table 15 : DC-bus filter losses when using 115/200 VAC input
The loss due to the surge limiting circuit is not included since it is only active during
start-up. A 12-pulse rectifying bridge would also here prove to be favorable as it
decreases the voltage ripple significantly doubling its frequency, relieving the capacitor
from much ripple currents which will require a smaller capacitor bank.
3.5.3. Output filter
The power loss in the output filter is depending on several factors where the largest is the
DC-bus voltage VDC, fsw, Ipeak and the maximum allowable dV/dt in the motor which can
be calculated using the simplified equation 3.5.3.1 from chapter 2.4.3.
VDC × I peak × f sw × π
=
4 × dV
dt
2
Poutput _ filter _ loss
(3.5.3.1)
Considering the maximum dV/dt for the motor rather fixed would make this model only
consist of fundamental parameters and straight to calculate.
Using the parameters for the model examined in this paper a power loss per phase is
estimated to be 10 W, giving a total of 30 W.
3.6.
Controller
The controller is responsible for an insignificant power loss in comparison to the high
power components. But as it is trivial to estimate the maximum loss there are no real
reason to neglect it. This loss can also be considered fixed regardless of the output power
or the switching frequency as it mainly depends on the internal clock frequency.
As it can be found in the datasheet the maximum loss is [email protected] but as explained
before a voltage regulator is needed, further contributing to an additional 0.8 W loss.
3.7.
Transistor drives
The power loss in the transistor drives consists of one rather fixed component and one
minor depending on the switching frequency. However if the transistor configuration
needs to be changed due to higher power it would alter the dynamic power loss as well.
Suppose the same transistor would be used in parallel to obtain higher power rating then
the dynamic power loss would increase with a factor corresponding to the number of
parallel transistors. Otherwise it would increase proportional to the switching frequency
as explained in chapter 2.7.
53
The 10kW inverter examined in this paper would accumulate about 5W of power loss due
to the three drives ( 3 × 1.5 W) and the dynamic 0.5 W assuming the switching frequency
is 10 kHz. Even if the switching frequency would be changed to 6 kHz or 14 kHz it
would not change the total significantly, (4.8 W, 5 W, 5.2 W).
A parallel coupled transistor would double this loss as well as requiring a stronger driver.
3.8.
Internal power supply
The power loss of the internal power supply is dependant on the devices it provides with
power as it usually has a rather fix percentage of loss. As the transistor drives have a high
peak current while low RMS current a buffering capacitor is favorable to use instead of
over sizing the power supply. Since the efficiency in the power supply is estimated to
80% the other 20% of the input power is pure loss dissipated in the supply.
As stated in chapter 3.7 and 3.8 the power loss from the controller and drives sums up to
7 W giving the power supply a loss of 1.75 W adding up to the total of 8.75 W put in to
the power supply.
54
4. Results
4.1.
Power density
The power density is defined as the ratio between the rated output power and the weight
of the device, the higher the better. The power density is very much related to the
efficiency of the device as thoroughly explained in this report. The weight of three
different power levels, 2000W, 10kW and 20kW will be estimated using the same
method as used in the paper for 10kW. Two different input voltages are used, the 400V
three phase input used in this paper and a 200V three phase input which is common in
aircrafts. The weight is divided in the respective segment of the inverter to pinpoint the
specific weight. As can be seen some parts grow with rated output while some parts
remain constant. The switching frequency is considered constant at 10kHz for this
evaluation.
Weight (g) @ 230/400VAC / 540VDC
Rated effect/component
2000W
Input filter
72
Rectifier
13
Dc-bus filter w. surge limiter 240
Power supply
70
Controller and driver
60
IGBT and FWD
54
Output filter
39
Internal wiring
130
Cooling
1200
Housing
630
Sum:
2525
10kW
393
30
780
70
60
54
219
240
3880
940
6666
20kW
720
97
1872
70
60
54
403
370
7230
1463
12339
Table 16 : Approximate separated weight for 230/400VAC applications
Weight (g) @ 115/200VAC / 270VDC
Rated effect/component
2000W
Input filter
98
Rectifier
30
Dc-bus filter w. surge limiter 90
Power supply
70
Controller and driver
60
IGBT and FWD
54
Output filter
53
Internal wiring
173
Cooling
1330
Housing
630
Sum:
2588
10kW
480
97
130
70
60
54
365
360
4160
940
6716
20kW
1230
165
585
70
60
54
623
470
8440
1463
12360
Table 17 : Approximate separated weight for 115/200VAC applications
55
4.2.
Scalability
The ability and simplicity to up-scale or down-scale a solution to fit the actual needs
without any unnecessary weight is the definition of scalability. Of course some weight is
added to but it is far from a linear relation. Some parts will remain fixed when adding or
removing output effect while some will be abundant.
From the tables in chapter 4.3. a mathematical approximation can be derived to estimate
the loss of an arbitrary converter with some basic input characteristics. The inputs would
be power in watts, switching frequency in hertz and the phase to phase voltage on the
input. The estimation has a maximum error according to the table of 7% and should be
precise enough to cover at least 20% outside the regions studied.
⎛ 400 ⎞
⎟⎟
Ploss (Pin , VACin , f sw ) = ⎜⎜
V
⎝ ACin ⎠
0.59
(121+ 0.0034( f
− 6000)
1.15
sw
+ 0.0084(Pin − 2000)
1.1515
)
(4.2.1)
The same approximation can be done for the estimated mass using the tables in chapter
4.1. Input parameters are the same and the result is given in grams.
⎛ 400 ⎞
⎟⎟
Mass(Pin ,VACin, f sw ) = ⎜⎜
⎝ VACin ⎠
0.0.24
(2290+ 0.27( f
sw
− 6000)
0.772
+ 0.305(Pin − 2000)
1.059
)
(4.2.2)
The method used to fit the equations to the values is to assume the relation is exponential
and then start from the lowest value. In this scenario the 200 VAC, 2 kW and 6 KHz
version is used as base. The increase in weight and loss is averaged on the values
considered as fixed when the value at hand is growing, in this case in three steps. Doing
this for all three input values and adding their components makes up (4.2.1) and (4.2.2).
4.3.
Efficiency
The estimated amount of loss compared to the rated output power. As can be seen in the
following tables a lot of combinations are given to estimate the power loss for
corresponding combination. For the lower voltage case the lower voltage version of the
transistor and FWD are used as they both have one. For the lower and higher effects
similar components but with different power ratings are taken of-the-shelf usually from
the same series and manufacturer as the components in the example.
Rated effect/component
@ 6kHz
Input filter
Rectifier
Dc-bus filter w. surge limiter
Power supply
Controller and driver
230/400VAC / 540VDC
2kW 10kW 20kW
23
111
220
7
45
85
2.6
27
54
1.5
1.5
1.5
2+4.8 2+4.8
2+4.8
56
115/200VAC / 270VDC
2kW
10kW 20kW
37
170
460
14
89.6
185
5
42
80
1.5
1.5
1.5
2+4.8
2+4.8
2+4.8
IGBT and FWD
Output filter
Snubber circuits
Sum of loss:
36
4
0.05
81
109
21
0.5
322
278
42
1.476
689
43
7
0.074
114.4
172
39
0.74
521
363
67
2.21
1165
Table 18 : Approximate separated losses @ 6 kHz switching frequency
Rated effect/component
@ 10kHz
Input filter
Rectifier
Dc-bus filter w. surge limiter
Power supply
Controller and driver
IGBT and FWD
Output filter
Snubber circuits
Sum of loss:
230/400VAC / 540VDC
2kW 10kW 20kW
23
111
220
7
45
85
3.2
33
67
1.75
1.75
1.75
2+5
2+5
2+5
44.6
145
355
5
30
56
0.083 0.83
2.46
92
373
794
115/200VAC / 270VDC
2kW
10kW 20kW
37
170
460
14
89.6
185
6.3
52
100
1.75
1.75
1.75
2+5
2+5
2+5
59
188
397
9
49
76
0.123
1.23
3.68
134
558
1230.8
Table 19 : Approximate separated losses @ 10 kHz switching frequency
Rated effect/component
@ 14kHz
Input filter
Rectifier
Dc-bus filter w. surge limiter
Power supply
Controller and driver
IGBT and FWD
Output filter
Snubber circuits
Sum of loss :
230/400VAC / 540VDC
2kW 10kW 20kW
23
111
220
7
45
85
3.8
41
81
1.85
1.85
1.85
2+5.3 2+5.3
2+5.3
53
175
433
8
47
93
0.117 1.167
3.44
104
429
924.6
115/200VAC / 270VDC
2kW
10kW 20kW
37
170
460
14
89.6
185
8.2
68
120
1.85
1.85
1.85
2+5.3
2+5.3
2+5.3
74
204
431
12
67
107
0.173
1.73
5.16
154.52 609.5
1317.3
Table 20 : Approximate separated losses @14 kHz switching frequency
4.4.
Reliability
The reliability is dependant on margins and precautions taken to ensure the longevity of
the device. Functions implemented to protect the device during errors or unwanted
conditions are a necessary factor to prevent permanent damage to the device. Also how
effective the components are cooled will result in a reliability factor as higher
temperature reduces lifetime in most semiconductors and especially electrolyte
capacitors. It may be needed to pinpoint every component that is subject to high power
loss and evaluate which temperature it will reach and how it will affect its lifetime.
Protective measures such as snubber circuits which will reduce voltage transients over
already very warm components can improve lifetime significantly as heat, junction
voltage and cosmic radiation are the main factors of reducing semiconductor lifetime
[21].
57
4.5.
Simulations
Some of the simulations in this master-thesis are simply to illustrate the qualitative
behavior of some circuits while some are just to measure noise and disturbances. A few
Simulations has also been done to verify the calculated values of some components. For
the snubber considerations only Pspice have been used to illustrate the effects a snubber
circuits can inflict on the voltage transients. Any tries to reduce power loss has been
disregarded as it is very time consuming to find components with values that actually can
be realized.
For component value verification Matlab and Simulink have been used, the layout of the
system can be viewed in appendix B. A fully working system using the surge current
limiter can be found in appendix A1 where the graphs shows the growing voltage of the
DC-bus, the switched voltage from an IGBT couple and the same low-pass filtered
voltage. Under the same circumstances the currents running through the diode bridge and
the transistor bridge have been measured and the results are presented in appendix A2.
First is an averaged form of the following diode currents. The average current seen in the
first graph eventually corresponds to the steady-state value as expected. However the
instantaneous current seen in the second graph is of larger interest as in the case of no
surge current limiter used these currents as seen in the top graph in appendix A4 rise to a
dangerous level. The current running through the IGBT bridge can be seen being
regardless of using the surge current limiter or not as seen in the bottom graph in
appendix A3 and A4.
In appendix C1 the frequency spectrum at the input terminal has been measured using the
model set up as in appendix B. The voltage spectrum consists of the fundamental
frequency and its harmonics without any filtering. In appendix C2 the current harmonics
have been measured. The very same voltage measurement has been done in appendix C3
but this time it is filtered. The most disturbing harmonics as well as the total harmonic
distortion has been reduced significantly. In appendix C4 the same voltage measurement
has been carried out but this time a 12-pulse rectifier is used. In addition the current
harmonics have been measured in appendix C5. As can be sent the harmonics are much
higher in frequency but much lower in magnitude making it much easier to filter if
needed, in this scenario it is not.
58
5. Conclusions
The most obvious conclusion from performing this master-thesis is that the selection of
switching transistors should be given some time as the amount of power loss can be
reduced significantly with a delicate choice of components. However, as seen in this
thesis there are other parts of the converter contributing to a vast amounts of loss and in
particular weight not foreseen in the beginning of the thesis. In the converter studied in
this master-thesis the filters add up to almost 25% of the total weight, although in some
other configurations it becomes more or less prominent. The power lost in the filters is in
the same size as in the switching devices making them worth further more attention.
Altogether it would make good choice to address the filters the same detailed approach as
the switching devices.
The choice of using a SiC FWD ended in slightly less calculated loss in the switching
devise due to the negligible reverse recovery energy; however this should be simulated
extensively under appropriate conditions to be verified. Although uncertain loss it can
easily be said that the SiC technology requires less cooling as it can work under very high
temperatures. However using a FWD not integrated in the package introduces additional
capacitance and inductance possibly slows down the switching process canceling the
reduction gained. In addition the need for snubbers circuits increase when having higher
stray inductance.
As it turns out, using cooling relying on passive convection only in an aircraft a high
altitude would be too inefficient and the built-in cooling system would be the only
choice; regardless of which coolant used. In an aircraft flying on low altitude on the other
hand the high velocity winds could be used to cool the equipment.
59
6. Future work
Several of the solutions explained in this master-thesis is today obsolete and can be
replaced with better performing and more sophisticated solutions. However most of these
solutions is by far too time consuming to evaluate and are excluded in favor of the less
sophisticated ones. More attention should be put to minimize the filter need as they
consume both power and space/weight. Additional time can be put to optimize and
evaluate the snubber circuits as they now only are evaluated in the sense of suppressing
voltage transients. All the wires with their respective stray inductance and capacitance
have to be estimated to avoid noise and transients. A light weight transformer could be
used to form a 12-pulse rectifier to remove the need for extensive filter on the input.
When having further knowledge of both the generator supplying power and the motor
consuming power more detailed filters could be designed. All parts also has to be verified
and tested extensively using both simulation and under real circumstances.
60
Bibliography
[1]
Martin Gårdman Andreas Johansson. Commissioning and Evaluation of an Inverter
Prototype. Dept. of industrial Electrical Engineering and Automation. Lund
University. 2007
[2]
Julius Luukko. Direct torque control of permanent magnet synchronous machinesanalysis and implementation. Lappeenranta 2000.
[3]
István Schmidt, Katalin Vincze, Károly Veszprémi and Balázs Seller. Adaptive
hysteresis current vector control of synchronous servo driver with different
tolerance areas. Department of electrical machines and drives. Budapest University
of technology and economics. June 29, 2000
[4]
Michael O’Neill. The benefits of using a Cree inc. IGBT/SiC Schottky co-pack in
AC inverter applications. Cree inc. September 2006.
[5]
Jim Richmond. Hard-switched silicon GBTs? Cut switching losses in half with
silicon carbide Schottky diodes. Cree Inc.
[6]
S. Vieillard, R. Meuret, High efficiency, high reliability 2 kW inverter for
aeronautical application. Hispano-Suiza, rond point René Ravaud, MoissyCamayel, France. 2007. ISBN: 9789075815108
[7]
Fuji IGBT modules application manual. Fuji electric device technology co., LTD.
February 2004.
[8]
Laszlo Balogh. Design and application guide for high speed MOSFET gate drive
circuits.
http://www.powersystems.eetchina.com/PDF/2007JUL/PSCOL_2007JUL26_DRO
P_TA_101.PDF
[9]
International Rectifier. Applications note AN-990. Application characterization of
IGBTs
[10] International Rectifier. Applications note AN-983. IGBT characteristics.
[11] Alf Alfredsson, Karl Axel Jaconbsson, Anders Rejminger. Elkrafthandboken,
Elmaskiner, Studentlitteratur, ISBN: 91-47-00066
[12] Mohan, Undeland, Robbins, Power Electronics, Wiley, ISBN:0-471-22693-9
[13] IGBT MODULE APPLICATION Manual Hitachi, Ltd. Ref.No.IGBT-01 (Rev.2)
[14] Mats Alaküla, Power Electronic Control, KFS Lund AB
61
[15] Designing LC-filters for AC-drives, Bravo Electric Components, Inc.
[16] Electrolytic capacitors application guide, Evox Rifa.
[17] AN9011High Input Voltage, Off-line Flyback Switching Power Supply using FSC
IGBT (SGL5N150UF), Fairchild semiconductor
[18] International Rectifier. Application note AN-1095. Design of the Inverter Output
Filter for Motor Drives with IRAMS Power Modules
[19] MIL-STD-704, Military Standard, Aircraft Electric Power Characteristics,
Department of Defense USA
[20] Snubber Considerations for IGBT Applications, International Rectifier
[21] Failure Rates of HiPak Modules Due to Cosmic Rays, AN-5SYA 2042-02ABB,
ABB, Nando Kaminski
62
Appendix A1 - Output Characteristics using Surge Current Limiter
Vdc
600
NO over-shoot
500
400
300
200
100
0
Vab inverter
600
400
200
0
-200
-400
-600
Vab Load
500
400
300
200
100
0
-100
-200
-300
-400
-500
0
Time offset: 0
0.005
0.01
0.015
0.02
0.025
Appendix A2 - Output Characteristics using NO Surge Current Limiter
Vdc
600
500
Slight over-shoot
400
300
200
100
0
Vab inverter
600
400
200
0
-200
-400
-600
Vab Load
500
400
300
200
100
0
-100
-200
-300
-400
-500
0
Time offset: 0
0.002
0.004
0.006
0.008
0.01
0.012
0.014
Appendix A3 - Rectifier/IGBT Currents using Surge Current Limiter
I Diodes 1 3 5 ave
18
16
14
12
10
8
6
4
Steady state current level
correspond to the calculated
value
2
0
I Diodes 1 3 5
40
35
30
25
20
15
10
5
0
-5
I IGBT 1 3 5
20
15
10
5
0
-5
-10
-15
-20
0
Time offset: 0
0.005
0.01
0.015
0.02
0.025
Appendix A4 - Rectifier/IGBT Currents using No Surge Current Limiter
I Diodes 1 3 5 ave
140
120
100
80
60
Plot available after
one cycle when
calculating RMS
40
20
0
I Diodes 1 3 5
300
Critical over-shoot
250
200
150
100
50
0
-50
I IGBT 1 3 5
20
15
10
5
0
-5
-10
-15
0
Time offset: 0
0.002
0.004
0.006
0.008
0.01
0.012
Multimeter
6
400V, 400 Hz
12 KVA
C
B
A
v
I IGBT 1 3 5
v
A-B
+
-
I Diodes 1 3 5
UU(E)
UU(E)
+
-
+
-
Discrete
RMS value
I Diodes 1 3 5 ave
3-phase scope
In
RMS
I Diodes 1 3 5
v
A-C
-
+
current scope
C
B
A
Rectifier
Conn2
surge limiter
Conn1
C
Discrete,
Ts = 4e-006 s.
L1
-
+
C
B
A
g
PWM
IGBT Inverter
v
Vdc
+
-
z
1
v
C
B
A
Discrete
PWM Generator
Pulses
Signal(s)
C
B
A
LC Filter
Vab_inv
+
-
c
b
a
A
B
m
Vabc_inv
Vd_ref (pu)
Vabc (pu)
Voltage Regulator
10Kw
400 V rms
667 Hz
Measure
C
B
A Vabc
+
-
v
Vab_load
C
A-B1
0.71
Vref (pu)
output scope
Appendix B - System
Simulation Model
Appendix C1 - Main Grid Voltage Harmonics without Filter
Appendix C
Mag (% of Fundamental)
0
5
10
15
20
25
30
35
-15
0.1
-10
-5
0
5
10
15
0
0.101
0.102
5
0.104
0.105
Time (s)
0.106
0.107
10
Harmonic order
15
Fundamental (400Hz) = 11.11 , THD= 34.18%
0.103
FFT window: 4 of 80 cycles of selected signal
20
0.108
0.109
25
Appendix C2 - Main Grid Current Harmonics without Filter
Mag
0
5
10
15
20
25
30
-600
0.1
-400
-200
0
200
400
600
0
2
0.105
4
0.11
0.12
Time (s)
0.125
0.13
6
8
10
Harmonic order
12
14
Fundamental (400Hz) = 561.1 , THD= 6.61%
0.115
FFT window: 16 of 80 cycles of selected signal
16
18
0.135
20
Appendix C3 - Main Grid Voltage Harmonics with Filter
Appendix C4 - Main Grid Voltage Harmonics with 12-Pulse Rectifier
Mag
0
1
2
3
4
5
6
7
8
9
10
-600
0.1
-400
-200
0
200
400
600
0
0.102
0.104
5
0.108
0.11
Time (s)
0.112
0.114
10
Harmonic order
15
Fundamental (400Hz) = 557.2 , THD= 2.32%
0.106
FFT window: 8 of 80 cycles of selected signal
20
0.116
0.118
25
Appendix C4 - Main Grid Voltage Harmonics with 12-Pulse Rectifier
Mag
0
0.05
0.1
0.15
0.2
0.25
-8
0.1
-6
-4
-2
0
2
4
6
8
0
0.101
0.102
5
0.104
0.105
Time (s)
0.106
0.107
10
Harmonic order
15
Fundamental (400Hz) = 6.57 , THD= 4.61%
0.103
FFT window: 4 of 80 cycles of selected signal
20
0.108
0.109
25
Appendix C5 - Main Grid Current Harmonics using 12-Pulse Rectifier
IRF
IRF
IRG4PH40UD
IRG4PH40K
APT
APT
APT
APT50GN120L2DQ2
APT50GF120JRD
APT50GF60BR
24
25
86 1200
InfineonSixPACK
FS15R12YT3 Fast
110 1200
InfineonSixPACK
FS15R12VT3
35
7 in one-pac 160 1200
FUJI
25
7MBR35UB120
25
7 in one-pac 115 1200
FUJI
7MBR25UA120
7 in one-pac 110 1200
FUJI
7MBR15SC120
35
7MBR25SA120
7 in one-pac 180 1200
FUJI
7MBR25SA140
35
FUJI
7MBR25SC120
7 in one-pac 180 1400
FUJI
FUJI
7MBI50N-120
35
75 50
75 50
50
600
7 in one-pac 400 1200
300
2,6
1,85
1,85
1,40
3,00 10-20kHz 1,07
1,65 2-38kHz
1,65 2-38kHz
1,35 2-38kHz
1,8 -38kHz
1,8 -38kHz
1,5 -42kHz
1,75 -42kHz
1,5 -40kHz
1,2 -24kHz
10-22kHz
0,80 10-21kHz
1,35
1,71
1,89
2,34
2,07
1,89
2,88
1,8
1,8 0,83
1,17 0,94
1,17 1,18
0,72 1,17
0,81 0,99
1,35 1,15
0,81 1,01
1,49 1,008 1,05
1,31 0,846 1,05
1,44
1,44
1,17
1,17
2,25
1,26
0,00
1,9
2,25
0,64
2,1
2,7
0,68
1,92
1,89
0,79
2,78
0,77
0,00
0,00
0,00
2,78
1,05
1
2,84
1,26
1,5
2,1
2,6
3,8
2,1
0,00
1,62
1,4
1,10 20-30kHz
3,25 8-44kHz
2,80 8-43kHz
1,20 8-42kHz
5
2,8
24,5
14
16
15,5
3,78
4,5
6,3
7,8
11,4
6,3
8,4
15
12,6
7,5
6,3
7,5
10,5
13
19
10,5
14
25
21
12,5
8,55 14,25
14,7
8,4
9,6
9,3
173,18
8,82 111,758
10,5 126,737
14,7 94,5579
18,2 147,848
26,6
14,7 175,735
19,6 136,843
35 188,969
29,4 172,213
17,5 120,288
19,95 157,157
34,3 170,708
19,6 160,117
22,4 119,513
21,7 124,812
3
5
5,7
6,3
9,5
10,5
7,41 12,35
6,57 10,95
32,4
19,8
13,5
22,5
7
112,06
158,7
31,5 143,836
34,65 172,623
27,72 182,316
27,09
45,36 206,625
27,09 158,795
47,25 194,145
13,3 84,3485
14,7 113,899
17,29 99,6341
15,33 104,602
12,208 11,53 19,22 26,901 148,739
11,817 11,58 19,31 27,027 146,695
9,532
12,745 14,85 24,75
17,331 11,88
13,848 11,61 19,35
13,848 19,44
13,848 11,61 19,35
11,282 20,25 33,75
8,3581
12,038
7,2804
10,188
13,785
23,164 8,505 14,18 19,845 206,684
14,076
16,623
9,4597
16,841
14,679
21,935
14,407
14,874
14,505
10,753
16,623
12,599
14,317
9,4597
10,686
194,855
193,027
197,836
232,023
229,836
205,14
284,385
205,235
275,145
107,148
139,099
129,274
130,882
124,06
240,704
126,878
144,737
119,758
179,048
218,78
200,935
170,443
248,969
222,613
150,288
191,357
229,508
193,717
157,913
162,012
240,971 Si
239,359 Si
251,836 Si
291,423 Si
277,356 Si
251,58 Si
362,145 Si
251,675 Si
356,145 Si
129,948 n/a
164,299 Si
158,914 Si
157,162 n/a
136,06 Si
274,724 SiC
141,998 Si
162,737 n/a
144,958 n/a
210,248 n/a
264,38 Si
226,135 Si
204,043 n/a
308,969 Si
273,013 Si
180,288 Si
225,557 Si
288,308 Si
227,317 Si
196,313 Si
199,212 Si
0,023
0,140
0,019
0,150
0,250
0,025
0,140
0,043
0,050
0,060
0,085
0,039
0,035
0,390
0,035
0,032
0,04
0,05
0,053
0,943 0,0315
1,2
1,15 0,0456
1,169
1,169
1,169 0,0333
1,285 0,0183
0,8
1,1
0,88
1
1,2
1,6
1,3
1,55
1
1,5
1,4
2
1,7
1,95
1,63
1,18
1,55
1,4
1,34
1
1,1
0,0778 0,6833 0,0525
0,0714 0,7286 0,0542
0,037
0,053
0,1
0,0654
0,0654
0,0654
0,034
0,034
0,052
0,019
0,04
0,063
0,13
0,06
0,07
0,033
0,075
0,06
0,095
0,04
0,03
0,045
0,035
0,07
0,04
0,06
0,033
0,039
1
1,0856
0,9
1
1
0,925
0,620
1,000
0,710
0,850
0,900
0,770
2,200
0,950
1,650
1,500
1,480
0,950
1,180
0,900
0,775
0,750
0,8375
0,8125
0,92857
Vd @15A
1,5
1,9
2,65
2,1
2,1
8-40kHz
8-41kHz
1,80
1,60
2,5
4,2
1,02
1,15
3,97
0,86
0,82
2,85
4,9
2,8
Err @125°C (mJ/cycle) 15A 600V
2,1
1,7
1,3
1,9
1,2
1,7
31 15 2,05
40 20
31 15 2,05
543 1200 134 66
460 1200
1,5
2,6
4-20kHz
4 1-40kHz
1,5 5-40kHz
5-40kHz
2,4 5-40kHz
2,4 4-20kHz
2,60 20-100kH
1,50 5-40kHz
1,7 4-20kHz
1,50 5-40kHz
0,00
Vt @15A
7 in one-pac 180 1200
TO-247
SOT-227
TO-264
600
99 50
80 40
521 1200 120 64
208
APT
2,3
3,4
2,3
2,7
1,5
2
2,6
2
2,3
30 15 2,53
41 21
40 20
41 21
45 24
45 24
51 28
80 40
78 42
60 30
3,2
Psw@6Khz
ISOTOP
CREE TO-220
APT50GF120JRDQ3
600
CID150660
208
IRF
595 1200
350 1200
160 1200
160 1200
300 1200
160 1200
200 1200
CREE 2 x TO-247-3 200 1200
TO-220AB
900
2 x CID100512
TO-247AA
TO-247AA
TO-247AC
TO-247AC
TO-247AC
TO-247AC
TO-247AC
200
200 1200
IRGB15B60KD
IRF
IRF
IRGP20B120UD-E
IRF
IRF
IRG4PH40U
IRGPS40B120U
IRF
IRG4PH50UD
TO-247AC
TO-247AC
595 1200
350 1200
300 1200
Vcesat @125°C & 15A
IRG4PSH71U
IRF
IRF
Package
IRG4PF50WD
Ptot @25°C
3,10
Psw@10@kHz
IRG4PH50KD
Vcemax
TO-274AA
Ic @25°C
1,70 5-40kHz
Psw@14kHz
TO-274AA
Ic @100°C
TO-274AC
Vf FWD @ 15A 125°C
IRF
IRF
IRGP30B120KD-E
Fsw
1,30 5-40kHz
Eon @125°C (mJ/cycle) (ns) 15A
600V
1,5
1,8
Eoff @125°C (mJ/cycle) 15A 600V
99 50
Pcond @ 15A 125°C 50% Dutycycle (W)
350 1200
Plosstot @6kHz
595 1200 120 60
Plosstot @10kHz
TO-274AA
Plosstot @14kHz
TO-274AA
FWD
IRF
IRF
Rt @15A
IRGPS40B120UD
IRF
Modelname IGBT
IRG4PSH71UD
Rd @ 15A
IRG4PSH71KD
Manufacturer
IRGPS60B120KD
Appendix D1 - IGBT chart 1
Ptot @25°C
Package
Manufacturer
Modelname IGBT (continue)
Infineon 7 in one-p 130
Infineon 7 in one-p 100
Infineon 7 in one-p 180
Infineon 7 in one-p 150
Infineon 7 in one-p 155
FP15R12W1T3
FP15R12W1T4_B3
FP15R12KE3G
FP15R12KS4C
FP25R12KE3
FP25R12KT3
145
Infineon SixPACK
Infineon 7 in one-p 105
FS25R12KT3
145
200
Infineon SixPACK
Vcemax
1200 40 n/a
1200 40 n/a
1200 40 n/a
1200 30 n/a
1200 25 n/a
1200 28 n/a
1200 25 n/a
1200 40 n/a
1200 35 n/a
1,3 n/a
1,4 n/a
1
2
4
2
2
1,3 n/a
2 n/a
0,9 n/a
0,9 n/a
2,1 n/a
2 1,95 n/a
2 1,35 n/a
2
2 1,35 n/a
2
Vf FWD @ 15A 125°C
BSM 25 GD 120 DN2 Infineon SixPACK
Ic @25°C
FS25R12KE3 G
Ic @100°C
1200 40 n/a
Vcesat @125°C & 15A
165
Fsw
Infineon SixPACK
Eon @125°C (mJ/cycle) (ns) 15A 600V
FS25R12YT3
Eoff @125°C (mJ/cycle) 15A 600V
1,35
3,24
1,62
1,26
1,26
0,9
0,99
0,63
1,8 1,17 1,215
1,71 1,35
1,8 0,99
1,98 1,35
1,62 1,17
1,98 1,26 0,945
1,35 1,58
2,07 1,35
1,35 1,98
1,35 1,53
0,81
1,08
1,04
20,9
1,28 8,518
0,90 9,954
1,04 24,54
1,09 13,05
1,46 14,69
1,27 12,28
0,89 9,608
0,98 15,54
0,96 9,778
0,91 10,51
0,99 11,04
Pcond @ 15A 125°C 50% Duty-cycle (W)
1,26 1,17
Err @125°C (mJ/cycle) 15A 600V
1,7 -84k 2,16 1,35
12,96
21,6
20,7
33,3
21,6
17,1
21,6
12,56 20,93
12,96
11,07 18,45
12,96
10,26
12,56 20,93
12,83 21,38
19,98
14,85 24,75
12,42
10,53 17,55
Psw@10@kHz
1,4 n/a
Psw@6Khz
3
177,5 219,62 Si
219 298,92 378,84 Si
214,4 266,24 Si
29,3 134,09 184,31 234,53 Si
30,24 142,89 194,73 246,57 Si
25,83 219,86 264,14 308,42 Si
30,24 162,56
23,94 158,45 199,49 240,53 Si
29,3 156,61 206,83 257,05 Si
29,93 139,95 191,25 242,55 Si
46,62
34,65 153,53 212,93 272,33 Si
28,98 143,08 192,76 242,44 Si
24,57 135,38
30,24 209,43 261,27 313,11 Si
Psw@14kHz
2
Plosstot @6kHz
25
Plosstot @10kHz
1200 45
Plosstot @14kHz
1200 25 n/a
FWD
205
0,9
1,07
1
2
0,9
0,028
0,93
0,045 0,875
0,12
0,073
0,083
0,061 0,983
0,042 0,875
0,057 1,588
0,032
0,042 1,025
0,054
0,142 1,025
Rt @15A
145
Vt @15A
Infineon SixPACK
1,017
0,985
0,8
1,7
1,01
0,95
0,073
0,9
0,033 0,8625
0,036
0,047
0,027
0,063 1,0125
0,037
0,031 0,9916
0,029
0,027 0,9438
0,03
0,096 0,3077
Rd @ 15A
FS25R12W1T4
Vd @15A
BSM 15 GD 120 DN2 Infineon SixPACK
Appendix D2 - IGBT chart 2
Plosstot @14kHz
794,7
Plosstot @10kHz
832,5
Semiconductor
870,3 Si
Appendix D3 - Mosfet/Diode chart
Conduction-loss @ 15A 125°C 50%
Duty-cycle (W)
0,21
Err @125°C (mJ/cycle)
9,45
Psw@6Khz
123
Psw@10@kHz
1,65
Psw@14kHz
15,75 22,05
Plosstot @6kHz
SiC
SiC
1,70 29nC
1,5
10 n/a
600
SiC
infineon TO-220-2
SDT10S60
75
1,70 38nC
1,5
16 n/a
600
infineon TO-220-2
IDT16S60C
136
SiC
2,50 61nC
2,60 28nC
1,6
1,6
10
5
10
22
CREE
Diode
Package
TO-220-2
CREE
Modelname MOSFET
Manufacturer
C2D05120
Manufacturer
Ptot
C2D10120
Package
SPP17N80C3 Infineon PG-TO220
Ptot
Vds
208
Vds
1200
Id 125°C
1200
Id 160°C
136
Vf 25°C @ 10A
800
Id @25°C
11
Id @100°C
17
Rdson @ 125 °C 15A
Vf 160°C @ 10A
0,6
Vf FWD @ 15A 125°C
312
Q
1
Switching Speed
Semiconductor
TO-220-2
Eon @125°C (mJ/cycle) (ns)
0,24
Eoff @125°C (mJ/cycle)
Appendix E1 - Decoupling Capacitor Snubber Circuit Schematics
Appendix E2 - Restricted Decoupling Capacitor Snubber Circuit Schematics
Appendix F - Layout of components within
enclosure
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