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Transcript

The Zero Voltage Switching(ZVS) Critical Conduction Mode(CRM) Buck Converter with Tapped-inductor Jong-Hu Park and B. H. Cho Department of Electrical Engineering Seoul National University, Seoul, Korea E-mail: [email protected] Abstract – This paper proposes a critical conduction mode buck converter that has soft-switching operation by tapped-inductor. In this converter, both the active switch and the diode have soft switching operation by the resonance between the switch parasitic capacitors and the filter inductor. And, the turn-off switching loss of the active switch is reduced by the decrease of the turn-off current. In addition, tapped-inductor changes the conduction operation, which makes some advantageous conditions that diminish the conduction loss and device stresses. Therefore, this topology, ultimately, achieves both the power and cost efficiency by the simple variation of the classical structure. For the proposed converter evaluation, this paper provides the operation analysis of the converter and the hardware verification by 50W prototype operating at maximum 70kHz. I. INTRODUCTION Critical conduction mode(CRM) control is a control strategy in which the active switch turns on when the inductor current falls into zero point to remove the freewheeling diode reverse recovery. This control operates on the boundary condition between continuous conduction mode(CCM) and discontinuous conduction mode(DCM) with variable switching frequency [1-2]. Even though this CRM control has some merits, the usage is restricted by the increasing switching loss of the active switch as the power capacity or the switching frequency goes up. The relatively small filter inductance of CRM converter, compared with CCM, leads to the high peak current of the active switch that augments the switching and conduction losses. For a low voltage, high current applications, the high switching current problem becomes more serious. Moreover, the high current rating semiconductor switch has a large parasitic junction capacitor in general, the discharging loss of the parasitic junction capacitor becomes significant at high switching frequency. In order to overcome these problems, this paper presents the tapped-inductor scheme for the CRM buck converter. Tapped inductor configurations have been applied to the conventional DC-DC converter in the previous researches [37]. Most non-isolated DC-DC converter circuits are based on the three basic topologies – the buck converter, the boost converter, and the buck-boost converter, and, the tappedinductor versions of these circuits have the benefits that the duty cycle of the converter can be adjusted to a desirable value by controlling the tapped winding ratio to prevent the extreme duty cycle at the high step-up or low step-down ratio [8]. This extra degree of design freedom enables the semiconductor devices to avoid a high-peak current, and, contributes to the reduction of switching loss and conduction loss of the CRM converters. In this paper, the ZVS CRM buck converter with the tapped-inductor is proposed. The converter alleviates the drawbacks of the conventional CRM buck converters. The proposed scheme also provides the ZVS operation for the switching devices. And, by optimal design of the winding ratio of the tapped-inductor, both the switching loss and the conduction loss can be minimized. The voltage stress of the diode is also reduced according to the tapped winding ratio. Hence, this converter is suitable for low step-down ratio applications employing MOSFETs. The following sections explain the principle of the proposed scheme and the operating parameter analysis mainly about the efficiency and device stress. And, finally, the results of the 50W prototype converter will be given for hardware verification. II. THE PRINCIPLE OF THE CRITICAL CONDUCTION MODE(CRM) OPERATION WITH SOFT-SWITCHING A. Idea The proposed tapped-inductor buck converter is shown in Fig. 1. The tapped winding ratio is: N=N1/N2 (N≥1), and N=1 is the conventional untapped buck converter case. Figure 2(a) and (b) show the Vs in Fig. 1 waveforms of the conventional and the proposed ZVS buck converter in CRM operation, respectively. As shown in the shaded area in Fig.2, there exists a resonant mode between the parasitic capacitances of the switching devices and the filter inductor in CRM operation. However, the resonant capacitor voltage of the untapped converter, Vs in Fig. 2(a), cannot exceed 2Vo, thus, the soft switching condition cannot be achieved in the untapped CRM buck for the input-output voltage conversion ratio lower than 0.5. Moreover, many emerging applications, such as the high intensity discharging lamp ballast and the battery charger, call for an extreme step-down ratio which makes the CRM circuit far from the optimal operating 0-7803-7769-9/03/$17.00 (C) 2003 IEEE Cd1 N1 Vs + Vin N2 Sw + Vs ZVS Vin RL 2Vo Vo Vdiode Cd2 - - Vs + - t Fig. 1 Proposed tapped-inductor buck converter condition[9]. In this paper, the idea to enhance the resonant voltage of the active switch into the input voltage level Vin by the tapped-inductor is suggested to satisfy the soft switching condition even when a step-down ratio below 0.5 as shown in Fig. 2(b). For the detail description, the equivalent circuit of the resonant mode is shown in Fig. 2(c) with assumption that the output filter capacitance is large enough to be considered as a voltage source, and the parasitic resistance of the resonant network is negligible. The resonant capacitor Cr accounts for the parallel of Cd2 and the reflected Cd1 in Fig. 1, and Lr refers to the inductance of N2. If the initial condition of the resonant mode is: (1) V C ( 0 ) = 0, I L ( 0 ) = 0 where Vc is Vdiode, and IL is the current flow of winding N2 in Fig.1. From this condition (1), the equations of Vc(t) and IL(t) in the resonant mode are derived as follows: V (2) V ( t ) = V − V cos ωt , I (t ) = O sin ωt C O O where, Z = Lr , ω = n Cr L 1 Zn . (3) LrCr For ZVS operation, the winding ratio N of the tappedinductor which steps up the secondary voltage Vc into primary voltage Vs, must satisfy the following condition: (4) max (V S ( t )) = V O + N ⋅ V O ≥ Vin . Thus, the soft switching condition expressed with voltage gain M and turn ratio N is established as: N ≥ 1 −1 . M (5) B. Operation mode analysis The proposed ZVS CRM converter in Fig. 1 has very similar operation to a conventional CRM buck circuit except for the resonant interval at the end of each switching period. The proposed circuit has 3 operation stages – active switch conduction stage, diode conduction stage, and resonant mode stage. The key waveforms of the converter are shown in Fig. 3. The resonant mode starts when the inductor current reaches zero. The time span of the diode conduction mode is the rest period of the active switch turn-off interval except the resonant mode. The resonant interval, Tr is derived from the t (a) (b) IL + Vc - Lr Vo Cr (c) Fig. 2. The voltage waveforms of Vs in Fig. 1, and the equivalent circuit of the resonant mode (Vin > 2Vo) (a) The CRM waveform with untapped inductor (b) The proposed ZVS CRM waveform (c) The equivalent circuit of the resonant mode following condition: V S (Tr ) = V O − N ⋅ V O cos( ω ⋅ Tr ) = Vin . (6) Then, the resonant interval time span, Tr, is: Tr = 1 ω M −1 cos −1 M ⋅N . (7) And, from the voltage-second balance, the voltage gain of the circuit is implicitly established as: fS Tr D (1 − M ) − M ⋅ N 1 − + M ⋅N ⋅D= 2 π T fn where, N ≥ 1 . −1 M ( N ⋅ M )2 − ( M − 1) 2 (8) And also the switching frequency can be derived implicitly using (8). However, the filter inductance Lr is relatively larger than resonant parasitic capacitance, thus IL(t) during the resonant period is negligible as in the discontinuous conduction mode. With this assumption IL(t)=0 (0≤ t ≤ Tr), we can obtain the following simple switching frequency equation: Tr (1 − M ) ⋅ 1 − RL T ⋅ f = L 2 ( − + 1) 2 MN M 2 2 (9) where, the L2 is the inductance of N2. From the inductor current waveform in Fig. 3, the instant the active switch turns off, the magnetic flux in the inductor keeps steady and the inductor turn ratio changes from N1+N2 into N2. Thus, the diode turn-on current jumps into N times of the switch turn-off current. This means that, for the same power transfer, the duty cycle of the switch should be wider than the untapped-case. And, consequently, the peak current 0-7803-7769-9/03/$17.00 (C) 2003 IEEE of the switch becomes lowered. This reduced peak current contributes to the decrease of the switching and the conduction losses of the active switch. In the next section, a qualitative analysis is performed with the aim of forecasting the converter behavior such as the efficiency and the device stress through the voltage conversion ratios. Sw On Off t Vin V s t N I sw,off III. OPERATION PARAMETER ANALYSIS t DTs (1-D)Ts-Tr Tr Fig. 3 Key waveforms of the proposed scheme sw : the gate signal of the active switch, Vs : the buck switch source port voltage, IL : the tapped inductor current A. Voltage gain: Fig. 4(a) Figure 4(a) shows the relation between voltage gain M and duty ratio D according to the various tapped inductor turns ratios. The Figure is derived from Eq. (10). At an extremely low M, the duty cycle of the operating point is increasing as N is increasing, which gives some benefits such as loss reduction and insensitivity of the voltage gain, M, to the duty cycle variation. M vs D I sw,off IL Using the operation parameter formulas derived in previous section, Fig. 4 shows the normalized parameter curves which are derived from dividing the proposed converter parameters by the conventional case(N=1). M = D D + (1 − D ) N ( N ≥ 1, D : duty cycle ) (10) B. Turn-off current of the active switch: Fig. 4(b) The turn-off switching current of the active switch, Isw,off , is derived as: M vs Ioff,norm 1 . M vs Isw,norm 1 2 1 N=1 N=1 0.9 0.9 0.9 0.8 0.8 N=1 0.7 0.5 N=3 0.4 N=4 0.6 0.5 0.6 N=3 0.5 N=4 0.3 0.4 0.4 0.3 0.3 0.2 0.1 0 0 0.1 0.2 0.3 0.4 0.5 D 0.6 0.7 0.8 N=2 0.7 N=3 2 Ioff,norm M N=2 Isw,norm 0.6 0.8 N=2 0.7 0.9 0.2 0 1 (a) Duty cycle vs. Gain 0.1 0.2 0.3 0.4 0.5 M 0.6 0.7 0.8 0.9 (b) Gain vs. switching current M vs Vdiode,norm 0.2 0 1 N=4 0.1 0.2 0.3 0.5 M 0.6 0.7 0.8 0.9 1 (c) Gain vs. active switch conduction loss M vs Vsw,norm 1 0.4 M vs Ploss,sw,norm 4 4 N=1 0.9 3.5 N=4 3.5 N=4 0.8 0.6 N=3 0.5 Ploss,sw,norm Vsw,norm Vdiode,norm 3 N=2 0.7 3 N=3 2.5 N=3 2.5 2 N=2 1.5 2 N=2 0.4 1 N=4 N=1 1.5 0.3 0.2 0 0.5 0.1 0.2 0.3 0.4 0.5 M 0.6 0.7 0.8 0.9 1 (d) Gain vs. diode voltage stress Fig. 4 1 0 N=1 0.1 0.2 0.3 0.4 0.5 M 0.6 0.7 0.8 0.9 1 (e) Gain vs. active switch voltage stress 0 0 0.1 0.2 0.3 0.4 0.6 0.7 0.8 0.9 (f) Gain vs. switching loss Operation parameter curves of the tapped-inductor CRM converter 0-7803-7769-9/03/$17.00 (C) 2003 IEEE 0.5 M 1 (11) 1− M I sw , off = 2 ⋅ I LOAD ⋅ M + ( N ≥ 1) N Vsw , norm And, the normalized parameter, Isw,off, norm is: = M+ 1− M N . (12) Figure 4(b) shows the result of Eq. (12). The drop of the turn-off switching current results in the reduction of the conduction loss as well as the switching loss. C. Conduction loss of the active switch: Fig. 4(c) The active switch conduction loss is proportional to the square of the rms conduction current, Isw,rms , which is: 2 2 I sw , rms M ( MN − M + 1) 4 I LOAD ⋅ N 3 = . (13) And, the normalized parameter Isw,rms, norm is : 2 I sw , rms , norm = ( MN − M + 1) . N (14) From the Fig. 4(c), the reduction of the rms value of the active switch current by the higher duty cycle lowers conduction loss in spite of the same average current as the untapped case. It is also shown that the higher N also effects on the reduction of the conduction loss. D. Conduction loss of the diode The diode conduction loss is approximately proportional to the average conduction current of the diode, Idiode,ave , which is in the tapped CRM converter: I diode, ave = I LOAD (1 − M ) . (15) Idiode,ave is independent of turns ratio N, thus the diode conduction loss is expected the same as that of the untapped CRM converter. E. Device stress: Fig. 4(d), Fig. 4(e) Tapped inductor also affects on the device stresses such as the diode voltage stress, Vdiode , and the active switch voltage stress, Vsw . The diode voltage stress drops by the voltage dividing of the tapped inductor, thus it is dependent on N as the following equation: 1 (16) V = V + (V − V ) . diode O N in O The normalized parameter Vdiode, norm is: Vdiode , norm = M+ 1 (1 − M ) . N = Vin + ( N − 1) ⋅ VO . (18) And, the normalized parameter Vsw, norm is: where ILOAD is the average load current. I sw , off , norm Vsw (17) And, the curve is shown in Fig. 4(d). On the other hand, the active switch stress is increasing as turns ratio N goes up. The relation with N is expressed in (18). = 1 + ( N − 1) M . (19) However, the Figure 4(e) shows that the voltage stress increase is not severe in low M range where the proposed tapped circuits are concerned in. F. Switching loss: Fig. 4(f) The switching loss of the ZVS CRM buck converter takes place in all the switching devices. The diode loss, however, is much minor than the active device loss because of the preclusion of the reverse recovery. On the active switch loss, the turn-off loss is dominant due to the ZVS operation of the switch. For the analysis, we approximate that the loss is proportional to the product of Isw,off and Vsw . Then, the normalized switching loss Psw,loss, norm is established as follows: Psw , loss , norm = I sw , off , norm × Vsw , norm = (1 − M + M ⋅ N )2 . (20) N And Fig. 4(f) displays the result. From the Figure, we know that in a viewpoint of the switching loss, the tapped converter with high N is advantageous to the low gain application. However, notice that the loss is increasing rapidly with voltage gain, thus there is some restriction to the voltage gain range applicable. IV. EXPERIMENTAL RESULTS The proposed ZVS CRM buck converter is implemented to the 50W high-intensity discharge(HID) lamp ballast application for the hardware experiment. The input voltage range of the ballast system is 220VDC ~ 320VDC, and the output voltage is 86VDC. The step-down ratio of the system is about 0.27~0.39. The switching frequency is maximum 70kHz. For these design considerations, the major components used for the hardware prototype are determined as table 1 presents. Table 1. The major components used for the hardware prototype Components Tapped L MOSFET Diode Parameters EI3329, N = 3.4, L1 = 2.3mH, L2 = 196uH IRFBC40 (600V, 6.2A) MUR820 (200V, 8A) Figure 5 and Fig. 6 show the experimental results. In Fig. 5, it is shown that the ZVS operation is achieved in the diode as well as in the MOSFET at the end of the resonant mode. There is no parasitic voltage ringing across the diode of the converter, Vdiode in Fig. 5, by the isolation of the tapped inductor from the turn-off oscillation of the MOSFET. A capacitive snubber parallel connected with the MOSFET can attenuate the ringing voltage of the MOSFET with no dissipation due to the ZVS operation. In Fig. 6, the previous untapped CRM buck converter shows the efficiency decreases 0-7803-7769-9/03/$17.00 (C) 2003 IEEE lower than 96% as the conversion ratio goes down, since a high input voltage drives the operating point into a low duty cycle and a high switching frequency. In contrast, the proposed converter keeps about 97% efficiency level with small variation through the input range. When the experiment was done in the ambient temperature, the temperature of the MOSFET that is TO-220 package in the proposed circuit rises up to 60 °C with no heat sink at the worst condition of efficiency, and in the previous case, the temperature rises up to 70 °C at the same condition. Thus, It is concluded that the efficiency gap between two converter prototypes comes mainly from difference in the active switch operation as shown in the result of the loss analysis in previous section. V. CONCLUSION In this paper, the ZVS CRM buck converter with the tapped-inductor is proposed. By adjusting the tapped winding ratio, the soft switching condition is satisfied, and the turn-off current of the active switch is reduced. Hence, the decrease of the total switching loss in the CRM scheme is achieved. In addition to the improvement of the switching loss, the increase of operating duty cycle at the same low voltage conversion as the conventional untapped version reduces the conduction loss, as well. In this paper, the operation principle of the converter, and the parameter analysis about the efficiency and device stress are included. The experimental verification with 50W hardware prototype is performed to evaluate the proposed topology and the result shows that the proposed converter has higher efficiency than the conventional hard-switching CRM buck type. REFERENCES [1] Felix A. Himmelstoss, Peter H. Wurm, “ Low-Loss Converters with High Step-up Conversion Ratio Working at the Border between Continuous and Discontinuous Mode,” The 7th IEEE International Conference on Electronics, Circuits and Systems, Volume: 2 , pp. 734 – 737, 2000. [2] Panov, Y., Jovanovic, M.M., “ Adaptive off-time control for variable-frequency, soft-switched flyback converter at light loads,” Power Electronics Specialists Conference, 30th Annual IEEE, Page(s): 457 -462 vol.1 , 1999. [3] M. Rico, J. Uceda, J. Sebastian, and F. Aldana, “ Static and Dynamic Modeling of Tapped-Inductor DC-to-DC Converters,” IEEE PESC, pp. 281-287, 1987. [4] Edry, D., Hadar, M., Mor, O., Ben-Yaakov, S., “ A SPICE compatible model of tapped-inductor PWM converters,” Applied Power Electronics Conference and Exposition Proceedings, Page(s): 1021 – 1027, vol.2 , 1994. [5] Wurm, P.H., Himmelstoss, F.A., “ Application of EXCEL for analyzing DC/DC converters,” Computers in Power Electronics, 6th Workshop on , Page(s): 113 – 118, 1998. Fig. 5 The major waveforms of the hardware prototype Measured efficiency (%) 99 98 97 96 95 Tapped - CRM 94 Previous CRM 93 92 220 240 260 280 300 320 Input Voltage(V) Fig. 6 The measured efficiency comparison between the proposed type and the previous type converters [6] Spiazzi, G., Tagliavia, D., Spampinato, S., “DC-DC flyback converters in the critical conduction mode: a reexamination ,” IEEE Industry Applications Conference, Page(s): 2426 –2432, vol.4, 2000. [7] Lio, J.-B., Lin, M.-S., Chen, D.Y., Feng, W.-S. ,” Singleswitch soft-switching flyback converter,” Electronics Letters , Page(s): 1429 – 1430, Volume: 32 Issue: 16 , 1 Aug. 1996. [8] Grant, D.A., Darroman, Y., “ Extending the tappedinductor DC-to-DC converter family,” Electronics Letters , Volume: 37 Issue: 3 , Page(s): 145 – 146, 1 Feb. 2001. [9] Qun Zhao, Fengfeng Tao, Yongxuan Hu, Lee, F.C., “Active-clamp DC/DC converters using magnetic switches ,” IEEE Applied Power Electronics Conference and Exposition, Page(s): 946 –952, vol.2 , 2001. 0-7803-7769-9/03/$17.00 (C) 2003 IEEE