Download Losses in Power Electronic Converters

Losses in Power Electronic Converters
Stephan Meier
Division of Electrical Machines and Power Electronics EME
Department of Electrical Engineering ETS
Royal Institute of Technology KTH
Teknikringen 33
SE-100 44 Stockholm
Abstract— This work is the proposed solution for Task 1,
Problem 1, in the Nordic PhD course on Wind Power, held
in Smøla, Norway, between June 5-11, 2005. It discusses the
converter losses and the expected costs of the back-to-back
converter in a doubly-fed induction generator (DFIG) in a wind
turbine application. Two different topologies of back-to-back
converters are considered: A conventional two-level converter
and a three-level diode-clamped converter.
I. I NTRODUCTION
During the past few years, variable-speed wind turbines
have become the dominant type among newly-installed units.
Variable-speed wind turbines are designed to achieve maximum aerodynamic efficiency over a wide range of wind speeds
by continuously adapting the rotational speed of the wind
turbine to the wind speed. The advantages of variable-speed
wind turbines are an increased energy capture, improved power
quality and reduced mechanical stress on the structure. In order
to achieve variable-speed operation of the wind turbine, the
electric system is getting more complicated. In recent years,
mainly back-to-back converters are being used in the power
conversion field for wind turbines. One solution is to use a fullscale back-to-back converter that allows full variable-speed
operation of the wind turbine at the cost of a large, expensive
and lossy frequency converter that is rated at nominal generator power. This configuration is used by e.g. Enercon. Another
solution is to equip the variable-speed wind turbine with a
DFIG. In the DFIG wind turbine configuration, the stator of the
wound-rotor induction generator is directly connected to the
collection grid whereas the rotor windings are connected to a
back-to-back converter over slip rings. However, this solution
does only provide a limited speed range, depending on the
rating of the frequency converter. A manufacturer using this
configuration is e.g. Vestas.
The advantage of applying back-to-back converters in the
power conversion field for wind turbines is that these converters are completely programmable and due to it, they are very
versatile. This allows different control strategies to control
the active power flow and to both provide reactive power to
the induction generator and to achieve the compensation of
reactive power on the line side. According to [1], the DFIG
system has the advantage that the back-to-back converter needs
only to be dimensioned with a fraction of the rated turbine
power depending on the required speed range. Both the costs
and the conduction and switching losses of the semiconductor
valves are approximately proportional to the converter rating
and are thus decreasing with the same proportion. Also the
converter filters and the filters for electromagnetic interference
(EMI) can be relaxed as they only have to be rated proportional
to the converter rating, which signifies an additional large cost
reduction.
The disadvantage of applying back-to-back converters is that
these electronic devices are relatively expensive and that they
introduce additional losses in the system due to the conduction
and switching losses of the semiconductor valves. Recently, a
new and promising technology was introduced, the multilevel
converters. These type of converters promise improvements
in the harmonic quality of the output voltage which is an
advantage because the output filters of the system can be
relaxed. But at first sight, these converters seem to increase
the cost and the losses of the converters, as the number of
components increases compared to the conventional two-level
converters.
Therefore, this work presents a study of the losses and
the expected costs of two different back-to-back converter
topologies; a conventional two-level converter and a threelevel diode-clamped converter. At first, the problem is defined
properly and it is determined what power flows that can be
expected in both the rotor-side and the line-side voltage source
converter (VSC). Then, the two considered topologies are
presented and the harmonic spectrum in the respective output
voltages are analyzed. Finally, a comprehensive simulation
of the losses in the back-to-back converter is presented. A
basic cost comparison and a summary of the main findings
concludes this work.
II. P ROBLEM
DEFINITION
The active and reactive power flows have to be determined
in order to know the operation status of the back-to-back
converter. Therefore, it is essential to have a generator model
and a basic control system for the two VSCs. The parameters
of the wound-rotor induction generator are given in p.u.-values
and it is very convenient to normalize the voltage-current
equations of the DFIG. In this section, it is also described how
the base values for the simulation were chosen and how the
variables have to be scaled during a transformation between
PSfrag replacements
different reference frames.
Pr
−Qs
Pr
Qr
A. Generator model
In order to determine the power flows, currents and voltages for different operating conditions, i.e. for different rotor
speeds, it is necessary to develop a generator model. The
wound-rotor induction generator used in the DFIG system
comprises a three-phase stator winding and a three-phase rotor
winding, which is fed via slip rings. The used generator
model is chosen according to [2], neglecting the stator and
rotor transients which are not important in this context. The
equations that describe the voltage-current relationship of a
doubly-fed induction generator are given in p.u.-values as:
P = Ps + Pr , Q = 0
Fig. 1.
uqr = −Rr iqr − sωs ((Lrσ + Lm ) idr + Lm ids )
TABLE I
Magnetising inductance Lm
Stator leakage inductance Lsσ
Rotor leakage inductance Lrσ
Stator resistance Rs
Rotor resistance Rr
Stator connection
Rotor connection
(1)
Value [p.u.]
4.0
0.1
0.1
0.005
0.005
Delta
Star
B. Simulation parameters and their normalization
For this work, it is assumed that the rated power SN of
the wind turbine is 1 MVA. The collection grid voltage UN
at the connection point is 690 V, which is a common choice
for wind turbines. The normalized p.u.-values of the woundrotor induction generator can be found in Table I. It is very
convenient to work with normalized values as the control
system and the design process get independent of the actual
generator size. The peak phase voltage and peak phase current
are chosen as the base values, base on which the other base
values of the model can be calculated as shown in Table II.
Ps = uds ids + uqs iqs
Pr = udr idr + uqr iqr
(2)
It has to be considered that all quantities are given in the rotating dq-reference frame, and that the stator windings are delta
connected while the rotor windings are star connected. The
transformation from the stationary three-phase abc-reference
frame to the rotating two-phase dq-reference frame via the
stationary two-phase αβ-reference frame is given as (valid for
both currents and voltages):
2
1
1
uα =
ua − ub − uc
3
2
2
!
√
√
2
3
3
uβ =
ub −
uc
(5)
3
2
2
The absolute values of the stator and rotor voltages, respective
currents, can be calculated as:
q
us = u2ds + u2qs
q
ur = u2dr + u2qr
q
is = i2qs + i2ds
q
ir = i2qr + i2dr
(3)
The power factors cos φ on the rotor and stator side are defined
as:
Ps
Ps
Ps
cos φs =
=p
=
2
2
Ss
us · i s
Ps + Q s
Pr
Pr
Pr
cos φr =
=p
=
2
2
Sr
ur · i r
Pr + Q r
WOUND - ROTOR INDUCTION GENERATOR .
Parameter
In these equations, a synchronous two-phase dq-reference
frame is used, that is fixed to the space vector of the stator
voltage. This is a convenient alternative because the DFIG
operates as a generator being fed with constant stator voltage
(in the dq-reference frame). Hence, the stator voltage and
current are given for line operation of the DFIG system. The
equations for determining active and reactive power flows,
which are defined according to Figure 1, are given as:
Qs = uqs ids − uds iqs
Qr = uqr idr − udr iqr
Active and reactive power flows in the DFIG system.
PARAMETERS OF THE
uds = −Rs ids + ωs ((Lsσ + Lm ) iqs + Lm iqr )
uqs = −Rs iqs − ωs ((Lsσ + Lm ) ids + Lm idr )
udr = −Rr idr + sωs ((Lrσ + Lm ) iqr + Lm iqs )
Ps , Q s
ud = uα cos θ − uβ sin θ
uq = uβ cos θ + uα sin θ
(6)
where θ is the angular position of the rotating dq-reference
frame relative to the stationary αβ-reference frame. However,
the dq-quantities have to be scaled in order to get the same
amplitudes as the phase quantities according to Table III [3].
(4)
2
TABLE II
1
P
M ODEL BASE VALUES .
Parameter
Equation
Value
Base voltage Ubase
= UN
Base power Sbase
Base current Ibase
= SN = 32 Ubase Ibase
2Sbase
= 3U
1 MVA
1.18 kA
Base impedance Zbase
=
0.48 Ω
Base angular frequency ωbase
= 2πfN
√
√2
3
V
PSfrag563.4
replacements
base
Ubase
Ibase
Active, reactive power [p.u.]
0.8
Ps
0.6
Qs
0.4
Pr
0.2
0
Qr
314 rad/s
−0.2
−0.3
TABLE III
Scaling factor
Stator voltages uds , uqs , us
2
√
3 3
2
3√
2 3
3
2
3
2
3
2
3
Rotor voltages udr , uqr , ur
Stator currents ids , iqs , is
Rotor currents idr , iqr , ir
Stator power Ps , Qs , Ss
Rotor power Pr , Qr , Sr
= 0.667
In order to get the operation conditions for different operation points, i.e. for different rotor speeds, a basic vector
control scheme was implemented. Its main purpose is to
guarantee stable operation and enable the independent control
of active and reactive power of the back-to-back converter.
The controller is using the generator model equations derived
in the previous section in the rotating dq-reference frame.
The desired rotor voltage command is determined in order to
control the active and reactive rotor power by controlling the
rotor currents. The line-side converter is controlling the DClink voltage and the reactive power of the total DFIG system,
which is assumed to have unity power factor, i.e. it is neither
absorbing nor generating reactive power (Q = 0).
0.2
0.3
Figure 3 shows the rotor and stator voltages and currents over
the required speed range. It can be seen that the stator voltage
is as expected 1 p.u. Also the stator and rotor currents are
constant over the whole speed range, while the rotor voltage
is approximately proportional to the absolute value of the slip
and becomes zero for zero slip.
In this study, it is assumed that the mechanical rotor speed
is required to have the possibility to change from 0.7 to 1.3
times the synchronous generator speed, which corresponds to
a slip range between -0.3 to +0.3. The slip s of the induction
generator is given as
ωs − ωmech
ωr
=
,
ωs
ωs
0.1
Figure 2 shows the active and reactive rotor and stator power
over the required speed range. It can be noticed that the active
rotor power Pr is flowing through the back-to-back converter,
as it cannot generate, consume or store active power (apart
from the losses that inherently appear). The total active power
generated by the doubly-fed induction generator is the sum
of the rotor and stator active power P = Ps + Pr . With the
chosen DFIG control scheme, the active stator power is kept
constant over the whole speed range while the rotor power
is proportional to the slip. In contrary to the active power,
the back-to-back converter can generate or consume reactive
power, which is utilized in order to get unity power factor at
the connection point of the wind turbine. It can be seen that the
back-to-back converter operates as a generator of active power
above synchronous speed and delivers active power to the grid.
At a slip of s = −0.3, the wind turbine delivers rated active
power to the collection grid. Contrary, below synchronous
speed, the back-to-back converter by-passes active power from
the grid into the rotor circuit and the active power delivered
to the grid becomes approximatively half the rated power at a
slip of s = 0.3.
1.155
0.667
0.667
0.667
C. DFIG vector control
s=
0
Slip
zero, which means that a pure DC current will flow in the rotor.
= 0.385
=
=
=
=
−0.1
Fig. 2. Active and reactive power of the rotor and stator as a function of
the slip.
S CALING FACTORS FOR REFERENCE FRAME TRANSFORMATIONS .
Parameter
−0.2
(7)
III. C ONSIDERED
where ωs is the electrical angular frequency of the stator
quantities (which is constant and equal to the base angular
frequency ωbase ), ωmech is the mechanical angular frequency
of the rotor shaft and ωr is the electrical angular frequency of
the rotor quantities. This equation is valid for an induction
generator with two poles (one pole pair). The number of
electrical poles in the induction generator does not influence
its electrical behavior but changes the requirement on the gear
ratio in the gear box of the wind turbine. It can be noticed
that the electrical angular rotor frequency at zero slip becomes
TOPOLOGIES
The considered topologies for the back-to-back converter
are a conventional two-level converter as shown in Figure 4 and a three-level diode-clamped converter as shown
in Figure 6. The two-level topology is widely used in VSC
transmission systems and in back-to-back converters in DFIG
wind turbines at a wide range of power levels. Figure 5
shows the output waveform of the two-level converter which is
either positive or negative. 1 p.u. voltage corresponds to half
the DC-link voltage. In order to improve the quality of the
voltage output, a pulse width modulation (PWM) switching
3
1
us
0.9
0.7
is
0.6
0.5
0.4
ir
0.3
0.2
ur
0.1
0
Fig. 3.
−0.3
−0.2
−0.1
0
Slip
0.1
0.2
Fig. 4.
0.3
Conventional two-level converter.
1
Voltage and current of the rotor and stator as a function of the slip.
0.5
Voltage [p.u.]
eplacements
Voltage, current [p.u.]
0.8
scheme is used that produces a waveform with a dominant
fundamental component with the compromise that significant
higher-order harmonics are also generated, as shown in the
harmonic spectrum of the two-level converter in Figure 5.
The applied PWM switching scheme is a carrier-based control
method with a switching frequency of 1050 Hz (frequency
modulation ratio p = 21). The amplitude modulation ratio
in Figure 5 is ma = 0.94, which corresponds to the operation
point of the line-side VSC in the back-to-back converter.
0
−0.5
−1
0
2
4
6
8
10
Time [ms]
12
14
16
18
20
0
10
20
30
40
50
60
Harmonic number
70
80
90
100
1
Amplitude [p.u.]
0.8
0.6
0.4
0.2
By splitting up the DC capacitor and the insulated gate
bipolar transistor (IGBT) valves and with the help of additional
diodes, a three-level diode-clamped converter as shown in
Figure 6 can be formed. The output waveform comprises three
voltage levels, i.e. 1 p.u., 0, - 1 p.u. as shown in Figure 7. 1 p.u.
voltage corresponds to half the DC-link voltage that is the
voltage above one of the bus-splitting capacitors. As for the
two-level converter, a carrier-based PWM switching scheme
with an identical frequency and amplitude modulation ratio is
appplied in order to be able to compare the results with the
two-level converter topology. Figure 7 shows the harmonic
content in the waveform, which has a considerably lower
total harmonic distortion (THD). It should be noticed that the
effective switching frequency of the IGBT valves is only half
the one in the two-level converter topology. This is due to the
splitting of the valves and the characteristics of the control
method.
0
Fig. 5. Output waveform and harmonic spectrum of the two-level converter.
Fig. 6.
The advantages and disadvantages of the two-, respectively
three-level converter topologies can be summerized according
to Table VI. The conduction and switching losses as well as the
converter costs and the capacitor size are further investigated
in this work.
Three-level diode-clamped converter.
TABLE IV
C OMPARISON BETWEEN TWO - AND
A. Choice of components
Table V shows the characteristics of the back-to-back converters and the choice of the IGBT semiconductor components
from Semikron [4] and the DC link capacitors from Evox Riva
[5]. Please refer to the corresponding datasheets for further
information about the chosen components.
4
THREE - LEVEL CONVERTERS .
Characteristic
Two-level
Three-level
Circuitry
Control
Capacitor size
IGBT duty
IGBT blocking voltage
Harmonic content
Switching losses
Footprint (size)
Very simple
Very simple
Small
Equal
Large
Large
High
Small
More complex
More problematic
Large
Different
Small (half)
Small
Relatively low
Somewhat larger
1
from the characteristic turn-on and turn-off energy (Eon ,
respectively Eof f ) given in the datasheets. Unfortunately, the
switching losses for the antiparallel diodes are not mentioned
and could therefore not be included in this study. Also the
losses from the reverse recovery energy Err have to be considered. A reverse recovery current is required in order to sweep
out the excess carriers in the anti-parallel diode and allow it to
block a negative polarity voltage. The switching losses are also
dependent on the switched current and the device temperature.
The switching losses Psw can be calculated by summing up
the switching events during a fundamental period according
to
X
X
X
Psw = f
Eon (Ice ) +
Eof f (Ice ) +
Err (Ice )
(9)
Voltage [p.u.]
0.5
0
−0.5
−1
0
2
4
6
8
10
Time [ms]
12
14
16
18
20
0
10
20
30
40
50
60
Harmonic number
70
80
90
100
1
Amplitude [p.u.]
0.8
0.6
0.4
0.2
0
A. Results of the loss comparison
Fig. 7. Output waveform and harmonic spectrum of the three-level diodeclamped converter.
The results of the loss comparison between the two- and
three-level converter topologies is shown in Table VI. Different
operation points corresponding to slip levels between -0.3 and
0.3 are investigated. The total losses are divided in switching
losses, IGBT conduction losses and diode conduction losses
and presented both for the rotor- and line-side converter. The
conclusions from Table VI can be summerized as follows:
• The total losses of the three-level converter are approximately 20 % bigger for all points of operation. This is
mainly due to the dominating conduction losses, which
are increasing by approximately 30 % compared to the
conventional two-level converter. The conduction losses
are contributing with over 90 % to the total losses.
• The switching losses of the three-level converter are
approximately 60 % smaller for all points of operation.
This is a huge improvement but does not influence the
total losses due to their relatively low significance at the
chosen switching frequency of 1050 Hz. However, for
increasing switching frequencies, the switching losses are
getting more important. Another advantage of the threelevel converter is that the low harmonic content allows to
decrease the switching frequency considerably compared
to the two-level converter, which will further decrease the
switching losses.
• It is also interesting to see how the distribution of the
conduction losses between the IGBT and their antiparallel diodes changes depending on the operation point
and the line- or rotor-side converter.
• It is also noticeable that the total losses are the smallest
when the DFIG system is operating near the synchronous
speed. The total losses are slightly increasing with an
increasing slip.
TABLE V
C HOICE OF COMPONENTS .
DC link voltage
1200 V
Semiconductor components [4]
IGBT
IGBT
IGBT
IGBT
module
module
module
module
(2-level
(2-level
(3-level
(3-level
rotor-side):
line-side):
rotor-side):
line-side):
Clamping diode module (3-level):
SKM
SKM
SKM
SKM
500GA123D
400GA123D
400GB066D
300GB066D
SKKD 205F
DC link capacitors [5]
2-level (3 series-capacitors à 400 V):
3-level (6 series-capacitors à 200 V):
PEH200VV447AM 4.7 mF
PEH169RV510VM 10 mF
IV. L OSSES
The losses are calculated in Matlab under the assumption
that the three-phase currents on the rotor- and line- side are
perfectly sinusoidal, which can be assumed as the current
ripple in average will not generate any additional losses. The
total losses consist of conduction and switching losses in the
IGBT and clamping diode modules.
The conduction losses Pcond depend on the on-state voltage
drop across the device and the current through it. They can
be calculated from the on-state threshold voltage Vce0 , the onstate slope resistance rce0 , and the device current Ice according
to
Z f1
2
Pcond = f ·
Vce0 · Ice (t) + rce0 · Ice
(t) dt (8)
t=0
V. C OST
Both the on-state slope resistance and the threshold voltage
depend on the device temperature and were chosen according
to the typical values given in the datasheets. The switching
losses consist of turn-on and turn-off losses of the IGBTs, the
anti-parallel diodes and the clamping diodes in the three-phase
converter topology. The switching losses can be calculated
COMPARISON
A cost comparison ist not simple and would require further
design consideration in order to get accurate results. However,
it is possible to estimate the thendency by watching at the
rating of the semiconductor devices and the size of the DClink capacitors.
5
TABLE VI
L OSS COMPARISON BETWEEN TWO - AND THREE - LEVEL CONVERTER TOPOLOGIES FOR DIFFERENT OPERATION POINTS .
Slip s
Shaft speed ωmech
0.3
0.7 ωs
0.2
0.8 ωs
0.1
0.9 ωs
0
ωs
-0.1
1.1 ωs
-0.2
1.2 ωs
-0.3
1.3 ωs
97.5
475.5
-0.71
1.1
475.5
N.A.
99.2
475.5
0.76
197.2
475.5
0.73
295.8
475.5
0.72
563.4
332.8
-0.18
563.4
328.0
0.0
563.4
333.9
0.18
563.4
349.3
0.34
563.4
374.1
0.48
Electrical phase quantities of the rotor-side converter
Voltage ur [V̂]
Current ir [Â]
cos φr
294.1
475.5
-0.72
196.1
475.5
-0.72
Electrical phase quantities of the line-side converter
Voltage us [V̂]
Current is [Â]
cos φs
563.4
374.1
-0.48
563.4
349.3
-0.34
Losses in the rotor-side converter
Topology
Switching losses [W]
IGBT conduction [W]
Diode conduction [W]
Total [W]
2level
3level
2level
3level
2level
3level
2level
3level
2level
3level
2level
3level
2level
3level
305
1419
572
2295
102
1661
885
2648
305
1313
648
2266
102
1538
1002
2642
305
1205
725
2235
103
1413
1121
2637
320
1130
824
2274
98
1329
1275
2702
305
991
879
2175
151
1161
1360
2673
305
888
953
2146
150
1041
1475
2666
305
784
1028
2117
150
921
1589
2660
2level
3level
2level
3level
2level
3level
2level
3level
2level
3level
2level
3level
2level
3level
250
560
944
1753
119
652
1351
2123
234
593
791
1619
109
694
1138
1941
226
646
670
1541
102
756
965
1823
221
733
578
1532
96
858
833
1788
226
855
511
1593
93
998
737
1828
234
1009
467
1710
94
1173
671
1939
249
1208
439
1897
95
1397
629
2121
211
531
205
541
194
531
244
539
Losses in the line-side converter
Topology
Switching losses [W]
IGBT conduction [W]
Diode conduction [W]
Total [W]
Total losses in the DFIG back-to-back converter
Switching losses [W]
Difference [%]
555
221
539
-60
-61
-61
-64
Conduction losses [W]
Difference [%]
3493
4550
+30
3346
4372
+31
3245
Total losses [W]
Difference [%]
4048
4771
+18
3885
4583
+18
3776
4460
+18
4255
3265
+31
-54
4296
3237
+32
3806
4257
+32
4490
+18
3768
4501
+19
244
-55
3317
554
-56
245
4361
+31
3460
4536
+31
3856
4605
+19
4014
4781
+19
level converter, it is not possible for the three-level converter.
Even the largest available capacitor with 10 mF does not limit
the voltage ripple to below 40 %. It can be noticed that the
DC-link voltage has to be actively controlled by the line-side
VSC in order to keep it in a reasonable range. A comparison
for the chosen configuration shows that the capacitor size
is twice as large for the three-level compared to the twolevel converter topology. Both implemented capacitors have
the same dimensions (75 mm diameter, 145 mm length), but
the number of required components differs with a factor two.
The rating of the semiconductor devices is comparable for the
two different converter topologies. The three-level converter,
however, has an additional clamping diode module for each
VSC. The costs for the gate drive and control system are
also increasing somewhat for the three-level converter, as the
number of IGBTs is twice the one in the two-level converter
and the control of mainly the DC capacitor voltage is more
complex as it is shown below.
The DC capacitor volume will also affect the costs for the two
converter topologies. It has to be calculated in order to limit
the voltage ripple to a comparable level. An acceptable voltage
ripple is 5 %. The size of the capacitance is then determined by
the capacitor current, which is shown in Figure 8 for the twoand three-level converters. It can be seen that the short-time
average current in the two-level converter is approximately
zero, unlike for the three-level converter, where it is varying
considerably. This is due to the different duty ratios of the
semiconductor devices. As expected, this fact has a strong
influence on the voltage ripple, as shown in Figure 9. While the
voltage ripple can easily be limited to below 5 % for the two-
In order to do an appropriate cost comparison, it would also
be essential not only to consider the initial costs but also the
costs due to increased or decreased system losses. However,
this is out of the scope of this work.
VI. C ONCLUSIONS
A conventional two-level and a three-level diode-clamped
converter have been introduced for the application in the backto-back converter of a DFIG wind turbine. A comprehensive
loss evaluation showed that the system losses are lower for the
two-level converter for any point of operation. This is valid
6
2−level converter
Capacitor current [A]
400
200
0
−200
−400
−600
0
0.002
0.004
0.006
0.008
0.01
Time [s]
0.012
0.014
0.016
0.018
0.02
0.014
0.016
0.018
0.02
3−level diode−clamped converter
Capacitor current [A]
400
200
0
−200
−400
−600
0
Fig. 8.
0.002
0.004
0.006
0.008
0.01
Time [s]
0.012
Capacitor current for the 2- and 3-level converter topologies.
2−level converter
DC−capacitor voltage [V]
1230
1220
1210
1200
1190
1180
1170
1160
0
0.002
0.004
0.006
0.008
0.01
Time [s]
0.012
0.014
0.016
0.018
0.02
3−level diode−clamped converter
DC−capacitor voltage [V]
750
700
650
600
550
500
450
0
0.01
0.02
0.03
0.04
0.05
0.06
0.07
Time [s]
Fig. 9.
Capacitor voltage for the 2- and 3-level converter topologies.
for the investigated switching frequency of 1050 Hz, where the
conduction losses are dominating over the switching losses. It
was also shown that the initial costs of the three-level converter
are somewhat increased due to the larger DC-link capacitors
required. The future will show if and in what applications
the obvious advantages of multi-level converters can stand
up to the simplicity and robustness of conventional two-level
converters.
R EFERENCES
[1] S. Müller, M. Deicke, R. W. de Doncker, Doubly Fed Induction Generator Systems for Wind Turbines, IEEE Industry Applications Magazine,
May/June 2002.
[2] Wind Power in Power Systems, Editor T. Ackermann, John Wiley & Sons,
Ltd. 2005.
[3] R. Pena, J. C. Clare, G. M. Asher, Doubly Fed Induction Generator
using Back-to-back PWM Converters and its Application to VariableSpeed Wind-Energy Generation, IEE Proc.-Electr. Power Appl., Vol. 143,
No. 3, May 1996.
[4] Semikron, http://www.semikron.com.
[5] Evox Riva, http://www.evox-rifa.com/europe/index.html
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