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ISSCC 2007 / SESSION 17 / ANALOG TECHNIQUES AND PLLs /173
17.7
A Double-Tail Latch-Type Voltage Sense Amplifier
with 18ps Setup+Hold Time
Daniel Schinkel, Eisse Mensink, Eric Klumperink, Ed van Tuijl,
Bram Nauta
input and output pads (for probe station measurement) was
placed on the same die. The layout of the double-tail SA is shown
in the inset of the chip micrograph in Fig. 17.7.7. An SR-latch is
connected to the output of the SA to create static output signals
without loss of timing information from the core of the SA. When
required, more advanced 'slave' stages could be used in applications [3].
University of Twente, Enschede, The Netherlands
Latch-type sense amplifiers, or sense amplifier based flip-flops, Figure 17.7.4 shows the measured relative delay under different
are very effective comparators. They achieve fast decisions due to conditions (the absolute delay is not measurable due to additiona strong positive feedback and their differential input enables a al delay from the output drivers). As intended, the minimal delay
low offset. Sense amplifiers (SA) are hence widely applied in, for is found at V.,, = l.lV At a Vc, of 0.6V, there is still only a 20ps
example, memories, A/D converters, data receivers and lately also increase in delay. The delay versus AV,, is 44ps/decade under
in on-chip transceivers [1,2]. Voltage-mode SA's, as shown in Fig. nominal conditions. In comparison, measurements in [4] on a con17.7.1, have become especially popular [3-5] because of their high ventional topology in 0.13Rtm CMOS with VDD= 1.5V show a delay
input impedance, full-swing output and absence of static power versus AVi7, of 100 to 170ps/dec and a 250ps increase in delay
consumption.
when V17< is lowered to 0.6V
However, the stack of transistors in Fig. 17.7.1 requires a large The offset in [4] is also very dependent on the V17>. and rises from
voltage headroom, which is problematic in low-voltage deep-sub- 8.5 to l9mV when the V>,, changes from 1.05 to 1.5V. For our
micron CMOS technologies. Furthermore, the speed and offset of design, measurements on 20 samples gave an offset of a,,= 8mV
this circuit are very dependent on the common-mode voltage of at both V., = 1.1V and Vc,, = 0.75V. If desired, area upscaling could
the input V,,,, [4], which is a problem in applications with wide further reduce the offset at the expense of power (P
2)
common-mode ranges, for example A/D converters.
Offset compensation schemes [5] are a good alternative if the
As an alternative, a double-tail sense amplifier is presented here, application allows for the added complexity. The power consumed
which uses one tail for the input stage and another for the latch- by the SA is 113fJ/decision when AVi is 5OmV (f[1k = 1GHz, VDD =
ing stage, as shown in Fig. 17.7.2. This topology has less stack- 1.2V, P = 113gtW f 1GHz, or 225gW @ 2 GHz), which drops to
ing and can therefore operate at lower supply voltages. The dou- 92fJ/decision for full-swing inputs.
ble tail enables both a large current in the latching stage (wide The SAs equivalent input noise was extracted, by measuring the
M12), for fast latching independent of the V,,,, and a small cur- average number of positive decisions versus AV,, as shown in Fig.
rent in the input stage (small M9), for low offset.
17.7.5. Fitting the measurements to a Gaussian cumulative disThe signal behavior of the double-tail SA is also shown in Fig. tribution gives an RMS noise voltage of V,,,,= 1.5mV.
17.7.2. During the reset phase (Clk = 0V), transistors M7 and M8 Setup and hold times are extracted from BER measurements
pre-charge the Di nodes to VDD, which in turn causes M10 and around the zero crossings of the full-swing input patterns, as
Mll to discharge the output nodes to ground (so there is no need shown in Fig. 17.7.6. No bit errors are measured outside an interfor dedicated reset transistors at the output nodes), After the val of 18ps, so the required setup+hold time is smaller than 18ps
reset phase, the tail transistors M9 and M12 turn on (Clk = VDD). (as input jitter is part of the 18ps). A conventional circuit in
At the Di nodes, the common-mode voltage then drops monotoni- 0.18gm CMOS [3] achieves 80ps, which would still be 40ps in
cally with a rate defined by ITMICDi and on top of this, an input 9Onm CMOS according to scaling theory. In the double-tail topoldependent differential voltage AVDi will build up. The intermedi- ogy, the setup+hold time could be further reduced with a wider
ate stage formed by Mi10 and Mll passes AVDi to the cross-coupled tail transistor M9, but at the expense of increased offset and
inverters and also provides additional shielding between the noise due to a shortening of the time that M5/M6 operate in satinput and output, with less kickback noise as a result. The invert- uration. Simulations predict that the current aperture time is
ers start to regenerate the voltage difference as soon as the com- already fast enough to sample data patterns of 4OGb/s, provided
mon-mode voltage at the Di nodes is no longer high enough for that interleaving is used to enable a suitably long regeneration
M10 and Mll to clamp the outputs to ground. The ideal operat- phase.
ing point (V'>,) and the timing of the various phases can be tuned
with the transistor sizes.
In conclusion, the double-tail topology has an added degree of
freedom that enables better optimization of the balance between
To compare the conventional and double-tail SAs, both circuits speed, offset, power and common-mode voltage. The circuit also
are simulated, with transistor dimensions scaled to get an offset has better isolation between input and output and is well suited
standard deviation of a,, = 13mV. The operating conditions are to operate at low supply voltages.
VDD = 1.2V and f,, = 3GHz, and the input has VC11= 1.1V. At this
high V>,, (found, for example, in memories), the conventional Acknowledgements:
authors thank Philips Research for chip fabrication, the Dutch
topology
topology needs reset tranLsistors at the Di nodes
[1,5] to rettniosthDThe
ensure Technology Foundation (STW, project TCS.5791) for funding and Gerard
that M5 and M6 at least start in saturation. The power consumed Wienk for assistance.
by both circuits is similar, about 40fJ/bit. Figure 17.7.3 shows the
delays of both circuits versus the differential input voltage. The References:
J. Rabaey, "Low-Swing On-Chip Signaling
[1] I. Zhang,Effectiveness
V. George and
feedback gives a logarithmic relation between the delay
and Robustness,"
IEEE T VLSI Systems, pp.
Techniques:
and Al">: 37ps per decade for the double-tail SA anLd 37 to '. 264-272, June, 2000.
43ps/dec for the conventional SA. The double-tail SA is both [2] D. Schinkel et al., "A 3Gb/s/ch Transceiver for 10-mm Uninterrupted
faster in general and the delay increases by only 7ps when V,,. RC-Limited Global On-Chip Interconnects," IEEE J. Solid-State Circuits,
297-306, Jan,, 2006.
drops to 0.7V, instead of the 25 to 60ps increase for the conven- pp.
[31 B.Measurements,"
Nikolic et al., "Improved Sense-Amplifier-Based Flip-Flop: Design
tional
simulated at
at VDD=
delay of
of the
thedou
topology. When
tional
When
topology.
VDD = 1Vthe
1 V, the delay
doU- and
IEEE J. Solid-State Circuits, pp. 876-884, June,
st,1imulatedt
ne
posAitivee
r
e
ble-tail SA is lps larger versus 29ps for the conventional SA.
Thne dLouble-tail SA was implementedt in a 1.2 V CMOSlU OOnm tech-
nology, as part of a low-swing on-chip data transceiver that operatesU aound1 V. = 1.1V. The 1, can hae, large variations due to,
for examnple, crosstalk effects. A double-tail SA with dedicated
314
200
2000
[41 B. Wicht at at., 'Yield anld Speed Optimnization of a Latch-Type Voltage
Sense Amplifier," IEEE J. Solid-Stnte circuits, pp. 1148-1158, JulLy, 2004.
[51 K.-Lid. Wong and C.-K.K. Yang, "Offset CompensatDion inl Comparators
with Minimum Input-Referred Supply Noise," IEEE J. Solid-State
circuits, pp. 837- 840, May, 2004.
IEEE11 Internaionlll Soid-lState Circuits Conference
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Authorized licensed use limited to: Ehsan Zhian Tabasy. Downloaded on November 15, 2009 at 04:17 from IEEE Xplore. Restrictions apply.
ISSCC 2007 1 February 13, 2007 /4:15 PM
VDD
VDD
M7
Cl-
L
M12
AVIN: 200.OmV
I\A8
.M
Out+u+
Ot
Out-
Mu
1.2
Clk-.-------Clk~~~~~~
VMM8
M7
SA,Vcm=1
0Convetional
-&- .1V1k0
M6
5
-
~~~~~~~~~~~~~~~~~-0.2 -__
Mlr M3
MIO
L
In+-I
1O.OmV
1.4
-----------~~~~~~~~~~~~~~
0':'D
Di
1 1.4
il
180
Dela.Ot
n0
M5
I
M6
Ou_t-
-0.2________n__....._............
In+-
800.Op
--l Covntoa SAVcDD7
t (s)
n
I 1
n .2n
Figure 17.71: Conventional latch-type voltage sense amplitier. The dotted transistors
Figure 17.72: Double-tail latch-type voltage sense amplitier and signal behavior.
examples ot common variations.
250
Conventional SA, Vcm=0.7V
180
180
Conventional SA, Vcm=l1 lV
AVin=5OmV, Vdd=1 .2V]
IF- Vcm=0.6V, Vdd=1 V
-+-Double-Tail SA, Vcm=0.7V
16
200 -------E Double-Tail SA, Vcm=l .1 V
Vdd=1.2V
are
--Vcm=0.75V,
Vcm=0.75V, Vdd0
----------------2V-----
140
27 12
cn
120
80
120
--------------------
CZ
CZ,
Xia
~
~
~
~~~~~,00
1
CD~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~C
>f>1
T0
CZ
-6
0
50 00
t18
103 E ~~~~~Vn
102
AVin (V)
(V)
10
0
2
0
-20
100
io
1
-4
--
20 --
001
-
102
101
AVin-Voffset (mV)
0.5
1.5
1
Vcm (V)
Figure 17.7.3: Simulated sense amplifier delays versus differential input voltage. The Figure 17.7.4: Measured relative delay versus differential and common-mode input
delay is the time between the clock edge and the instant when AOut crosses 1/2 VDD. voltages.
I
0.8
'
measured
Gaussian, cy =1 .5mV
0.8
,1
I~~
I
~
0 1 0 P attern
PRBS Pattern
;1
/
f
10.-2
10
~~~~
~
~~~~~~~~~~~~~ /11
0.6
EE
0~~~~~~~~~~~~~~~i1
1,0.4r
-6
081
-
0.2
I/
-6
-4
-2
0
2
AVin - Voffset (mV)
1
4
6
Figure 17.7.5: Measured cumulative noise distribution and tit to Gaussian distribution.
;+,t ,~~~~~~~~~~1
l
10
1 8ps setup + hold time
-610
-605
-600
-595
Clock skew (ps)
-590
-585
Figure 17.7.6: Bit error rate versus clock skew, at tcIk= 1GHiz.
Conztinued onz Patge 605
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