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Analog and Mixed Signal Circuits On Digital CMOS Processes Jerry Twomey, Member IEEE Abstract - CMOS processes that have been developed primarily for logic are now increasingly used for analog and mixed signal applications. Process restrictions, wide parametric variance, and noisy application environments lead to special efforts in circuit design/architecture, layout techniques, and optimizing the use of a limited process. This paper addresses a combination of techniques that provide a system development approach for mixed signal circuitry on digital CMOS processes. Index Terms - Mixed analog-digital design, MOS integrated circuits, CMOS analog integrated circuits, analog integrated circuits. I. INTRODUCTION: The motivations for using mixed signal designs on CMOS are multiple. A dominant portion of IC’s now produced are CMOS. Emphasis on high levels of system integration creates wide demand for IC’s that can process both analog signals and digital logic. High production volumes of CMOS motivate a preference for process compatible designs. Cost of CMOS wafer processing is a major consideration. These wafers typically have 7 to 9 masks in their lithography/process, whereas a more versatile design environment such as BiCMOS can have 17 or more. This is reflected in wafer cost. • Methodology in layout and mask design • Empirical evaluation of silicon Each of these areas is addressed, and cross references on topics are provided. II. PROCESS EVALUATION AND USE: Restrictions of a Digital CMOS Process: A digital CMOS process (DCP) optimizes parameters for logic functionality: switching speed, low voltage power supplies, submicron geometry, and high component density. Considerations that address analog and mixed signal circuits are not emphasized in the definition of process parameters. Process elements commonly used in analog signal processing are often left out to keep process costs low. A DCP restricts the process to those elements necessary for logic design. A full set of bipolar elements are not available; resistor structures are limited to process layers used in CMOS transistors. Capacitors are often implemented with the gate capacitance of CMOS transistors. [1,2,3] Process Characterization and Models: Silicon needs analysis for items not usually addressed in DCP models. These include: Mixed signal designs on CMOS requires the use of multiple techniques to achieve success. Items needing special attention include: Capacitors may have significant parasitic elements that provides substrate coupling. These need to be accounted for and included in simulation analysis. • Designing with a restrictive process • Noise isolation • Compensation for wide process variance • Decisions regarding digital/analog architecture Resistors made from isolation wells can be bias sensitive and have substrate leakage current. Resistance can vary as a function of the DC bias to the substrate. These resistors also have parasitic capacitance coupling to the substrate that needs to be part of the element model. Figure 1 briefly illustrates these items. The procedure for development can be divided into several areas: • Optimal use and evaluation of process • Techniques in circuit architecture Page: 1 Capacitor Physical Layout Poly Gate N-Well Substrate N-Well Resistor A N-Well Substrate Equivalent Circuit Poly Gate N-Well Substrate B A B Consequently, voltage headroom, in linear circuits, can limit how the circuit is implemented. CMOS Mixed Signal Elements: Substrate Figure 1. Parasitic elements. Tolerable variance in process parameters is more stringent for analog applications. Variation of elements across multiple wafers and fabrication cycles requires monitoring. Changes in threshold voltages, matching of elements on a common wafer need to be carefully analyzed, so the designer knows the variance that must be compensated for. Minimum geometry elements are used for logic designs. These are often not acceptable for analog signal processing. Parametric variance is generally higher for smaller devices. This includes both active and passive elements. Lithography variations on minimum geometry resistors will cause higher variances than larger bodies. These effects need to be analyzed so informed selections can be made about element geometry. Transistor models and simulation tools can have problems not seen in digital simulations. Simulator accuracy of charge conservation and linearity of transistor models are frequently sources of error. Examination of changes in transistor performance due to short channel effects, threshold variance, and body effect, should be done. [4,5,6] Power Supply, Threshold Voltages, and Biasing Circuits: As DCP’s go to smaller geometry transistors, power supply voltage must also decrease. Submicron CMOS is now using 3.3 V or lower for system power. Logic circuits are biased in the triode region or turned off. Analog circuits require bias in the saturation region, which has a higher drain to source voltage. Analog circuits frequently require three to five transistors, connected drain to source, between power and ground. Page: 2 Threshold voltages restrict the number of transistors that can be properly biased. Sometimes a higher supply voltage is appropriate. However, processes can have low breakdown voltages, which restricts the power supply. Circuit elements that are not normally used on DCP’s can be developed. This includes: Substrate tied PNP transistors can be done on many CMOS processes. [7] These devices have been successfully used in Gilbert cell multipliers and matched pairs in differential amplifiers.[8] Isolation wells for CMOS devices can be forward biased and used as diodes. Although tied to the substrate, they are suitable for the development of thermally compensated bandgap references. Capacitor structures are available from the gates of CMOS transistors or adjacent metal layer capacitors. Ground tied capacitors can be developed using diffusion and isolation wells to the substrate. Designs on Transistor Arrays: Some DCP’s use transistor arrays that are standard to multiple IC’s. These transistors are minimum geometry, suitable to logic designs. These “gate arrays” use a unique set of metal layers to define circuitry. Analog designs on gate array processes can be deficient, because the designer cannot optimize the layout. However success can be achieved in certain applications. The primary restriction of the gate array is its fixed geometry. Variable channel widths and length can be done using transistor arrays. (see Figure 2) DC bias results are similar, but distributed capacitance in the layout can degenerate performance. Parallel transistors to increase channel width are frequently done; series arrays to increase effective channel length are not commonly used. Process parameters for such elements need characterization to show viability. [2] Drain Drain Gate Bulk Bulk Bulk Gate Bulk Source Source Channel width extension Channel length extension Figure 2. Use of fixed geometry transistors. Designs with transistor arrays have greater success at lower frequencies. However, selective layouts, shielding, and noise isolation efforts cannot be done on these devices. Transistor arrays do not allow optimal designs. Designs that can generate custom layouts are preferential. Analog signal processing is more susceptible to noise and process variances. Complexity of digital implementations is sometimes increased, but die area of digital implementation is often smaller. Use of a digital equivalent on a DCP is recommended whenever possible. Analog Cells of Primary Importance: Due to architecture and portability considerations, a set of robust designs for converters between digital and analog are of primary importance. Converter accuracy needs to be maintained on a process that can have a large amount of variance. [9,10,11,12,13] Other analog cells necessary include: • Variable gain control cells: Provides amplitude range for the converter inputs. Gilbert cell multipliers or stepped gain amplifiers under digital control are both feasible. [7,14] • Continuous time filters for: ADC inputs, to prevent aliasing due to Nyquist rate limitations. DAC outputs, to minimize quantization noise on converter outputs. [15,16,17,18,19,20] III. CONSIDERATIONS IN CIRCUIT ARCHITECTURE: Design Portability: Portability of a design to a new set of process parameters becomes important due to the rapid evolution of DCP’s. These processes are undergoing reduction of transistors size, lowering gate threshold voltages, and smaller power supply voltages. At the gate level, synchronous logic designs are largely immune to these changes; analog circuitry is not. Digital designs allow easier portability between processes and has become largely automated, based upon functional definition in some form of hardware description language. Analog designs require more analysis and validation. Analog or Digital Architecture: Decisions on circuit architecture depend largely on the ability to quantize the signal. If acceptable signal conversions between analog and digital can be performed, digital signal processing can usually provide a suitable transfer function. Accuracy and sampling rates of converters become the limiting factors of this strategy. Ground Noise and Distribution of Control Bias: Ground reference across the IC will have dynamic variance due to substrate currents and inductance of the metal layer ground connections. This results in noise from switching transients between grounds across the IC. Differential probing of ground on opposite sides of a logic IC illustrates the problem. Presence of ground noise dictates many of the design mechanisms used on these IC’s. Distribution of bias controls is commonly done by two methods: • Distribution of reference current. • Distribution of reference voltage. Small induced voltages can lead to a large amount of current modulation in a current mirror. The equivalent circuit model is shown in Figure 3. Page: 3 Reference Current Distribution of a reference voltage VSS Transistors separated by distance on the IC VSS Accuracy and Modulation of Current Mirror is dependent upon potentials developed across the substrate. VDD Distribution of a reference current VSS External filter capacitors with inductive component Noise between separated grounds. Reference Current Transistors separated by distance on the IC Power bondwire and package inductance External VDD Mirror Current Internal VDD Package Capacitors Internal Filter Capacitors Ground bondwire and package inductance External VSS Internal VSS Figure 4. Equivalent Network for a power pin on an IC. Mirror Current Noise between separated grounds. VSS External filter capacitors need to be examined for the frequency where they become ineffective. Due to physical structure, all capacitors have reactive components that cause variance from an ideal response. Inductive resonance of ceramic surface mount capacitors is shown in Figure 5. Above resonance, impedance of the capacitor is dominated by its inductance, and is not an effective filter. Figure 3. Distribution of control bias across the IC. Power Isolation: Logic causes high frequency noise on the power supply. Isolated power supplies for analog circuits should be considered mandatory. Power supplies will not always be quiet due to internal noise coupling. Some noise will be present on all nodes, and a multitude of techniques can be used to reduce the net effect of this. This includes circuit architecture internal to the IC, circuitry external to the IC, and internal die layout. [21,23] Capacitor self resonance for ceramic surface mount capacitors Self-resonant frequency characteristics 10 Self-resonant frequency (MHz) Distribution of a current across the IC avoids current source modulation. The bias current is channeled through a diode connected device at the local reference ground. This provides a local voltage reference for control of the current source. 1000 C1608 10 C2012 C3216 10 1 1 10 100 1000 10000 Capacitance (pF) 100000 Distributed Filtering: Noise coupling is a distributed process. Isolation and filtering, to be most effective, need to also be distributed throughout the system. This involves filtering both power and signal/control nodes. Low frequency power filtering can be external to the IC. However, the IC package and bond wire has inductance that degenerates performance of external filters at high frequencies. Figure 4 shows the equivalent network for an IC package pin. Page: 4 Figure 5. Equivalent network for a ceramic surface mount capacitor. Distributed values of filter capacitors allow overlap of resonance points, minimizing self resonance effects. External filters need close proximity to the IC, to minimize inductance of the connection. Logic with rise/fall transients under 1ns lead to spectral noise that defies external filters. Capacitors on the die become necessary for high frequency filtering. Classic Nested VDD VDD Power filtering for devices that use constant or low currents may benefit from RC sections placed between the power supply and the circuit. This will give a quieter power source. Loss of voltage headroom due to the filter, or voltage variations due to current surges needs review for viability. Long Channel Device Noise Filtering of Control Signals: In addition to power filtering, use of filters on control signals will reduce noise. Figure 6 shows a common implementation. This gives both differential and common mode filtering relative to the local ground. The response of the filter is ineffective to the spectral content of the control signal. However, the filter provide attenuation for the clocking frequencies of the logic. Reference Current Mirror Current Reference Currents Mirror Current Note: Classic Cascode requires 2*Vth to bias properly. Nested Cascode gives performance improvement of cascode, and can use less bias headroom. Figure 7. Comparison of Classic Cascode and Nested Cascode. Differential Circuits: VSS CONTROL_A + CONTROL_B VSS Note: 1. Spectral content of “CONTROL A/B” is below the filter’s frequency at which it starts to attenuate. 2. Switching noise, normally high frequency, is prevented from entering the amplifier. Figure 6. Use of distributed filtering on control signals. Active filters are often not effective here. High frequency noise can couple through the parasitic capacitances of an active filter. RC filters are simple to implement, but effective in limiting logic switching noise. Systems operating in a noisy environment will greatly benefit by using differential signals to minimize noise coupling. Differential signals between stages, with the lines routed together, will have common mode noise, which can be removed by the receiver. A differential transmitter and receiver can be used, or, a ground referenced transmitter and a differential receiver. The differential receiver processes both the control signal and the transmitter’s ground reference. (Figure 8) Both methods allow the common mode noise reduction of a differential signal path, and remove ground noise effects. Fully differential system: - Cascode Isolation: Current sources from the power supply will have a greater isolation from noise with the use of a cascode configuration. Cascode structures exhibit higher impedance, and more closely approaches the ideal current source that is desired. Headroom restrictions can inhibit cascode use due to the voltage required for biasing. Classic cascode structures use twice the threshold voltage for biasing. Nested cascode structures (Figure 7) can reduce bias headroom losses. + + + - Amplifiers separated by distance on the IC - Differential receiver, processes control signal and remote ground: + - + Amplifiers separated by distance on the IC - Figure 8. Passing of Differential Signals. Page: 5 Differential receivers should be designed for a large amount of common mode signal rejection. 9(a) Bonding Pads Signals off Chip, and Isolation for Control/Observation: Circuits going outside the IC will usually have increased noise coupling. Designs should review the need to exit the IC and minimize the use of external signals. Any noise sensitive signal should have a minimal amount of routing attached. Common examples are: VCO inputs in phase locked loops, comparator inputs, and the inputs to op-amps. These can have a large amount of gain and be very noise sensitive. Frequently the need is for signal observation and not signal control. In these cases, other methods to avoid offchip noise coupling can be pursued. Figure 9(a) shows the problems associated with passing the signal off chip. External noise coupled to the pin cause signal noise. Figure 9(b) passes the signal internal to the IC and uses a buffer amplifier to allow external observation. Figure 9(c) allows signal observation, and the capability to connect to the internal node. However, digital control allows the node to be isolated from external noise if the observation pin is not in use. These methods improve noise performance while allowing signal control/observation. High Gain and High Impedance Circuit Nodes: Connections outside the IC should require a large amount of voltage swing to achieve the desired control. Input circuits with a large amount of gain will be very noise sensitive. Noise present at the bond pads will couple into the receiver. 9(b) Bonding Pads Signal is passed internally. Buffer amplifier allows external observation without adding noise to the signal. 9(c) Bonding Pads Switch Signal is passed internally, can control node through external bond pad if needed. Can turn off outside connection when not needed. Figure 9. Passing signals external to the IC. VSIGNAL Z-source VRECEIVE Z-parasitic Common examples of this are inputs of op-amps and comparators. Op-amp inputs should remain internal to the IC where possible. Comparators should be bandwidth limited or include hysterisis in the design for noise immunity. High impedance nodes will be susceptible to noise coupling. Most MOS circuits use gates of transistors as amplifier inputs. Gate inputs are high impedance and prone to noise coupling. In these cases, the impedance of the circuit that drives the gate should be reviewed. The high frequency impedance of the node needs to remain low for noise immunity. Figure 10 illustrates the concept. Page: 6 VNOISE As: Z-source << Z-parasitic Vnoise minimizes at Vreceive node. Figure 10. Effect of source impedance on noise received. Signals Between Analog/Digital Parts of the IC: Digital controls need to be passed in/out of the analog part of the IC. Some linear systems must remain functional while controls change. These systems need to exclude switching noise in the analog section. Figure 11 shows a three stage isolation method that works reasonably well. Calibration and Alignment: Digital Environment Signal A Analog Environment Q Q Signal B Differential Low Pass Filter Signal C Analog Circuit with Nonswitch Response Signals: Signal A Signal B Signal C Figure 11. Passing of Digital control into Analog environment with minimal noise effects. Limit the slew rate of the digital control signal and provide a differential output. Pass the differential signals through filters with low bandwidth, and referenced to the local analog ground. Make sure that the receiver takes the “soft switched” control and responds in a linear fashion. Input to a comparator would immediately generate more switching noise. A multitude of techniques for offset, gain, and operating point adjustment are available to the designer. A partial listing includes: • Dynamic calibration. • Static calibration under digital control. • Static calibration while testing. A simple example of each is included for illustration: Dynamic Calibration - Figure 12 shows a clocked comparator, AC coupled, with the comparator offset being cancelled by the voltage on the coupling capacitors. Figure 12(a) shows the comparator in “precharge” mode. Inputs are connected to a common voltage, and outputs are fed back to drive the coupling capacitors. This forces the inputs to the comparators crossover point. The capacitors charge up to the offset voltage. Figure 12(b) shows the comparator in “compare” mode. The input is connected to the comparator through a capacitor voltage that stores the input offset voltage. 12(a) This provides the passing of digital control into the analog environment with a minimal amount of noise. Outputs of the differential control are routed as a pair, so transients are complementary, and provide a minimal noise source. IN_PLUS Precharge Mode OUT_PLUS + + VDD/2 Offset - - OUT_MINUS IN_MINUS Element Matching and Offsets: Junction mismatch effects manifest themselves as offsets in amplifiers, and imprecise current mirror matching. Circuit mismatches are more prevalent in DCP devices because threshold voltage variances are larger than those of bipolar junctions. Process control in DCP’s can tolerate wide variances while producing acceptable logic circuits. Larger geometry elements and use of centroid differential amplifiers reduce some variations, but size increase does not always suffice. Larger geometry causes bandwidth reduction due to junction capacitance, or higher currents must be used to retain bandwidth. To avoid these problems, process variances can frequently be compensated for with calibration and alignment circuits. 12(b) IN_PLUS Compare Mode OUT_PLUS + + VDD/2 Offset - - OUT_MINUS IN_MINUS Figure 12. Dynamic calibration/alignment - Offset cancellation in a differential comparator. Static Calibration Under Digital Control - Circuits go through a digitally controlled alignment process for compensation. An example of op-amp gain adjustment is shown in Figure 13. Gain control is a function of resistor ratios. One resistor has a switched R-2R ladder to allow fractional gain adjustments. A digital counter is incremented until the comparator state changes. This Page: 7 value is maintained while the linear circuit is used. A controlled, definable gain can be created. V_SIGNAL To: Alignment Comparator V_REFERENCE Metal masks which includes small geometry fuses can be forcibly opened by a voltage forced by the IC tester. (Figure 15) The resulting high current opens a fuse link. This process of “link blowing” allows permanent adjustments while on the test system. Ground Bus + R R - R Normal Signal Path 2R 2R 2R To Switch Control 2R Power Bus Metal Probe Pad VSS Fuse Link Resistor Bottom Top Resistor Figure 13. Static Calibration - Op Amp Gain Adjustment Circuit. This type of system can be applied to compensate for offsets, gain variance, current source matching, etc. Static Calibration While Testing - The two prior methods can compensate for many process variations. These techniques are comparative, and suitable for many applications. However, definition of a high accuracy voltage requires a metering system as a reference standard. Bandgap references can provide thermal compensation, but will have errors due to offsets in the differential amplifier. Figure 14 outlines a bandgap reference that has static calibration under digital control. BOND_PAD VDD VDD RTOP RTOP BOND_PAD FUSE_LINK FUSE_LINK RBOTTOM VSS RBOTTOM VSS + BANDGAP_OUT Figure 15. Bottom resistor is present to allow forcing of logic high to switch control. Elevation to higher voltage will open link and leave just the top resistor as a pullup. This technique can have reliability problems with the fuse link remaining permanently open. A voltage step will usually open a properly designed link reliably. Slow elevation of voltage until the link opens is much less reliable. Laser trimming is not generally deemed as cost effective in high volume production and techniques such as “zener zapping” are not usually available on DCP’s. Multiple other alignment/calibration methods exist. [24,25,26] Of primary importance is the need to recognize that the wide variance of a DCP requires these elements to produce accurate and consistent functionality. Amplifier Bandwidth: - VSS Figure 14. Static Calibration while Testing - Bandgap with Fusable Link Trimming. Digital control from the IC tester determines the proper digital pattern for a calibrated bandgap. This pattern then requires permanent storage in the IC. Page: 8 Linear circuits with proximity to digital signals require designs that do not respond to switching noise. Active gain stages should be designed with a spectral response suitable only to the signals of interest. Many designs include circuits with bandwidth above the desired signal. Although noise can be eliminated by bandwidth limiting/filtering of just the output circuits, it does cause propagation of noise among internal circuits, and onto power and ground lines. Bandwidth limiting of amplifiers offers the added bonus of power reduction, due to the correlation between lower currents and lower bandwidth. Electrostatic Discharge (ESD) Protection: Diode protection is usually available for ESD potentials below ground. Design of input elements should consider larger geometry devices, allowing current distribution over a wider junction area. Resistors in series with input pins will help to provide ESD current limiting. Latching ESD protection devices used for overvoltage protection on digital input lines are applicable. [27] IV. LAYOUT AND MASK DESIGN: Shielding: Additional noise reduction can be achieved in metal mask design through judicious use of shielding. Use of metal layer shields around analog signals is effective. Figure 16 shows a shield on all sides which is electrically connected by vias and grounded. Passing a signal as a shielded differential pair will lower effects due to common mode noise and variations between local grounds. Substrate Isolation and Guard Rings: Substrate coupling is common to all elements on the IC die. Noise transients generated in digital areas will be present across the chip. Guard rings consist of metal layer connections which are connected to substrate taps. Depending on process characteristics, low resistance diffusions are often used under guard rings to improve isolation. The guard ring creates a barrier to noise coupling. To be most effective, guard rings need to be placed around the noise source, and around circuits which need noise shielding. The effect desired is to reduce the noise from the source, and also reduce the noise at the receiver. A single guard ring will help noise and ground stability somewhat, but multiple rings, with the distributed resistance of the substrate between rings will work better. The reasons for this are illustrated in Figure 17. The single guard ring allows potentials at the perimeter to be connected directly to the other side of the ring. The double ring structure allows noise shielding, but noise couples to the second ring only through the resistance of the substrate. Split Guard Ring Top View Wide Guard Ring Two layers of metal: shield is placed towards substrate. Signal - Signal - Signal + Cross Sectional View Signal + One dimensional model of substrate resistance and guard ring isolation. Three layers of metal: fully enclosed signal path. Figure 16. Differential channel shield. Noise isolation needs to be optimized at the receiver. For this reason the shield should be connected to a ground reference at the receiver only. Shielding of a nondifferential signal will still be susceptible to ground noise. Substrate Resistance Substrate Resistance Figure 17. Split guard rings vs. wide guard rings. Although models for substrate noise have been attempted, accurate models are questionable due to the large number of parasitic elements present. Empirical methods to directly measure noise magnitude on the substrate are available. [22] Placement of Pins and Bonding Pads: Grouping of analog signals away from digital pins is suggested. Digital input/output cells generate large noise transients due to the currents necessary to drive external loads. Ground pins between digital and analog pins are desirable. Page: 9 Grouping of logic signals will also help. Any logic signals that are near the analog pins should be static digital controls. Clocks and high frequency data paths should be kept away. Mixed signal cells, should be placed such that the analog side of the cell is beside the bonding pads that will have analog signals. This minimizes routing of analog signals, and gives less opportunity for noise coupling. Distribution of Filter Capacitance: V. EVALUATION OF SILICON: Filter capacitors can take up die space, but, are needed for noise performance. However, filter capacitors can often be added without affecting die size. As layout designs near completion, the designs should be reviewed for empty areas that are not actively used. These can be filled with filter capacitors. Although a circuit may function properly in simulation, proper functionality can only be validated through silicon testing. Transistor models developed for logic may have sufficient accuracy, but analog designs often require more precise models. Simulation needs to be used as a tool for sensible design. A large amount of die space is used for metal layer routing. Most capacitors are in base layers, underneath the metal. This allows designers to put distributed filter capacitance underneath any area that is only used for metal routing. Evaluation testing needs to be performed over extremes of temperature and power. Performance over process variance can be evaluated using IC’s from different wafers and multiple wafer lots. Power buses for distribution of power and ground are frequently routed beside each other. These can be placed on top of each other. This provides a power filter capacitor between metal layers, and reduces die area. Also, a filter capacitor can be placed under the power bus areas. Design validation should be in a similar environment to that of the final application. This includes a noisy digital environment outside the cell. Figure 18 suggests a test environment. The analog test cell is surrounded by a group of logic cells that are chained together under a common clock. The logic cells are underneath the control of an external clock, which can be swept in frequency. Sizeable power filter capacitors can be included using these methods. Exclusion of Noisy Signal Sources: Routing of digital signal over analog cells will increase noise coupling. Two methods to avoid this are suggested: Some layout tools have blocking layers, to prevent routing. A guard ring at the cell perimeter using all metal layers in its design will make crossover routing impossible. As part of a cell library, the cell and IC designers are often different. For this reason, design of the mixed signal cell should anticipate, and try to circumvent noisy signal routing. Layout and Placement of Mixed Signal Cells: Routing of logic controls out one side of the cell, and analog signals out the opposite side is advised. Page: 10 Analog Cell + - Bond pads at edge of chip Figure 18. Test environment: Analog cell with noisy digital environment around it. This allows functional testing, with noise injection and an empirical evaluation of noise immunity. VI. CONCLUSION: A number of items key to analog design on a DCP have been touched upon. Including: Process Characterization and CAD Models: [4] Y. Tsividis et. al., “MOSFET Modeling for Analog Circuit CAD: Problems and Prospects.” IEEE J. Solid-State Circuits, vol. 29, No. 3, pp. 210-216, March 1994. • Optimal use of a restrictive process • Process evaluation • Threshold and bias limitations • Designs on gate arrays • System architecture considerations • Cells of primary importance for a library • Distributing control bias Bipolar Transistors on CMOS: • Power Isolation and distributed filtering • Circuits for noise isolation [7] T. Pan et. al., “A 50-dB Variable Gain Amplifier Using Parasitic Bipolar Transistors in CMOS,” IEEE J. Solid-State Circuits, vol. 24, No. 4, pp. 951-961, August, 1989. • Control signals that do not disturb functionality • Offsets and mismatching concerns • Calibration and alignment • Layout methods to optimize performance • Silicon evaluation A single document cannot provide in depth coverage of these topics. However, all of these items require consideration for successful designs. VII. ACKNOWLEDGMENTS: My thanks to the members of several groups at LSI Logic: Mixed Signal Design, Mask Design/Layout, Technical Publications, and Modeling/Process Development. Also, my thanks to Christine Ingalls for her help and support. REFERENCES CMOS System Design Considerations: [1] Y. Tsividis, “Analog MOS Integrated Circuits - Certain New Ideas, Trends, and Obstacles,” IEEE J. Solid-State Circuits, vol SC-2, No. 3, pp. 317-321, June 1987. [2] P. Duchene, et. al., “Analog Circuit Implementation on CMOS Semi-Custom Arrays,” IEEE J. Solid-State Circuits, vol. 28, pp. 872-874, July 1993. [3] P. Gray et. al., “Design Considerations for a High Performance 3-um CMOS Analog Standard-Cell Library,” IEEE J. Solid-State Circuits, vol. SC-22, No. 2, pp. 181-189, April 1987. [5] H. Haddara, “Characterization Methods for Submicron Mosfets”, 1995, Kluwer Academic Press. [6] K. Lakshimikumar et. al., “Characterization and Modeling of Mismatch in MOS Transistors for Precision Analog Design,” IEEE J. Solid-State Circuits, vol. SC-2, No. 6, pp. 1057-1066, December, 1986. [8] E. Vittoz, “MOS Transistors Operated in the Lateral Bipolar Mode and Their Applications in CMOS Technology,” IEEE J. Solid-State Circuits, vol. SC-18, No. 3, pp. 273-279, June, 1983. Data Converter Technology: [9] A. Karanicolas, et. al., “A 15-b 1-Msample/s Digitally SelfCalibrated Pipeline ADC,” IEEE J. Solid-State Circuits, vol. 28, No. 12, pp. 1207-1215, December, 1993. [10] B. Razavi, et. al., “A 12-b 5-Msample/s Two-Step CMOS A/D Converter,” IEEE J. Solid-State Circuits, vol. 27, No. 12, pp. 1667-1677, December, 1992. [11] M. Flynn, et. al., “CMOS Folding A/D Converters with Current-Mode Interpolation,” IEEE J. Solid-State Circuits, vol. 31, No. 9, pp. 1248-1257, September, 1996. [12] A. Venes, et. al., “An 80-MHz, 80-mW, 8-b CMOS Folding A/D Converter with Distributed Track-and-Hold Preprocessing,” IEEE J. Solid-State Circuits, vol. 31, No. 12, pp. 1846-1853, December, 1996. [13] P. Yu, et. al., “A 2.5-V, 12-b, 5-Msample/s Pipelined CMOS ADC,” IEEE J. Solid-State Circuits, vol. 31, No. 12, pp. 1854-1861, December, 1996. Variable Gain Amplifier Design: [14] J. Rijns, “CMOS Low-Distortion High-Frequency Variable-Gain Amplifier,” IEEE J. Solid-State Circuits, vol. 31, No. 7, pp. 1029-1034, July, 1996. Continuous Time Filter Design: [15] B. Stefaneli et. al., “A 2-um CMOS Fifth-Order Low-Pass Continuous-Time Filter for Video-Frequency Applications,” IEEE J. Solid-State Circuits, vol. 28, No. 7, pp. 713-718, July 1993. Page: 11 [16] Y. Tsividis, “Integrated Continuous-Time Filter Design An Overview,” IEEE J. Solid-State Circuits, vol. 29, No. 3, pp. 166-176, March 1994. [17] C. Wu et. al., “The Design of CMOS Continuous-Time VHF Current and Voltage-Mode Lowpass Filters with Q-Enhancement Circuits,” IEEE J. Solid-State Circuits, vol. 31, No. 5, pp. 614-624, May, 1996. [18] A. Durham et. al., “High Linearity Continuous-Time Filter in 5V VLSI CMOS,” IEEE J. Solid-State Circuits, vol. 27, No. 9, pp. 1270-1276, September, 1992. [19] T. Kwan et. al., “An Adaptive Analog Continuous-Time CMOS Biquadratic Filter,” IEEE J. Solid-State Circuits, vol. 26, No. 6, pp. 859 - 867, June, 1991. [20] Z. Chang, et. al., “A Highly Linear CMOS Gm-C Bandpass Filter with On-Chip Frequency Tuning,” IEEE J. Solid-State Circuits, vol. 32, No. 3, pp. 388-397, March, 1997. [26] C. Mensink, et. al., “A CMOS ‘Soft-Switched’ Transconductor and Its Application in Gain Control and Filters,” IEEE J. Solid-State Circuits, vol. 32, No. 7, pp. 989-997, July, 1997. ESD Protection: [27] M.-D. Ker, “ESD Protection for CMOS Output Buffer by Using Modified LVTSCR Devices with High Trigger Current,” IEEE J. Solid-State Circuits, vol. 32, No. 8, pp. 1293-1296, August, 1997. General Purpose CMOS Design References: [28] P. Allen and D. Holberg, “CMOS Analog Circuit Design”, 1987, Publisher: Holt Rinehart and Winston, ISBN: 0-03-006587-9. Noise Characterization and Control: [29] R. Gregorian and G. Temes, “Analog MOS Integrated Circuits for Signal Processing”, 1986, Publisher: John Wiley and Sons, ISBN: 0-471-0979-7. [21] M. Ingels et. al., “Design Strategies and Decoupling Techniques for Reducing the Effects of Electrical Interference in Mixed-Mode IC’s,” IEEE J. Solid-State Circuits, vol. 32, pp. 1136-1141, July 1997. [30] R. Unbehauen and A. Cichocki, “MOS Switched Capacitor and Continuous-Time Integrated Circuits and Systems”, 1989, Publisher: Springer-Verlag, ISBN: 3-540-50599-7. [22] K. Makie-Fukuda, et. al., “Measurement of Digital Noise in Mixed-Signal Integrated Circuits,” IEEE J. Solid-State Circuits, vol. 30, No. 2, pp. 87 - 92, February 1995. [31] A. Grebene, “Bipolar and MOS Analog Integrated Circuit Design”, 1984, Publisher: John Wiley and Sons, ISBN: 0-471-08529-4. [23] J. Olmstead, et. al., “Noise Problems in Mixed AnalogDigital Integrated Circuits”, IEEE 1987 Custom Integrated Circuits Conference, pp. 659-662. AUTHOR BIOGRAPHY: Alignment and Calibration Circuits: [24] J. Atherton et. al., “An Offset Reduction Technique for Use with CMOS Integrated Comparators and Amplifiers,” IEEE J. Solid-State Circuits, vol. 27, No. 8, pp. 1168 -1175, August 1992. [25] T. Gabera, et. al., “Digitally Adjustable Resistors in CMOS for High Performance Applications,” IEEE J. SolidState Circuits, vol. 27, No. 8, pp. 1176-1185, August 1992. Page: 12 Jerry Twomey received a BS and MS in electrical engineering from Worcester Polytechnic Institute in 1979 and 1982. He has done mixed signal IC design for mass storage and data communication systems. He is presently at LSI Logic, Milpitas California, as part of their Mixed Signal Design group.