Download Interactive computer aided design of permanent magnet DC motors

IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 31, NO. 4, JULYIAUGUST 1995
933
Interactive Computer Aided Design
of Permanent Magnet DC Motors
David A. Staton, Malcolm I. McGilp, and Timothy J. E. Miller, Senior Member, IEEE
Abstract-A summary of the design theory associated with dc
commutator motors is presented. This is followed by a detailed
description of a Computer Aided Design program written specifically for this class of motor. Tbe software engineering content
is extremely important: Once the engineering design equations
are developed and validated, the productivity of the design
engineer depends critically on the efficiency and user-friendliness
of the software itself. The program described in the main text
permits the integration of the motor design with the design of the
electronic controller, together with the simulation of the wbole
system. When sizing motors for a particular application, speed
of execution and the flexibility to evaluate a wide range of design
options and parameter variations are essential, while absolute
accuracy is only of secondary importance.
I. INTRODUCTION
B
EFORE the wide availability of inexpensive computing
facilities, motor design was carried out using simplified
analytical and lumped circuit calculation techniques, consolidated by empirical and experimental data to account for
saturation, leakage, etc. [ 11-[5]. The original Computer Aided
Design (CAD) programs developed for motor design were
direct translations of the traditional procedures. However, it
was soon realized that the greater calculation speed could be
more fully utilized by developing improved lumped circuit
models to account directly for saturation and flux leakage
paths [6]-[ 101. Over the last decade, there has been a massive
increase in the amount of research carried out on developing
CAD software for the design of electrical machines. Much of
this research has been dedicated to improving the accuracy
of the machine models used in the packages, with particular
emphasis being placed on the use of finite element analysis.
Powerful programs based on the finite-element method, capable of electromagnetic, thermal, and mechanical analysis are
increasinglyused to improve old designs or generate new ones.
Generic simulation packages complement these tools with
the simulation of the power electronic and digital or analog
controllers. In this environment, there is an important role
for simpler “sizing” software to prepare preliminary designs,
and it is all to the good if such sizing programs integrate
a range of capabilities covering electromagnetic, thermal, and
control aspects. While accuracy is not the primary requirement,
speed of execution is paramount, together with the flexibility
Paper IPCSD 95-08, approved by the Electric Machines Committee of the
IEEE Industry Applications Society for presentation at the 1993 IEEE Industry
Applications Society Annual Meeting, Toronto, Ontario, Canada, October 3-8.
Manuscript released for publication January 20, 1995.
The authors are with the SPEED Laboratory, Department of Electrical and
Electronic Engineering, University of Glasgow, Glasgow G I 2 8LT, Scotland.
IEEE Log Number 941 I 123.
to evaluate a wide range of design options and parameter
variations. In this paper, a sizing package for dc commutator
motors, PC-DCM, will be described.
PC-DCM is the newest of the family of CAD packages written specifically to run on a PC [ 111, [ 121, the other packages
being for the design of switched reluctance motors (PC-SRD)
and brushless dc motors (PC-BDC). All the packages have
similar user-interfaces, file-handling, and use common databases. This makes it straightforward for a designer to move
from one package to the next. The software engineering
content is extremely important: Once the engineering design
equations are developed and validated, the productivity of
the design engineer depends critically on the efficiency and
user-friendliness of the software itself. The CAD packages
developed take full advantage of the latest graphical interface
and file handling techniques.
11. DESIGNTHEORY
A. The D 2 L Equation
The fundamental design equation for many types of machine
is the so-called D 2 L equation, which relates the gross output
torque of a motor to the armature volume, specific magnetic
loading (B), and specific electric loading ( Q )
T =T / ~ D ~ L , Q B
where
T gross output torque (Nm),
Q specific electric loading (Nm),
B specific magnetic loading (T),
D, armature diameter (m),
La armature length (m).
This expression holds the key relationships between the
mechanical, electrical, and magnetic design variables. For a
given torque (or power at a set speed), the armature volume
can only be reduced if the values of B or Q are increased.
Alternatively, for a given motor frame, the torque can only be
increased if B , Q. or the armature volume is increased.
Specific Magnetic Loading (B): The specific magnetic
loading is the mean flux density in the airgap, and is equal to
where B, is the airgap flux density beneath a magnet pole,
and ,Om is the magnet arc (electrical radians).
0093-9994/95$04.00 0 1995 IEEE
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS. VOL. 31, NO. 4, JULYIAUGUST 1995
934
6)
MAGNET
MAGNET
1
Fig. 1. Magnetic lump circuit (two-pole motor).
FLUX INCREASE DUE TO MAGNET OVERHANG
60
50
1
-
-MULLARDS DATA
_ _ _ _ 3-D FE DATA
\
Dab
\
ARMATURE
1)
6
40-
2B
30-
8%
m-
-
3.14
157
1.13
P
Fig. 3. 3-d finite element model of magnetlframe overhang [ 6 ] .
1.c4
s
0.78
LINEAR
NON-LINEAR
10 -
1
1.2
1A
1.6
1.8
2
DEMAGNETISATIONCURVE
MAGNET/ARMATURE LENGTH RATIO
Fig. 2. Flux increase due to magnet overhang.
The value of B, is related to the average flux density in
the magnets (B,) by the ratio of the magnet to airgap crosssectional area. In order to account for flux focusing within the
magnet, the magnet area is calculated at a radius one third the
distance from the inner surface of the magnet (i.e., a radius
equal to D,/2
1, lm/3).The magnet area is also adjusted
to account for magnet leakage flux. I, is the radial magnet
thickness and I, is the radial airgap length.
In order to calculate the average flux density within the
magnet, the lumped-circuit shown in Fig. 1 must be solved.
Due to the low remanence of ferrite materials, it is common
practice to let the magnet overhang the armature by around
20% in order to increase the flux-per-pole by up to 15%.
Fig. 2 plots the increase in flux due to a magnet overhanging
the armature. This data was compiled by Mullards Ltd. [4]
from measurements on a range of ferrite motors. In order
to substantiate this data, a series of three-dimensional finite
element calculations have been carried out [6], [7]. Fig. 3
shows a typical mesh and flux plot taken from the analysis.
While the use of magnet overhang is usually cost effective for
low-priced ferrite materials, it is not normally used with more
expensive rare earth magnets.
It is also common to let the magnetic section of the frame
overhang the magnet and armature in the axial plane as shown
in Figs. 3 and 8. The overhanging frame is used to relieve the
flux density in the active section of the frame, as shown in
Fig. 3. This allows a thinner frame section to be used. It is
also beneficial for mounting the brush gear, provides space
for the end windings, and allows less bulky end shields to be
fitted to the motor.
+ +
H w
Hk
MAGNETISING FORCE (Nm)
Fig. 4. PM demagnetization characteristic and load-line.
A permanent magnet is characterized by its demagnetization
characteristic (see Fig. 4), i.e., its B-H characteristic in the
second quadrant. The equation for the linear section of the
B-H characteristic is
Bm = P o P v H m
+ B,
(3)
where
B, magnet remanent flux density,
B, magnet working flux density,
H, magnet working magnetizing force, and
p, magnet recoil permeability.
If the iron sections of the magnetic circuit are not heavily
saturated, the magnet flux can be calculated using a simple
linear-load line technique. The load line is plotted on the
demagnetizationcharacteristic of Fig. 4. It has a gradient equal
to
(4)
where 1; is the extended airgap, A , is the airgap area beneath
a magnet pole, and A, is the magnet area.
935
STATON et al. : INTERACTIVE COMPUTER AIDED DESIGN OF P E R M A ~ N TMAGNET DC MOTORS
1
The extended airgap is used to account for the fact that the
effective reluctance of a slotted airgap is larger than the actual
airgap and is taken in to account by applying the analytical
work of Carter to calculate an airgap extension factor [l], [ 131.
The intercept of the load-line with the demagnetization
characteristic is given by
1mAgIAm
B , = Br
prlb lmAg/Am '
+
l.W
9
0.80
t
o.60
(5)
m
0.40
0.w
For a given magnet material, in order to calculate the specific
magnetic loading ( B ) it is necessary to know the magnet arc,
H (k")
airgap length, and magnet thickness. In order to maximize
B , the magnet arc should be as large as possible, although Fig. 5. Typical PM demagnetization characteristics.
an upper limit will usually be set (typically 120' to 140')
by the need to allow a sufficiently wide neutral zone such
that the conductors undergoing commutation have no induced small, the armature teeth will usually be thin and there will
rotational EMF. The airgap length is usually determined by be large slots. This results in a comparatively low value of B
mechanical tolerances and is typically 0.5 to 1.O mm. The main and a high value of Q. In a rare earth magnet motor, whose
factors that affect the choice of magnet thickness are demag- remanence is large, the armature teeth will usually be wide
netization withstand (see later section), mechanical constraints and small slots. This results in a comparatively large value of
(i.e., the magnet arc will be fragile if 1, is too small) and cost B and a low value of Q.
(rare earth magnets are extremely expensive, therefore 1, will
usually be small).
B. Magnet Materials
If the iron sections of the magnetic circuit become saturated,
Fig. 5 compares typical demagnetization characteristics of
the load-line is no longer linear as depicted in Fig. 4, which the major magnet material types available today. The metal
results in a reduction in B m . In this case, the simple formula alloy materials are seldom used in motors of this type, since
given in (5) is no longer valid, and an iterative technique must their highly nonlinear demagnetization characteristics and low
be used in order to solve the magnetic circuit.
coercivities give them a low resistance to demagnetization.The
Specijc Electric Loading (Qj: The specific electric loading essentially linear demagnetizationcharacteristicsand relatively
is the ampere-conductors per meter of armature periphery
high coercivities of ferrite and rare earth magnets make them
ideally suited to motor applications. There has been continuous
Q = - ZIa
development of ferrite materials since their introduction in
xaD,
1953, such that they are nearing their theoretical optimum [14]
where Ia is the armature current (amperes), a is the number and are well established as the most cost effective material for
of armature winding parallel paths, and Z is the total number many types of motors. A new generation of potentially lower
of conductors in the armature.
cost rare earth Neodymium-Iron-Boron (Nd-Fe-B) materials
The allowable specific electric loading can either be esti- has recently emerged, having a larger energy-densitythan older
mated by the designer from previous experience, or calculated Samarium-Cobalt (Sm-CO) materials. Current grades of Ndfrom an assessment of the achievable conductor current den- Fe-B have a limited operating temperature range due to their
sity, J , based on thermal considerations, and the total copper large temperature coefficients of remanence and coercivity.
area
Currently Nd-Fe-B is still much more costly than ferrite. SmCOis even more expensive and will continue to find use only in
the most technically demanding applications requiring precise
performance over extended temperature ranges.
where A c U is the total copper area in rotor slots and KFF is
the current form factor.
The area available for copper in the slotted armature usually C. Additional Design Constraints
only amounts to 30% of the actual slot area due the conductor
Thermal Constraints: The specific torque is limited ultiand slot liner insulation and wedges and a realizable packing mately by the permitted temperature rise of the most temdensity. The current form factor is included in the equation to perature sensitive element within the motor which is usually
account for the fact that torque is dependent upon the average determined by the winding insulation class. The major sources
armature current but the losses, and thus thermal aspects are of heat are copper losses in the windings, eddy current and
dependent on the rms current.
hysteresis losses in the rotor steel, friction and windage losses,
Relationship Between B and Q: The values of B and Q are and brush contact losses. The subsequent temperature rise is
not independent, since, in general, higher levels of B demand determined by the thermal dissipation of the motor frame
more iron to carry the flux while higher levels of Q demand ( K D P ) In
. PC-DCM: Friction (PFR)and windage (PwI)
increased slot areas, both competing for space, particularly in losses must be estimated on the basis of previous experience;
the armature. In a ferrite magnet motor, whose remanence is iron loss (PFE)
is calculated using specific loss curves for
0
-900
-800
-700
-6W
-500
-4W
-3W
-2W
-IC0
0
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 31, NO. 4. JULYIAUGUST 1995
936
the lamination material (curves resident in the steel database)
and component flux density levels output by the magnetic
lump circuit model; brush loss calculated (PBR)from the
brush current and brush voltage drop (taken from the brush
database); and copper loss (Pc,,) calculated from the rms value
of armature current and winding resistance (Ra).The total
frame surface area available for dissipation (ADP)is calculated
and the winding temperature rise (TR)estimated using the
equation
TR =
+
(Pcu+ PFE+ PBR PFR+ PWI)
.
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M o r cross-sation editor
2 lun t i u i a t i o n
11
3-
4 Output deslgn sheet
5 Steel aat.b&
6 lbgnet database
7 b u s h database
T TorquefSpced charactcrist i c
G Simulation graphs
U Uinding design
S Sizing roitine
I h i n t laminations and grnphs
I
8 User preferences
(8)
PFR is calculated from the input value of TFR (friction
loss torque). Pm71 is calculated from the input values of
WFO,RPMo, NWF and the speed at which the calculation
is performed (WM) according to the equation
9 Return to DOS
Fig. 6 . PC-DCM main menu screen.
(9)
RPMo is the speed at which the windage power loss W,, is
specified.
Demagnetization Constraints: Under a load condition, the
load-line is displaced as shown in Fig. 4. The trailing pole tips
of the magnets are subject to a demagnetizing field and the
leading pole tips to a promagnetizing field. Solving Ampere's
Law for the magnet circuit results in the following expression,
which calculates the maximum level of armature current (Iddm)
which can be supplied to the motor before the magnet working
point at the trailing edge of the magnet is driven beyond the
linear section of the demagnetization characteristic (specified
by Hk in Fig. 4)
I
I
I
Fig. 7. Cross-section editor (radial).
Commutation Constraints: The value of reactance voltage
is the main parameter used to judge the ability of a motor to
commutate satisfactorily. This is the voltage induced in a coil
undergoing commutation, when the current is reversed, which
opposes the change in current. The magnitude of the generated
EMF is proportional to rate of change of current (speed and
current magnitude dependent) and the inductance (dependent
upon the number of turns and the magnetic circuit associated
with a coil in a slot) of the coil undergoing commutation. A
design rule is that this EMF should not be greater than 3 V
for satisfactory commutation. The easiest solution that can be
implemented to reduce the reactance voltage is to increase
the commutator segments (subdivide the coil into sections).
This can be achieved in PC-DCM by increasing the number
of coil sides per layer, but remembering to reduce the turns
per coil proportionally. In addition to this transformer EMF,
there may also be a rotational induced EMF due to the leakage
flux on the neutral axis of the machine. The rotational EMF
is minimized by limiting the magnet arc to between 120" and
140". The combined transformer and rotational EMF can cause
poor commutation, and lead to arcing at the brush contacts and
rapid brush wear, or in severe cases commutation failure.
Mechanical Constraints: Mechanical limitations are imposed by manufacturing considerations, and application
requirements. Typically, manufacturing limitations will set
minimum and maximum magnet thicknesses, maximum
armature length, minimum tooth thickness, etc. Typical
application limitations are standard shaft heights, limits on the
rotor inertia, and overall frame diameterflength constraints.
111. THE PC-DCM DESIGN
PACKAGE
A. General Procedure for Using PC-DCM
On entering PC-DCM, the main menu options listed in
Fig. 6 are available to the user. The general procedure for
using the package is to define the motor geometry using
a cross-section editor. This is not only used to define the
geometry, but also to view and edit the motor cross section,
both in the radial and axial planes (Figs. 7 and 8 show radial
and axial cross sections for a small 120 W, 3000 r/min
ferrite motor). This geometry can also be input using the
template editor (see Fig. 9), which is used to set the winding
STATON er al.: INTERACTIVE COMPUTER AIDED DESIGN OF PERMANENT MAGNET DC MOTORS
937
m.m
cam
I0.m
a.m
2o.m
6.030
1o.m
0.W
1a
3.m
m
2
I
72.391
l.Oo0
Fig. 8.
Fig. 10. PC-DCM winding editor.
Cross-section editor (axial).
Tu
SORng
hrn
Width
LCon
SD
3.508
28.008
68.008
6.808
lotTypc Sericirc
LPrane
Lend1
kndt
15.808
StP
0.958
CSL
1
1
ILcyth
40.800
10.008
2.008
109.808
39.808
SOD
Lnafnet
Lend3
8.400
78.808
20.000
n
1i ndi nf Paraneterr
t
UpIypc
tap
PronlRcg
Prow
1
TC
YDia
62
8.458
lhrou
HBrush
b
2
ontrol Parmeters
RPll
D U C ~
VSupply
1268.809
5e.808
108.908
HrChop
2869.808
29.008
Pisptn
nur
18.00B
2
Xlnd
1.809
Drive
Chopper
2. bo
6bo
am
12:1
ser option Parameters
Haplnp
28.808
1.808
Ydplmp
me
zsomoe
1.990
XPe
1.808
XLenk
IOUnits
Losrlr
XSht
Hctric
9.820
1.8W
1 . m
Fig. 9. PC-DCM template editor.
(the graphical winding editor must be used if the winding is
nonstandard), control, and other parameters.
A graphical winding editor, shown in Fig. 10, is provided
to generate and display winding layouts. The winding is
generated automatically if it has a standard lap or wave
format. Alternatively, custom windings can be "wound" using
a cursor. The program automatically builds a coil list table
as the "winding" proceeds, and during the simulation phase it
calculates the back-EMF waveform, resistance, and inductance
of the complete winding and the waveforms of flux-density in
the teeth and yoke as the rotor rotates. The latter waveforms
are used for the computation of core-losses.
The run simulation option is used to calculate the performance at a singly defined operating point on the torque-speed
characteristic. A view simulation graph facility is provided
to view the waveforms of supply voltage, back-EMF, and
armature voltage and current (see Figs. 11 and 12). An outpur
design sheet facility is used to examine the performance in
fine detail. All the input and calculated design parameters
are listed. In all, over 120 parameters are included and
are divided into Dimensional. Winding, Control, Material
Fig. 11. Simulation v-z wavefonns (Z-kHzchopper drive).
I
U60
I
1.m
1.m
2.M
xm
3.m
I
Fig. 12. Simulation v-i waveforms (semi-converter drive).
Data, Magnet Design, Simulation Output and Miscellaneous
categories. Fig. 13 shows an example of a small section of a
typical output design sheet.
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 31, NO. 4, JULYIAUGUST 1995
938
In the initial stages of a design it may be beneficial to
carry out a sizing exercise based on the D 2 L equation and
calculations of B and Q. A sizing option is provided for this
purpose.
----___---
krlgn:
2 e . w Dcp.c
Lia
0.33 T
d
Q.219 T
x
1.WT
cw
k
1.B9Q
B. Databases
wlrtion Output:-----td
8 . ~ 1HM
2.380 II
18.851 dim
ron)
4.273 U
rush)
1.681 U
d a l 3667.882 rpm
I1
18.124 13
sh
2.3W A
pRirc 19Q.232 Dep.C
MF
RPH
lrms
err
UWrict)
U(Totrl)
2714.088 rpm
102.216 U
73.211 X
5.684 U
17.024 U
28.149 U
5.118 U
7 . 2 ~n m ^ z
2.w m
Tnl
TStsll
kckUlF
Idcmg
8.w nm
5.w nu
85.761 U
28.253 A
i.em
iemo U
urns
108.088 U
2.3@ A
L m
98.088 U
isccllancaur:----------
Fig. 13. PC-DCM output design sheet.
Four databases are used within PC-DCM. There is a main
design database and three material databases. The design
database files contain all the dimensional, winding, and control
parameters required to specify a design. The file also stores
information required to form a link to the appropriate material
databases and the selected material datafiles.
The three material databases contain steel, magnet, and
brush data. The steel datafiles contain B / H and iron loss
curve-fit data. The magnet database contains data required
to model the demagnetization curve together with a limiting
value of magnetizing force ( H k ) beyond which irreversible
demagnetization will result. The brush database contains a
curve fit of the variation in brush voltage drop with brush
current density. All material data is input via specific template
editors and graphical routines are provided for viewing the
appropriate curve fits.
A feature of all the databases is that they contain version
information and future releases of PC-DCM will be able to
read old datafiles and make the relevant changes to the data.
C. Calculations Perlformed by PC-DCM
A variety of drive types are integrated into the package,
which are:
a) dc voltage source
b) PWM chopper
c) Semi-converter phase controller
d) Full-converter phase controller
In calculating the performance of the motor and drive
Fig. 14. Torque-speed characteristic (2-kHz chopper drive).
combination, the following general calculations are performed:
1) Calculate Motor Dimensions: The important dimensional parameters such as airgap length and slot area
are calculated.
2 ) Calculate Airgap Extension: Carter’s coefficient is used
to calculate the effective airgap taking into account
slotting.
3 ) Solve Magnetic Circuit: The magnet working point is
calculated, accounting for the nonlinear magnet circuit,
end effects (magnet and frame overhang) and leakage
(see Section 11-A). The flux-per-pole and component
flux density levels are also calculated.
4) Calculate Winding Parameters: The mean-length-pertum, armature resistance, armature inductance, and inductance of the commutating coils are calculated.
5) Back-EMF Calculation: This is calculated from a
knowledge of the winding design, flux-per-pole, and
speed.
Fig. 15. Torque-speed characteristic (semi-converter drive).
6) Calculate Voltage and Current Waveforms: The supply
voltage is dependent upon the selected drive type (dc,
Rather than just calculating the performance at a singly
chopper, or phase control). The armature current is caldefined point on the torque-speed characteristic, it is possible
culated knowing the drive details, back-EMF waveform,
to calculate and display the entire torque-speed characteristic
armature resistance and inductance, and the speed of
as shown in Figs. 14 and 15.
rotation.
STATON er al.: INTERACTIVE COMPUTER AIDED DESIGN OF PERMANENT MAGNET DC MOTORS
7 ) Calculate Copper Loss: This is calculated from the
armature rms current and winding resistance.
8 ) Calculate Component Weights: The iron, copper, and
magnet weights are calculated. The individual weights
of the armature teeth and core are also calculated and
are used in the iron loss calculation.
9) CaZculate Iron Loss: This is calculated from a knowledge of the component weights, flux density levels and
speed. The variation in wattskilograms with flux density
and frequency is modeled and the relevant coefficients
for the selected steel type are held in the steel database
files.
10) Calculate Friction and Windage Loss: The friction loss
torque is an input variable in the template editor. The
windage loss ( W W I )is calculated using (9).
11) Calculate the Performance: The torque, power, and
efficiency are calculated at the defined speed (RPM).
12) CaEculate the Demagnetization Withstand: The maximum value of current that can be supplied to the
armature before the onset of irreversible demagnetization is calculated using (10).
13) Predict the Temperature Rise: The temperature rise of
the armature winding is estimated from a knowledge
of the total loss, the frame surface area and a value
of dissipation ( K D ~Ultimately,
).
the temperature rise
limits the continuously rated output torque of the motor.
14) Analyze the Commutation Performance: The commutation performance is analysed by calculating the reactance voltage of the coil undergoing commutation
(Section 11-C).
D. Performance at a Singly Defined Operating Point
The Run Simulation option is used to calculate the performance at singly defined operating point on the torque-speed
characteristic. The armature voltage and current waveforms
are calculated and can be viewed. Fig. 11 shows voltage and
current waveforms for the motor shown in Fig. 7, when driven
from a 2-kHz chopper running at 50% duty cycle. The two
characteristics shown in the top section of the graph are the
50% duty cycle supply voltage waveform and the back-EMF
waveform. The characteristic shown in the central section of
the graph is the armature voltage waveform, which is identical
to the supply voltage waveform in this case. This is because the
armature current is continuous with time (no zero component),
which is shown in the bottom section of the graph. If the
armature current is discontinuous with time (having a zero
component), the armature voltage is equal to back-EMF during
periods when both the supply voltage and armature current
are zero. This is in fact the case in Fig. 12, which shows the
voltage and current waveforms for the motor shown in Fig. 7,
operating from a semi-converter phase controller. The phase
controller has a 90" firing angle (turn-on angle) and the same
average voltage and speed as the previous chopper calculation.
E. The Torque Speed Characteristic
When calculating the torque-speed characteristic of the
motor and drive, a range of voltages (dc supply), duty cycles
939
(chopper), or firing angles (converters) can be selected and
multiple characteristics plotted on the same graph. Fig. 14
shows the torque-speed characteristic for the motor shown in
Fig. 7 when supplied from a 100-V, 2-kHz chopper drive with
the duty cycle ranging from 50% to 100% in 10% increments.
The characteristics are virtually linear and equally spaced.
The slight deviation from a purely linear characteristic is due
to the fact that friction, windage, and iron losses vary with
speed, and brush loss varies with armature current. In order
to account for armature reaction, a magnetic lumped circuit
much more complex than that shown in Fig. 1 is required,
and this results in a multi-fold increase in computation time.
Armature reaction is not calculated in the present version of
PC-DCM as the speed of response must be fast enough for
true interactive use. However, with the increased availability
of high-speed 486 E ' s , it may be possible to include a more
complex lumped circuit model in the future. In Fig. 14, there
is a discontinuity near no-load due to the armature current
becoming discontinuous [ 151.
Fig. 15 shows the torque-speed characteristics for the same
motor as used in Fig. 14, but operated from a 157 V-Peak
(100 V-Mean at 0" firing angle), 50-Hz Semi-converter phase
controller. Calculations were performed at firing angles equal
to O", 15", 30°, 4 5 O , 60°, 75", and 90". In this case, the
torque-speed characteristics are far from linear over much of
their section. This is due to the fact that the armature current
is discontinuous over much of the torque-speed characteristic
[15], as shown in Fig. 12. The discontinuous current is mainly
due to the low frequency of the drive, but is also a function of
the armature inductance, which is relatively low in permanent
magnet motors due to the large effective airgap. The spacing
between individual torque-speed characteristics is also far from
constant and is due to the nonlinear variation in average
voltage with firing angle.
F. The Sizing Routine
When starting a completely new design, the designer has the
option of running a sizing routine. A sizing template editor is
used to define the parameters required by the routine. These
include: the required torque and speed, the frame diameter and
length constraints, the allowable temperature rise, the frame
dissipation, component flux density levels, frame and magnet
overhang factors, expected full-load current form-factor, and
demagnetization constraints. The appropriate magnet, steel,
and brush materials must also be selected. The sizing algorithm
is then run.
The sizing routine is a nested-loop iterative solver, with
the outer loop setting current loading, itself dependent upon
the allowable temperature rise, total losses, and the frame
dissipation figure. Central to the operation of the routine is
the D 2 L equation. In the first iteration, rough estimates of B
and Q are made. Using (1) and a value of D,/L,, the first
estimates of D, and L, are calculated. The magnetic circuit
shown in Fig. 1 determines the tooth to slot width ratio and the
frame dimensions. The magnet thickness is determined by the
demagnetization constraints. Once the motor dimensions have
been set, the winding design and losses can be calculated.
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 31, NO. 4, JULYIAUGUST 1995
940
Gowing the total loss, an estimate of the temperature rise is
then made. If this is larger than the specified requirement, Q
is adjusted appropriately- In the second iteration, the
calculated values of B and Q are fed into (1) and the new
values of D~ and L, calculated (with account taken for the
frame constraints). Within a few iterations, a stable solution
should be obtained. At this stage, the user should check the
design using the full computational capabilities of PC-DCM.
[ l l ] T. J. E. Miller and M. I. McGilp, “Nonlinear theory of the switched
reluctance motor for rapid computer-aided design,” Proc. IEE, vol. 137,
Pt.B, no. 6, pp. 337-347, Nov. 1990.
“High-speed CAD for brushless motor drives,” EPE Firenze,
[12] -,
vol. 3, pp. 435439, 1991.
1131 G. Qishan and G . Hongzaan, “Slotted permanent magnet machines,”
Electrical Machines and Power Syst., pp. 273-284, Oct. 1985.
(141 J, Banige, G , m i t e , and C, Tufts, ‘ m e design and application of
ceramic magnets for dc permanent magnet motors,” Magnets In-YourFuture, Oct. 1986.
[ 151 P. C. Sen, Thyristor DC Drives. New York: Wiley-Intencience, I98 I .
Iv. DISCUSSION
AND CONCLUSION
The software-engineeringcontent of motor design continues
to expand. Powerful programs based on the finite-element
method, capable of electromagnetic, thermal, and mechanical
analysis are increasingly used to improve old designs or
generate new ones. Generic simulation packages complement
these tools with the simulation of the power electronic and
digital or analog controllers. In this environment there is
an important role for simpler “sizing” software to prepare
preliminary designs, and it is all to the good if such sizing
programs integrate a range of capabilities covering the electromagnetic as well as the control aspects. While accuracy is
not the primary requirement, speed of execution is paramount,
together with the flexibility to evaluate a wide range of design
options and parameter variations.
The dc commutator motor design program PC-DCM includes many of the basic design calculations normally performed in preparing a new design. The speed and ease with
which design variations can be undertaken gives a completely
new dimension to the design process, and it compresses the
time to produce a design from days into hours.
REFERENCES
[I] M. G. Say, Direct Current Machines. New York: Pitman, 1980.
[2] A. F. Puchstein, The Design of Small DC Motors. New York: Wiley,
1961.
[3] A. E. Clayton, The Performance and Design of DC Machines. New
York: Pitman, 1959.
[4] I. J. Williams, “Why permanent magnet? An introduction to a new range
of permanent magnet dc motors,’’ G.E.C.J. for Ind., vol. 7, no. 3, Oct.
1983.
J. R. Ireland, Ceramic Permanent Magnet Motors-Electrical and Magnetic Design and Applications. New York McGraw-Hill, 1968.
D. A. Staton, “CAD of permanent magnet dc motors for industrial
drives,” Ph.D. thesis, University of Sheffield, England, Aug. 1988.
I. J. Williams, T. S. Birch, D. Howe, and D. A. Staton, “Computeraided design of permanent magnet dc motors,” Contr./Morors/Drives
Con&, 1985, pp. 29-35.
181 D. A. Staton, M. I. McGilp, T. J. E. Miller, and G. Gray, “High-speed
PC-based CAD for motor design,” EPE Brighton, vol. 6, pp. 2 6 3 1 ,
1993.
M. A. Jabbar, “Design and operational aspects of high-speed appliance
motors using ceramic magnets, electric machines and drives,” IEE Cof,
London, Publication no. 282, pp. 311-315, Nov. 1987.
D. B. Jones, “Computer simulations compare brush and brushless dc
motor designs for high power density applications,” PCIM, pp. 20-34,
June 1988.
David A. Staton was born in Chesterfield, England,
on July 29, 1961. He received the B.Sc. (Hons.)
degree in electrical and electronic engineering
from Trent Polytechnic, Nottingham, England, in
1983, and the Ph.D. degree from the University of
Sheffield, England, in 1988.
From 1977 to 1984, he was employed by
British Coal, who sponsored him while he was
undertaking the B.Sc. degree. While at the
University of Sheffield, he developed CAD software
for permanent-magnet dc motors in collaboration
with GEC Electromotors Ltd. From 1988 to 1989, he was with Thorn EM1
Central Research Laboratories, and was engaged in the design of motors for
the Kenwood range of food processors. Since 1989, he has been employed as
a research fellow in the SPEED Laboratory, University of Glasgow, Scotland.
His research interests are in the computer-aided design of permanent-magnet
and reluctance motors.
Malcolm I. McGilp was born in Helensburgh,
Scotland, on March 12, 1965. He received the
B.Eng. (Hons.) degree in electronic systems and
microcomputer engineering from the University of
Glasgow, Scotland, in 1987.
He has been a research assistant in the SPEED
Laboratory, University of Glasgow, since 1987,
mainly working on computer-aided design of
electric drives and power electronic circuit analysis.
%?’
Currently, he is working on creating engineercomputer interfaces for SPEED software and for
interfacing SPEED software to other engineering applications.
Timothy J. E. Miller (M’7&SM’82) ) is a native of
Wigan, U.K. He was educated at Atlantic College,
and the University of Glasgow and the University
of Leeds.
. He had a student apprenticeship with Tube Investments Ltd. From 1979 to 1986, he was an Electrical
Engineer and Program Manager at General Electric
Corporate Research and Development, Schenectady,
NY. His industrial experience includes periods with
GEC (UK), British Gas, ana International Research
and Development. He is currently Lucas Professor
in Power Electronics, and founder and Director of the SPEED Laboratory,
University of Glasgow, Scotland. He is the author of 105 publications in the
fields of motors, drives, power systems, and power electronics.
Dr. Miller is a Fellow of the Royal Society of Edinburgh, Fellow of the
IEE, and a member of Ivy CC, Glasgow.