* Your assessment is very important for improving the workof artificial intelligence, which forms the content of this project
Download Ultra Linear Low-Loss Varactor Diode Configurations for Adaptive RF Systems
Power over Ethernet wikipedia , lookup
Variable-frequency drive wikipedia , lookup
History of electric power transmission wikipedia , lookup
Current source wikipedia , lookup
Mathematics of radio engineering wikipedia , lookup
Nominal impedance wikipedia , lookup
Electrical engineering wikipedia , lookup
Resistive opto-isolator wikipedia , lookup
Switched-mode power supply wikipedia , lookup
Voltage optimisation wikipedia , lookup
Stray voltage wikipedia , lookup
Alternating current wikipedia , lookup
Surge protector wikipedia , lookup
Rectiverter wikipedia , lookup
Anastasios Venetsanopoulos wikipedia , lookup
Buck converter wikipedia , lookup
Power MOSFET wikipedia , lookup
Mains electricity wikipedia , lookup
Electronic engineering wikipedia , lookup
IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 57, NO. 1, JANUARY 2009 205 Ultra Linear Low-Loss Varactor Diode Configurations for Adaptive RF Systems Cong Huang, Student Member, IEEE, Koen Buisman, Student Member, IEEE, Mauro Marchetti, Student Member, IEEE, Lis K. Nanver, Member, IEEE, Francesco Sarubbi, Milos̆ Popadić, Tom L. M. Scholtes, Hugo Schellevis, Lawrence E. Larson, Fellow, IEEE, and Leo C. N. de Vreede, Senior Member, IEEE Abstract—Two linear low-loss varactor configurations for tunable RF applications are compared. The wide tone-spacing varactor stack provides the best linearity for signals with relative large tone spacing like receiver jammer situations. The narrow tone-spacing varactor stack offers the highest linearity for in-band-modulated signals, and is better suited to adaptive transmitters. Both structures make use of a varactor with an exponential ( ) relation, and so the different requirements of transmit and receive chains can be addressed in one technology. Both configurations have been realized in a silicon-on-glass techat 1.95 GHz is from 40 to 200 over a nology. The measured capacitance tuning range of 3.5 with the maximum control voltage of 12 V. The measured OIP3 of both structures are roughly 60 dBm. Index Terms—Adaptive systems, band switching, impedance matching, low distortion, tunable filters, tuners, varactors. I. INTRODUCTION R ECENTLY, reconfigurable RF systems have received considerable attention, resulting in an extensive search for suitable adaptive RF components, which facilitate band switching, antenna mismatch correction and amplifier load tuning for improved efficiency. To meet the strict requirements of these wireless applications, an ideal tunable element will exhibit low loss, low dc power consumption, high linearity, high ruggedness, wide tuning range, high reliability, low cost, low area usage, and is continuously tunable with a high speed. Microelectromechanical systems (MEMS) based switches [1], [2] can provide very low losses while consuming little dc power. When switching in fixed capacitors, very large capacitance ratios, very low loss and very high linearity can be accomplished. However, this approach is most useful when low-switching speed is acceptable and continuous tuning is not W typically required. Also “hot switching” with RF power Manuscript received May 07, 2008; revised July 11, 2008. First published December 12, 2008; current version published January 08, 2009. This work was supported by NXP, Delft Institute of Microsystems and Nanoelectronics (DIMES), and the MEMPHIS Project. C. Huang, K. Buisman, M. Marchetti, L. K. Nanver, F. Sarubbi, M. Popadić, T. L. M. Scholtes, H. Schellevis, and L. C. N. de Vreede are with the Delft Institute of Microsystems and Nanoelectronics (DIMES), Delft University of Technology, 2628CT Delft, The Netherlands (e-mail: [email protected]; [email protected]). L. E. Larson is with the Center for Wireless Communications, Department of Electrical and Computer Engineering, University of California at San Diego, La Jolla, CA 92092 USA (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TMTT.2008.2008978 needs to be avoided to maintain good reliability. In general, MEMS switches require high actuation voltages (15–80 V) and are sensitive to reliability issues, although this has exhibited significant improvement recently. The MEMS varactor [2]–[4] is considered to be less sensitive to reliability issues, and specific implementations can provide for continuous tuning. Nevertheless, these continuous tunable implementations can suffer from intermodulation distortion, since the position of the membrane can move with changes in the modulation envelope of the applied RF voltage [3], [5]. Moreover, its relatively small tuning range or high drive voltage constrains its RF performance. Also, its relatively small capacitance density makes it difficult to implement large valued capacitors. For this reason its application is mostly found above 5 GHz. Other proposed tuning techniques, based on voltage-variable dielectrics, show a much higher capacitance density, while a moderate tuning range at a relatively low voltage can be achieved with a good quality factor. However, the poor linearity properties [6], [7] due to the inherently nonlinear nature of the ferroelectric materials limit their application. Their integration compatibility is also limited. To avoid the limitations of MEMS and BST-based tunable capacitors, low distortion semiconductor varactor based solutions were proposed [8]–[14]. In this study, we focus on the implementation of two extremely linear varactor diode configurations with complementary linearity properties in a single varactor diode technology. These ultra-low distortion components can be utilized for the realization of tunable RF circuits, e.g., matching networks, filters [15], [16], phase shifters [17], [18] etc. Both varactor configurations use antiseries varactor diode configurations, where the diodes share the same exponential depletion capacitance relation. However, the proposed structures differ in their harmonic termination and varactor area ratios, resulting in a fundamentally different linearity behavior versus tone spacing. The first varactor configuration exhibits the highest linearity for modulated signals with wide tone spacing, and is hereafter named the wide tone-spacing varactor stack. This structure is a direct derivation of the earlier proposed high tuning range varactor stack [8]. The second structure has the highest linearity for narrowband modulated signals and is named the narrow tone-spacing varactor stack, and was discussed in [19]. For receiver applications the input signal is typically small in ampli30 dBm) relaxing the in-band linearity requirement. tude ( Therefore, for the receiver, the most troublesome distortion 0018-9480/$25.00 © 2008 IEEE Authorized licensed use limited to: Univ of Calif San Diego. Downloaded on March 8, 2009 at 23:17 from IEEE Xplore. Restrictions apply. 206 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 57, NO. 1, JANUARY 2009 comes from strong out-of-band interferers whose undesired mixing products can fall in-band and pollute the reception of the desired signal. Note that in practical situations these interfering or jammer signals are typically separated from the desired signal by more than 100 kHz. As result the linearity for signals with larger tone spacing is most important. Considering the above, the wide tone-spacing varactor stack is a very attractive candidate for adaptive receivers. On the other hand, the linearity properties of the narrow tone-spacing varactor stack are best suited for the transmit path. Here, complex modulated signals with a high output power need to be handled without introducing channel-to-channel interference or increasing the error vector magnitude due to intermodulation distortion. Since the theory of the narrow tone-spacing varactor stack has been already extensively studied in [19], in this study, we focus in Section II on the analysis of the wide tone-spacing varactor stack, yielding its required varactor area ratio for third-order intermodulation (IM3) cancellation. The remaining fifth-order intermodulation (IM5) distortion for the wide tone-spacing varactor stack has been quantified by a compact expression, while the influence of the center-tap impedances on the linearity as a function of tone spacing has been analyzed. To experimentally verify the theory and compare the resulting performances, both topologies (wide tone-spacing varactor stack and narrow tonespacing varactor stack) have been implemented on the same silicon-on-glass wafer, yielding the first practical realization of the antiseries antiparallel linear varactor topology. The resulting and their structures have been characterized for linearity performance is reported in Section III. Section IV concludes this paper with a comparison of the two varactor linearization techniques. II. WIDE-TONE SPACING VARACTOR STACKS Fig. 1. Schematic of the antiparallel/antiseries connection of the diodes with exponential C (V ) relation. of the diode areas is set to . For simplicity, we assume the frequencies of the two-tone voltage source are iden. The resulting expression for the IM3 is tical (1), shown at the bottom of this page, where (2a) (2b) (2c) are the capacitance Taylor coefficients of each varactor diode with being the reverse applied voltage with a positive value, is the source impedance, is the complex RF center of the two-tone signal, and represents the frequency voltage amplitude of the source signal. Since we aim for integration in the same technology as the narrow tone-spacing varactor stack [19], the proposed configuration will share the doping prorelation [19], i.e., file with the related A. Third- and Even-Order Distortion Cancellation The high-linearity wide tone-spacing varactor stack configuration with four diodes in the antiseries/antiparallel connection is a special case of the high tuning range varactor stack proposed in [8] and is shown in Fig. 1. We consider the IM3 current , which through the source impedance at frequency we will determine using Volterra analysis [20]. The center-tap impedance is set to infinity for all frequency and the ratio components present in the circuit (3) where and are the doping profile dependent coefficients [19]. In (3), is the zero-bias capacitance, while determines how rapidly the capacitance changes with reverse bias voltage . By substituting (3) into (1) and (2), (1) can be simplified as (4), shown at the bottom of this page, which yields the required diode area ratio for IM3 cancellation, namely, . Note that this cancellation condition is independent of (1) (4) Authorized licensed use limited to: Univ of Calif San Diego. Downloaded on March 8, 2009 at 23:17 from IEEE Xplore. Restrictions apply. HUANG et al.: ULTRA LINEAR LOW-LOSS VARACTOR DIODE CONFIGURATIONS FOR ADAPTIVE RF SYSTEMS 207 solving for the fifth-order Volterra series, we assume for the center-tap connections and . Next, we relation (3) into the IM5 formulation. To substitute the simplify the analysis, we assume and , providing us the IM5 products that appear at and , namely, (7) where Fig. 2. Circuit for the computation of second-order kernels of the circuit of Fig. 1. the source impedance and the particular values of and . The antiparallel/antiseries topology enforces opposite phases of the even-order nonlinear current sources so no even-order components leave the varactor structure. To illustrate this point, consider the second-order Volterra series of the schematic shown in Fig. 2. The second-order Volterra kernel for the voltage at input node is is the exponential coefficient of the relation (3) and is the voltage amplitude of the two-tone test signal at the fundamental frequencies. This situation represents the worst , all fifth-order current case condition since, for will flow through the source impedance, which approximates a short-circuit condition. Based on (7), the fifth-order intercept point (IIP5) can be expressed as V (8) The resulting linearity is independent of the reverse bias voltage. In order to be consistent with further discussions, we , which is the peak amplitude of the replace with two-tone input voltage signal. Hence, (7) can be rewritten as (9) (5) where varies from 0.028 to In practical cases, the value of 0.54 V [19], yielding IIP5 values from 5.1 to 98.4 V. Note that since the IM5 drops at the rate of 80 dB per decade, outstanding linearity can be achieved for even modest IIP5 values. C. Influence of Center-Tap Impedance on Linearity (6a) and (6b) are the second-order nonlinear current sources. From (5), it is clear that as long as the mismatch of and is small, will be close to zero, and no second-order distortion current flows into the source impedance. Therefore, no second-order voltage components will develop across the varactor structure and secondary mixing (which can lead to IM3) is avoided; this yields an IM3 cancellation condition independent of source impedance. A more elaborate analysis shows that all even-order distortion components vanish, resulting in a tunable capacitor with no residual distortion with an order lower than five. B. Influence of Fifth-Order Distortion on Linearity Although this approach is very effective in cancelling the IM3 and all even-order intermodulation, IM5 is still present. When In Section II-A, it was assumed that the center-tap impedance is infinite for all frequency components so it has no influence on the RF operation. In practical situations, this requires that the center-tap impedances should be much higher than the ac impedances of the varactors themselves. This requirement is difficult to fulfill for the baseband frequency component of when the tone spacing approaches a two-tone signal zero. For the resulting baseband frequency, the capacitive reactance of the varactors increases without bound as the tone spacing approaches zero. Consequently, there is a lower frequency limit of tone spacing where the third-order distortion cancellation is violated; hence, this configuration is called the wide tone-spacing varactor stack. In practical implementations of the wide tone-spacing varactor stack, similar to what has been done for the uniformly doped distortion-free varactor stack configuration in [8] and [10], an integrated resistor [see Fig. 3(a)] can be used for the dc biasing networks of the center-tap. For a wide tone-spacing varactor stack configuration as depicted in Fig. 1, using resistive center-tap connections, the simulated distortion components at as function of tone spacing are depicted in Fig. 4, which confirms the conclusion above. When the tone spacing is relatively small, the third-order distortion is no longer canceled, and the IM3 distortion has a value of Authorized licensed use limited to: Univ of Calif San Diego. Downloaded on March 8, 2009 at 23:17 from IEEE Xplore. Restrictions apply. 208 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 57, NO. 1, JANUARY 2009 Fig. 3. (a) Wide tone-spacing varactor stack configuration with resistor bias. (b) Wide tone-spacing varactor stack configuration with resistor and antiparallel diode bias for linearity improvement at low-tone spacing. (c) Narrow tone-spacing varactor stack configuration with inductive center-tap connection. 0 Fig. 4. Simulated capacitive current distortion component at 2f f as a func= 1 V, f = tion of input signal tone spacing (V = 2:5 V, V 2 GHz) for a wide tone-spacing varactor stack with an effective zero bias capacitance of 4.2 pF (a = 0:358 V ; X = 2 3; C = 0:1 pF) and single diode with the same zero bias capacitance and a . 0p (10) With the increase of tone spacing, the impedance of the varactor drops and at a certain frequency (above 4 MHz), it becomes much smaller than the center-tap impedance, yielding the improved linearity as predicted by (9). For the resistive center-tap configuration, the corner frequency (marked in Fig. 4), where the linearity starts to approach the fifth-order distortion dominated regime, can be approximated by (11) where is the center-tap resistance. As one can observe, the corner frequency is inversely proportional to the product of the center-tap resistance and the varactor capacitance. In order to improve the linearity performance at low-tone spacing, the center-tap configuration [shown in Fig. 3(b)] with a series resistor and antiparallel diode bias was proposed in [8] and [10], whose effect can be observed in Fig. 4. If the antiparallel diodes are not forward biased, they provide very high impedance extending the high linearity operation of the wide tone-spacing varactor stack to very low-tone spacing. Note that for this configuration, the dc leakage current of the reverse-biased diodes will limit the linearity at ultra-low tone spacing since a small forward bias of the center-tap diodes will . For slightly higher tone yield a drop in their ac impedance of the antiparallel diode spacings, the zero-bias capacitance pair limits the IM3 to a small constant value, consequently it is best to use small diodes in the center-tap connection. Above the corner frequency (in this case approximately 40 MHz), defined by (11), fifth-order distortion constrains the linearity. Note that the corner frequency is ten times larger than the single resistor (100 k ) is ten times smaller, which case [see Fig. 3(a)] since is consistent with (11). Fig. 4 shows that the wide tone-spacing varactor stack clearly offers superior linearity over the single diode (with comparable effective capacitance), especially at relatively large tone spacing. D. Comparison Between Single Diode, Wide Tone-Spacing Varactor Stack, and Narrow Tone-Spacing Varactor Stack The narrow tone-spacing varactor stack and wide tone-spacing varactor stack are compared with single diodes in Table I in terms of topology, harmonic conditions, area ratio required for IM3 cancellation, total area for a given capacitance (relative to the single diode), maximum RF voltage, and linearity. It indicates that both varactor stacks, with suitable harmonic terminations, dramatically outperform the single diode in terms of linearity since all distortion components with an order less than five are cancelled, while the residual fifth-order is extremely low. For the varactor distortion at stacks, the antiseries configuration will distribute the applied RF voltage over the two diodes. Therefore, the maximum , which is constrained by the allowable RF voltage forward bias and breakdown of the individual diodes, will be correspondingly increased, yielding higher power handling and linearity. The only cost of the antiseries connection is the extra chip area needed to implement the same capacitance value. However, due to the high capacitance density of semiconductor Authorized licensed use limited to: Univ of Calif San Diego. Downloaded on March 8, 2009 at 23:17 from IEEE Xplore. Restrictions apply. HUANG et al.: ULTRA LINEAR LOW-LOSS VARACTOR DIODE CONFIGURATIONS FOR ADAPTIVE RF SYSTEMS 209 TABLE I PERFORMANCE COMPARISON BETWEEN SINGLE DIODE, WIDE TONE-SPACING VARACTOR STACK, AND NARROW TONE-SPACING VARACTOR STACK 0 Fig. 5. Measured C V and Q control voltage (V ) at 1.95 GHz. 0V dependence as function of reverse Fig. 6. Microphotograph of the 10-pF-wide tone-spacing varactor stack used in the linearity measurements with 500-k center-tap resistor. varactors (typically 0.5–2 nF/mm ) this is not a problem for most RF applications. III. EXPERIMENTAL RESULTS To confirm the theory of Section II, the wide tone-spacing varactor stack was implemented, along with the narrow 0 Fig. 7. Measured distortion components at 2f f versus output power for single diode and the wide tone-spacing varactor stack with the center-tap resistance of 500 k (for both cases: the effective zero bias capacitance = 10 pF, = 2 GHz, a = 0:0896 V , and V = 5 V). f tone-spacing varactor stack, using the dedicated silicon-on-glass technology [21]–[23] at the Delft University of Technology, Delft, The Netherlands. The starting material is a 4-in silicon-on-insulator wafer on which the specific varactor profile has been grown epitaxially. In this test run, we aimed for varactor devices with a breakdown voltage of 20 V, capacitance tuning range of 6:1, and epi-layer thicknesses of 1.1 m were realized. For these varactors MEDICI [24] simulations predicted a quality factor of 50 at zero bias. A. Measurement of the and Dependence As mentioned earlier, the advantages of the wide tone-spacing varactor stack over the distortion-free varactor stack (with its uniform doping profile) are its higher capacitance tuning range and its compatibility with the narrow tone-spacing varactor stack, which has superior linearity with narrow tone spacing. From the technology point of view, the implementation of the doping profile, and the resulting exponential relationship, is critical for achieving optimum performance. To Authorized licensed use limited to: Univ of Calif San Diego. Downloaded on March 8, 2009 at 23:17 from IEEE Xplore. Restrictions apply. 210 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 57, NO. 1, JANUARY 2009 0 Fig. 8. Measured distortion components at 2f f versus output power for the wide tone-spacing varactor stack with proper area ratio (X = 3:7) and wide tone-spacing varactor stack with 50% area ratio offset (X = 1:9) (for = 2 GHz, both cases: the effective zero bias capacitance = 10 pF, f a = 0:0896 V , and V = 5 V). 0 Fig. 10. Measured distortion components at 2f f versus output power for the narrow tone-spacing varactor stack (the effective zero bias capacitance = 10 = 2 GHz, a = 0:0896 V , pF, inductance at terminations = 20 nH, f and V = 5 V). Fig. 9. Microphotograph of the 10-pF narrow tone-spacing varactor stack used in the linearity measurements (inductance at terminations = 20 nH with the self resonation frequency above 6 GHz). verify the relation, we measured the capacitance density of a single diode with a zero bias capacitance of 1 pF. Its and behaviors are shown in Fig. 5. The behavior exhibits a straight line in the logarithmic plot as a function of reverse voltage, indicating a near-ideal exponential relationship. The measured quality factor is 43 at zero bias. However, at 11.5 V, the has already increased to 192, indicating the advantage of copper plating. Above 11.5 V, the varactor suffers from a steep increase in leakage current due to imperfections in the epilayer; this restricts its useable voltage range, and consequently, also the upper limit for . B. Linearity Measurement of the Wide Tone-Spacing Varactor Stack To verify the linearity of the wide tone-spacing varactor stack, we have measured the shunt wide tone-spacing varactor stack with an effective zero bias capacitance of 10 pF in a two-port configuration using a 500-k resistive center-tap connection (see Fig. 6). Fig. 11. Source–pull setup used to evaluate the linearity of narrow tone-spacing varactor stack for different impedances of the surrounding circuitry (effective = zero bias capacitance = 10 pF, inductance at terminations = 20 nH, f 2 GHz, a = 0:0896 V , and V = 5 V). The linearity testing is performed using a two-tone signal GHz) with varying tone spacing [25]. Fig. 7 plots ( as a function the measured distortion components at of output power for different values of tone spacing. It can be observed that the linearity improves with the increase of tone spacing and a superior linearity over that of the single diode is obtained for a tone spacing higher than 10 MHz. Note that the linearity at high tone spacing is comparable to that of the measurement setup as shown for the tone spacing of 10, 30, and 80 MHz in Fig. 7. Since at these linearity levels it becomes difficult to separate the nonlinearities resulting from the varactors from that of the measurement setup, a conservative boundary of trust is marked as “measurement limitation” in Fig. 7 and the Authorized licensed use limited to: Univ of Calif San Diego. Downloaded on March 8, 2009 at 23:17 from IEEE Xplore. Restrictions apply. HUANG et al.: ULTRA LINEAR LOW-LOSS VARACTOR DIODE CONFIGURATIONS FOR ADAPTIVE RF SYSTEMS 211 Fig. 12. Simulated linearity contours and measured linearty points as function of the source impedances of the remaining IM5 components in the source current at 2f f in dBc for different values of tone spacing. The envelope peak voltage over the varactor stack is kept constant at 8V (effective zero bias capacitance = = 2 GHz, a = 0:0896 V , and V = 5 V). (a) Tone spacing = 1 MHz. (b) Tone spacing = 10 MHz. 10 pF, inductance at terminations = 20 nH, f (c) Tone spacing = 40 MHz. (d) Tone spacing = 52 MHz. 0 following figures. Consequently, conservatively speaking, the OIP3 will be larger than 56 dBm for a tone spacing larger than 10 MHz. Note that we use here OIP3 rather than IIP3 since a significant part of the input power will be reflected by the varactor shunt impedance. Therefore, the output power level is a better measure of the actual RF voltage over the varactors. It is this voltage that determines the resulting IM3/IM5 levels [see (9) and (10)]. or Normally, it is handy to consider the components as function of power (in dBc) to verify if the third-order distortion is truly cancelled and only IM5 distortion remains. Note that for this condition, a 4:1 slope should be found. However, in our experiments the distortion level is so low that this phenomenon is difficult to verify experimentally. Fortunately, we also implemented varactor stacks with an area ratio of 1.9 instead of 3.7, which we have tested for their linearity as reference. These devices have a 50% area ratio offset with respect to the ideal ratio needed for IM3 cancellation. By comparing the linearity of the ideal wide tone-spacing varactor stack with these structures, the IM3 cancellation was verified. As shown in Fig. 8, the correctly dimensioned wide tone-spacing varactor stack clearly offers superior performance over the wide tone-spacing varactor stack with 50% area ratio offset, especially for the tone spacing larger than 10 MHz. Authorized licensed use limited to: Univ of Calif San Diego. Downloaded on March 8, 2009 at 23:17 from IEEE Xplore. Restrictions apply. 212 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 57, NO. 1, JANUARY 2009 C. Linearity Measurement of the Narrow Tone-Spacing Varactor Stack The wide tone-spacing varactor stack configuration offers a complementary linearity solution to the narrow tone-spacing varactor stack [19]. When diode leakage currents are small, the center-tap impedance conditions are relative easily implemented through the use of a high valued resistor and potentially antiparallel diodes. Such a network can be implemented in a relatively small area. Therefore, for most situations, the complete wide tone-spacing varactor stack structure can be made more compact than the narrow tone-spacing varactor stack, which requires low impedance baseband connections, often implemented as inductors. As addressed in [19], the narrow tone-spacing varactor stack offers the highest linearity for narrowband signals and has been implemented, in this sutdy, on the same wafer as the wide tonespacing varactor stack. Note that this structure is less sensitive to diode leakage currents and is more suitable for fast modulation of the center-tap voltage. Due to the more favorable splitting of the RF voltage over the antiseries varactor diodes, the narrow tone-spacing varactor stack power handling is better than that of the wide tone-spacing varactor stack. It is these properties that make it an excellent choice for the implementation of adaptive transmitters, capacitive modulators or even dynamic loadline amplifiers. To test the linearity of the narrow tone-spacing varactor stack and compare it to the wide tone-spacing varactor stack, a twoport configuration of a shunt 10-pF narrow tone-spacing varactor stack with 20-nH terminations is used (Fig. 9). This configGHz) uration is measured using a two-tone signal ( with varying tone spacing. Fig. 10 plots the measured distoras a function of output power for tion component at different values of tone spacing. It can be observed that the linearity improves with decreasing tone spacing. Again, the measurement accuracy is limited by the measurement system nondBm linearities, while conservatively speaking, the up to 30-MHz tone spacing. In order to confirm that the narrow tone-spacing varactor stack can provide superior linearity for any surrounding circuit condition, the source–pull setup of Fig. 11 is used. In these exis adjusted with the periments, the two-tone source voltage tunable source impedance such that for all loading conditions stack is kept the voltage amplitude over the varactor close to 4 V for each tone yielding an envelope peak voltage of 8 V. Since the varactor stack is reverse biased at 5 V and the RF voltage will split over the varactor diodes, the diodes are operated close to their maximum voltage swing, but will remain reverse biased at all times. The frequency components of the resulting source current are used to monitor the linearity of the varactor. Due to the fact that the inductors satisfy the low-impedance requirements for the baseband signals, source independent IM3 cancellation takes place. The remaining IM5 in dBc, relative components in the external current at to the fundamental, has been simulated as function of the source impedance and plotted as contours of constant distortion for different values of tone spacing (Fig. 12). From these results, we conclude that the narrow tone-spacing varactor stack element 0 Fig. 13. Measured and simulated distortion component at 2f f as function of tone spacing at a power level of 15 dBm (for both cases: effective zero bias = 2 GHz, a = 0:0896 V , and V = 5 V). capacitance = 10 pF, f The narrow tone-spacing varactor stack has two inductors of 20 nH at terminations (see Fig. 9), while the center-tap resistance of the wide tone-spacing varactor stack is 500 k (see Fig. 6). in this experiment provides an dBc for any source impedance up to 10 MHz bandwidth. At higher tone spacing (40 and 52 MHz), the linearity degrades as expected. Since the linearity of the varactor outperforms that of the measurement setup in most of the conditions, only several points can be accurately measured for a tone spacing of 40 and 52 MHz. These measured results are also indicated in Fig. 12(c) and (d) and confirm the trend of the simulated linearity. Note that a slightly inductive source yields the “worst” linearity since, in combination with the capacitive varactor stack, it provides a close to perfect ac short causing all available nonlinear current to flow through the source. D. Overview of the Narrow Tone-Spacing Varactor Stack and Wide Tone-Spacing Varactor Stack In order to complete our overview for the linearity of the narrow tone-spacing varactor stack and wide tone-spacing varactor stack, we plot the measured and simulated distortion comas a function of tone spacing at an output ponent at power level of 15 dBm (Fig. 13). According to the simulation results, the wide tone-spacing varactor stack starts to offer better linearity than the narrow tone-spacing varactor stack above 5 MHz tone spacing for the center-tap impedances indicated. Note that the implementation of the center-tap impedance determines the linearity bandwidth. For example, when using a higher impedance for the wide tonespacing varactor stack center tap (now 500 k ), the high linearity operation can be extended to much lower frequencies. Taking the influence of the leakage current into account, the measurement results of the wide tone-spacing varactor stack Authorized licensed use limited to: Univ of Calif San Diego. Downloaded on March 8, 2009 at 23:17 from IEEE Xplore. Restrictions apply. HUANG et al.: ULTRA LINEAR LOW-LOSS VARACTOR DIODE CONFIGURATIONS FOR ADAPTIVE RF SYSTEMS match the simulation quite well. It can be observed that for the wide tone-spacing varactor stack structure the presence of the leakage currents will cause a modulation of the center-tap voltage for narrow tone spacing, resulting in linearity degradation. Even if so, the combination of the narrow tone-spacing varactor stack and wide tone-spacing varactor stack provide a superior linearity (below the measurement limitation: 80 dBc) for the whole range of the tone spacing, making them very attractive candidates for many RF applications. IV. CONCLUSION For the first time, two extremely linear varactor configurations for tunable RF applications have been successfully implemented in a single silicon-on-glass process technology. When considering their linearity versus tone spacing, the wide tone-spacing varactor stack and the narrow tone-spacing varactor stack design approaches offer complementary linearity behavior, with the wide tone-spacing varactor stack performing best with wide-tone spacing and the narrow tone-spacing varactor stack performing best with narrow tone spacing. This makes them very suitable to handle the different requirements of adaptive receivers and transmitters in a single varactor technology. Due to the almost perfect control of the doping profile relation and ohmic contacts, a close to ideal exponential behavior and quality factor were realized. This accurate yielded a measured linearity performance of dBm dBm for the wide tone-spacing varactor stack and for the narrow tone-spacing varactor stack, which is compatible with the requirements of most demanding wireless applications. Their usability in practical circuit conditions was demonstrated through source–pull simulations and measurements, illustrating that high linearity can be maintained in all cases. Their integration capabilities, compact size, high reliability, low-cost implementation, high speed, high quality factor, and free to choose tuning and control voltage range make them very interesting components for the implementation of ultra-linear high-performance adaptive RF systems. ACKNOWLEDGMENT The authors acknowledge S. Milosavljević and W. de Boer for processing the varactor samples, both with Delft Institute of Microsystems and Nanoelectronics (DIMES), Delft, The Netherlands. The authors would like to thank R. Jos, F. van Straten, and P. Lok, all with NXP, Nijmegen, The Netherlands, A. de Graauw and R. Woltjer, both with NXP, Eindhoven, The Netherlands, and P. Zampardi, Skyworks, Newbury Park, CA, for their encouragement and support. REFERENCES [1] J. B. Muldavin and G. M. Rebeiz, “High-isolation CPW MEMS shunt switches—Part 1: Modeling,” IEEE Trans. Microw. Theory Tech., vol. 48, no. 6, pp. 1045–1052, Jun. 2000. [2] L. Dussopt and G. M. Rebeiz, “Intermodulation distortion and power handling in RF MEMS switches, varactors and tunable filters,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 4, pp. 1247–1256, Apr. 2003. [3] D. Girbau, N. Otegi, L. Pradell, and A. Lazaro, “Study of intermodulation in RF MEMS variable capacitors,” IEEE Trans. Microw. Theory Tech., vol. 54, no. 3, pp. 1120–1130, Mar. 2006. [4] Y. Lu, L. P. B. Katehi, and D. Peroulis, “High-power MEMS varactor and impedance tuners for millimeter-wave applications,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 11, pp. 3672–3678, Nov. 2005. 213 [5] D. Girbau, N. Otegi, L. Pradell, and A. Lazaro, “Generation of third and higher-order intermodulation products in MEMS capacitors, and their effects,” in Proc. 35th Eur. Microw. Conf., Oct. 2005, pp. 593–596. [6] D. R. Chase, L. Y. Chen, and R. A. York, “Modeling the capacitive nonlinearity in thin-film BST varactors,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 10, pp. 3215–3220, Oct. 2005. [7] J. S. Fu, X. A. Zhu, D. Y. Chen, J. D. Phillips, and A. Mortazawi, “A linearity improvement technique for thin-film barium strontium titanate capacitors,” in IEEE MTT-S Int. Microw. Symp. Dig., San Francisco, CA, Jun. 2006, pp. 560–563. [8] K. Buisman, L. C. N. de Vreede, L. E. Larson, M. Spirito, A. Akhnoukh, T. L. M. Scholtes, and L. K. Nanver, “Distortion free varactor diode topologies for RF adaptivity,” in IEEE MTT-S Int. Microw. Symp. Dig., Long Beach, CA, Jun. 2005, pp. 157–160. [9] K. Buisman, L. C. N. de Vreede, L. E. Larson, M. Spirito, A. Akhnoukh, Y. Lin, X. Liu, and L. K. Nanver, “Low-distortion, low-loss varactor-based adaptive matching networks, implemented in a silicon-on-glass technology,” in Proc. Radio Freq. IC Symp., Long Beach, CA, Jun. 2005, pp. 389–392. [10] K. Buisman, C. Huang, A. Akhnoukh, M. Marchetti, L. C. N. de Vreede, L. E. Larson, and L. K. Nanver, “Varactor topologies for RF adaptivity with improved power handling and linearity,” in IEEE MTT-S Int. Microw. Symp. Dig., Honolulu, HI, Jun. 2007, pp. 319–322. [11] R. G. Meyer and M. L. Stephens, “Distortion in variable-capacitance diodes,” IEEE J. Solid-State Circuits, vol. SSC-10, no. 2, pp. 47–55, Feb. 1975. [12] Q. Han, K. Inagaki, and T. Ohira, “Perturbation analysis and experimental verification on intermodulation and harmonic distortion in an anti-series varactor pair,” IEICE Trans. Electron., vol. E88-C, no. 1, pp. 89–97, Jan. 2005. [13] M. Akaike, T. Ohira, K. Inagaki, and Q. Han, “An analysis of nonlinear terms in capacitance–voltage characteristic for anti-series-connected varactor–diode pair,” Int. J. RF Microw., vol. CAE-14, pp. 274–282, 2004. [14] Q. Han, K. Inagaki, R. Schlub, T. Ohira, and M. Akaike, “Harmonic distortion suppression technique for varactor-loaded parasitic radiator antennas,” IEICE Trans. Electron., vol. E85-C, no. 12, pp. 2015–2020, Dec. 2002. [15] K. Entesari and G. M. Rebeiz, “A differential 4-bit 6.5-10-GHz RF MEMS tunable filter,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 3, pp. 1103–1110, Mar. 2005. [16] K. Entesari and G. M. Rebeiz, “A 12-18-GHz three pole RF MEMS tunable filter,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 8, pp. 2566–2571, Aug. 2005. [17] F. Ellinger, H. Jackel, and W. Bachtold, “Varactor-loaded transmission-line phase shifter at C -band using lumped elements,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 4, pp. 1135–1140, Apr. 2003. [18] T. M. Hancock and G. M. Rebeiz, “A 12-GHz SiGe phase shifter with integrated LNA,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 3, pp. 977–983, Mar. 2005. [19] C. Huang, L. C. N. de Vreede, F. Sarubbi, M. Popadić, K. Buisman, J. Qureshi, M. Marchetti, A. Akhnoukh, T. L. M. Scholtes, L. E. Larson, and L. K. Nanver, “Enabling low-distortion varactors for adaptive transmitters,” IEEE Trans. Microw. Theory Tech., vol. 56, no. 5, pp. 1149–1163, May 2008. [20] P. Wambacq and W. M. Sansen, Distortion Analysis of Analog Integrated Circuit. Norwell, MA: Kluwer, 1998, ch. 5. [21] K. Buisman, L. K. Nanver, T. L. M. Scholtes, H. Schellevis, and L. C. N. de Vreede, “High performance varactor diodes integrated in a silicon-on-glass technology,” in Proc. 35th Eur. Solid-State Device Res. Conf., Grenoble, France, Oct. 2005, pp. 389–392. [22] L. K. Nanver, H. Schellevis, T. L. M. Scholtes, L. La Spina, G. Lorito, F. Sarubbi, V. Gonda, M. Popadic, K. Buisman, L. C. N. de Vreede, C. Huang, S. Milosavljevic, and E. J. G. Goudena, “Silicon-on-glass technology for RF and microwave device fabrication,” in Proc. 8th Int. Solid-State Integr. Circuit Technol. Conf., Shanghai, China, Oct. 2006, pp. 162–165. [23] R. Dekker, P. G. M. Baltus, and H. G. R. Maas, “Substrate transfer for RF technologies,” IEEE Trans. Electron Devices, vol. 50, no. 3, pp. 747–759, Mar. 2003. [24] “Medici Two-Dimensional Device Simulation Program User Manual,” ver. 2002.4, Synopsys Inc., Mountain View, CA, Feb. 2003. [25] M. Spirito, M. J. Pelk, F. van Rijs, S. J. C. H. Theeuwen, D. Hartskeerl, and L. C. N. de Vreede, “Active harmonic load-pull for on-wafer out-of-band device linearity optimization,” IEEE Trans. Microw. Theory Tech., vol. 54, no. 12, pp. 4225–4236, Dec. 2006. Authorized licensed use limited to: Univ of Calif San Diego. Downloaded on March 8, 2009 at 23:17 from IEEE Xplore. Restrictions apply. 214 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 57, NO. 1, JANUARY 2009 Cong Huang (S’07) was born in Shanghai, China, in 1980. He received the B.S. degree in microelectronics from Fudan University, Shanghai, China, in 2002, the M.Sc. degree (with honors) in microelectronics from the Delft University of Technology, Delft, The Netherlands, in 2005, the M.Sc. degree in microelectronics from Fudan University, Shanghai, China, in 2005, and is currently working toward the Ph.D. degree at the Delft Institute of Microsystems and Nanoelectronics (DIMES), Delft University of Technology. His master’s research concerned low phase-noise voltage-controlled oscillators and piezoelectric-material-based cantilevers for MEMS applications. Since 2005, he has been with DIMES, where he is involved with high-performance varactors for RF adaptivity and the development of RF integrated circuits (RFICs) for next-generation wireless systems. Koen Buisman (S’05) received the M.Sc. degree in microelectronics from Delft University of Technology, Delft, The Netherlands, in 2004, and is currently working toward the Ph.D. degree at Delft University of Technology. Since 2004, he has been with the Delft Institute of Microsystems and Nanoelectronics (DIMES), Delft University of Technology. He was involved in the development of a pulsed dc and RF measurement system and the development of a custom in-house DIMES technology for high-performance “distortion-free” varactors. He has authored or coauthored over 20 papers. His research interests are varactors for RF adaptivity, nonlinear device characterization, and compact modeling of heterojunction bipolar transistors. Mauro Marchetti (S’07) received the M.Sc. degree (cum laude) in electrical engineering from the University of Naples Federico II, Naples, Italy, in 2006, and is currently working toward the Ph.D. degree at Delft Institute of Microsystems and Nanoelectronics (DIMES), Delft University of Technology, The Netherlands. He is currently involved in the development and implementation of advanced characterization setups for RF amplifiers. Lis K. Nanver (S’80–M’83) received the M.Sc. degree in physics from the University of Aarhus, Aarhus, Denmark, in 1979, the Dr.ing. degree from the Ecole Nationale Supérieure des Télécommunications, Paris, France, in 1982, and the Ph.D. degree from the Delft University of Technology, Delft, The Netherlands, in 1987. While with the Ecole Nationale Supérieure des Télécommunications, she was involved with the simulation of CCD structures. While with DIMES, she was involved with the development of a medium-frequency BIFET process. In 1988, she joined the Delft Institute of Microsystems and Nanoelectronics (DIMES) Integrated Circuit (IC) Process Research Sector as a Bipolar Process Research Manager. She became an Associate Professor and then Professor with the Faculty of Electrical Engineering, Mathematics and Computer Science, Delft University of Technology, detached at the DIMES Technology Center in 1994 and 2001, respectively. Within the Laboratory of Electrical Components, Technology and Materials, she manages research on the integration of silicon devices, mainly for RF, microwave, or smart sensor applications. This research involves technologies such as epitaxy by chemical-vapor or metal-induced solid-phase deposition, excimer laser processing, and substrate transfer techniques. Prof. Nanver has served on the committees of ESSDERC, BCTM, IWJT, RTP, and SBMicro. Francesco Sarubbi was born in Rome, Italy, in 1977. He received the M.Sc. degree (with honors) in electronics engineering from the University of Naples Federico II, Naples, Italy, in 2004, and is currently working toward the Ph.D. degree at Delft University of Technology, Delft, The Netherlands. He completed his Master’s thesis on the electrical characterization and modeling of heavy doping effects in 4H-silicon carbide with the Department of Microelectronics and Information Technology, Royal Institute of Technology, Stockholm, Sweden. In 2005, he joined the Faculty of Electrical Engineering, Mathematics and Computer Science, Laboratory of Electronic Components, Technology and Materials, Delft Institute of Microsystems and Nanoelectronics (DIMES), Delft University of Technology. His research interests include characterization of shallow junctions formed by using atmospheric/low-pressure chemical vapor deposition (AP/LPCVD) for the development of advanced Si-based RF devices, and DUV/EUV radiation detectors. Milos̆ Popadić was born in Belgrade, Serbia, in 1981. He received the M.Sc. degree in electrical engineering from the University of Belgrade, Belgrade, Serbia, in 2005, and is currently working toward the Ph.D. degree at Delft University of Technology, Delft, The Netherlands. In 2005, he joined Delft University of Technology, where he is involved with the electrical properties of contacts and ultra-shallow junctions. Tom L. M. Scholtes received the HBO Engineering degree in chemistry from the Hogere Laboratorium School, Sittard, The Netherlands, in 1984, the HTS Engineering degree in applied physics from the Hogere Technische School, Heerlen, The Netherlands, in 1988, and the post HBO degree in environmental technology from the Hogere Technische School, Dordrecht, The Netherlands, 1994. From 1988 to 1990, he was with the consultancy bureau Cauberg Huygen. In 1990, he joined the Laboratory of Optics, Faculty of Applied Physics, Delft University of Technology, Delft, The Netherlands, where he was involved with the processing of InP–In–GaAsP devices. In 1996, he joined the Delft Institute for Microsystems and Nanoelectronics (DIMES), Delft University of Technology, as a Process Technologist with the Integrated Circuit (IC) Process Research Sector attached to the Laboratory of Electrical Components, Technology and Materials, Faculty of Electrical Engineering, Mathematics and Computer Science, Delft University of Technology. His research interests are technologies such as advanced chemical vapor deposited (CVD) epitaxy, ion implantation, and excimer laser annealing for the fabrication of silicon-based semiconductor devices. Hugo Schellevis received the Chemical Engineer degree from the Institute of Technology in Amsterdam, The Netherlands, in 1986, and .the M.Sc. degree in chemical engineering from the University of Amsterdam, The Netherlands, in 1996. In 1987, he joined the Integrated Circuit (IC) Process Rsearch Sector, Delft Institute for Microsystems and Nanoelectronics (DIMES), Delft University of Technology, Delft, The Netherlands, as a Process Engineer. His main interest is currently the development of metallization processes and substrate transfer technology. Authorized licensed use limited to: Univ of Calif San Diego. Downloaded on March 8, 2009 at 23:17 from IEEE Xplore. Restrictions apply. HUANG et al.: ULTRA LINEAR LOW-LOSS VARACTOR DIODE CONFIGURATIONS FOR ADAPTIVE RF SYSTEMS Lawrence E. Larson (S’82–M’86–SM’90–F’00) received the B.S. and M.Eng. degrees in electrical engineering from Cornell University, Ithaca, NY, in 1979 and 1980, respectively, and the Ph.D. degree in electrical engineering and M.B.A. degree from the University of California at Los Angeles (UCLA), in 1986 and 1996, respectively. From 1980 to 1996, he was with Hughes Research Laboratories, Malibu, CA, where he directed the development of high-frequency microelectronics in GaAs, InP, and Si-SiGe, and MEMS technologies. In 1996, he joined the faculty of the University of California at San Diego (UCSD), La Jolla, where he is the Inaugural Holder of the Communications Industry Chair. He is currently Director of the UCSD Center for Wireless Communications. During the 2000–2001 academic years, he was on leave with IBM Research, San Diego, CA, where he directed the development of RF integrated circuits (RFICs) for third-generation (3G) applications. During the 2004–2005 academic year, he was a Visiting Professor with the Technical University of Delft, Delft, The Netherlands. He has authored or coauthored over 250 papers. He holds 31 U.S. patents. Dr. Larson was the recipient of the 1995 Hughes Electronics Sector Patent Award for his research on RF MEMS technology. He was corecipient of the 1996 Lawrence A. Hyland Patent Award of Hughes Electronics for his research on low-noise millimeter-wave HEMTs, the 1999 IBM Microelectronics Excellence Award for his research in Si–SiGe heterojunction bipolar transistor technology, and the 2003 IEEE Custom Integrated Circuits Conference Best Invited Paper Award. 215 Leo C. N. de Vreede (M’01–SM’04) was born in Delft, The Netherlands, in 1965. He received the B.S. degree in electrical engineering from The Hague Polytechnic, The Hague, The Netherlands, in 1988, and the Ph.D. degree from the Delft University of Technology, Delft, The Netherlands, in 1996. In 1988, he joined the Laboratory of Telecommunication and Remote Sensing Technology, Department of Electrical Engineering, Delft University of Technology. From 1988 to 1990, he was involved in the characterization and physical modeling of CMC capacitors. From 1990 to 1996, he was involved with modeling and design aspects of HF silicon ICs for wideband communication systems. In 1996, he became an Assistant Professor with the Delft University of Technology, during which time he was involved with the nonlinear distortion behavior of bipolar transistors with the Delft Institute of Microsystems and Nanoelectronics (DIMES). In Winter 1998–1999, he was a guest of the High-Speed Device Group, University of San Diego, La Jolla, CA. In 1999, he became an Associate Professor responsible for the Microwave Components Group, Delft University of Technology. Since that time, he has been involved with RF solutions for improved linearity and RF performance at the device, circuit, and system levels. He has coauthored over 60 IEEE refereed conference and journal papers. He holds several patents. His current interest is technology optimization and circuit concepts for adaptive wireless systems. Dr. de Vreede was the recipient of the 2008 IEEE Microwave Prize. Authorized licensed use limited to: Univ of Calif San Diego. Downloaded on March 8, 2009 at 23:17 from IEEE Xplore. Restrictions apply.