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Ben-Gurion University Faculty of Natural Sciences Department of Physics Undergraduate Project in Physics: Precision measurements with alkali vapor cells High frequency induction heating Submitted by: Maxim Sokol Advisor: Dr. David Groswasser Content: 1. Abstract 2 2. Theoretical background - Vapor cells 3 2.1 Introduction - Vapor cells 2.2 Fine and hyperfine structure of the Rb atom 2.3 Alkali vapor cells 2.4 Interaction with radiation field and decoherence effects 3. Theoretical background – Induction heating 3.1 Introduction – Induction heating 3.2 Induction heating 3.2.1 How does induction heating works? 3.2.2 Practical implementation 4. System design 3 3 4 5 6 6 7 7 8 9 4.1 Power amplifier design 4.2 Work-coil design 4.3 Parallel resonant tank design 4.4 Heat transfer and heating time 9 10 13 15 5. Measurement, configuration and results 5.1 Equipment 5.2 Heat transfer and heating time 5.3 Results - heating time 15 15 17 18 6. References 19 7. APEX PA119CE/PA119CEA datasheet 20 1 1. Abstract Precision measurements based on light interaction with alkali vapor are important in various metrology fields (atomic clocks, magnetometers etc.). Often, the sensitivity of these measurements is limited by decoherence time of the alkali atoms in the vapor cell. It was found that the two main decoherence mechanisms are atom-atom collisions between alkali atoms and atom-wall collisions. The purpose of this project is to reduce decoherence rate due to atom-wall collision by paraffin coating of the inner vapor cell’s walls. The preparation process of such paraffin coated vapor cells is complex and is carried out only by a limited number of skilled experts in the world. The AtomChip group initiated a program to develop such a process in our university. A part of the vapor cell production process is to depose the paraffin on the cell walls. Paraffin at room temperature is solid. In order to deposit a homogenous layer it has to be heated and evaporated on the cell walls. However, the cell is evacuated so that the heat transfer from outside the cell to the paraffin sample is problematic. We decided to solve this problem by using induction heating (Induction heating is the process of heating an electrically conducting object, usually a metal by electromagnetic induction). Our task in this project is to characterize, design and build a small induction heating system that will be used in the vapor cell filling facility build by the AtomChip group. (1) (2) (3) Figure 1. 1: Vapor-cell filling system, including the turbo pump (1), buffer gas 2 in fig. 4.2.1. & fig. 4.2.2. manifold (2) and glass manifold (3) see detailed view 2. Theoretical background - Vapor cells 2.1. Introduction - Vapor cells Alkali atoms are ideal systems for precision measurements. They have a single electron at the outer shells (S). This electronic structure is very convenient for calculations of the atomic spectrum and transitions. In addition, their ground state is split into two sub-states with very ultra-narrow natural spectral width. The energy difference between these two sub-levels (typically refers to as the “hyperfine split”) corresponds to frequencies in the microwave range. As the lifetime of these levels is ridiculously long, the natural width of these sub-levels is very narrow. Therefore, the hyperfine transition frequencies can be used as highly accurate frequency standards. In fact, these standards are so accurate that one of them is used to define the second: The natural frequency of the hyperfine split in 133Cs defines the SI second: The second is the duration of 9,192,631,770 periods of the radiation at a frequency corresponding to the transition between the two hyperfine levels of the ground state of the 133Cs atom. Another alkali element that is typically used as a frequency standard is 87Rb, with a hyperfine split corresponding to 6,834,682.610.90429Hz. 2.2. Fine and hyperfine structure of the Rb atom Rubidium – Rb, is an alkali metal in atom configuration of [Kr]5S1, existing typically in nature as 72% 85Rb and 28% 87Rb. 87Rb atom has a fine structure of J (electron total angular momentum) equals to 1/2 in it ground state. By interacting an atom with laser of wavelength ~780nm, an electron can be excited from 5S1/2 to 5P1/2 (D2 transition), or by interaction with laser of wavelength ~795nm it might be excited to 5P3/2 (D1 transition). Those energy splits are a result of an interaction between the electron spin and the electrons' orbital angular momentum, often called the Fine split. The hyperfine structure occurs due to the interaction of the nuclear magnetic dipole moment in the magnetic field generated by the electrons, and the interaction of the nuclear electric quadruple moment in the electric field gradient due to the Figure 2.2.1: 87 Rb fine and hyperfine split diagram distribution of charge with the atom. 3 2.3. Alkali vapor cells The Technology of vapor cell manufacturing already exists for several decades. Such cells are typically made of an evacuated tube made of optical quality glass with a drop of alkali metal inside. From basic thermodynamic equations the vaporpressure of the alkali metal inside the cell can be calculated as function of the cell temperature. In practice, in cells that are produced by the AtomChip group the Rb vapor pressure is deliberately kept bellow the solid-vapor equilibrium value. Figure 2.3.1: Vapor pressure of 87Rb. For many years alkali vapor cells comprising Lithium (Li), Sodium (Na), Potassium (K), Rubidium (Rb), Cesium (Cs), and Francium (Fr) are used as atomic references in spectroscopy. Cs and Rb are also commonly used in atomic clocks as frequency standards. In recent years there is a growing interest in Na, K, Rb and Cs cells also for optical magnetometery measurements. Optical magnetic sensors based on alkali vapor are the most sensitive probes of magnetic fields reaching extreme sensitivities of sub fT/Hz1/2! 4 2.4. Interaction with radiation field and the decoherence effect The accuracy of any measurement based atom-light interaction is dependent on the coherence time of the atomic ensemble. If the phase evolution in the sample is not homogenous, each atom will obtain a different phase, and during the measurement will give a different signal. As a result, the overall output signal will be “garbles”, or even averaged out to zero. In vapor cells, the leading decoherence processes that are responsible for loss of phase between the atoms are collisions between alkali atoms and collisions of alkali atoms with the glass walls of the cell. It was found that decoherence due to collisions between alkali atoms may be significantly reduced by adding inert buffer gasses such as He, Ar, Xe or N2 to the cell. Under such condition the collision frequency between alkali atoms is significantly reduced while collisions with the buffer gas do not change the Figure 2.4.1: spin state of the alkali atom. In addition, it was found Rubidium vapor cell. that when coating the glass walls with a thin layer of paraffin or a similar derivative, the coherence time is increased by more the a factor 1000. This is since alkali-paraffin collisions are "spin-preserving" and do not affect the phase evolution of the alkali atom. However, the coating process of vapor cells with paraffin is long and tricky. As a result, the production yield is very low and only a handful of research groups worldwide apply it. Obviously, the problem of collisions with the cell walls is increased in miniaturized vapor cells. Thus, there is an increasing motivation for developing a reliable paraffin coating procedure. 5 3. Theoretical background – Induction heating 3.1. Introduction – Induction heating Induction heating is a non-contact heating process. It uses high power AC fields to resonantly heat materials that are electrically conductive. Since it is non-contact, the heating process does not contaminate the material being heated. It is also very efficient since the heat is actually generated inside the workpiece. For these reasons induction heating lends itself to some unique applications in industry, for example: heating electrically conductive material in a clean, efficient and controlled manner. As we describe in section 1 and section 2.3, devices based on rubidium vapor cells are very sensitive to atom-wall interactions which can shift the rubidium hyperfine transition frequency and even depolarize the atoms (decoherence). Previous studies have shown that long polarization times may be achieved by coating the inner surface of the vapor cell which containing the rubidium with a "separating material". Also studies have shown that in cell with tetracontane - C40H82 (component of standard paraffin) coated inner walls, 87Rb atom can bounces even 105 times from the surface without losing its polarization as shown in figure 3.1.1. The tetracontane coating process is very complicated. The tetracontane is evaporated to homogenously coat the walls, but this must happen under vacuum and that's why an induction heating system is required. Figure 3.1.1: Zeeman resonances at B=0.28 gauss from uncoated and tetracontane coated Pyrex cells. 6 3.2. Induction heating 3.2.1. How does induction heating works? A source of high frequency electric field is used to drive a large alternating current through the work-coil. The current flow through the work-coil generates a very intense and rapidly changing magnetic field in the space within the work-coil. The workpiece to be heated is placed within this intense alternating magnetic field. The alternating magnetic field induces a current flow in the conductive workpiece. The arrangement of the work-coil and the workpiece can be thought of as an electrical transformer (Figure 3.2.1.1). The work-coil is like the primary where electrical energy is fed in, and the workpiece is like a single turn secondary that is short-circuited. Ideal transformer equation: (1) Vs N s I p Vp N p I s This causes tremendous currents to flow through the workpiece. Those currents are known as eddy currents. Figure 3.2.1.1: The ideal transformer as a circuit element. Now, depending on the nature of the workpiece material, number of effects happen: First, these eddy currents flow against the electrical resistivity of the metal, generating localized heat. This heating occurs with both magnetic and non-magnetic parts, and is often referred to as the "Joule effect", referring to Joule's first law: (2) Q I 2 R t Where Q is the amount of heat produced, I is the current flowing through the part (workpiece), R is the electrical resistance of the part, and t = time. Secondly, due to the high frequency used in induction heating applications gives rise to a phenomenon called skin effect. This skin effect forces the alternating current to flow in a thin layer towards the surface of the workpiece. The skin effect increases the effective resistance of the metal to the passage of the large current. Therefore it greatly increases the heating effect caused by the current induced in the workpiece. (3) 1 0 r f 503 r f Where δ is the skin depth, ρ it the resistivity of the medium, μr is the relative permeability of the medium and f is the frequency of the wave. 7 Finally, additional heat is produced within magnetic parts through hysteresis - The intense alternating magnetic field inside the work-coil repeatedly magnetizes and de-magnetizes the iron crystals. This rapid flipping of the magnetic domains causes considerable friction and heating inside the material. Heating due to this mechanism is known as Hysteresis loss, and is greatest for materials that have a large area inside their B-H curve. This can be a large contributing factor to the heat generated during induction heating, but only takes place inside ferrous materials. 3.2.2. Practical implementation The work-coil is usually connected to a resonant tank circuit. This has a number of advantages. Firstly, it makes either the current or the voltage waveform become sinusoidal. This minimizes losses in the amplifier circuit. The sinusoidal waveform at the work-coil also represents a more pure signal and causes less RF (Radio Frequency) interference to nearby equipment. We will see and discuss about two main resonant schemes: Series resonant tank circuit: The work-coil is made to resonate at the intended operating frequency by means of a capacitor placed in series with it. This causes the current through the work-coil to be sinusoidal. The series resonance also magnifies the voltage across the work-coil, far higher than the output voltage of the amplifier alone. The amplifier sees a sinusoidal load current but it must carry the full current that flows in the work-coil. These high currents make the design of the amplifier very complex. Parallel resonant tank circuit: The work-coil is made to resonate at the intended operating frequency by means of a capacitor placed in parallel with it. This causes the current through the work-coil to be sinusoidal. The parallel resonance also magnifies the current through the work coil, far higher than the output current capability of the amplifier alone. The amplifier sees a sinusoidal load current. However, in this case it only has to carry the part of the load current that actually does real work. The amplifier does not have to carry the full circulating current in the work-coil. Thus makes the design of the amplifier less complex. Figure 3.2.2.1: From left to right - Series resonant tank circuit, Parallel resonant tank circuit. 8 4. System design 4.1. Power amplifier design One of our main demands was high and stable frequency BW (bandwidth), in order not to limit ourselves in the resonant tank circuit design. The second issue was the power output capability. In order to keep relatively high power outputs at high frequencies a high slew-rate is a must. By taking to consideration all the said below, workpiece dimensions, material nature and the desired heating time. We came to decision to base the amplifier on the APEX PA119CE/PA119CEA (datasheet is attached) power operational amplifier chip. All the circuit design and all the frequencies simulations were performed by OrCAD PSpice 10.5 and all the real-time simulations were performed by National Instruments Circuit Design Suite 11.0. Figure 4.1.1: Power amplifier - main circuit. ORcAD PSpice 10.5. 9 (1) (2) Figure 4.1.2: (3) 4.2. Workamplifier coil design Assembled including the PCB (1), heatsink (2) and fan (3). During the work-coil design two main demands were always in front of us. The first was the coil dimensions – the work-coil should fit into the vacuum system (see fig. 4.2.2) as was described in sections 1 and section 2.4 (see fig. 4.2.1). Second, in order to get high currents, the reactive impedance must to be low. (4) X L j L L XL j Work-coil position Figure 4.2.1: A diagram of the glass manifolds used for cell production and filling in the AtomChip lab. When XL is the reactive impedance and L is the coil induction. From equation (4) we can immediately see that a low inductance was needed. 10 Glass Manifold Workpiece Work-coil Figure 4.2.2: Work-coil and workpiece position on a prototype glass manifold. Work-coil dimensions: r – radius of the coil l – length of coil N – numbers of turns 5mm 10cm 70 The inductance formula for a cylindrical coil is: (5) L 0 N 2 r 2 l ; 0 4 107 [ H ] m From equation (5) the coil inductance can be calculated: 4 107 2 702 (5 103 )2 L 4.83 H . 0.1 From measurements that we performed, the "real" inductance is L 4.6 H . 11 Workpiece Figure 4.2.3: Work-coil ( L 4.6 H ) during a heat process with workpiece inside. Figure 4.2.4: Workpiece (made of steel) model. 12 4.3. Parallel resonant tank design As described in section 3.2.2, a parallel resonant tank was chosen. On one hand, low reactive impedance and a deeper skin penetration are needed (lower frequency – section 3.2.1) in order to get high currents. On the other hand, we want also to get heat generation from the magnetic hysteresis effect (higher frequency – section 3.2.1). Using ORcAD PSpice 10.5 for frequency sweep simulations a 200-300 kHz range was chosen. Figure 4.3.1: ORcAD PSpice frequency sweep simulations circuit. We chose capacitors with values suitable to this frequency range. Due to the fact that temperature has a significant effect on capacitor parameters we decide to use few capacitors in a serial/parallel mode in order to decrease heat generation and increase heat dissipation. X L X C 2 f L Resonant frequency calculation: f resonant 13 1 2 LC 1 2 f C Graph 4.3.1: Work coil voltage as a function of frequency. V (in) =2v – without amplifier. Graph 4.3.2: Vout/Vin as a function of frequency. V (in) =2v – without amplifier. 14 It's easy to see from graphs 4.3.1 and 4.3.2 that when the workpiece is inserted inside the work coil there is a voltage drop on the work coil. This voltage drop can be attributed to a power transfer from the work coil into the workpiece. Another interesting note is that the workpiece slightly shifts the resonant frequency and degrades the Q factor (graph 4.2.1). Therefore, the input frequency should be adjusted in order to get maximum power after the workpiece has been positioned inside the work coil. 5. Measurement, configuration and results 5.1. Equipment As described in previous theoretical section - in order to determine the transmitted power we need to measure the difference in the supplied current and voltage over the work coil. The measurement of this current was done by the main power supply's internal current meter. Figure 5.1.1: Main power supply's internal current and voltage meter In section 4.3 we saw from graph 4.3.1 that when the workpiece insert inside the work coil there is a slightly shift in the resonant frequency. In order to determine exactly the resonant frequency we used an oscilloscope – model Fluke 199c. This oscilloscope model has a dual input with separated grounds. This feature made the measurement process very convenient and gave as the possibility to simultaneously measure the voltage over the work-coil and the amplifier output in parallel to keep the amplifier steady as possible. Another oscilloscope – model Tektronix TDS210 was used for measuring the input signal. 15 Figure 5.1.2: Fluke 199c - oscilloscope Figure 5.1.3: Tektronix TDS210 - oscilloscope The last measurement device used in this project is the temperature meter – model Fluke 187. The temperature meter was attached directly to the workpiece with a bimetal wire. Figure 5.1.4: Fluke 187 - temperature meter Figure 5.1.5: General view of the system - amplifier, resonant tank, power supply, signal generator, two oscilloscopes and the temperature meter. 16 5.2. Heat transfer and heating time In order to calculate the energy needed for the heating procedure we can use the next equation: (6) E m T Cp E is the energy, m is the workpiece mass, T is the temperature and Cp is the specific heat capacity. When we find the energy needed in order to heat the workpiece we can calculate the time needed for this procedure: (7) T E P Where, P is the power. The mass of the workpiece is about 15gr. Average heat capacity for steels is 0.46 [kJ/Kg∙K]. Temperature that we would like to achieve is about 200 °C => ΔT=175 °C or °K. E 1220 J By measuring the voltage and the difference in the current at the amplifier output, we can calculate the power involved in the heating procedure: (8) PNet V I 10 w Now the time for the process can be calculated: T 1220 122sec 2.03min 10 This number is really close to the time measured (see section 6), that stands about 2.25min. 17 5.3. Results - heating time As we describe in section 5.2 the heating time is proportional to the energy received by the workpiece. Many repeated measurements were done (more than twenty) and the results were steady and constant: Graph 5.3.1: Heating time of the workpiece as a function of time. As we can see from graph 5.3.1 there is a slightly curve shift. That curve shift occurs due to high currents inside the work coil, and those currents heat the work coil. Temperature has a strong influence on material properties like electric parameters, which change the coil inductance and resistance. When the coil temperature stabilizes at 120 °C the curve gets stable too. In section 5.2 we semi-theoretically (the electrical power was measured) calculated the heating time: Tcalc 2.03sec and from real measurements we get a very close result: Tmesured 2.25sec . 18 6. References: [1] – D. F. Phillips, A. Boca and R. L. Walsworth, "Evaporative Coating of Rb Maser Cell", http://cfa-www.harvard.edu/~dphil/work/coat.pdf (1999). [2] – D. A. Steck "Rubidium 87 D Line Data", revision 1.6. Source – http://steck.us/alkalidata (2003). [3] – S. Cartalevaa, T. Karaulanova, N. Petrova, D. Slavova, K. Vasevaa, A. Yaneva, M. Mijailovicb, Z. Grujicb, "All-Optical Magnetometer Based on Resonant Excitation of Rubidium Atoms by Frequency Modulated Diode Laser Light", ACTA Phys. Pol. A 112, 871 (2007). [4] – Bison G, Castagna N, Hofer A, et al, "A room temperature 19-channel magnetic field mapping device for cardiac signals", Appl. Phys. Lett. 95, 173701 (2009). [5] – Affolderbach, C., Stahler, M., Knappe, S. & Wynands, R., "An all-optical, high sensitivity magnetic gradiometer", Appl. Phys. B 75, 605–612 (2002). [6] – R. Michael Garvery, "Atomic Frequency Standards", ITSF 06 (2006) [7] – M. Shuker, O. Firstenberg, R. Pugatch, A. Ron, and N. Davidson, "Storing images in worm atomic vapor", Phys. Rev. Lett. 100, 223601 (2008). [8] – M. Klein, I. Novikova, D.F. Phillips, and R.L. Walsworth, “Slow light in paraffincoated Rb vapor cells”, Journal of Modern Optics 53, 2583 (2006). [9] – V. Rudnev, D. Loveless, R. Cook, M. Black, "Handbook of Induction Heating", New York: Marcel Dekker (2003). [10] – E.J.Davies, "Conduction and induction heating", IEE Power Engineering Series II, Peter Peregrinus Ltd, (1990). [11] – M. Fishenden, O.A. Saunders "An Introduction to Heat Transfer" Oxford University Press. Oxford, (1950). 19 PA119CE • PA119CEA PA119CE • PA119CEA Product IPnr no od vuac t i oI nn n o v a t i o n F r o m PA119CE, PA119CEA From Video Power Operational Amplifier FEATURES • VERY FAST SLEW RATE — 900 V/µs • POWER MOS TECHNOLOGY — 4A peak rating • LOW INTERNAL LOSSES — 0.75V at 2A • PROTECTED OUTPUT STAGE — Thermal Shutoff • WIDE SUPPLY RANGE — ±15V TO ±40V APPLICATIONS 8-pin TO-3 PACKAGE STYLE CE • VIDEO DISTRIBUTION AND AMPLIFICATION • HIGH SPEED DEFLECTION CIRCUITS • POWER TRANSDUCERS UP TO 5 MHz • MODULATION OF RF POWER STAGES • POWER LED OR LASER DIODE EXCITATION TYPICAL APPLICATION DESCRIPTION The PA119 is a high voltage, high current operational amplifier optimized to drive a variety of loads from DC through the video frequency range. Excellent input accuracy is achieved with a dual monolithic FET input transistor which is cascoded by two high voltage transistors to provide outstanding common mode characteristics. All internal current and voltage levels are referenced to a zener diode biased on by a current source. As a result, the PA119 exhibits superior DC and AC stability over a wide supply and temperature range. High speed and freedom from second breakdown is assured by a complementary power MOS output stage. For optimum linearity, especially at low levels, the power MOS transistors are biased in a class A/B mode. Thermal shutoff provides full protection against overheating and limits the heatsink requirements to dissipate the internal power losses under normal operating conditions. A built-in current limit of 0.5A can be increased with the addition of two external resistors. Transient inductive load kickback protection is provided by two internal clamping diodes. External phase compensation allows the user maximum flexibility in obtaining the optimum slew rate and gain bandwidth product at all gain settings. A heatsink of proper rating is recommended. This hybrid circuit utilizes thick film (cermet) resistors, ceramic capacitors, and silicon semiconductor chips to maximize reliability, minimize size, and give top performance. Ultrasonically bonded aluminum wires provide reliable interconnections at all operating temperatures. The 8-pin TO-3 package is hermetically sealed and electrically isolated. The use of compressible thermal washers and/or improper mounting torque will void the product warranty. Please see “General Operating Considerations”. TYPICAL APPLICATION +40V ±5mA 1K DAC 110Ω EQUIVALENT SCHEMATIC 3 Q1 Up to 4A Q5 Q9 Q11 Q15 Q13 Q12 Q16 1 D1 Q19 Q20 5 Q17B Q17A Q21 4 Q22 Q23 Q24 7 Q25 D2 6 EXTERNAL CONNECTIONS RCL+ +V 2 3 1 4 +IN 5.6pF TOP VIEW 5 –IN 8 6 –40V PA119 AS FAST POWER DRIVER www.cirrus.com Q8 Q10 –V PA119U 2 Q7 8 ±32.5V PA119 Q2 Q4 Q3 500Ω RCL+ RCL– This fast power driver utilizes the 900V/µs slew rate of the PA119 and provides a unique interface with a current output DAC. By using the DAC’s internal 1KΩ feedback resistor, temperature drift errors are minimized, since the temperature drift coefficients of the internal current source and the internal feedback resistor of the DAC are closely matched. Gain of VOUT to IIN is –6.5/mA. The DAC’s internal 1K resistor together with the external 500Ω and 110Ω form a “tee network” in the feedback path around the PA119. This effective resistance equals 6.5KΩ . Therefore the entire circuit can be modeled as 6.5KΩ feedback resistor from output to inverting input and a 5mA current source into the inverting input of the PA119. Now we see the familiar current to voltage conversion for a DAC where VOUT = –IIN x RFEEDBACK. Copyright © Cirrus Logic, Inc. 2010 (All Rights Reserved) PHASE COMPENSATION OUT GAIN CC CC 1 10 100 1000 330pF 22pF 2.2pF none 7 RCL– FEB 20101 APEX − PA119UREVC PA119CE • PA119CEA ABSOLUTE MAXIMUM RATINGS Product Innovation From SUPPLY VOLTAGE, +VS to –VS OUTPUT CURRENT, within SOA POWER DISSIPATION, internal INPUT VOLTAGE, differential INPUT VOLTAGE, common mode TEMPERATURE, pin solder — 10 sec TEMPERATURE, junction1 TEMPERATURE, storage OPERATING TEMPERATURE RANGE, case SPECIFICATIONS PARAMETER TEST CONDITIONS 2 INPUT OFFSET VOLTAGE, initial OFFSET VOLTAGE, vs. temperature OFFSET VOLTAGE, vs. supply OFFSET VOLTAGE, vs. power BIAS CURRENT, initial BIAS CURRENT, vs. supply OFFSET CURRENT, initial INPUT IMPEDANCE, DC INPUT CAPACITANCE COMMON MODE VOLTAGE RANGE3 COMMON MODE REJECTION, DC TC = 25°C TC = 25°C to +85°C TC = 25°C TC = 25°C to +85°C TC = 25°C TC = 25°C TC = 25°C TC = 25°C TC = 25°C TC = 25°C to +85°C TC = 25°C to +85°C, VCM = ±20V GAIN OPEN LOOP GAIN at 10Hz OPEN LOOP GAIN at 10Hz GAIN BANDWIDTH PRODUCT at 1MHz POWER BANDWIDTH, AV = 100 POWER BANDWIDTH, AV = 1 TC = 25°C, RL = 1KΩ TC = 25°C, RL = 15Ω TC = 25°C, CC = 2.2pF TC = 25°C, CC = 2.2pF TC = 25°C, CC = 330pF OUTPUT VOLTAGE SWING3 VOLTAGE SWING3 VOLTAGE SWING3 SETTLING TIME to .1% SETTLING TIME to .01% SLEW RATE, AV = 100 SLEW RATE, AV = 10 TC = 25°C, IO = 4A TC = 25°C to +85°C, IO = 2A TC = 25°C to +85°C, IO = 78mA TC = 25°C, 2V step TC = 25°C, 2V step TC = 25°C, CC = 2.2pF TC = 25°C, CC = 22pF POWER SUPPLY VOLTAGE CURRENT, quiescent TC = 25°C to +85°C TC = 25°C THERMAL RESISTANCE, AC, junction to case4 RESISTANCE, DC, junction to case RESISTANCE, junction to air TEMPERATURE RANGE, case TC = 25°C to +85°C, F > 60Hz TC = 25°C to +85°C, F < 60Hz TC = 25°C to +85°C Meets full range specifications NOTES: * 1. 2. 3. 4. CAUTION 2 MIN PA119 TYP ±.5 10 10 20 10 .01 5 1011 6 ±VS–15 ±VS–12 70 104 74 111 88 100 3.5 250 ±VS–5 ±VS–1.5 ±VS–3 ±VS–.75 ±VS–1 ±VS–.5 .3 1.2 600 900 650 ±15 –25 80V 5A 75W 40V ±VS 300°C 175°C –65 to 150°C –55 to 125°C PA119A MAX MIN TYP MAX UNITS ±.35 ±.75 5 15 * * 5 50 * 3 25 * * * * mV µV/°C µV/V µV/W pA pA/V pA MΩ pF V dB * * * * * * dB dB MHz MHz kHz * * * 750 * * * * * * * V V V µs µs V/µs V/µs * * V mA ±3 30 200 100 * * ±35 100 ±40 * 120 * * 1.46 1.84 30 1.64 * * 2.0 * * * +85 * * °C/W °C/W °C/W °C The specification of PA119A is identical to the specification for PA119 in applicable column to the left. Long term operation at the maximum junction temperature will result in reduced product life. Derate internal power dissipation to achieve high MTTF. The power supply voltage for all specifications is the TYP rating unless noted as a test condition. +VS and –VS denote the positive and negative supply rail respectively. Total VS is measured from +VS to –VS. Rating applies if the output current alternates between both output transistors at a rate faster than 60Hz. The internal substrate contains beryllia (BeO). Do not break the seal. If accidentally broken, do not crush, machine, or subject to temperatures in excess of 850°C to avoid generating toxic fumes. PA119U PA119CE • PA119CEA POWER DERATING 70 CURRENT LIMIT, ILIM (A) CL 20 10 0 0 200 OUTPUT VOLTAGE, VO (V) SLEW RATE, (V/s) 400 100 80 60 40 10M 100M 10K 100K 1M FREQUENCY, F (Hz) POWER SUPPLY REJECTION, PSR (dB) COMMOM MODE REJECTION, CMR (dB) COMMON MODE REJECTION 120 PA119U 0 –30 –50 0 2 4 6 10 20 40 60 100 200 400 COMPENSATION CAPACITOR, CC (pF) 1K 10 –20 40 20 VIN = 2V AV = 10 tr = 10ns RL = 15W 20 –10 100 80 21 15 11 | +VS | + | –VS | = 80V 8 100K 200K 600K1M 2M 4M 8M FREQUENCY, F (Hz) 50 100 150 200 250 300 TIME, t (ns) POWER SUPPLY REJECTION 100 80 60 40 20 0 1K 10K 100K 1M 10M 100M FREQUENCY, F (Hz) 20M INPUT NOISE PULSE RESPONSE 30 RL = 15W 5 30 F SLEW RATE VS. COMP. 1000 800 600 1 2 3 4 OUTPUT CURRENT, IO (A) RL = 15W 41 2pF –V 58 .2p 1.0 0 POWER RESPONSE 80 +V 0.5 30 40 50 60 70 80 TOTAL SUPPLY VOLTAGE, VS (V) =2 100 1K 10K 100K 1M 10M 100M FREQUENCY, F (Hz) .6 =2 0 OUTPUT VOLTAGE SWING .8 CC F RCL = ∞ .5 0pF 0p F 1.0 1.0 = 33 33 20 2p pF RCL = 1.2W 1.2 CC 40 2. 22 1.5 1.4 CC 60 2.0 1.5 VOLTAGE DROP FROM SUPPLY (V) 80 .27 W 0 –50 –25 0 25 50 75 100 125 CASE TEMPERATURE, TC (C) 25 50 75 100 125 150 CASE TEMPERATURE, TC (C) SMALL SIGNAL RESPONSE 100 2.5 OUTPUT VOLTAGE, VO (VPP ) 30 =0 QUIESCENT CURRENT 1.6 INPUT NOISE VOLTAGE, VN (nV/ √ Hz) 50 40 OPEN LOOP GAIN, AOL (dB) R 3.0 60 –20 CURRENT LIMIT 3.5 30 COMMON MODE VOLTAGE, VCM (VP–P) INTERNAL POWER DISSIPATION, P(W) 80 NORMALIZED QUIESCENT CURRENT, IQ (X) Product Innovation From 70 20 15 10 7 5 3 10 1K 100 10K 100K FREQUENCY, F (Hz) 1M COMMON MODE VOLTAGE 65 60 55 50 45 40 10 100 1K 10K 100K 1M 10M FREQUENCY, F (Hz) 3 PA119CE • PA119CEA Product Innovation From GENERAL Please read Application Note 1 "General Operating Considerations" which covers stability, supplies, heat sinking, mounting, current limit, SOA interpretation, and specification interpretation. Visit www.Cirrus.com for design tools that help automate tasks such as calculations for stability, internal power dissipation, current limit; heat sink selection; Apex Precision Power’s complete Application Notes library; Technical Seminar Workbook; and Evaluation Kits. CURRENT LIMIT Q2 (and Q25) limit output current by turning on and removing gate drive when voltage on pin 2 (pin 7) exceeds .65V differential from the positive (negative) supply rail. With internal resistors equal to 1.2Ω, current limits are approximately 0.5A with no external current limit resistors. With the addition of external resistors current limit will be: ILIM = .65V RCL +.54A To determine values of external current limit resistors: RCL = .65V ICL – .54A PHASE COMPENSATION At low gain settings, an external compensation capacitor is required to insure stability. In addition to the resistive feedback network, roll off or integrating capacitors must also be considered when determining gain settings. The capacitance values listed in the external connection diagram, along with good high frequency layout practice, will insure stability. Interpolate values for intermediate gain settings. 1. The current handling capability of the MOSFET geometry and the wire bonds. 2. The junction temperature of the output MOSFETs. The SOA curves combine the effect of these limits and allow for internal thermal delays. For a given application, the direction and magnitude of the output current should be calculated or measured and checked against the SOA curves. This is simple for resistive loads but more complex for reactive and EMF generating loads. The following guidelines may save extensive analytical efforts: 1. Capacitive and inductive loads up to the following maximums are safe: CAPACITIVE LOAD ±VS 40V .1µF 11mH 30V 500µF 24mH 20V 2500µF 75mH 15V ∞ 100mH 2. Safe short circuit combinations of voltage and current are limited to a power level of 100W. 3. The output stage is protected against transient flyback. However, for protection against sustained, high energy flyback, external fast-recovery diodes should be used. SUPPLY CURRENT The PA119 features a class A/B driver stage to charge and discharge gate capacitance of Q7 and Q19. As these currents approach 0.5A, the savings of quiescent current over that of a class A driver stage is considerable. However, supply current drawn by the PA119, even with no load, varies with slew rate of the output signal as shown below. SAFE OPERATING AREA (SOA) SOA 10 5 4 3 ST DY t= 30 s s ST AT TC=25°C E 10 20 30 40 50 80 100 INTERNAL VOLTAGE DROP SUPPLY TO OUTPUT, VS-VO (V) 4 SUPPLY CURRENT VOUT = 60VP-P SINE RL = 500 Ω 300 200 100 0 30K 10 0m 0m EA 2 1 t= 400 SUPPLY CURRENT, IS (mA) OUTPUT CURRENT FROM +VS or -VS The MOSFET output stage of this power operational amplifier has two distinct limitations: INDUCTIVE LOAD 100K 300K 1M 3M FREQUENCY, F (Hz) 10M OUTPUT LEADS Keep the output leads as short as possible. In the video frequency range, even a few inches of wire have significant inductances, raising the interconnection impedance and limiting the output current slew rate. Furthermore, the skin effect increases the resistance of heavy wires at high frequencies. Multistrand Litz Wire is recommended to carry large video currents with low losses. PA119U PA119CE • PA119CEA Product Innovation From THERMAL SHUTDOWN STABILITY The thermal protection circuit shuts off the amplifier when the substrate temperature exceeds approximately 150°C. This allows the heatsink selection to be based on normal operating conditions while protecting the amplifier against excessive junction temperature during temporary fault conditions. Thermal protection is a fairly slow-acting circuit and therefore does not protect the amplifier against transient SOA violations (areas outside of the steady state boundary). It is designed to protect against short-term fault conditions that result in high power dissipation within the amplifier. If the conditions that cause thermal shutdown are not removed, the amplifier will oscillate in and out of shutdown. This will result in high peak power stresses, destroy signal integrity, and reduce the reliability of the device. Due to its large bandwidth, the PA119 is more likely to oscillate than lower bandwidth power operational amplifiers. To prevent oscillations a reasonable phrase margin must be maintained by: 1. Selection of the proper phase compensation capacitor. Use the values given in the table under external connections and interpolate if necessary.The phase margin can be increased by using a larger capacitor at the expense of slew rate. Total physical length (pins of the PA119, capacitor leads plus printed circuit traces) should be limited to a maximum of 3.5 inches. 2. Keep the external sumpoint stray capacitance to ground at a minimum and the sumpoint load resistance (input and feedback resistors in parallel) below 500Ω. Larger sumpoint load resistances can be used with increased phase compensation and/or by bypassing the feedback resistor. 3. Connect the case to any AC ground potential. 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