Download Godoy, P.A., S. Chung, T.W. Barton, D.J. Perreault, and J.L. Dawson, “A Highly Efficient 1.95-GHz, 18-W Asymmetric Multilevel Outphasing Transmitter for Wideband Applications,” 2011 International Microwave Symposium , pp. 1-4, June 2011.

2011 International Microwave Symposium, pp. 1-4, June 2011.
A Highly Efficient 1.95-GHz, 18-W Asymmetric Multilevel Outphasing
Transmitter for Wideband Applications
Philip A. Godoy, SungWon Chung, Taylor W. Barton, David J. Perreault, and Joel L. Dawson
Microsystems Technology Laboratories, Massachusetts Institute of Technology, Cambridge, MA 02139, USA
Abstract— A 1.95-GHz asymmetric multilevel outphasing
(AMO) transmitter with class-E GaN power amplifiers (PAs)
and discrete supply modulators is presented. AMO transmitters achieve improved efficiency over envelope tracking (ET)
transmitters by replacing the continuous supply modulator with
a discrete supply modulator implemented with a fast digital
switching network. Outphasing modulation is used to provide the
required fine output envelope control. A 4-level supply modulator
is implemented that allows for fast and efficient discrete envelope
modulation with up to 28-V supply voltages using low-voltage
gate drivers and time-alignment logic. With two class-E GaN
PAs that achieve 62.5% power-added efficiency (PAE) at 40dBm peak output power, the AMO transmitter delivers 42.6dBm peak output power at 1.95-GHz. For a 16-QAM signal at
36-dBm output power, the transmitter achieves 44.2/42.8/41.4%
average system efficiency and 2.0/2.1/3.1% EVM for 10/20/40MHz channel bandwidth, respectively.
Index Terms— power amplifier, base station, outphasing, LINC,
discrete supply modulator, digital predistortion
Multilevel DC/DC
AMO
Modulator
With
Predistortion
A
ࢥ
Time Delay
Alignment
A1
A2
ࢥ1
ࢥ2
Combiner
PA1
Phase
Modulator
PA2
Predistorter
Training
Fig. 1. Asymmetric multilevel outphasing (AMO) architecture with discrete
supply modulator.
Vdd
EL=90°
@3fc
EL=140°
EL=90°
EL=25°
EL=58°
Vin
10pF
EL=14°
10pF
EL=90°
@3fc
I. I NTRODUCTION
Achieving high efficiency for wideband, high peak-toaverage power ratio (PAPR) signals is one of the most difficult
problems for modern wireless transmitters. As one of the
most promising high-efficiency PAs, an envelope tracking (ET)
PA [1], [2] has been demonstrated with excellent efficiency
with 3.84-MHz bandwidth WCDMA signals. ET PAs use a
switching power converter to continuously modulate the supply
of a PA to improve efficiency at significant power backoff.
The tradeoff that these systems face is ultimately between
modulation bandwidth and efficiency in the power converter;
in general, the higher the modulation bandwidth, the lower
the efficiency. By including a feedforward path in the supply
modulator, a continuous supply modulator for ET PAs was
demonstrated to modulate 10-MHz signals with more than
50% PAE [1]. As the bandwidth increases, however, it becomes
increasingly difficult for the supply modulator to maintain high
efficiency with high power handling capability.
Fig. 1 shows the asymmetric multilevel outphasing (AMO)
architecture with discrete supply modulator [3], [4], which
has the potential to allow even larger baseband bandwidths
while maintaining high average efficiency and high linearity.
The supply modulator for AMO transmitters only provides
a discrete set of envelope amplitudes for the PAs. Fine
envelope amplitude modulation is achieved by outphasing
modulation. Unlike ET PAs, the discrete supply modulator can
operate open-loop without feedback or oversampling. Thus,
high-effficiency and wide-bandwidth supply modulation are
allowed. In this paper, we demonstrate the highest-efficiency
AMO implementation reported in the literature.
Switch
Switch
EL=28°
EL=20°
Vout
10pF
50
GaN HEMT
EL=90°
EL=90°
@2fc
EL=90°
@2fc
EL=57°
10pF
Vbias
Fig. 2.
Circuit schematic of the 10-W 1.95-GHz class-E GaN PA.
II. C LASS -E G A N P OWER A MPLIFIER
Fig. 2 shows the circuit schematic of the 10-W 1.95-GHz
class-E GaN PA designed for the AMO transmitter prototype. The PA was designed using the methodology described
in [5]. The harmonic terminations were implemented using
quarter-wave open-circuit stubs. Load-pull and source-pull
simulations were performed in the Agilent advanced design
simulator (ADS) to characterize the optimum output and
input impedances for the Cree GaN CGH40010 transistor.
The maximum PAE from simulation, including losses from
printed circuit board (PCB) traces and discrete components,
was 80.1% with 39.4-dBm output power and 12.8-dB gain
at 𝑉𝐷𝐷 of 28 V. Fig. 3 shows the drain voltage and current
waveforms after deembedding package parasitics, exhibiting
the zero-voltage switching characteristics of class-E operation.
Fig. 4 shows the fabricated class-E GaN HEMT PA, made
with a low-loss Rogers 4350B dielectric. The PA was designed
80
70
60
50
40
30
20
10
0
-10
-20
4.0
Voltage
Current
V1
V2
V3
V4
3.5
3.0
4.7 V
2.5
2.0
1.5
1.0
0.5
Current [A]
Voltage [V]
2011 International Microwave Symposium, pp. 1-4, June 2011.
Isolating
DC−DC
Converter
Vsup
5V
0.0
-0.5
-1.0
0
200
400
600
800
1000
Isolation
Barrier
Isolation
Barrier
Isolation
Barrier
Isolation
Barrier
Fine Delay
Alignment
Fine Delay
Alignment
Fine Delay
Alignment
Fine Delay
Alignment
Time [ps]
3.3V
Fig. 3. Simulated drain voltage and current for the class-E GaN PA after
deembedding package parasitics, indicating class-E operation. The design
frequency was 2.14 GHz.
VDD
Fundamental
input match
Fig. 6.
4-level discrete supply modulator.
3rd harmonic
/4
III. D ISCRETE S UPPLY M ODULATOR
VOUT
VIN
/4
Fundamental
output match
2nd harmonic
Vbias
Photograph of the fabricated class-E GaN PA.
90
Efficiency [%]
80
45
Pout
Drain Eff.
40
70
35
60
30
50
PAE
Pout
PAE
40
25
Drain Eff.
Gain
30
20
15
Gain
20
10
10
5
0
0
10
12
14
16
18
20
22
24
26
Output Power [dBm] / Gain [dB]
Fig. 4.
28
Supply Voltage [V]
Fig. 5. Measured PAE, drain efficiency, output power, and gain of the class-E
GaN PA at 1.95 GHz.
for a target operating frequency of 2.14 GHz, but after fabrication the optimum frequency was found to be 1.95 GHz. Fig. 5
shows the measured PAE, drain efficiency, output power, and
gain of the implemented Class-E GaN PA at 1.95 GHz. At
the 40-dBm peak output power, the PAE is 62.5% with 13dB gain. The peak PAE is 67% with 20-V supply voltage.
Although drain efficiency is better at lower supply voltage, the
PAE is degraded due to the fixed input power used to drive
the PA. The PA operating frequency was chosen for maximum
peak output power.
Fig. 6 shows the circuit schematic of the 4-level discrete
supply modulator implemented in this work. Silicon LDMOS
transistors from Polyfet (L8821P) were used for the supply
voltage switches, and reverse current flow was prevented by using SS16 Schottky diodes from Fairchild Semiconductor. The
gates of the power supply switches are driven with NC7WZ04
inverters from Fairchild Semiconductor. A 1-W isolated, unregulated dc-dc converter (TI model DCP010505BP) provides
the 5-V power supply for the inverters driving the supply
switch gates, whose reference ground potential is determined
by the supply modulator amplitude setting. This low-voltage
configuration reduces the power loss associated with driving
the gates of the power supply switches, which must block
voltages up to 28 V. Another source of power loss in the
discrete supply modulator is due to the diode voltage drop,
which is acceptable in base-station applications where the
supply voltages are as high as 28 V. The inverter gate drivers
are driven by Si8423 digital isolators from Silicon Labs, which
are in turn driven by MC100EP196B programmable delay
chips from ON-Semiconductor. The delay chips implement
fine delay adjustment for time alignment between the supply
switch control signals as well as time alignment between the
amplitude and phase paths.
Fig. 7 shows the measured step response of the AMO
transmitter output envelope when the supply voltage is changed
from 7.5 V to 13 V for both PAs. The envelope settles in less
than 15 ns. The nonzero settling time is caused by the parasitic inductances and capacitances of the discrete components
and the PCB traces. Although nonzero settling time of the
supply modulator introduces error-vector magnitude (EVM)
and spectrum degradation, 30–40-dBc adjacent channel power
ratio (ACPR) with up to 40-MHz channel bandwidth can be
obtained without any step response calibration or correction.
Because the phase modulator typically provides much faster
settling time, when higher ACPR is desired, a phase-domain
2011 International Microwave Symposium, pp. 1-4, June 2011.
Phase
Modulator
VDD=7.5V
Baseband
I−channel
DAC
VDD=13V
Coarse
Delay
FPGA
Fine Delay
Alignment
BPF
DAC
DAC
2GHz LO
Coarse
Delay
Fine Delay
Alignment
V4
V3
Discrete
Supply
V4
V3
V2
V1
Modulator
V2
V1
Circulator
PA
Fig. 7. Measured step response of the transmitter output envelope when the
supply voltage is changed from 7.5 V to 13 V.
Combiner
Preamp
Phase
Modulator
AMO Modulation
/ Predistortion
DAC
2GHz LO
Baseband
Q−channel
Coupler
Circulator
BPF
Preamp
PA
Coupler
(a)
envelope-correction technique could be implemented.
IV. T RANSMITTER M EASUREMENT R ESULTS
(b)
Fig. 8. Prototype implementation of the 18-W 1.95-GHz AMO transmitter
with discrete supply modulators and class-E GaN PAs. (a) block diagram, (b)
photograph.
0.009
80
PDF
LINC
AMOͲ2
AMOͲ4
VDD
60
0.006
40
0.003
20
0
Ͳ16
Ͳ14
Ͳ12
Ͳ10
Ͳ8
Ͳ6
Ͳ4
NormalizedOutputPower(dB)
Ͳ2
TotalEfficiency(%)
0.012
ProbabilityDensity
Fig. 8 shows the prototype 18-W, 1.95-GHz AMO transmitter, which uses two class-E GaN PAs and two discrete
supply modulators. Digitally pre-distorted baseband data was
generated by MATLAB after characterizing the AMO transmitter with a narrowband training signal. The two phase
modulators were each implemented by a 16-bit dual-channel
DAC evaluation board from Analog Devices (AD9779A). An
FPGA provides the digital inputs to the phase modulators
and the discrete supply modulators, and also performs coarse
time alignment between the phase and amplitude paths. Fine
time alignment is performed by programmable delay lines in
each supply modulator. The supply modulators provide two
independent supply voltages selected from four different power
supplies to the two class-E PAs.
Fig. 9 shows the measured AMO transmitter total efficiency
vs. output power as the supply voltage to both PAs is varied.
Here we define the total efficiency as 𝜂𝑡𝑜𝑡 = 𝑃𝑜𝑢𝑡 /(𝑃𝐷𝐶 +
𝑃𝑖𝑛 ). It should be noted that the efficiency numbers include
the insertion loss from the isolating power combiner, which is
approximately 0.4 dB. For a given number of supply voltage
levels, this data can be used to find the optimum supply
voltages that maximize the efficiency for a given probability
density function (PDF) [3]. In this work, we test the system
with a 16-QAM signal with a PAPR of 6.5 dB, the PDF
of which is shown in Fig. 9. The corresponding optimum
efficiency curves for both 2 and 4 available supply voltages
are also shown in the figure.
Fig. 10 shows the measured static nonlinearity of the AMO
transmitter output amplitude and phase depending on the
supply voltage combination and the outphasing angle. There
are 7 different curves, each for a different combination of
supply voltages for the two outphased PAs. A lookup table
(LUT) constructed from the data in Fig. 10 implements the
digital predistortion (DPD) to correct the static nonlinearity.
Fig. 11 shows the average system efficiency, ACPR, and
EVM of the prototype AMO transmitter for the 16-QAM signal
0
0
Fig. 9. Measured efficiency vs. output power for the AMO prototype, along
with the optimized amplitude levels for a 16-QAM signal with 6.5-dB PAPR.
The PDF of the 16-QAM signal is also shown.
transmission with various channel bandwidths from 2.5 to 40
MHz. For comparison purposes, we also show the results for
the LINC (linear amplification of nonlinear components) case,
when only a single supply voltage is used for both PAs. The
2011 International Microwave Symposium, pp. 1-4, June 2011.
10
DAC
LINC
LINC w/ DPD
AMO4 w/ DPD
Output Spectrum [dBr]
0
-10
-20
-30
-40
-50
-60
-70
1.935
Fig. 12.
ACPR2 / AMO-4
ACPR2 / LINC
0
5
10 15 20 25 30 35 40
Channel Bandwidth [MHz]
4.0
45
3.5
40
3.0
35
2.5
30
2.0
Effic. / AMO-4
Effic. / LINC
EVM / AMO-4
EVM / LINC
25
20
15
1.5
1.960
1.965
Transmit spectrum for the 2.5-MHz bandwidth 16-QAM signal.
DAC
LINC
LINC w/ DPD
AMO4 w/ DPD
0
-10
-20
-30
-40
-50
-60
-70
1.0
1.80
0.5
10
1.945
1.950
1.955
Frequency [GHz]
10
Output Spectrum [dBr]
ACPR1 / AMO-4
ACPR1 / LINC
50
EVM [%]
-30
-35
-40
-45
-50
-55
-60
-30
-35
-40
-45
-50
-55
-60
Average System Efficiency [%]
ACPR [dBc]
ACPR [dBc]
Fig. 10.
Measured static nonlinearity of the AMO transmitter output
amplitude and phase. Each curve corresponds to a different combination of
supply voltages for the two outphased PAs.
1.940
1.85
1.90
1.95
2.00
Frequency [GHz]
2.05
2.10
0.0
0
5 10 15 20 25 30 35 40
Channel Bandwidth [MHz]
Fig. 11. Performance comparison for LINC and 4-level AMO with 6.5-dB
PAPR 16-QAM transmission for various channel bandwidths.
sampling rate of the digital data was set to 10x the symbol
rate for all channel bandwidths except for 40 MHz, in which
case the sampling rate was 200MHz (5x the symbol rate).
This is due to the speed limitation of the DACs used in
the phase modulators. The average system efficiency numbers
include the dynamic power consumption from the discrete
supply modulators and is defined as 𝜂𝑠𝑦𝑠 = 𝑃𝑜𝑢𝑡 /(𝑃𝐷𝐶 +
𝑃𝑖𝑛 + 𝑃𝑆𝑢𝑝𝑀 𝑜𝑑 ). From the plots, we can see a significant
improvement in the efficiency for 4-level AMO vs. LINC.
However, the ACPR and EVM are degraded because of the
occurrence of glitches in the output envelope due to the finite
settling time of the discrete supply modulators. It should be
noted that the efficiency of the AMO transmission only begins
to slowly degrade when the signal bandwidth exceeds 10 MHz.
This demonstrates the relatively high efficiency of the discrete
supply modulators compared to continuous supply modulators
for wideband signal transmission.
Fig. 12 and Fig. 13 show the spectrum of the 16-QAM signal
transmission at 2.5-MSym/s and 40-MSym/s, respectively. The
curves labeled “DAC” show the spectrum using just the phase
modulators without the PAs. At 20-MHz, 37-dB ACPR1 and
2.2% EVM is achieved with 42.8% average system efficiency.
The LINC transmission at the same symbol rate achieves 38dB ACPR1, 2.1% EVM, and 14.6% efficiency. The efficiency
improvement with 4-level AMO vs. LINC is almost 3x.
Fig. 13.
Transmit spectrum for the 40-MHz bandwidth 16-QAM signal.
V. C ONCLUSION
We demonstrate a 1.95-GHz asymmetric multilevel outphasing (AMO) transmitter for 3G/4G base-station applications.
With two Class-E GaN PAs that achieve 62.5% PAE at 40-dBm
peak output power, the AMO transmitter delivers 42.6-dBm
peak output power at 1.95 GHz. For a 16-QAM signal at 36dBm output power, the transmitter achieves 44.2/42.8/41.4%
average system efficiency and 2.0/2.1/3.1% EVM for 10/20/40MHz channel bandwidth, respectively. The results illustrate the
potential of the AMO architecture for both high-efficiency and
wide-bandwidth power amplification.
R EFERENCES
[1] C. Hsia et al., “Wideband high efficiency digitally-assisted envelope
amplifier with dual switching stages for radio base-station envelope
tracking power amplifiers,” in IEEE MTT-S Int’l Microwave Symp. Dig.,
May 2010, pp. 672–675.
[2] J. Moon et al., “Doherty amplifier with envelope tracking for high
efficiency,” in IEEE MTT-S Int’l Microwave Symp. Dig., May 2010, pp.
1086–1089.
[3] S. Chung et al., “Asymmetric multilevel outphasing architecture for multistandard transmitters,” in IEEE Radio Freq. Int. Circ. Symp. Dig. Tech.
Papers, June 2009.
[4] J. Hur et al., “Highly efficient uneven multi-level LINC transmitter,”
Electronics Lett., vol. 45, no. 16, pp. 837–838, July 2009.
[5] D. T.-T. Wu et al., “First-pass design of high efficiency power amplifiers
using accurate large signal models,” in IEEE Wireless and Microwave
Tech. Conf., 2010, pp. 1–4.