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Transmitter and Receiver Circuits MODULATOR CIRCUITS Solid-state AM modulators are usually biased class C, and the output RF signal amplitude is modulated by causing the collector supply voltage to vary with the modulation signal. As illustrated in Figure 6-19, the collector voltage is varied by transformer-coupling (TA) the audio modulation signal in series with Vcc. The RF choke (RFC) and RF bypass capacitor (RF-BP) isolate the carrier from the audio section and power supply line. 1 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Transmitter and Receiver Circuits Figure 6-19. Transistor AM modulators. Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan 2 Transmitter and Receiver Circuits If the RF amplifier of Figure 6-19 had a constant supply voltage (nomodulation), the collector vote e would be Vcc. With RF input, the collector voltage can swing almost to ground (less Vcc(sat)) on the negative swing and to 2V, on the positive swing, thereby maintaining an average Vcc. For 100% modulation, however, the peak supply voltage reaches 2Vcc, and therefore the positive peak RF swing can reach 4Vcc, as seen in the collector waveform of Fig. 6-19. As a result, the choice of transistor for the AM modulator must include high breakdown voltage. In particular, BV > 4Vcc (6-29) Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan 3 Transmitter and Receiver Circuits Since an amplitude-modulated signal must maintain amplitude linearity, any amplifier following the modulator must be class A or, as is usually the case in transmitters, class B push-pull. Figure 6-20 shows an all-solid-state transmitter output with a tuned push-pull amplifier final. R1 can be adjusted to bias Q3 and Q4 for class A or class B (actually, almost class AB) operation. Sometimes R1 is replaced by a diode that will help trackout temperature variations of VBE for Q3 and Q4. Diodes D1 and D2 form a network to allow the driver Q1 to be modulated on positive modulation peaks when Vc exceeds Vcc. Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan 4 Transmitter and Receiver Circuits Figure 6-20. All-solid-state AM modulator/transmitter. 5 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Nonlinearities in RF Mixers An RF mixer requires a nonlinear circuit component. Two broad categories cover the range of nonlinearities: 1). the general nonlinearity that is expressed mathematically by a power series; 2). the nonlinearity produced in switching or sampling mixers. 6 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Nonlinearities in RF Mixers The output versus input nonlinearity of any device can be expressed mathematically by a power series such as io = Io +avi +bvi2 +cvi3 +… +nvin (7-1) illustrated in Figure 7-6. Part (a) shows a mixer circuit with the LO and RF voltages transformer-coupled to a high-frequency diode. The currents produced in the diode produce an output voltage vo a ioZ in the tuned-output circuit impedance Z. 7 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Nonlinearities in RF Mixers Figure 7-6b shows the nonlinear input-output V-I curve of the diode. Id is the bias current developed for the purpose of placing the diode at the best operating point for optimum-mixer performance. Curves 1, 2, and 3 show the linear (avi), 2nd-order (bvi2), and 3rd-order (cvi3) nonlinearities at the bias point Id. The constant a is the slope of the io curve at the bias point, and similarly, constants b and c are scaling factors for the first two nonlinearities. For a simple 2nd-order nonlinear device such as a FET, io = Id + avi + bvi2 (7-2) 8 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Nonlinearities in RF Mixers Figure 7-6. Mixer circuit (a) and diode characteristic curve (b) showing nonlinear input-output relationship. 9 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Nonlinearities in RF Mixers The inputs vLO and vRF are linearly added by transformer T1 to produce vi = vLO + vRF = coswLOt + AcoswRFt. Substituting for vi in Equation 7-2 gives (7-3) The fourth term, which is a result of the 2nd-order (square law) nonlinearity, can be expanded by a trigonometric identity as bcos2(wLOt) = (b/2)(1+cos(2wLOt)) = (b/2) + (b/2)cos(2wLOt) This shows that some dc and LO second-harmonic currents are produced by distortion. Likewise the last term of Equ. (7-3) will yield dc and RF second-harmonic currents. Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan 10 Nonlinearities in RF Mixers These, together with the scaled input current (terms 2 and 3 of Equ. (7-3)), yield io = Id + (input signal and 2nd harmonic current) + 2bA.cos(wLOt).cos(wRFt) , (7-4) where Id = (Io +b/2 +bA2/2). The last term of Equation (7-4) is the mixer product of interest, 2bA.cos(wLOt) x cos(wRFt). 11 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Nonlinearities in RF Mixers By trigonometric identity, 2bA.cos(wLOt) x cos(wRFt) = bA.cos(wLO + wRF)t + bA.cos(wLO - wRF)t which shows that the product of two input sinusoidal signals will yield sum and difference frequency currents with amplitudes of one-half of the product term. Replacing the product in Equ. (7-4) with this result gives the total circuit current io. If the parallel LC circuit of Figure 7-6 is tuned to the difference frequency wLO - wRF = 2p(fLO - fRF), then the output voltage developed across the high-impedance tank will be proportional to the RF input voltage but will have an intermediate frequency of fIF = fLO - fRF. Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan 12 RF Mixer Circuits Figure 7-8. Mixer circuits (a)&(b). 13 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan RF Mixer Circuits In the simple diode mixer of (1), the LO is capacitively coupled. C1 is always small in value because the LO frequency is almost always higher than he RF; more importantly, the high impedance of the small C1 allows for isolation of the oscillator circuit from the mixer, while T1 provides for impedance matching of the mixer to the RF circuit. Mixer (2) is an active mixer in which a net conversion gain is achieved. The base-emitter junction is driven into nonlinear or switching operation by the large LO signal. Mixing occurs in the input junction, and the transistor current gain and tuned-collector circuit produce power gain at IF. The overall circuit gain is 6 dB less than the circuit would produce with an IF signal as an input. Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan 14 RF Mixer Circuits Figure 7-8. Mixer circuits (c). Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan 15 RF Mixer Circuits In Mixer (3) makes use of a dual-gate MOSFET to achieve good isolation between the LO and RF input circuits. Also illustrated in this VHF mixer for a TV tuner (RF front-end) is a circuit connection for automatic gain control (AGC). 16 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan RF Mixer Circuits Figure 7-8. Mixer circuits (d). 17 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan RF Mixer Circuits Integrated circuits, such as the CA (or LM) 3028A which operates to 120 MHz, also give good isolation in this mixer configuration. As seen schematically in mixer (4), the 3028A has a differential amplifier with a (normally) constant current source Q1 which is modulated by the LO. Thus, sum and difference frequencies are produced. This IC has numerous applications, including AM modulator and cascade amplifier with AGC. 18 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan RF Mixer Circuits Figure 7-8 Mixer circuits (e)&(f). 19 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan RF Mixer Circuits Circuit (5) is the double-balanced mixer which achieves isolation between each of the three ports. Circuit (6) illustrates the frequency converter. The IF tuned circuit has very low impedance at the much higher LO frequency (due to C1), and C2 provides a feedback path for the local oscillator. Also, the oscillator tuned circuit is essentially a shortcircuit to the IF due to L1; therefore the emitter circuit has low impedance for good gain. 20 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Balanced Mixers Application 21 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Balanced Mixer A primary application of a mixer is to shift the spectrum of a signal, usually down in frequency. This application is typically encountered in equipment such as receivers, spectrum analyzers, and radio repeaters. The careful balancing of the Schottky barrier diodes in the 10514 and 10534 Mixers provides excellent suppression of the local oscillator and input frequencies at the output port. This usually eliminates the need for a BPF following the mixer. Figure 1 shows typical mixer performance with -3 dBm input at the RF (or "R") port and +7 dBm input at the local oscillator (or "L") port. Conversion loss is typically less than 6 dB. Excess noise contributed by the diodes is negligible above 50 kHz, so noise figure is essentially equal to conversion loss. 22 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Balanced Modulator The balanced modulator application is an extension of the frequency conversion mode just described. Here, however, the signal of interest is shifted up from base band and centered on a suppressed carrier, typically at radio frequencies. Both time and frequency domain presentations of the balanced modulation output are shown in Figure 2. Balanced modulator operation is useful, along with the proper filters, for generating the proper frequency components for SSB-SC systems. It is also an inexpensive way to synthesize a two-tone RF signal with stable frequency spacing from an audio and an RF signal source. Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan 23 Balanced Modulator Figure 2. Frequency and time domain presentations of balanced modulator output. Note suppression of carrier and absence of higher order sidebands. 24 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Amplitude Modulator The balanced modulator output can easily be converted to a conventional amplitude modulated format by reinserting the carrier at the output of the mixer. This is illustrated in Figure 3. Another, although less desirable, way of re-establishing the carrier is to unbalance the unit by applying a direct current bias along with the modulation signal to the "X" port. Generally, however, lower distortion and better control of modulation index result if the carrier is reinserted after the mixer. 25 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Amplitude Modulator Fig. 3. AM signal constructed by reinserting carrier on balanced modulator output. Time domain display compared modulation signal with AM signal envelope. Block diagram illustrated method for synthesizing such a signal. 26 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Pulse Modulator The broad bandwidth of the 10514 and 10534 Mixers allows the current controlled attenuation mode to be extended to pulse modulation. Figure 4 shows both time and frequency domain presentations of a 100 ns wide burst of a 50 MHz carrier. Here power during burst is +4 dBm. Clean bursts with only minimal video feedthrough are provided for RF levels between 0 and +10 dBm by excellent bandwidth and balance. Generally envelope rise time and shape are preserved if frequency spectrum of desired RF output is within bandwidth of "L" and "R" ports (500 MHz for HP10514, and 150 MHz for HP-10534). Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan 27 Pulse Modulator Figure 4. Frequency and time domain presentations of pulse modulated 50 MHz carrier. Pulse burst is 0.1 ms wide; repetition rate is 1 MHz. Note fast, clean turn-on and turn-off characteristics provided by broad bandwidth of control, or “X” port. Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan 28 Phase Detector With the same frequency present at “R” and “L” ports, the dc output from the “X” port indicates the phase difference between the two input signals. As this phase difference varies from 0° to 90° to 180°, the output will vary from a maximum negative value to zero to a maximum positive value, following a cosine function. A typical plot of this response is shown in Figure 5. The specified low frequency (1/f) noise, not usually given for mixers, makes the HP-10514 and 10534 particularly useful as phase detectors in phase lock loops. They are also valuable in determining the short term, relative stability of high quality signal sources. Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan 29 Phase Detector Figure 5. Plot shows typical dc output from “X” port when HP-10514/534 is used as phase detector. Low 1/f noise qualifies these mixers for use in phase locked sources requiring low fm or phase noise. The accompanying block diagram represents such a stabilized source. 30 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Current Controlled Attenuator The HP-10514 and 10534 Mixers can also be used as variable attenuators under the control of a direct current input. The transfer function between “L” and “R” ports depends on the current entering or leaving the normal difference frequency (or "x") port. As shown in Figure 6, a control current of 10 mA will give minimum attenuation, while greater than 40 dB isolation occurs at zero current. Note the wide range of linear attenuation and the low minimum loss. 31 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Current Controlled Attenuator Figure 6. Typical attenuation L to R vs. dc "X" control current; fL = 30 MHz at 0 dBm. Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan 32 Mixing Modulated Signals When a modulated signal is up or down converted in a properly designed mixer the output will contain the same modulation as at the input. For example, suppose a carrier fR is amplitudemodulated by a sinusoid of frequency fm. The time domain and frequency spectrum for this AM signal are sketched in Figure 7-10 as the input to a mixer. The (ideal) LO spectrum is also sketched with fL > fR. This is high-side injection. 33 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Mixing Modulated Signals The mixer output spectrum may be sketched by computing the sum and difference frequencies between the LO and each of the RF spectral components. Assigning numerical values to fR, fm, and fL will make this easy, and if you want to portray accurate amplitudes, make the amplitudes of each sum and difference output component one-half that of the RF input spectrum. The results for the IF spectrum are shown at the filter output. Please notice that the time sketch has the same AM index and envelope rate (fm) and shape. Only the average frequency (fIF) and amplitudes (1/2) are different from the RF input. The one-half amplitude (-6 dB) is ideal for a second-order mixer nonlinearity; actual mixer efficiencies will vary. 34 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Mixing Modulated Signals Figure 7-10. Down-conversion of AM signal (high-side LO injection to mixer). Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan 35 Tuning the Broadcast Receiver Figure 7-11 is the schematic of a simple but representative AM superheterodyne receiver available in kit form from Graymark Incorporated. The separate circuits do not track correctly the RF resonant frequency and the LO frequency to produce exactly 455 kHz. While the RF is being tuned from 540 to 1600 kHz (3:1 frequency range), the LO must tune from 995 to 2055 kHz (2:1 frequency range). 36 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Tuning the Broadcast Receiver A compromise front-end alignment scheme is used in order to avoid sideband (therefore audio) distortion. One procedure used for receiver alignment is as follows (refer to Figure 7-11): 1). Disable the LO with a short to ground from the negative lead of C2. 2). AC-couple a very low-level 455-kHz signal to the base of Q2 and tune T2 through T4 for a maximum level at the output of T4. This aligns the narrowband IF amplifiers (use a sweep generator if available). 3). Alternatively, AM the input generator and tune for the maximum at the detector output. Keep the output as low as possible by reducing the signal generator power to avoid saturation. 37 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Tuning the Broadcast Receiver 4). Remove the LO short; set the signal generator to 540 kHz and the radio tuning dial (C1 to the low end of its range. Tune T1 for 995 kHz using a counter, or maximize the receiver output at T4 or the detector output (with AM). 5). Set the generator to 1600 kHz and C1 to the high end of its range. Since C1 will have a minimum capacitance, the RF and LO trimmers C1A and C1B can be tuned for fLO = 2055 kHz and maximum receiver output. Keep the signal generator power low to avoid saturation. 38 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan Tuning the Broadcast Receiver Fig. 7-11. AM superheterodyne receiver. 39 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan AGC CIRCUITS AGC system is shown in our prototype AM receiver in Figure 7-11, where the detector diode is reversed. The controlled amplifier is an NPN transistor so that, as the AGC voltage increases (more negative voltages), the Q3 bias current decreases. When the IF amplifier gain is controlled by a reduction of current, the system is known as reverse-AGC. Reverse-AGC is used in satellite and space transponders or battery-operated receivers where minimum power consumption is important. Otherwise, in ground stations and home broadcast systems (AM, FM, and TV), forward-AGC is used. 40 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan AGC CIRCUITS With forward-AGC the amplifier bias point is set at IF, as shown in Figure 7-26. As the received signal strength increases the AGC voltage increases. The fed-back AGC voltage increases the already hightransistor bias current (Ip ~ 10 to 20 mA) so that the base region is flooded with current, thereby spoiling the transistor current gain b. Forward-AGC has the advantage of increasing the amplifier power-handling capability as gain is reduced in order to maintain the best overload characteristic on strong signals when it is most needed. 41 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan AGC CIRCUITS Figure 7-26. Transistor gain versus bias current. Bias points for forward-and reverse-AGC. 42 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan AGC CIRCUITS Figure 7-27 shows an example of a circuit arrangement that would have to operate as forward-AGC. With no input signal the detector diode will be reverse-biased due to the positive voltage established by the R1-R2 voltage divider. These two resistors, along with R3, set Q1 for high gain with collector current in the range of IF in Figure 7-26. As the positive peaks of the received signal at the detector increase, the base bias voltage on Q1 increases. Hence, the bias must be must be set near IF for the gain to be reduced for increasing input signal. Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan 43 AGC CIRCUITS FIGURE 7-27. AGC network. Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan 44 AGC CIRCUITS Assume that the R1-R2 voltage divider sets the bias point and AGC line to R1Vcc/(R1+R2) = V1 (= VAGC) Not only does this establish the Q1 base bias voltage, but it also sets AGC threshold. This is because the IF input to the detector must reach V1 + Vd on peaks in order for the diode to start conducting and the AGC voltage to begin rising. Vd is the diode cut-in voltage drop, which we can assume to be approximately 0.2V for germanium diodes. 45 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan AGC CIRCUITS If V1 is already established by the choice of R1, R2 and Vcc, then the detector input power required for the onset of AGC (threshold) must be Pi (threshold) = Vpk2/2Zi = (Vi + 0.2)2/Rdc (7-44) Rdc is the dc resistance from the diode cathode to ground, which is, for this circuit, (7-45) where (Ri)Q1 = (b+1)R3 is the dc input resistance of Q1. 46 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan AGC CIRCUITS AGC of a transistor is a very nonlinear function. For reverse-AGC the reduction in base voltage reduces emitter current and, therefore, voltage gain due to the increase in re. This is approximately linear until VBE goes below about 0.55V. However, IF amplifiers are designed to amplify the input power. At the same time that voltage gain is reduced due to current-starving of the transistor, the current gain is also decreasing---this is in itself a nonlinear function. Forward AGC relies on spoiling the b by flooding the transistor base region with current. Therefore, putting a value on AGC threshold is mostly empirical, which often includes, a potentiometer in the circuit. 47 Prof. J.F. Huang, Fiber-Optic Communication Lab. National Cheng Kung University, Taiwan