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Gerhard Mercator Universität Gesamthochschule Duisburg Department of Optoelectronics Annual Report 1996/97 Gerhard-Mercator-Universität Gesamthochschule Duisburg Fachbereich Elektrotechnik Fachgebiet Optoelektronik ZHO Lotharstr. 55 D - 47057 Duisburg Germany Head: Prof. Dr. rer. nat. D. Jäger Tel: +49-203-379-2340 Fax: +49-203-379-2409 Email: URL: [email protected] http://www.oe.uni-duisburg.de Editor: R. Buß 4 The Center for Solid-State Electronics and Optoelectronis (ZHO) After only nine month of construction the topping-out-ceremony of the new Center for Solid-State Electronics and Optoelectronics (Zentrum für Halbleitertechnik und Optoelektronik, ZHO) has been celebrated on June, 4th, 1997. The center is planned to be the new home of the Department of Optoelectronics and the SolidState Electronics Department in the beginning of October 1998. It consitst of two parts: The cleanroom building with an area of approx. 470 m2 and the building for the offices and laboratories with an area of approx. 1200 m2. Pictures of the Center for Solid-State Electronics and Optoelectronics taken in Dec. 1997 5 Table of Contents 1 Foreword 7 2 Members of the Department 9 3 Research 3.1 Optical Networks 3.1.1 3.1.2 3.1.3 3.1.4 3.1.5 High-speed, high-power travelling-wave photodetectors Fabrication and characterization of a travelling-wave photodetector Simulation of the microwave generation of a travelling-wave photodetector Determination of RF-equivalent circuit elements of travelling-wave photodetectors using network analysis Polarization insensitive waveguide modulators on InP 3.2 Optical Interconnects and Processors 3.2.1 3.2.2 3.2.3 3.2.4 3.2.5 3.2.6 Neurotechnology: Retina Implant Analysis of the optical energy and signal transfer module for an artificial vision prosthesis Development of an optical signal and energy transmission system Infrared data link for rotating display-systems Nonlinear hybrid GaAs/AlGaAs multilayer-heterostructures for high-speed information processing 8 x 8 LED arrays integrated with 64 channel Si-driver circuits 3.3 Millimeterwave Electronics 3.3.1 3.3.2 3.3.3 Picosecond pulse generation on monolithic nonlinear transmission lines using high-speed InP-HFET diodes Millimeter wave power generation nonlinear transmission lines Nonlinear RTD circuits for high-speed A/D conversion 3.4 Optical Sensor Systems 3.4.1 3.4.2 3.4.3 MQW-Electroabsorption-Modulator for application in a fiberoptic fieldsensor Photovoltaic cells for fiber optic EMC - Sensor power supply Time- and frequency-domain electro-optic field mapping of nonlinear transmission lines 11 11 11 14 17 20 25 29 29 32 35 38 41 48 51 51 55 59 62 62 64 66 6 TABLE OF CONTENTS 3.4.4 3.4.5 Characterization of monolithic microwave integrated circuits by heterodyne electro-optic sampling Development of an experimental setup for field probe measurements on nonlinear transmission lines 3.5 Technologies for Optoelectronic Components and Systems 3.5.1 3.5.2 3.5.3 3.5.4 Development of a measurement system for the optical characterization of full-colour-LED-displays Opinion poll on the evaluation of the legibility of LED-based displays Evaluation of possible improvements to enhance the UV-power efficiency of a xenon flashlamp system Construction of a flip chip device for bonding integrated circuits 4 Teaching activities 71 74 77 77 80 83 89 93 4.1 Lectures, excercises, and practical studies 93 4.2 Seminars and colloquia 95 4.3 Doctoral, Diploma, and Graduate theses 100 5 Publications and presentations 103 6 Guide to the Department of Optoelectronics 107 7 1 Foreword The Fachgebiet Optoelektronik at the Gerhard-Mercator-Universität Duisburg has a tradition of research and teaching excellence dating back to its establishment in 1989/90. This Report provides a summary of major research involvements and teaching activities reviewing also publications and presentations by the members of the institute . The last two years were characterized by a further-broadening of our scientific networks and additional projects funded by various external institutions. A first key research area is microwave photonics with special emphasis on travelling-wave photodetectors, electro-absorption modulators and nonlinear optoelectronic devices, where system aspects played a continuously increasing role. Special emphasis has further been laid upon the two-dimensional electro-optical characterization of monolithic microwave integrated circuits and high-speed devices. As a third topic, nonlinear optics and optoelectronics in III-V-heterostructures and pulse compression in nonlinear MMICs were studied in detail. Additionally to our activities within the Sonderforschungsbereich 254, mayor funding has been provided by the Retina Implant project (EPIRET) and the collaborative programme on the development of an EMC-field sensor, where our Fachgebiet acts as the coordinator. We are already proud of looking back to a relevant exhibition during the Laser 97 Fair in Munich. Another remarkable event was the International Topical Meeting on Microwave Photonics (MWP) held at the moated castle Schloß Hugenpoet in September 1997 which was organized by our Fachgebiet, D. Jäger being simultaneously the Chair of the International Steering Committee on MWP (details on next page). As a result of their remarkable research work, Dr.-Ing. G. David received a fellowship of the Alexander von Humboldt-Stiftung, funding a twoyears stay at the University of Michigan. Moreover, Dr.-Ing. A. Stöhr was awarded a grant to carry out research work at the Communications Research Laboratories, Ministry of Posts & Telecommunications in Tokyo. Further, D. Jäger received the title Professor Onorific from the University of Brasov/Romania and became the Chair of the German IEEE/LEOS Chapter. Besides the usual and obligatory courses, the Fachgebiet Optoelektronik offered in 1997 a new lecture Einführung in die Multimediatechnik Technologien, Systeme, Anwendungen. Moreover, the Institute was involved in the Duisburg Summerschool for Women, the Tag der Forschung and the organisation of research activities on photonic bandgap materials in the framework of the Forum Materialforschung in our university. Finally, we note with great pleasure that T. Alder and D. Kalinowski have been awarded the University Price for excellent diploma theses. I wish to thank all friends inside and outside the university for their continuous encouragement and assistance. Also, I would like to express my sincere thanks to all members of the Institute for their efforts and contributions to our success in optoelectronics. Duisburg, September 1998 8 T he 1997 International Topical Meeting on Microwave Photonics (MWP97) has been held from September 3 through 5 in the historical buildings of the majestic 17th century moated Castle Schloß Hugenpoet, situated in the beautiful Ruhrtal countryside in the south of Essen near Duisburg. This 7 th Topical Meeting in the series on this subject followed those in Cernay-la-Ville, France (1994), Keystone, U.S.A. (1995), and Kyoto, Japan (1996). It was the first one held in Germany and has been organized by the GerhardMercator-Universität-Duisburg. On September 3, the Meeting started with a Workshop entitled Photonic technologies for phased array antennas where 6 invited papers addressed recent results in this continously growing field of research. In the Plenary Session on September 4, 3 invited speakers, Dr. R. Heidemann, Alcatel SEL AG, Stuttgart, Germany, Dr. M.J. Wale, GEC-Marconi Materials Technology Ltd, Northampton, U.K., and Dr. D. Novak, The University of Melbourne, Australia, presented lectures on the topic Microwave Photonics: Present and Future. The regular conference program consisted of 7 sessions on topics such as Optical generation of microwave signals, Optoelectronic modulators, mixers, and receivers, Microwave photonic systems, Fibre radio networks, Modelling in microwave photonics, and Microwave photonics for measurements. Each session has been opened by invited senior technologists from France, Hong Kong, Japan, U.K., U.S.A., and Germany having provided additional impetus to this multi-disciplinary research area of microwave photonics. The whole program included furtheron a video session via internet with KDD in Tokyo, Japan, a poster session and was completed by a postdeadline session. More than 100 papers have been submitted from 13 countries showing the increasing interest of scientists and engineers in this area. After careful evaluation the Technical Program Committee has recommended 69 papers - 54 oral and 15 poster - for presentation. Additionally, 4 papers have been selected during the conference for the postdeadline session. 140 scientists and engineers have registered for the Topical Meeting. In addition to the technical program, a Partners Program, a Welcome/Barbecue Party and a Gala Dinner have been organized. The Meeting has been sponsored by the Deutsche Forschungsgemeinschaft (Bonn, Germany), Hewlett-Packard GmbH (Ratingen, Germany), Institut für Mobil- und Satellitenfunktechnik GmbH (Kamp-Lintfort, Germany), Lucent Technologies (Allentown, P.A., U.S.A.), and the Gerhard-Mercator-Universität Duisburg. Moreover it has been cooperatively sponsored by the IEEE MTT-S and LEOS including the German Chapters. The organizers of MWP97 look back to a very fruitful conference and look forward to the upcoming meetings MWP98 in Princeton, N.J., U.S.A., and MWP99 in Melbourne, Australia. 9 2 Members of the Department Department of Optoelectronics ZHO, Lotharstr. 55 47057 Duisburg, Germany fon: +49 203 379-2340 fax: +49 203 379-2409 Head of the Department Jäger, Dieter Prof. Dr. rer. nat. Secretary Gappa, Ulrike Tempel, Karin Optoelectronics SFB 254 Scientists Alles, Martin Alder, Thomas Braasch, Thorsten Buß, Rüdiger David, Gerhard Groß, Matthias Heinzelmann, Robert Hülsewede, Ralf Jäger, Irina Kalinowski, Dirk Knigge, Steffen Kremer, Ralf Redlich, Stefan Schmidt, Manuel Stöhr, Andreas Wingen, Georg Zumkley, Stefan Dipl.-Ing. Dipl. Ing. Dipl.-Phys. Dipl.-Ing. Dr.-Ing. Dipl.-Phys. Dipl.-Ing. Dipl.-Phys. Ph. D. Dipl.-Ing. Dr.-Ing. Dr.-Ing. Dipl.-Ing. Dipl.-Phys. Dr.-Ing. Dipl.-Phys. Dr. rer. nat. Guest Scientists Dragoman, Mircea Johnson, Roger Lee, Chi Mezentsev, Vladimir Wendrix, Veronique Prof. Dr. Dipl.-Ing. Prof. Dr. Ph. D. Dipl.-Ing. Technicians Mang, Sabine Schedwill, Veronique Slomka, Heinz Students Appenrodt, Nils Balci, Senay Baumeister, Thomas Berger, Oliver Boscher, Guido Brings, Ludger Bussek, Peter Christoffers, Niels Einweck, Michaela Engel, Thomas Ervens, Jutta Hedtke, Ralph Heinzdorf, Michael Jabs, Mirco Kampermann, Claus Kreuder, Andreas Lüdeke, André Lotz, Oliver Manh-Duc, Ngo Meininger, Mark Moeck, Jens-Peter Neuhaus, Birgit Ponellis, Bernd Reintjes, Stefanie Rogall, Michael Spiegeler, Britta Wenning, Michael Weimann, Uwe Ing. grad. 10 3 RESEARCH 3.1 Optical Networks 3 Research 3.1 Optical Networks 3.1.1 High-speed, high-power travelling-wave photodetectors M. ALLES R ecently, new communication systems combining the advantages of wireless transmission and fiber optics have been proposed. Applications are transmission of traffic information, multimedia, or Internet access. These systems operate usually at an optical wavelength of 1.55µm at frequencies up to 60GHz. Since the photodetector should generate as much electrical power as possible, the travelling-wave photodetector under investigation has to fulfill these requirements. Introduction Novel wireless millimeterwave communication systems have been proposed by several groups, see for example [1-4]. These systems use fiber optics to transmit the signals over large distances. Fiber optic cables have very low attenuation of about 0.8dB per km, which is also independent from the signal frequency, whereas loss of coaxial cables is about 100dB per km. An optical source generates a heterodyne signal at 1.55µm, where the difference frequency of the two optical carriers corresponds to the electrical millimeterwave frequency. The data signal is modulated on one optical carrier. The optical heterodyne signal is distributed to an antenna station using usual fibre optic components. At this antenna station, the photodetector converts the optical signal to a millimeter- 11 wave, which is amplified and transmitted to a remote station. These communication systems should be used for traffic information, digital video, multimedia, or Internet access. This system approach leads to some important requirements for the photodetector. The photodetector has to work at an optical wavelength of standard fiber optics, i.e. at 1.3/1.55µm. In a further point, the photodetector should be capable to operate in the high frequency regime to generate electrical signals in the millimeterwave regime. Additionally, the photodetector has to generate as much electrical output power as possible to reduce the requirements of the electrical amplifier used in the antenna station. Usually, high-speed photodetectors are fabricated as lumped devices which are RC-time limited. This means that high bandwidths can only be reached if the device size is scaled down to the micrometer regime. Dimensions of photodetectors operating at 100GHz are about 10µm2 [5]. Due to the small device size the photodetectors can only operate at low optical input powers to avoid saturation effects in the small volume. This limitation can be overcome if the travelling-wave concept is considered for the development of high-speed photodetectors [6]. High-speed travelling-wave photodetectors The travelling-wave photodetector is fabricated as an optical waveguide which is coupled to an electrical waveguide due to an optical absorption layer. The capacitance of the electrical waveguide is compensated by the inductance of the transmission line resulting in an electrical bandwidth which is not RC-time limited. This concept avoids scaling down the device dimensions. In contrast, the travelling-wave photodetector can be fabricated as a distributed device in order to reduce optical saturation effects. 12 3 RESEARCH The structure of the fabricated travelling-wave photodetector is depicted in Fig. 1. The device consists of an active region and a taper at the end of the structure, used for hybrid integration with electrical millimeterwave amplifiers. The photodetector is MBE-grown on a semi-insulating InP-wafer for operation in the 1.3/1.55µm regime. An InGaAlAs layer is used as an optical waveguide. An InGaAs absorbing layer, leakage coupled to the waveguide, generates electron-hole pairs. Finally, an InAlAs-layer as a cladding layer and an InGaAs/InGaAlAs superlattice as a Schottky-barrier enhancement layer are grown. Electrical waveguiding is achieved using a coplanar transmission line. The outer conductors form ohmic contacts to the n+ doped region of the optical waveguide, whereas the center conductor is fabricated as a Schottky contact. The depletion layer underneath the center con- ductor separates the photo-generated electronhole pairs in the absorption region. Since the travelling-wave photodetector uses the interaction of optical and electrical waves, the optical and the electrical phase velocity should be equal (phase-matching condition). This can be achieved due to the fact, that the Schottky-contact generates slow-wave effects on the electrical transmission line [7]. The efficiency of the travelling-wave photodetector can be calculated numerically using a distributed equivalent circuit model for generation and propagation of electrical waves on the coplanar transmission lines (see Calculation of the electrical millimeterwave generation of a travelling-wave photodetector in this annual report). A distributed current source describes the impressed photocurrent per unit length due to electron-hole generation in the depletion layer. The numerical calculation of the output power leads Fig. 1: Sketch of the travelling-wave photodetector. Popt and Pel are the input optical and output electrical powers, 3.1 Optical Networks to an electrical millimeterwave power of 18.4dBm at 40GHz for a travelling-wave photodetector with an active length of 1mm. The optical wavelength is 1.3µm and the input power is 0dBm per optical carrier. The electrical output power of the travellingwave photodetector has been measured using an optical heterodyne setup. Two tunable 1.3µm Nd:YAG-lasers generate an optical heterodyne signal with beating frequencies in the millimeterwave regime. The generated electrical signal is measured using a coplanar on-wafer probe and a spectrum analyzer. The fabricated travelling-wave photodetector leads to an electrical output power of -19.7dBm at a reverse bias voltage of 12V and a frequency of 40GHz using the heterodyne measurement setup, which is in good agreement with the theoretically determined value. Since the capacitance of this device is 1.2pF, the resulting RC -time constant in a 50W system would lead to a 3dB-frequency of about 2.5 GHz, which is far below the measured frequency. This shows the validity of the travelling-wave concept to overcome RC-time limitation. 13 ing the epilayer and for fabrication of the travelling-wave photodetector. References [1] R. Heidemann, R. Hofstetter, H. Schmuck, 60GHz fibre-optic distribution technology for traffic information and multimedia, IEEE, MTT-S and LEOS Topical Meeting on Optical Microwave Interactions, Proc. pp. 133-136, Abbaye des Vaux de Cernay, 1994 [2] J. Park, K.Y. Lau, Millimetre-wave (39GHz) fibre-wireless transmission of broadband multichannel compressed digital video, Electron. Lett., pp. 474-476, Vol. 32, 1996 [3] D. Wake, C.R. Lima, P.A. Davies, Transmission of 60-GHz signals over 100km of optical fibre using a dual-mode semiconductor laser sources, IEEE Photon. Techn. Lett., pp. 578580, Vol. 8, 1996 [4] E. Boch, High bandwidth mm-wave indoor local area networks, Microwave Journal, pp. 152158, 1996 [5] K. Kato, A. Kozen, Y. Muramoto, Y. Itaya, T. Nagatsuma, M. Yaita, 110-GHz, 50%-efficiency mushroom-mesa waveguide p-i-n photodiode for a 1.55-µm wavelength, IEEE Photon. Conclusion Future wireless millimeterwave communication systems using optical heterodyne techniques are described. The requirements for highspeed photodetectors are discussed. Since high bandwidth and large electrical output power are needed simultaneously, travelling-wave photodetectors are under investigation. Up to now, an electrical output power of -19.7dBm at a frequency of 40 GHz could be measured. Acknowledgment The author would like to thank U. Auer (Fachgebiet Halbleitertechnik/-technologie) for grow- Techn. Lett., pp. 719-721, vol. 6, 1994 [6] D. Jäger, Optical Information technology, ed. S.D. Smith and R.F. Neale, Springer-Verlag, pp. 328-333, 1193 [7] D. Jäger, Slow-wave propagation along variable Schottky contact microstrip line, IEEE Trans. Microwave Theory and Techn., pp. 566-573, vol. 24, 1976 14 3 RESEARCH 3.1.2 Fabrication and characterization of a travelling-wave photodetector Fabrication of travelling-wave photodetectors For the fabrication of a travelling-wave photodetector several etch steps, a polyimide step V. WENDRIX AND M. ALLES and metalization steps are needed [2]. The MBEgrown wafers, which contain the necessary layecently, high-speed travelling-wave ers are depicted in Fig. 1 are fabricated in the photodetectors are under investigaDepartment of Optoelectronics. The travellingtion as optoelectronic power converters used wave photodetector contains in general three in future communication systems. In this different layers grown on a semi-insulating InP work the fabrication of high-speed travellingwafer. wave photodetectors is described. The fabAn InGaAlAs layer acts as optical waveguide, ricated devices are characterized using an InGaAs quantum well layer provides optical standard measurement techniques. The elecabsorption, and, finally, an InAlAs layer is used trical millimeterwave generation is deteras a cladding layer. The metalization of the travmined using an optical heterodyne setup with elling-wave photodetector is fabricated as an two 1.3µm Nd:YAG lasers. electrical coplanar waveguide. The center conductor forms a Schottky contact to the InAlAs Introduction layer. The outer metalization is evaporated on For future communication systems combinthe InGaAlAs-layer. This metalization is alloyed ing fiber optic links with wireless transmission in order to form ohmic contacts to the n-doped techniques, high-speed photodetectors are semiconductor. The taper is fabricated at the needed. These photodetectors should be used output end of the travelling-wave photodetecfor hybrid integration with millimeterwave amtor. In order to reduce millimeterwave attenuaplifiers. A taper at the output end of the phototion and for a characteristic impedance of 50W, detector facilitates flipchip- or wire-bonding with the metalization of the taper has to be fabricatadditional devices. The fabrication of travellinged directly on the semi-insulating InP-wafer. wave photodetectors with tapers is described in Therefore, an insulation of the edge of the methis work [1]. sas is necessary to prevent a short circuit between the center conductor and the outer metalization. The processing of the travellingwave photodetector starts with two InAlAs etch steps. The etching defines the InGaAs SQW/MQW lateral dimension of the optical InGaAlAs : Si waveguide and of the absorbing layInP er, cf. Fig. 2. R Fig. 1: Travelling-wave photodetector layer structure. 3.2 Optical Networks InP 15 (a) (b) Fig. 2: Etching of the two mesas, (a) cross-section, (b) top-view. InP (a) (b) Fig. 3: Polyimide step, (a) cross-section, (b) top-view. InP (a) Fig. 4: Evaporation of the metalization for the ohmic contacts, (a) cross-section, (b) top-view. (b) 408), Fig. 3. The polyimide is processed as a negative light sensitive resist and developed with a polyimide developer. A hard bake process at 350°C makes the polyimide resistant for the following process steps. In the next step, the metalization for the ohmic contacts is evaporated on top of the n-doped InGaAlAs-layer, Fig. 4. The metalization consists of Ge (30nm), Ni (5nm), and Au (300nm). The metallic layers are alloyed at about 550°C. This leads to ohmic contacts with low impedances between the metalization and the semiconductor. The fabrication of the travelling-wave photodetector finishes with the metalization of the center conductor and the taper, Fig. 5. Since the center conductor should form a Schottky contact to the semiconductor, it is necessary to evaporate Pt/Ti/ Pt/Au or Cr/Au. Measurements The characterization of the travellingwave photodetector is first done using current-voltage and capacity voltage measurements. The current-voltage measurement gives information about the InP (a) (b) functionality of the Schottky-contact diode Fig. 5: Processing of center conductor and taper, (a) formed between the center conductor and cross-section, (b) top-view. the outer metalization. As can be seen from Fig. 6, the fabricated devices show The fabrication of the mesa structure is done the typical current-voltage characteristic of a diusing wet chemical etching with a liquid etchant ode. consisting of H3PO 4:H 2O2 :H2 O (1:1:40). This In forward direction the current raises expoetch system has almost no effect on InP. nentially with increasing voltage. With reversed The insulation of the mesa edge is done usbias, the diode shows a high dark current which ing a non-conducting polyimide step (Probimid increases with the voltage applied to the device. The build-in voltage of this diode is about 0.6V. 16 3 RESEARCH with the area of the depletion layer A, the dielectric constant er e0, the density of donor dopants ND, the build-in voltage UB, the applied bias voltage U, Boltzmanns constant k, the temperature T, and the electron charge q. If the capacity is known, it is possible to calculate the density of donor dopants: 100 10 1 0.1 0.01 ND = − 0.001 -2 -1 0 U (V) 1 2 Fig. 6: Current-voltage measurement of a travelling- The capacitance-voltage measurements allows the determination of the doping level of the fabricated structures. The capacitance C of a depletion layer is given by: C=A Laser 1 lens Laser 2 lens ε r ε0 N D kT 2 − U B − U − q 2 ( dU qε r ε 0 A 2 d 1 C 2 ) The capacity-voltage measurement leads to a doping level of about 1018cm-3 in the InGaAlAs-layer and less than 1017cm-3 in the InGaAs and the InAlAs-layer. Optical heterodyne setup For the measurement of the optoelectronic conversion efficiency, a heterodyne setup with two Nd:YAG-lasers operating at 1.3µm is used. The wavelength of the lasers is adjustable by detuning the temperature of the laser head. Up to now, the two lasers illuminate the travellingwave photodetector via free space and a microscope lens. To facilitate the optical coupling to the Spektrumphotodetector the use of analyzer optical fibers has been investigated. This measurement setup is shown in Fig. 7. DUT Each laser is coupled on-wafer Bias-Tee to a monomode fiber. probe Both fibers are coupled to a third monomode fiMultiber using a GRIN-lens. meter Finally, this fiber is direct- Fig. 7: Heterodyne measurement setup using monomode fibers. 3.1 Optical Networks ly coupled to the device under test (DUT). The electrical measurement of the optoelectronically generated millimeterwave is achieved using a coplanar on-wafer probe, a bias-tee to separate the high-frequency and the dc-signals, a multimeter for photocurrent, and an spectrum analyzer to measure the amplitude of the millimeterwave in frequency domain. Conclusion In this report, the fabrication of travelling-wave photodetectors is described. Photodetectors have been processed successfully. Characterization of these devices is done using currentvoltage and capacity-voltage measurements. Finally, the heterodyne measurement setup has been investigated in order to facilitate optical coupling to the photodetectors using monomode fibers. References [1] V. Wendrix, Fabricatie en karakterisatie van een 17 3.1.3 Simulation of the microwave generation of a travelling-wave photodetector A. LUEDEKE AND M. ALLES I n this report the photoelectronic microwave generation of a coplanar InGaAlAs/InP Schottky-contact travelling-wave photodetector (TWPD) is analyzed by numerical solution of the wave-equations. Computational simulations of the electrical behavior of the travelling-wave photodetector are carried out using an equivalent circuit model. Introduction The electrical behavior of the travelling-wave photodetector can be described using the equivalent circuit model of Fig. 1 [1]. The travellingwave photodetector is fabricated as an electrical millimeter waveguide. Therefore all elements are per unit length. Traveling-Wave Photodetector, Diploma thesis, Department Toegepaste Natuurkunde, Vrije Universiteit Brussel in cooperation with V R’m dc L’ Fachgebiet Optoelektronik, Gerhard-MercatorUniversität Duisburg, 1996 [2] R. Haupt Experimentelle Untersuchungen zur Integration von Schottky-Kontakt-Varaktordioden R’hl für den Einsatz in periodischen Leitungs- G’rlz C’rlz strukturen, Graduate thesis, Fachgebiet I’p C’L Optoelektronik, Gerhard.Mercator-Universität Duisburg, 1995 C’b G’b Fig. 1: Equivalent circuit model of the travelling-wave photodetector. All elements are per unit length. 18 3 RESEARCH The impedances Rm and Rhl describe the longitudinal ohmic losses in the metalization and the semiconductor, respectively. The inductance of the electrical waveguide is taken into account with the inductance L. The conductance Grlz and the capacity Crlz consider the Schottky-contact depletion layer. The two elements Cb and Gb describe the behavior of the bulk material. An additional capacity CL is introduced for the electric field in air above the photodetector. The impressed current source IPh describes the optoelectric conversion in the absorbing quantum well layer. ’ ’ ’ ’ ’ ’ ∂ 2u ’ Y1 ⋅Y 2 + Y1 ⋅Y 3 + Y 2 ⋅Y 3 = u ⋅W ⋅ ∂ z2 Y 1’ + Y ’2 I ’p ( z ) ⋅ + W ’ ⋅ Y ’2 where W ist the impedance and Y1 , Y2 and Y 3 the admittances of the transmission line. I p(z) is given by I ’p ( z) = q ⋅ ηopt ⋅ (1 − R ) ⋅ α opt ⋅ Popt h ⋅ν ⋅e Numerical solution The wave-equations of the travelling-wave photodetector are derived by using a summarized equivalent circuit model, shown in Fig. 2, which based on the model above. The equation of the complex voltage amplitude along the transmission line can be shown as − (α opt + jβopt ) ⋅ z (2) where hopt is the internal quantum efficiency, aopt is the optical absorption coefficient, hn is the photon energy, Popt is the incident light power, R is the reflection of interface device/air and bopt is the optical phase coefficient. i + ∂ i dz ∂z W ’dz i (1) Y 1’ + Y ’2 iB iA u1 Y 1’dz I ’p ( z) dz u3 u Y ’3 dz u u+∂ ∂z i A + I ’p ( z) dz Y ’2 dz u2 Fig. 2: Summarized equivalent circuit model of the travelling-wave photodetector. All elements are per unit length. 3.1 Optical Networks 19 Using the finite differences method [2], the numerical solution of equation (1) is given by u k +1 + u k −1 − h 2 ⋅ I ’p ( z k ) ⋅ uk = 2 + h ⋅W ⋅ 2 ’ W ’ ⋅ Y 2’ Y1 + Y 2 ’ ’ Y 1’ ⋅ Y 2’ + Y 1’ ⋅ Y 3’ + Y 2’ ⋅ Y 3’ Y 1’ + Y 2’ (3) with h = b−a N (4) where a is the beginning and b the end of the transmission line. N is the number of discrete points, where the voltage is calculated (0 £ k £ N). The loads at the two ends of transmission line are defined as Z1and Z2, the characteristic resistance of the transmission line is Z. According to microwave theory there exist reflections r1 and r2 at the two ends z = 0 and z = l of the device. The boundary condition for this problem is de- termined by the current voltage relation at the specific load resistance. u ( z = 0) = u 0 = Z 1 + r1 1 ⋅ ⋅ ⋅ [ u1 − u − 1 ] W ’ 1 − r 1 2h (5) u (z = l ) = u N = − Z 1+ r2 1 ⋅ ⋅ ⋅ [u N +1 − u N −1 ] W ’ 1 − r 2 2h (6) The solution of equation (3) is computationally calculated. Voltage (mV) Simulation For the simulation of photoelectrical microwave generation a frequency of 40GHz is taken. The wavelength of the optical sources is 1.3µm and the light power of the two beams are both 1mW. In the following calculation the quantum efficiency h opt of photoelectrical conversion is assumed to be 1 and the incident 40 light energy is fully and uniformly 35 coupled to the active layer, so the 30 optical reflection R is 0. Fig. 3 shows the voltage dis25 tribution along z direction. The 20 reflection factor at z = 0mm is 1 15 and at z = 1mm the reflection is 0. 10 The dashed line shows the re5 sult by using a simplified equivalent circuit model, in which Cb, Gb 0 0 0,2 0,4 0,6 0,8 1 and CL are neglected. An existz (mm) ing simulation program, which calculates the solution analyticalFig. 3: Voltage distribution along z direction for a simplified ly, is based on this model. The (interrupted line) and the fully equivalent circuit model. 20 3 RESEARCH Voltage (mV) Conclusion This report presents the numerical solution of the photoelectrical microwave generation of a coplanar travelling-wave photodetector. The wave equations are solved by the finite differences method. The results of the simulations have shown, that the efficiency is reduced by 10% relative to using a simplified equivalent circuit model and that a further optimization is possible. 60 50 40 30 20 10 0 0 0,1 0,2 0,3 0,4 0,5 z (mm) References [1] Fig. 4: Voltage disribution along z direction for Z 2 = 50W D. Jäger, R. Kremer, Trav- elling-wave optoelectronic devices (dashed line) and Z2 = Z (solid line) for a device length of for microwave applications, Proc. IEEE, MTT-S and LEOS Topical Meeting on Optical Microwave Interactions, pp. comparison with the numerical simulation, which is based on the complete equivalent circuit model, shows, that the efficiency is reduced by 10%. One possibility to optimize the device relative to the output-voltage is to reduce the length of the transmission line. In the following simulation the length of the device is set to 0.5mm. Fig. 4 shows the results by using Z 2 = Z (solid line) and Z2 = 50W (dashed line). In case of adaptation the efficiency is raised by 44% relative to a device length of 1mm. In case of mismatching there is a high voltage at the end of the device, but the efficiency has not raised because of higher load at the device´s end. Another possibility of optimization is to reduce the value of Rm. Simulations have proved that the efficiency is raised by 20%, if Rm is set at 0W (device length is 1mm). 11-14, 1994, France [2] D. Marsal, Finite Differenzen und Elemente: numerische Lösung von Variationsproblemen und partiellen Differentialgleichungen, SpringerVerlag, Berlin/Heidelberg, 1989 3.1.4 Determination of RF-equivalent circuit elements of travelling-wave photodetectors using network analysis O. BERGER AND M. ALLES I n this report, the determination of the elements of the equivalent circuit model of coplanar waveguides is described. Since the longitudinal and the transverse complex 3.1 Optical Networks impedance of the equivalent circuit model can be calculated from the characteristic impedance and the propagation coefficient measured with a network analyzer, it is possible to compute the equivalent circuit straight forward. This method has been used to determine the equivalent circuit model of travelling-wave photodetectors. Results and comparison to theory and other measurements are shown. 21 cuit model directly from network analyser measurements. Equivalent Circuit Model The coplanar waveguide structure of the travelling-wave photodetector can be described using the distributed equivalent circuit model shown in Fig.1 [1]. Note that all elements are per unit length. The impedances RM and R describe the longitudinal ohmic losses in the metalization and the semiconductor, respectiveIntroduction ly. The inductance of the electrical waveguide Recently, 60GHz-travelling-wave photodetecis considered by the inductance L. The depletors are under development in the Fachgebiet tion layer of the Schottky-contact is taken into Optoelektronik. For characterization and further account with the conductance G and the caoptimization of the device, network analyzer RFpacitance C while GB and CB characterize the measurements are analyzed in a new way in bulk material of the semiconductor. The addiorder to determine the high-frequency equivational capacitance CL is associated with the lent circuit elements of the device. The impleelectrical field in air above the device. Finally, mented method determines the equivalent cirthe impressed current source IPh is introduced to describe the optoelectronic conversion within the absorbing layer. To determine the equivalent circuit elements it is necessary to make some simplifications. Usually, network analyser measurements take place without optical illumination of the device, therefore IPh can be neglected. The two capacitances C L and C B have much less influence on the behavior of the device in comparison to the Schottky-capacitance C and can be neglected. One can conclude, that in the longitudinal and in the transverse part of the equivalent circuit three elements are to be considered: two real impedances and one Fig. 1: Equivalent circuit model of the travelling-wave photodeimaginary one. The longitudinal tector. 22 3 RESEARCH and the transverse part of the equivalent circuit can be described separately with W ’= R ’( R’M + jω L’) R ’+ R’M + jω L’ and . G’ (G’+ jωC’) Y ’= B G’B + G’+ jωC ’ Using separate Smith charts for the longitudinal and the transversal part of the equivalent circuit to display the frequency dependence of W and Y, semicircles with two intersections with the real axis for ω = 0 and theory low-frequency measurements high-frequency measurements Ω R ’M mm 6.58 6.21 13.8 Ω R ’ mm 707 - 887 S G’ mm 6.7×10 -14 750×10 -12 1.3×10-3 874 1.1 2.54 nH L’ mm 0.466 - 0.62 pF C’ mm 1.39 1.17 0.91 S G ’B mm ω → ∞ can be determined. The intersections with the real axis are Tab. 1: Values for the elements of the equivalent circuit model of a travelling-wave photodetector. Z ’(ω = 0) = R ’R ’M , R ’+ R ’M Z ’(ω → ∞ ) = R ’ for the longitudinal part and Y ’(ω = 0) = G’G ’B , G ’+ G’B Y ’(ω → ∞) = G’B for the transverse part. Determination of the equivalent circuit elements The network analyser measures S-parameters from a device under test (DUT). The characteristic impedance Z and the propagation coefficient g are determined from this data. The new method calculates the impedance of the longitudinal part and the admittance of the transverse part using the relations W = g Z and Y = g / Z. The frequency dependence of both parts is displayed in separate Smith charts. An statistic-based algorithm fits a semicircle to the measured data. The intersections with the real axis are used to determine the real elements of the equivalent circuit model. With these results, it is possible to calculate the imaginary elements L and C for each frequency. This method has been implemented to an easy-tohandle windows-program with graphic features. The calculation results are displayed instantly on-screen. Results The equivalent circuit elements calculated using this method have been compared with results of low-frequency measurements and theoretical determined values [2], Tab. 1. As can 3.1 Optical Networks be seen from this table, the measurements are in good agreement with theoretically determined values. Only the two admittances show different values. In case of G the network analyzer measurement leads to a higher value. An explanation is, that the accuracy of the network analyzer makes it impossible to measure these low admittances. The theoretical value of the bulk admittance GR is much larger than the measured values. A reason is, that the metal-semiconductor resistance is neglected in the derivation of GB. Taking measurements at various bias voltages one can see that only the elements C and G regarding the depletion layer show major bias dependence while the other ones keep constant over the bias voltage. Fig.2 shows the characteristic impedance, Fig. 3 the phase and attenuation coefficient calculated from the measured data. The real part of the characteristic impedance rises at frequencies above 35GHz indicating that probably the major part of the electrical quasi-TEM wave on the structure changes from TEM to TM. The smithcharts with the measured data and the semicircle calculated as an approximation for the longitudinal part and the transversal part of the equivalent circuit are shown in Fig. 4. As is visible, the frequency dependence of the measured data leads to semicircles, indicating that the device under test can be described with the simplified equivalent circuit model described above. 23 40 30 re{Z} 20 10 0 im{Z} −10 −20 −30 0 10 20 frequency (GHz) 30 40 Fig. 2: Characteristic impedance of a travelling-wave photodetector. 4 4 3 3 2 2 1 1 0 0 10 20 30 frequency (GHz) Fig. 3: Phase coefficient and attenuation coefficient of a travelling-wave photodetector. 0 40 24 3 RESEARCH (b) (a) frequency frequency Fig. 4: Smithcharts for the longitudinal part (a) and for the transversal part (b). The frequency dependence of the inductance L and the capacitance C, shown in Fig. 5, can be determined directly with the program. Due to a suggested change of the quasi-TEM-Mode to a TM-mode, the inductance rises and the capacitance decreases with higher frequencies. Conclusions A method to determine the RF-equivalent elements directly from network analyser measurements has been developed. This method approximates the frequency dependence of the 1.0 measurement data in a Smith chart graphically without any further knowledge about the values to be determined. The measured results fit well with both theoretically determined and low-frequency measured values. It is also possible to examine the frequency dependence of the inductance and the capacitance. Acknowledgement The author would like to thank U. Auer (Fachgebiet Halbleitertechnik/-technologie) for growing the epilayer and for fabrication of the travelling-wave photodetector. 1.0 References [1] M. Alles, T. Braasch, D. Jäger, 0.8 0.8 High-speed coplanar Schottky travelling-wave photodetectors, Int. Conf. 0.6 0.6 on Integrated Photonics Research, Proc. pp. 380-383, Boston, USA, 1996 0.4 0.4 [2] O. Berger, Bestimmung der HFErsatzschaltbildelemente von Photo- 0.2 0.2 detektoren mit Hilfe der Netzwerkanalyse, Graduate thesis, 0.0 0 10 20 30 frequency (GHz) Fig. 5: Inductance and capacitance versus frequency. 0.0 40 Fachgebiet Optoelektronik, GerhardMercator-Universität Duisburg, 1997 3.1 Optical Networks 3.1.5 Polarization insensitive waveguide modulators on InP T. ALDER AND R. HEINZELMANN E lectroabsorption modulators using strained multiple quantum well (MQW) structure have been designed, fabricated and characterized. Utilizing the Quantum Confined Stark Effect (QCSE) due to high electric field underneath a Schottky-electrode, the absorption coefficient of the optical waveguide can be changed. The use of strained quantum wells enables an operation of the device, with almost no sensitivity to different polarisation. With this device an on/ off-ratio of 18.5dB has been achieved. 25 Introduction In general, the absorption change in a MQW structure is strongly polarization dependent [1]. From the viewpoint of system applications, a polarization insensitive or at least polarization independent modulator is desirable. Appropriate structures can be designed using quantum wells with tensile strain [2-3]. In this paper electroabsorption waveguide modulators using a strained InGaAs/InAlAs MQW structure in the electrooptical active region will be presented. Device structure and principle of operation A schematic diagram of the modulator structure is shown in Fig. 1. The modulators investigated utilize a nin-structure containing Si-doped InAlAs top and bottom cladding layers with thickness of 570nm and 1120nm, respectively. Fig. 1: Schematic diagram and cross section of the modulator. 26 3 RESEARCH The doping concentration of the bottom cladding layer is ND = 1×1017cm-3, whereas that of the top cladding layer is ND = 1×10 16cm-3. The non intentionally doped guide consists of 19 × 6nm thick InGaAs MQWs separated by 19 × 7.7nm thick InAlAs barriers. The structure was grown using the MBE machine of the Department of Optoelectronics. To examine the dependence of the device behaviour on contact geometry, a number of devices were fabricated with chrome-gold-Schottky-electrodes of different width, ranging from 8µm to 16µm. The Schottky-electrodes were manufactured by thermal evaporation of chrome and gold in ultra high vacuum. The waveguide structure was formed using wet chemical etching after evaporation of the Schottky contacts. The second contact, shown in Fig. 1 is carried out as Ohmic contact and was manufactured by thermal evaporation of germanium, nickel and gold in ultra high vacuum. As a reverse bias is applied to the Schottkyelectrode, there will be a high electric field underneath the Schottky-contact within the depletion region. This increases the absorption coefficient of the guide due to the quantum confined Stark effect (QCSE). In this way, the optical output power can be controlled electrically. Experimental results Fig. 2 shows the nearfield-pattern and the lateral profile of the optical waveguide mode at different reverse biases. From this figure, it can be seen, that the output becomes weaker as the applied reverse bias is increased. This behavior is due to the increased absorption coefficient in the optically guiding region. As the previous result shows, it is possible to control the optical output power by a reverse bias. In the following, systematic results on transmission changes will be presented. In Fig. 3 the transmission is plotted as a function of different reverse biases. From this figure, it can be seen, that within the range from -4.3V to -7.4V the transmission changes almost linear with the applied bias. The ratio between maximum and minimum transmission is 18.5dB. Furthermore it can be seen, that the change in transmission from 0V to about -4V is very low. This indicates that quantum wells were grown smaller than they were designed. Fig. 4 shows the transmission as a function of reverse bias for TE- and TM-polarization. It is evident from the figure, that for TE- and TM-po- 100 position 0V position -2V position -4V position -6V position -8V position transmission [%] 80 60 = 1,2mm 40 w = 14µm -10V 20 0 -10 Fig. 2: Nearfield-pattern and lateral profile of the optical waveguide mode at different reverse biases. -8 -6 -4 voltage [V] -2 Fig. 3: Transmission as a function of different reverse biases. 3.1 Optical Networks 27 maximum transmission change [dB] transmission [a.u.] larization the change in transmission is almost equal. While for TE1 polarization an on/off-ratio of 18,5 dB 18.5dB could be measured a 0,8 slightly smaller value of 17.2dB 17,2 dB appeared for TM-polarized light. 0,6 Considering only the linear range (the voltage range be0,4 tween -4.3V and -7.4V bias voltage) of the transmission characTE - polarisation TM - polarisation teristic, for TE- as well as 0,2 TM-polarization an on/off-ratio of 8.2dB is measured. 0 -10 -8 -6 -4 -2 To examine the dependence voltage [V] on device dimensions modulators with different Schottky-electrode Fig. 4: Transmission as a function of different reverse biases for widths from 8µm to 16µm were TE- and TM-polarisation. investigated. The results are shown in Fig. 5. As can be seen, there is no recognizable influence from contact ther for TE-polarization nor for TM-polarization. width on the maximum transmission change, neiThis is of major importance, as with a change in the contact width the propagation properties of the electrical waveguide can be fit to those of 20 the optical waveguide, for highspeed operation the travelling16 wave concept [4] can be applied. 12 TE - polarisation TM - polarisation 8 4 0 7 9 11 13 contact width [µm] 15 Fig. 5: Maximum transmission change as a function of different reverse biases for TE- and TM-polarization. 17 Conclusions Electroabsorption waveguide modulators based on strained InGaAs/InAlAs-MQW have been designed, fabricated and characterized. A maximum on/off-ratio of 18.7dB has been achieved. It could be shown, that the polarisation influence on the transmission behaviour was small, due to the influence of the strain in the quantum well region. Additionally 28 3 RESEARCH no influence of the contact dimensions on the transmission was observed. References [1] T. Aizawa, K. G. Ravikumar, R. Yamauchi, Polarisation Independent Refractive Index Change In InGaAs/InGaAsP Tensile Strained Quantum Well , Electronics Letters, Vol. 29, No. 1, pp. 21 - 22, January 1993 [2] H. W. Wan, T. C. Chong, S. J. Chua, Considerations For Polarisation Insensitive Optical Switching and Modulation Using Strained InGaAs/ InAlAs Quantum Well Structure, IEEE Photonics. Techn. Lett., Vol. 3, No. 8, pp. 730 732, August 1991 [3] J. Shimizu, T. Hiroshima, A. Ajisawa, M. Sugimoto, Y. Ohta, Measurement of the polarisation dependence of field induced refractive index change in GaAs/AlAs multiple quantum well structures, Appl. Phys. Lett., Vol. 53, No. 2, pp. 86 - 88, 1988 [4] D. Jäger, R. Kremer, and A. Stöhr, Travellingwave optoelectronic devices for microwave applications, IEEE MTT-S 1995 International Microwave Symposium, Vol. 1, pp. 163-166, 1995 (invited paper) 3.2 Optical Interconnects and Processors 29 3.2 Optical Interconnects and Processors retina layer, and a wireless signal and energy transfer from RE to RS. The task of the Department of Optoelectronics in this project is the development of a device for optoelectronic signal and energy transfer into the eye. To achieve this, a prototype consisting of a laserdiode as transmitter and a receiver consisting of a monolithically integrated photovoltaic cell array and a photodiode, together with driving and receiving electronics, was manufactured. 3.2.1 Neurotechnology: Retina Implant M. GROSS AND R. BUSS T he Department of Optoelectronics is a member of a consortium of 14 German expert groups, working on the project EPI-RET: Retina Implant. This interdisciplinary project, funded by the Federal Ministry for Education, Science, Research and Technology (BMBF) in Germany, is developing a retina implant. This device is a neural prosthesis, designed for patients blinded by a disease where the outer retinal layer degenerates (retinitis pigmentosa or macula degeneration). It consists of three parts: a so-called retina encoder (RE) outside the eye, simulating the function of the retina, the retina stimulator (RS), a microchip placed on the retina with electrodes stimulating the ganglion cells in the outer Fig. 1: Signal and energy transmission into the eye Introduction The signal and energy transmission line described here is part of a technical system functioning as a vision aid for blind people who have lost their vision due to retinal degenerations, especially retinitis pigmentosa [1]. An often appearing kind of blindness is the partially degeneration of the retina, e.g. the disease retinitis pigmentosa, which is leading to blindness through following steps: The typical begin is the loss of the rod photoreceptors, causing night blindness. Next the cone photoreceptors are 30 dying off, beginning at the outer perimeter of vision. This leads to a tunnel vision and finally to total blindness, when the cones in the fovea are lost. However, while the photoreceptors are dying off, the nerve cells in the retina and subsequent parts of the central visual system are remaining mostly intact [1]. This leads to the possibility of developing a visual prosthesis with the ability to replace main parts of the retina that gives sight back to the visually impaired [2]. System description The whole system sketched in Fig. 1 consists of three main parts: > a retina encoder (RE), consisting of a CMOS camera and an artificial neural network (encoder) for image data processing, > a retina stimulator (RS), a flexible chip, epiretinal affixed, with µ-electrodes on the back, > a wireless signal and energy transmission line from the retina encoder to the retina stimulator. The system works as follows: First the high dynamic range CMOS camera generates a picture. This dataset is then reduced by an artificial neural network and transformed into digitally coded pulse trains (nerve signals), to which the ganglion cells can react. This corresponds to the data reduction from 120 million photoreceptors to 1 million ganglion cells by the human retina. The dataset is then optically transmitted at a rate of 1 Mbit/s to a microcontact foil on the retina (receiver), where eye movements of up to +/- 15°, measured from looking straight ahead have to be compensated. Together with this information transfer an optical bias is transmitted, supplying the driving circuit with 5 mW electrical power. The retina stimulator is a soft microcontact foil which is implanted adjacent to the 3 RESEARCH ganglion cell layer on the outer retinal limit. The µ-electrodes are stimulating the ganglion cells of the retina, thus transmitting the signals via the optic nerve to the visual cortex in the brain. Results The signal and energy transmission has been realized in a first prototype using a laser diode as transmitter and a photovoltaic cell array together with a photodiode as receivers for energy and signals, respectively. In the first step we designed the parts for the optical transmission, considering the boundary conditions given by the human eye and the technical demands of the whole device. For surgical reasons optical fibres cannot be used to connect transmitter and receiver directly. Therefore, a light source (e.g. a pigtailed laser diode) has to be fixed in front of the eye, transmitting the signal and energy onto the retina, using free space optics. A micro-lens system was developed mapping the light homogeneously on the retina in a spot with a diameter of about 5 mm. This assures that the receiver is illuminated for eye movements of up to +/ - 15°. The material used for the receiver is GaAs, mainly to achieve high conversion efficiencies with the photovoltaic cell array [2]. This has several advantages: 1. The fibre has an gaussian beam profile, while laser diodes have strong astigmatism that has to be corrected with a microoptic in front of the eye. 2. The heat that the laser diode produces is led away very easily. 3. The high frequency modulation of the laser diode for the signal transmission is made far away of the eye avoiding problems with electromagnetic compliance. Latest results are shown in Fig. 2 and Fig.3: Fig. 2 schematically depicts the system design. 3.2 Optical Interconnects and Processors 31 Fig. 2: System design In Fig. 3(a) the I-V characteristics of a single photovoltaic cell and of an array with 5 cells connected in series are plotted. At a wavelength of l = 800 nm this array delivers up to 5 mW electrical power with a conversion efficiency of about 23%, which turns out to be a great improvement as compared with the results published in [3]. It should be noted, however, that this efficiency was obtained without any antireflection coating. Experiments have shown that the efficiency can be increased up to almost 31% by encapsulating the cell array with a biocompatible antireflection coating consisting of SiO2 /Si3N4 multilayers. Moreover, Fig. 3(b) shows measurements of the signal transmission: The signals at the output of the encoder, curve (1), are transmitted optically into the eye at a rate of 1 Mbit/s. The output of the receiver is plotted in curve (2) of Fig. 3(b) together with the recovered clock, curve (3), in Fig. 3(b). Conclusion In this report the progress in the work for an optoelectronic signal- and energy transmission line for use in a visual prosthesis is presented. A concept is developed and a prototype is described. This prototype currently has the capability of delivering 5 mW electrical power together with digitally coded signals at a rate of 1 Mbit/s simultaneously, thus meeting the current sys- Fig. 3: (a) I-V characteristic of photovoltaic cells (PVCs), (b) digitally coded signals before (1) and after (2) transmission, and (3) recovered clock signal. 32 3 RESEARCH tem requirements. However, the optical link described here is capable of transmission rates up to 1 Gbit/s, suitable for optically powered highspeed data links. Acknowledgement The authors would like to thank the Federal Ministry for Education, Science, Research and Technology for financial support and all members of the EPI-RET team for fruitful discussions. References [1] R. Eckmiller Retina implants with adaptive retina encoders, Proc. of the 1996 RESNA Research Symp., Salt Lake City, pp. 21-24, 1996 [2] M. Groß, T. Alder, R. Buß, R. Heinzelmann, M. Meininger, and D. Jäger, Micro Photovoltaic Cell Array for Energy Transmission into the Human Introduction The Retina Implant project was founded in 1995 as part of a young and interdisciplinary area of research: Neurotechnology. The goal of this ten year project is the development of both an artificial eye implant (retina stimulator), stimulating the ganglion cells of the human retina from patients, who lost their eyesight due retinitis pigmentosa or macula degeneration and an encoder transforming the signals coming from a video system like human retina does. One of the coming tasks is to develop a system transporting the electrical power for this implant and the signals from the output of the encoder into the eye. This system analysis shows the technical preferences for this transport by IR-rays. Eye, Proc. of the 14th European Photovoltaic Solar Energy Conference, Barcelona, Spain, vol. 1, pp. 1165-67, 1997 [3] J. Rizzo, J. Wyatt, Silicon retinal implant to aid patients suffering from certain forms of blindness, Proc. of the 1996 RESNA Research Symp., Salt Lake City, pp. 1-3, 1996 3.2.2 Analysis of the optical energy and signal transfer module for an artificial vision prosthesis T. BAUMEISTER, M. G ROSS, AND R. BUß W ithin the scope of the Retina Implant Project supported by the German government the possibility of a wireless transfer of signal and energy into the eye of human patients was analyzed. Criteria catalogue The base of each scientific analysis is a criteria catalogue being a decisive help for the evaluation of possible alternatives. The following criteria were found: 1. The fundamental criteria are the dimensions of the implant. Due to surgical reasons the maximum length is limited to 1.5 mm. 2. The Efficiency of the power transmission is a criterion of great importance for any implanted system, because of the absence of possibility to cool any part of system inside the eye. 3. The reliability is fundamental too, because it is nearly impossible to repair any failure and the exchange of the whole system is more dangerous for the patient as the first time implantation. 4. The biocompatibility is one more very important criterion for the long time function of any 3.2 Optical Interconnects and Processors 5. 6. 7. 8. implant. This may be given by a biocompatible coating of any material, but there is the risk of damage during the implantation and fixation of the innerocular part. Further, we should be aware of the influence of overgrowing the receiving part of the signal and energy transport system. The receiver can´t be placed at all places on the retina, because the fixation of it may damage axons of stimulated ganglion cells. The technical availability and the need of development of parts of the system are important due to the cost and the time to market of the system. The possibility of extension, especially the number of stimulating electrodes, is an important criterion for the future. The acceptance of the whole system by the patient and his milieu is quiet important too. System analysis First the need of bandwidth and power for the stimulation of a given amount of stimulating electrodes were analyzed. The first version of the retina stimulator will consist of an array of twenty stimulating electrodes. The bandwidth needed for the stimulation with twenty electrodes is nearly 100kbit/s, for 400 electrodes we found a bandwidth of approximately 25Mbit/s. This calculation of the bandwidth includes the scheme shown in Fig. 1, the number of bits needed to address the electrodes, the number of bits needed for encoding the stimulating pulseform, the rate of neuro impulses, and fac- 33 tor needed for encoding the signal by wireless transmisson. The power needed using a single amplitude modulated IR-laser for simultaneous transport of energy and signals is 190mW of optical power in a worst case analysis with twenty electrodes stimulating in one time frame. The worst case is defined here by a constant electrical stimulation power of 750µW at the electrodes. Next, the possibilities of signal encoding were analyzed. The whole signal of amplitude modulated laser beam includes an AC signal, the encoded signal for stimulation and addressing the electrodes, and a DC offset for the energy transfer. The CMI-code (see Fig.2) also known as 2AMI-1-code or modified FSK-code is the best compromise between the technical expense of signal and clock recovering and the need of a DC free signal in this application. This coding is mentioned with a factor of two in analysis of bandwidth [1]. Considering this we found the best way to transfer the energy and signals is to Fig. 1: Stimulation scheme 34 3 RESEARCH Fig. 2: Calculation of the power needed use a single amplitude modulated IR-laser diode. Fig. 3: Example of a DC-free encoded signal added with an offset for transmision of power Analysis of the optics To avoid a complicated, heavy and cost intensive eyetracking system focussing the laserbeam exactly on the position of the energy and signal receiving photodiode, we decided to illuminate an area on the retina great enough to compensate eyemovements about 10° in each direction. The position of this area is vertical ahead the macula. At this position the beam is least influenced by the eye lid and the greatest eye movement possible. The best way to lower the reflection of the beam at the iris and thereby to expand the possible angle of movement is to focus the beam in the hole of the iris. For a nearly uniform illumination of area, needed for an uniform supply of power of the stimulation electronic, we have developed a ring-shaped focus of the beam. Thus, we solved the problem how to 3.2 Optical Interconnects and Processors 35 days industrial optics including one standard micro lens. Fig. 4: Example of CMI encoding get a uniform illumination on the screen of a good imaging optical system i.e. human eye. To solve this problem we simulated the whole optical path with the ray tracing program ZEMAX SE for human eye and additional for the rabbit eye, see Fig. 5. For the simulation of the human eye we used a slightly modified model of the eye from Helmholtz. We completed this model with the axes of movement [2]. For the eye of the rabbit we used a similar model [3]. The optics consists of a two lens beam expander and one complex collecting lens to get the above mentioned ring shaped focus (see Fig.5). The optics is not modular, i.e. it is not possible to use parts of the optics designed for the human eye in the optical system for the rabbit eye. The optical source can be the end of an optical fiber or a laser diode with a micro lens that corrects the astigmatism of the laser. The lenses used in our model are all in the range of to- Rabbit eye Conclusion Here is shown, that, principally, it is possible to transfer enough power and a signal with sufficient bandwidth into the eye by using an IR light beam. Further it is shown, that it is preferable to use a single amplitude modulated laser diode according to the above mentioned catalogue of criteria. This paper shows that a transport of power and signal from the signal processing unit outside the eye to the implanted microelectronics by optical means can be realized without the need of an eye tracking system. 3.2.3 Development of an optical signal and energy transmission system R. HEDTKE AND M. G ROSS A n optical transmission system to power and provide a retinal implant with energy and digitally coded information is developed. Human eye Fig. 5: Optical analysis of the rabbit eye and the human eye Introduction The EPI-RET: Retina Implant project is part of the Neurotechnology Program of the Federal Ministry for Education, Science, Research and Technology (BMBF). The implant system is evolved in co-operation with several interdisciplinary project partners as a vision aid for people who 3 RESEARCH 23 POPT OPTICAL OUTPUT-POWER (POPT) 36 TIME TIME I I th DRIVING CURRENT (I ) Fig. 1: Principle of the modulation of the laser diode are suffering from retinal degenerative defects like retinitis pigmentosa and macula degeneration. With the help of this optical transmission system the retinal implant is provided with the needed information and energy. Transmitter The driving current, as shown in Fig.2, is regulated by a current control unit to secure a constant Receiver The modulated laser light hits the photodiode and the photovoltaic cell array (see Fig.3). The photodiode generates the detector signal which is transformed to TTL-level by a signal processing unit. A clock recovery unit is used to generate the clock signal. The other part of the laser CURRENT ADJUSTMENT ACTUAL CURRENT VALUE CURRENT CONTROL RESISTOR AND SIGNAL CONVERSION PHOTO DIODE AND SIGNAL CONVERSION DR IVING C URRENT DRIVING UNIT SIGNAL MODULATOR + MOD ULATION C URR ENT Principle For the transmission to the retinal implant the information is modulated onto the laser light, which transmits the energy. The kind of modulation that is used is shown in Fig. 1. A continuos energy transmission into the implant is guaranteed by modulating the driving current around a mean operating point (OP). The driving current and the optical outputpower for the laser diode is sketched for an stimulation with a rectangular-signal. optical output power of the laser diode. This control unit compares the preset current value with the actual and provides the control signal for the driving unit. To generate the actual current value a photodiode integrated into the laser diode [1] (alternatively: an external shunt-resistor) is used. With the help of a signal conversion the measured signal is processed for the following current control unit. The modulation of the driving current occurs in the previously described way. ALTERNATIVE GENERATION OF ACTUAL CURRENT VALUE LASER DIODE Fig. 2: Block diagram of the transmitter O PTICAL OUTPUT-POW ER 3.2 Optical Interconnects and Processors 37 CLOCK RECOVERVY MODULATED LASER LIGHT PHOTO DIODE DETECTOR SIGNAL CLOCK SIGNAL PROCESSING SIGNAL PHOTOVOLTAIC CELL ARRAY ENERGY Fig. 3: Block diagram of the receiver light is converted into electrical energy by the photovoltaic cell array [2]. Measured results Fig. 4 shows the driving current of the laserdiode for a stimulation with a 1 MHz rectangular wave. The used mean operating point is 400 mA [3] or rather 190 mW optical power. To code the signal the Manchester code [4] is used. The main advantage of this coding are the well-balanced relation of logical high an low states (important for a constant energy transmission) and the easy recovery of the clock signal. The transmitted and the received Manchester coded signal and the recovered clock are shown in Fig. 5. 800 I/mA 600 400 TRANSMITTED SIGNAL RECEIVED SIGNAL CLOCK 0 5 10 15 20 t/µs Fig. 5: In- and output signals of the transmission system Conclusion At the Fachgebiet Optoelektronik (GerhardMercator-Universität Duisburg) an optical signal and energy transmission system was developed to provide a retinal implant with digitally coded signals. To realize this aim a transmission and a receiving unit were designed. References 200 [1] SDL, Laser Diode Operators Manual & Techni0 0 1 2 3 4 t/µs 5 cal Notes, SDL Inc., San Jose/USA, 1994 [2] M. Meininger, Entwicklung photovoltaischer Fig. 4: Modulated driving current for the Zellen zur Energieversorgung einer künstlichen laserdiode Sehprothese (Retina Implantat) Diploma thesis, 38 3 RESEARCH Fachgebiet Optoelektronik, Universität Duisburg, 1997 [3] T. Baumeister, Systemanalyse des optischen Energie- und Signalübertragungsmoduls für eine (dongle) for notebook-PCs without an builtin infrared interface and a special data transmission software protocol is described in the second part of the report. künstliche Sehprothese (Retina Implantat) Diploma thesis, Fachgebiet Optoelektronik, Universität Duisburg, 1996 [4] R. Mäusl, Digitale Modulationsverfahren, Hüthing, Heidelberg, 1991 3.2.4 Infrared data link for rotating display-systems U. WEIMANN AND R. B Uß T he magicball is a recently designed rotating LED display system for texts and graphics. In this report two methods of wireless programming of the magicball by using either an infrared remote control or a notebook-PC are presented. In the first instance, a polymethylmethacrylate (PMMA)body is designed and a new software is developed allowing the programming via an IR-remote control unit. An infrared interface Introduction The magicball display-system is used as an eye-catcher in stores or at exhibitions amongst others. Its design and operation principles are shown in Fig. 1. The 16 LEDs are fixed at the end of the rotating arm. The microcontroller creates moving titles on the surface of the magicball by switching the LEDs on and off individually. The microcontroller and the LEDs on the rotating carrier and arm are powered by a generator situated in the socket of the magicball. The data transmission was formerly achieved by a serial PC cable and rubbing contacts. By designing a wireless infrared data link between the programming unit and the display, programming of the magicball is simplified and allows customizing. In addition, the life-span of the product is increased while running costs decrease. IR Data Link: Remote Control => Magicball The data transmission via infrared remote control unit is unidirectional from the programming unit to the display. The infrared signal of the remote control is biphase coded , similar to the standard RC 5 code, and consists of a pre-signal and the main signal [1]. A photodetector - with an integrated preamplifier, a demodulator and a filter - is used for detecting the infrared signal of the remote control unit [2]. Since the receiver module is placed on the Fig. 1: Side and top view of the magicball-display-system 3.2 Optical Interconnects and Processors remote IR -bea m control 39 15 mm epoxy glue IR-Detektor 10 mm Fig. 2: Exemplary infrared signal beam 15 mm PMMA-body IR Data Link: Notebook <=> Magicball For the data transmission to the magicball and vice versa , the notebook-PC needs an infrared interface (built-in or additional) which operates according to the SIR (Serial Infrared) standard of the IrDA (Infrared Data Association) [3]. The SIR standard enables a data transmission range of up to 3 m, at angles of up to 15° and at a data rate of 115.2 kbit/s. For notebook-PCs without a built-in infrared interface a dongle fitting to the serial port of a PC has been designed. The encoder/decoder chip and the transceiver module with an additional IR-LED for wider transmission ranges support the infrared data trans- rotating parts of the magicball, an optical system (e.g. a special mirror) is necessary to ensure an uninterrupted connection between transmitter and receiver. This is why a dynamically balanced body made of polymethylmethacrylate (PMMA) has been developed. Fig. 2 shows the path of an exemplary infrared ray from the reFig. 3: Block diagramm of the developed infrared dongle mote control unit through the PMMA-body onto the detector. The loss of inmission based on the SIR standard. The contensity caused by the reflection and transmisverters allow a power supply of the dongle with sion involving the PMMA-body is less than 10%. the serial RS232-port. The principle block diaTests show that the configuration consisting of gram of the dongle is shown in Fig.3. the remote control unit, PMMA-body and phoIn contrast to the remote control unit configutodetector enables a transmission range of up ration, the notebook - magicball infrared data to 10 m. In addition to the hardware components, link is bidirectional. Because of the bidirectionspecific software has been developed to proality and the lower radiation intensity using the cess the incoming infrared bi-phase coded sigSIR standard, the PMMA-body cannot be used nal. After sampling the signal the microcontrolhere. Therefore, a simple transmitter-receiver ler reassembles the keycode of the remote system has been chosen, omitting the optical control unit and displays the new character on system used before. This configuration is illusthe surface of the magicball. 40 3 RESEARCH Conclusion In this report, two transmitter methods of the reception area wireless programem iss ming of a magSe ion nd ew ang receiver ink le icball via infrared el data transmission are discussed. In Fig. 4: Transmitter - rotating receiver - configuration the first instance, programming is trated in Fig.4 . Because of the symmetry of the done with an IR remote control using a PMMA infrared data link, transmitter and receiver as body with integrated photodetector as an optishown on the illustration are interchangable. The cal medium on top of the rotating arm. In the bi-phased character of the data link remains in second instance, a customized program has place. been developed to enable a bidirectional IR data Due to the rotation of the detector and belink between a notebook-PC and the magicball cause of the angle of detection/emission, we without an additional body. Many tests have obtain a periodically recurring optical contact shown the possibilty of infrared data transmisbetween transmitter and receiver. Because of a sion on such a rotating receiver system. rotation of about 3600 rpm and a detection anIn terms of practicability, the remote control gle of 15° the optical contact time (time window) allows a wider transmission range and a lower is nearly 1 ms, while the non-contact time is 15 level of noise interference. On the other hand ms. The software we have developed takes adthe comfortable editor for the notebook-PC is a vantage of the resulting time window, thus enbig advantage for the second solution, especially abling the data exchange between the notebook if large ammounts of text has to be transmitted. and magicball. The software program first partitions the information into blocks of data with Acknowledgement equal size depending on the time window. After We would like to thank the LUMINO Licht Elebeing received, the single data blocks are reasktronik GmbH for the possibilty of working on sembled. The data security of the transmission the display system magicball and the support is safeguarded by parity bits on the one hand during this work. and by feedback of magicball on the other hand ( acknowledge / non-acknowledge). For examReferences ple, a bad data block effects a non-acknowledge [1] Adaptive Micro Systems Infrared Communication Theory of Operations Abstract, 13.06.88 response and the block will be transmitted again. [2] TEMIC Semiconductors TFMx IR Detector The software for this type of data transmission Photomodules Design Guide, June 1996 and the infrared dongle have been successfully [3] St. Williams, I. Millar The IrDA Platform HP tested for functionality. receiver path n t io tec e de angl Labaratories, Bristol, 1996 3.2 Optical Interconnects and Processors 3.2.5 Nonlinear hybrid GaAs/AlGaAs multilayer-heterostructures for highspeed information processing C. KAMPERMANN, A. KREUDER, AND S. REDLICH I n this report we present theoretical and experimental results on the nonlinear optical, electrical, electrooptical and optoelectronic properties of hybrid GaAs/AlGaAs multilayer heterostructures. These structures exhibit fast nonlinear properties and high sensitivity which can be used for highspeed information processing in microwavephotonics. Introduction In recent years optical nonlinearity and bistability in multilayer heterostructures (MLHS) have received increasing attention because of their potential use in all-optical high-speed information processing systems. Applications are foreseen in the areas of photodetectors and modulators with internal amplification as well as fast optical switching and memory devices. In 1991 He et.al. [3] achieved all-optical bistability in a 30 period GaAs/AlAs structure at an optical intensity of 10kW/cm² [1]. Switching intensities in Fig. 1: Sketch of the device 41 the range of kW/cm², however, are orders of magnitude too high for optical information processing. In the presence of an applied electric field perpendicular to the layers of a MLHS, the Franz-Keldysh effect together with the accumulation of photocarriers in the GaAs layers and the voltage dependence of the current through the device are used to decrease the switching intensities by five orders of magnitude [2]. These kind of hybrid MLHS exhibit the lowest switching intensities in comparison to other device concepts. Appropriate designed MLHS also exhibit s-shaped negativ differential conductivity (SNDC), based on bistability between tunneling and thermionic emission across the heterobarriers. Calculations and experiments have shown that selfsustained voltage oscillations up to 100 GHz occur, if the MLHS is driven in an external resonator. In this report we present theoretical and experimental results concerning these novel kind of devices. Device Structure Fig. 1 shows the cross section of the device containing a periodical GaAs/ Al0.45Ga 0.55As MLHS. The MLHS consists of 20 bilayers with nominal thicknesses of 58 nm (GaAs) and 69 nm (AlGaAs). The layers are grown by usual 42 3 RESEARCH MBE on s.i. GaAs substrates. A 1000nm n+GaAs contact layer is introduced for sufficient high values of the bulk conductance. A number of different process techniques such as wet etching and evaporation are used to form the device. Conventional photolithography is applied to pattern structures on the wafer. The transmission line is made of a TiPtAu multilayer metalization, which is applied to the wafer by evaporation. To prevent a short circuit between the center conductor of the transmission line and the contact layer at the bottom of the MLHS the edges of the mesas are coated with polyimide. A voltage can be applied to the coplanar transmission line to get a high electric field concentrated in the MLHS. With coplanar transmission lines used as electrical contacts, operation up to millimeterwave frequencies is possible. The optical input and output ports are defined by a via hole in the center conductor of the transmission line. Theory To simulate the device it is necessary to analyse the optical wave propagation as well as the transport and the accumulation of charge carriers in the MLHS. Additionally, the electrooptical and the optoelectronic interactions between the optical and the electrical subsystems, namely the Franz-Keldysh effect and the generation of photocarriers in the GaAs layers have to be considered. Optical properties For an optical wave propagating in a periodically layered medium like a MLHS it has been shown that, there exists an optical resonance effect when the optical wavelength is close to the optical stopband. This means that the intensity distribution in the MLHS depends on the optical wavelength and strongly increases when approaching the resonance. This effect is essential for the behaviour of the device and we had to take this into consideration for our simulations. In the linear case one can use the transfer-matrix method (TMM) to calculate the intensity distribution in the layered medium. In the nonlinear case of the MLHS the standard TMM cannot be applied because of the intensity dependent refractive index of the GaAs layers. To overcome this problem we used a generalised Fig. 2: (a) Linear reflectivity spectrum of a InGaAlAs/InAlAs MLHS and average optical intensity in the InGaAlAs layers of the structure.( b) Reflectivity over incident optical intensity of the same MLHS for two different wavelength. 3.2 Optical Interconnects and Processors form of the TMM [3] where the GaAs layers are divided into a number of sublayers. Assuming that the optical intensity in each sublayer is constant one can determine the intensity dependent refractive indices by using the boundary conditions of the wave amplitudes in two adjacent layers. Then, specifying the field at the end of the structure one can apply the standard TMM to calculate the nonlinear characteristics of the MLHS. Figure 2(a) shows the linear reflectivity spectrum of a MLHS consisting of 40 pairs of InGaAlAs/InAlAs bilayers, lattice matched to InP. As can be seen at a glance, by using the InGaAlAs/InP system one can shift the operation wavelength of the device towards 1.5µm, an important wavelength for applications in optical communication systems. Like demonstrated for the AlGaAs/GaAs system by He et.al. all-optical bistability based on the nonlinear refractive coefficient n2 without an applied electric field can also be observed. The nonlinear reflectivity-versus-intensity characteristics of the structure, calculated with the above method, are shown in Fig.2(b). The curves are calculated for two different wavelengths at the long-wavelength side of the stopband. Both are Z-shaped and exhibit a bistable hysteresis loop. As mentioned above we have experimentally shown that by applying an electric field perpendicular to the layers of the MLHS switching intensities below 10mW/cm² can be achieved. Therefore, besides the optical properties, the electronic, optoelectronic and electrooptical properties of the MLHS are from special interest. Electrical properties Our model of the transport and the accumulation of charge carriers in the MLHS is based on an analytical model of a heterostructure hot electron diode (HHED) by Wacker et.al.[4]. The 43 HHED shows S-shaped negative differential resistance (NDR) or differential gain and consists of two undoped adjacent heterolayers (GaAs/ AlGaAs) with ohmic contacts. In this structure two conduction mechanisms exist: At low fields the current is limited by tunneling through the AlGaAs layer (low conductance state on theFig.3(a)). At higher fields the charge carriers are heated up to sufficiently high energies so that thermionic emission over the barrier becomes dominant (high conductance state on the Fig. 3(b)). The extremely fast transition between these conduction modes leads to NDR or differential gain. We extended the model of Wacker et.al. to calculate the electronic properties of MLHS. The physical processes of the charge transport in the MLHS are sketched in Fig. 4(a). As an additional effect, the cooling of the charge carriers, which means the capture by the GaAs wells is included. The numerically obtained current-density-voltage characteristics (see Fig. 4(b)) are in good agreement with the results of Monte-Carlo simulations published by Reklaitis [5]. Fig. 4(b) further elucidates a pronounced Sshaped NDR for (GaAs 100nm / AlGaAs 70nm) and for thicknesses used in our device structure merely a preliminary stage of NDR. W a) W GaAs AlGaAs b) GaAs AlGaAs Fig. 3: Schematic conduction band structure of a GaAs/AlGaAs heterostructure with a perpendicular electric field. The two possible conduction states are shown. 44 3 RESEARCH current density in kA/c of the refractive indi1600 ces in the GaAs layEmission ers. It has been shown 1200 TOTAL CURRENT that the same change 800 Heating of the refractive index 400 can be achieved at 0 W much lower optical in2,0 3,0 4,0 5,0 6,0 a) b) voltage in V tensities as for the intrinsic optical nonlinFig. 4: (a) Schematic view of the conduction band structure of a MLHS with earity. Including these a perpendicular electric field and the physical processes of the charge mechanisms the feedtransport. (b) Current density-vs.-voltage characteristic of a heterostructure back, which deterfor two different layer thicknesses. mines the optical bistability of the hybrid MLHS can be described Interaction as follows: An incident optical intensity leads to Fig.5 shows schematically the system of a an optical intensity in the MLHS, where a part of hybrid MLHS [5]. The optical and the electrical the light will be absorped. The generated carrisubsystems are coupled by two interaction ers give rise to a photocurrent. This in turn leads mechanisms, the generation of photocarriers to a change of the voltage drop across the GaAs (optoelectronic) and the Franz-Keldysh effect layers and a change of the refractive indices of (electrooptical). The photocarriers are generatthis material. This variation finally changes the ed by optical absorption in the GaAs layers. Due reflectivity of the hybrid MLHS and in turn the to the nonlinear electrical properties of the MLHS absorped optical intensity. Thus, a feedback loop a small photocurrent leads to a strong variation exists. The interaction mechanisms and the of the internal voltage distribution and, by the feedback loop are also implemented in our modFranz-Keldysh effect (FKE), to a large change el of the hybrid MLHS. Thus, the experimentelly observed device characteristics including the opi0 tical, electrooptical, optoelectronic and optically induced electrical bistability of a hybrid MLHS could be verified (Fig. 6). Cooling GaAs 100nm / 0.45 Al Ga0.55As 70nm GaAs 58nm / Al 0.45Ga0.55As 69nm C Z V0 Fig. 5: Cross section of the MLHS with I the current flow and P the optical wave propagating through the device. Experimental results Comprehensive measurements of the optoelectronic properties of hybrid MLHS have shown a high optical sensitivity of these devices. At a reverse bias of V=30V we have measured photocurrents of around I=1mA and dark currents of merely 20nA (see Fig. 7). A bias voltage also changes the reflectivity, as shown on theFig. 8(a) , where the electroop- 3.2 Optical Interconnects and Processors 45 current in A tical modulation near the bandgap wavelength is due to the Franz-Keldysh effect. A modulation contrast of about 6dB could be reached at a voltage change Fig. 6: (a) Measured current-voltage characteristic of a hybrid MLHS under of 20V. Time and illumination. (b) Calculated I-V curve of the same device and operation pafrequency dorameters. main measurements were carried out to investigate the dynamic cut-off frequencies of up to 420 MHz, measured properties of the MLHS. We have determined from MLHS devices based on another contact geometry. The frequency response of these devices is RC limited, therefore higher cut-off fre-2 10 quencies should be reached by using the transdark -3 10 Popt = 1mW mission line design. The coupling of the -4 10 -5 electrooptical modulation and the optoelectron10 -6 ic properties of the device leads to optical bista10 -7 10 bility, which has been measured at optical in-8 10 tensities below 10mW/cm² (Fig. 9). -30 -20 -10 0 10 20 30 voltage in V Conclusion In this report new theoretical and experimental results concerning hybrid MLHS are present- Fig. 7: Measured current-voltage character- 6 -10 V -20 V 4 2 0 -2 860 880 wavelength in nm 900 rel. modulation in dB modulation contrast in istics in the dark and illuminated case. 4 A = (600µm)² 0 -4 -8 10 5 10 6 10 7 10 8 frequency in Hz Fig. 8: (a) Modulation contrast over wavelength at different reverse biases. (b) Relative modulation characteristic as a function of frequency. 46 3 RESEARCH 3.2 Optical Interconnects and Processors 47 48 3 RESEARCH Fig. 2: Photograph of the array with one LED illuminated . above 1 MHz are to be expected. In comparison, the results differ only slightly from those obtained with commercially available superluminescence diodes. 64 Channel Silicon Driver Circuit The TTL-compatible silicon chip was designed to power the above mentioned array of 8 x 8 LEDs, with each LED driven independently by an output current adjustable in the range of 0 - 10 mA. Based on the requirement of a maximum input current for the IC of 1 mA, a current amplification with a factor of 10 must be achieved. The realized circuit consists of the following main components: (i) two 3 to 8 demultiplexers for binary coded x- and y-selection of each LED driver, (ii) a current mirror to tune the gain of 10 and to shorten the input I in if Izero is set to zero, and (iii) 64 independently selectable LED drivers containing xy-selection units and a capacitor providing a constant output current Iout during regeneration cycles. In Fig. 3 the layout of the chip, consisting of 64 identical amplifier cells, bondpads, and demultiplexing circuits, is sketched. Each cell requires an area of 200 * 200 µm², leading to a total square surface of 1.6 * 1.6 mm² and consequently to a pixel density of 127 DPI. Together with control circuitry and pads for external wire bonding, a total chip dimension of only 2.3 * 1.6 mm² is achieved. The electrical characterization of the silicon circuit by measuring the pulse response with a sampling oscilloscope was leading to rise and fall times less than 250 ns. This results in a cutoff frequency of fc ³ 1.45 MHz. Due to the fact that the I-V characteristic referred to the input of the circuit shows strong non-linear behaviour, an interface circuit (voltage driven current source) was applied, leading to a decrease of the cut-off frequency. Together with a D/A converter computer board the system shown in Fig. 4 was built, providing a good linearity between input voltage and output current. Fig. 3: Layout of silicon integrated circuit with detail of LED drivers. Hybrid integration In Fig. 5 the final device consisting of the silicon driver IC bonded to the LED array is depicted. A composition of almost 3.2 Optical Interconnects and Processors 49 photonic IC, are established using wire bonding technique. Fig. 4: Computer controlled silicon driver circuit. eutectic solder (60 wt % Sn, 40 wt % Pb) is evaporated onto the metal contacts of the LED array. After the reflow process, at 200° C for 10 seconds, both the array and silicon driver IC are adjusted and bonded together. Since the PbSn layer thickness is much smaller than electroplated PbSn due to the evaporation process, this bonding technique is a mixture of thermocompression and soldering. Following the flipchip process the ground contact for the LED array, together with connections for packaging of the Applications With this system presented here several applications can be realized. One example is a special kind of vision aid for blind persons with a blurred cornea. In Fig. 6 one possible realization of this vision aid is sketched. Under various circumstances (accidents where the cornea is damaged, e.g in explosions or by erosion due to acid) a number of people loose their sight although their ret- Fig. 6: Vision aid for people with blurred cornea. ina is fully intact. A photodetector array converts images into digital information wirelessly transmitted to a miniature display like our proposed model, implanted into the lens. This display projects a very simple image onto the retina, offering a primitive vision. Fig. 5: Cross-sectional view of the silicon driver circuit bonded to the LED array. Conclusion Experimental investigations of both the silicon circuit and the LED array show cut-off frequen- 50 3 RESEARCH cies beyond 1 MHz, leading to the conclusion that this hybrid integrated circuit is a promising subsystem not only for parallel optical information processing systems but also for a novel application of photonic integrated circuits in the field of neurotechnology. Acknowledgements This work was financially supported by the Federal Ministry for Education, Science, Research, and Technology (BMBF) in the frame of the EPI-RET: Retina Implant project under contract number 01 IN 501 G. The authors would like to thank G. Sixt (TEMIC Telefunken, Heilbronn) for providing the GaAsP/GaP wafer and R. Klinke (Fraunhofer Institut-IMS, Duisburg) for helping designing the silicon chip. Thanks goes also to K. Heimann (Uni-Augenklinik, Köln) for giving an insight into several ophtalmological problems. References [1] H.F. Bare et al., IEEE Photon. Technol. Lett., 5, 2, pp.172, 93 [2] G.W. Turner et al., IEEE Photon. Technol. Lett., 3, 8, pp.761, 91 [3] A.J. Moseley et al., Electron. Lett., 27, 17, pp.1566, 91 [4] K. Werner, IEEE Spectrum, pp.30-39, Jul. 94 [5] W.R. Imler, et al., IEEE Trans. Compon., Hybr. Manufact. Technol., 15, 6, pp.977, 92 [6] Y. Nitta et al., IEEE Photon. Technol. Lett., 4, 3, pp.247, 92 [7] H. Yonezu et al., Electron. Lett., 25, 10, pp.670, 89 [8] M.A. Brooke et al., Optics & Photonics News, pp.26, Jun. 93 [9] M. Wale et al., IEEE Circuits & Devices, pp.25, Nov. 92 3.3 Millimeterwave Electronics 51 3.3 Millimeterwave Electronics L/2 R/2 Ik 3.3.1 Picosecond pulse generation on monolithic nonlinear transmission lines using high-speed InPHFET diodes R. HÜLSEWEDE C(V) Vk G Fig. 2: Equivalent circuit for one element of the NLTL E lectrical pulses with transients less than 5 ps are generated and compressed on monolithic InP-HFET diode nonlinear transmission lines. The transients are measured by time domain electro-optic sampling technique and the waveforms show good agreement with numerical results. Additionally, frequency domain measurements and numerical simulations reveal that the nonlinearities work with frequencies higher than 400GHz for 20µm x 20µm InP-HFET diodes. Instead of costly ion implantation technology a chemical recess is used to isolate the active structures. Introduction Monolithic nonlinear transmission lines (NLTL) are circuits with an alternating arrangement of coplanar waveguides and Schottky diodes as shown in Fig. 1. The capacitance-voltage characteristic of the Schottky diodes in combination with the low pass filter characteristic of the periodic structure leads to the generation of shock waves and the formation of pulses with (sub-) picosecond transients [1,2]. Therefore, these circuits are important for novel measurement and characterization methods for new high-speed devices. For numerical simulations the equivalent circuit as shown in Fig.2 is used leading to a difference equation for current and voltage at each element of the NLTL. Applying a transition to a differential equation one obtains the following wave equation which considers separately the influences of the nonlinearity of the diodes, the periodic structure and the losses of the transmission line (for details see [3,4]): ∂V C(V ) ∂V + = 1 − ∂x C0 ∂ t + D D D D Fig. 1: Monolithic NLTL with periodic array of Schottky diodes R/2 L/2 C0 ∂ 2V L ⋅ C0 ∂ 3V R V − + G ∂t2 L 12 ∂ t 3 (1) Here the nonlinearity of the diodes is considered by the normalized capacitance-voltage dependence C(V)/C 0 , where C0 is the capacitance at the operating point of InP-HFET diode In order to improve the nonlinearity of the Schottky diodes in NLTLs d-doped diodes based 52 3 RESEARCH C(V)/Co 10 Schottky contact Ohm contact InGaAs-channel InAlAs InGaAlAs InP δ -Si Fig. 3: Schematic profile of an InP-HFET diode (for details see [5]) on InP-HFET layer structures are used ( see Fig.3 and [5]). High electron concentration in the d-doped layer (4.9 1012cm-2), maximum mobility (10900 Vs/cm-2), and the 2 dimensional electron gas (2-DEG) in the InGaAs channel are special features of this layer structure at T=300K. The strong nonlinearity of the HFET-layer structure is shown in Fig. 4 where the normalized capacitance-voltage characteristic of an InP-HFET diode is sketched. Over the 0.5V bias range around the working-point a 2200% change of the capacitance is achieved. This is a 20x greater nonlinearity than d-doped GaAs Schottky diodes used in NLTLs described in [6]. One reason for this strong nonlinear behaviour is the depletion of the 2-DEG underneath the negative biased Schottky contact. Using InP-HFET diodes in NLTLs the nonlinear interaction of the propagating waves is increased and thus the line-losses are decreased due to shortening of the line length. Another advantage is the application of a C4H6O4,H2O2,NH3 recess [7] for electrical isolation of the InP-HFET diodes in NLTLs. Thus, no costly ion implantation process is needed and no preparation of an accelerator is required. 1 0.1 0.01 -3 -2 -1 0 Bias voltage(V) Fig. 4: Normalized capacitance-voltage characteristic of InP-HFET diodes InP-HFET NLTL In a first step a 10 diode periodic InP-HFET NLTL was fabricated in order to verify experimental and numerical results. For that purpose frequency domain measurements along the center conductor are shown in Fig. 5 (see also [8] and P. Bussek et al., Time- and frequency domain electro-optic field mapping of nonlinear transmission lines, in this annual report).The electro-optic signal of the excited input wave (15GHz, 25dBm) decreases from input to output of the NLTL, whereas the generated harmonic signals at 30GHz, 45GHz and 60GHz increase. Using the nonlinearity of InP-HFET diodes (Fig.4) and a FFT the nonlinear wave propagation on this NLTL is simulated. The result is shown by the grey lines in Fig. 5. With respect to the -128dB noise level and the +/-5dB accuracy of the sampling signal both results are in good agreement. This agreement and the numerical value for C0/G = 3,5 10-13s indicates that the nonlinearity works with frequencies higher than 400GHz for the 20µm ´ 20µm InP-HFET diodes. Thereupon different NLTLs are simulated based on these perceptions in order to generate one 3.3 Millimeterwave Electronics 53 Signal voltage (V) Signal (dBm) (a)-(d). The frequency of the input signal is 6.5GHz with an amplitude of 3.5V (measured at 50W). Clear-80 15GHz ly the steeping of a shock wave Input signal: 15GHz,27dBm -90 (k=11) and the generation of a sinexperiment simulation -100 30GHz gle pulse (k=31) with FWHM of -110 10ps and a fall time of about 5ps 45GHz is observed. Thus, picosecond -120 60GHz pulse generation on NLTL using -130 high speed InP-HFET diodes is -140 shown for the first time. Addition2500 2000 0 500 1500 1000 x (µm) ally, the numerical results (Fig.6) at the corresponding points of Fig. 5: Generation of harmonic signals on periodic InP-HFET measurements are plotted in Fig. 7 NLTL. The structure of NLTL is sketched at top of this figure. (grey lines). The agreement of (Data of simulation: L = 120 pH , C0 = 1.6 pF, waveforms is satisfying demon− 13 R / L = 2 ⋅1010 s −1 , C 0 / G = 3.5 ⋅ 10 s ) strating that the fundamental mechanisms of nonlinear wave single pulse per period of the sinusoidal input sigpropagation on NLTLs have been considered in nal. The simulation in Fig.6a demonstrates the equation (1). generation and compression of single pulses out of the exiting input signal (6.5GHz, 2.5V) on a Conclusion graded NLTL with increasing values of L and C0 In this work the generation and compression in direction of the propagating microwave. The of picosecond pulses on InP-HFET NLTL is demtransient with minimum 10-90% fall time of 4ps onstrated. Advances have been achieved by and an amplitude of 1.8V is shown in Fig. 6b. applying high speed InP-HFET diodes exhibitAfter fabrication of this graded InPHFET NLTL using self aligned opSignal voltage 1 tical contact lithography process(b) 0 es [7] the electro-optic sampling set-up was modified making time 1V -1 x domain measurements (see P. -2 Bussek et al., Time- and frequen0 80 120 40 cy domain electro-optic field map10ps Time (ps) t (b) (a) ping of nonlinear transmission lines, in this annual report). In Fig. 6: Simulation of pulse generation on a graded InP-HFET Fig. 7 a top view of the graded NLTL; (a) development of a sinusoidal signal along the NLTL, InP-HFET NLTL is figured includ(b) transient with minimum fall time (Data of simulation: ing the four points of measurement L = 960 pH , C = 1.76 pF, R / L = 2 ⋅1010 s −1 , 0 10 − 1 C 0 / G = 3.5 ⋅ 10 − 13 s , grading: 0.89αx , α = 8 .4 ⋅ 10 s ) 54 3 RESEARCH (a) (b) (c) signal voltage (a.u.) input (a) -80 -80 k = 11 -40 0 time (ps) 40 (c) -80 k = 15 -40 0 time (ps) signal voltage (a.u.) (b) (d) signal voltage (a.u.) signal voltage (a.u.) output 40 k=7 0 -40 time (ps) (d) -80 40 k = 31 -40 0 time (ps) experiment 40 simulation Fig. 7: Pulse compression on graded InP-HFET NLTL (top view of the processed NLTL in the upper left side of this figure, (a)-(d): points of electro-optic measurements) ing strong nonlinearities. The numerical simulation has been improved by considering separately the influence of nonlinearity, periodic structure and losses to nonlinear wave propagation on NLTLs. optoelectronics, Proc. IEEE, Vol. 82, No. 7, 1994, pp. 1037-1059 [2] D. Jäger, Characteristics of travelling waves along nonlinear transmission lines for monolithic integrated circuits: A review, Int. J. Electron., Vol. 58, 1985, pp. 649-669 Acknowledgement The author would like to thank Dipl.-Phys. U. Auer (Fachgebiet Halbleitertechnik / -technologie) for processing of the transmission lines and Dr. D. v.d.Weide (at that time: Max-Planck-Institut für Festkörperforschung, Stuttgart) for lending a suitable mask for optical contact lithography processes. [3] D. Jäger, Pulse generation and compression on nonlinear transmission lines, workshop on Picosecond and Femtosecond Electromagnetic Pulses: Analysis and Applications, MTT-S Symp. Dig., 1993, pp. 37-57 [4] R. Hülsewede et al, CAD of pulse compression on nonlinear transmission lines, Proc. MIOP 95, Sindelfingen, 1995, pp. 511-515 [5] U. Auer et al, InP based HFETs with high qual- References ity short period InAlAs/InGaAs Superlattice [1] M.J.W. Rodwell et al, Active and nonlinear wave Channel Layers, J. o. Crystal growth, vol. 146, propagation devices in ultrafast electronics and 1995 3.3 Millimeterwave Electronics [6] D.W. van der Weide, Delta-doped Schottky diode nonlinear trans-mission lines for 480-fs, 3.5V transients, Appl. Phys. Lett. Vol. 65, No. 7, 1994, pp.881-883 [7] C. Heedt et al, On the Optimisation and Reliability of Ohmic- and Schottky Contacts to InAlAs/ InGaAs HFET, Proc. 4th InP & Related Materials Conference, Newport, USA, 1992 [8] Report on the Special Collaborative Programm SFB 254, 1993-1995, Gerhard-MercatorUniversität - GH - Duisburg, 1995 3.3.2 Millimeter wave power generation on nonlinear transmission lines R. HÜLSEWEDE , V. K. MEZENTSEV, AND I. V. R YJENKOVA I n this paper nonlinear transmission lines are described which are used to generate millimeterwave signals with high efficiencies. In particular, arrays of monolithic varactor diodes loading a coplanar waveguide are studied which can be applied for travelling wave harmonic generation where special phase matching and filter structures give rise to high conversion efficiencies. A second transmission line consisting of any array of resonant tunneling diodes is used as a distributed active device which can generate millimeterwave power at frequencies as determined by a resonance condition of the resonator structure under study. In this paper theoretical and numerical results are presented based upon experimental data. 55 Introduction: The generation of millimeter waves by harmonic frequency generation and active wave propagation along nonlinear transmission lines (NLTLs) has recently become a subject of major research activities [1-4]. However, the power efficiencies achieved so far are small because millimeterwave power is converted into undesired frequency components when the dispersion and filter characteristics of the NLTL are not designed in a suitable way. In this paper, firstly we describe the bi-modal NLTL which uses concepts of nonlinear optics aiming towards achieving phase matching condition between the frequency components under study [5,6]. In particular, we study a bi-modal NLTL where, as an example, the phase velocity of the second or third harmonic equals that of the fundamental wave and where other components are suppressed by a suitable filter structure leading to a distinct cut-off frequency [7-9]. Secondly, we discuss the characteristics of a travelling-wave tunneling-diode transmission line resonator capable of generating high power millimeter wave signals [7-9]. In Fig. 1, the basic structure of an NLTL is sketched consisting of a suitable array of nonlinear devices D in a passive coplanar waveguide [2,3]. As nonlinear elements, we have studied Schottky diodes, quantum barrier var- D D D D Fig. 1: Sketch of a nonlinear transmission line 56 3 RESEARCH actor structures (QBV), as well as resonant tunneling diodes (RTDs). The second characteristic feature of the circuit in Fig. 1 is the dispersion which is mainly determined by the arrangement of the diodes, which can be periodic, bi-periodic, graded etc. The dispersion itself controls the phase velocities of different spectral components and hence the strength of interaction and the superposition in the time domain. Reflections at input and output ports further determine the resonance behavior of the whole structure. Harmonic frequency generation: We have studied nonlinear wave propagation along NLTLs of Fig. 2. In a first example the parameters used are those of an experimental device on InP-HFET substrate as discussed in [10]. Input frequency and power are 100GHz and 11dBm as delivered into a small-signal characteristic impedance of 50W. As a numerical tool we have used a continuum approximation on the basis of a corresponding nonlinear evolution equation as described in [10,11] and compared the results with a CAD model based on a discrete representation of the NLTL , cf.[10]. The results of our numerical calculations are plotted in Fig. 3 showing the spatial distributions of the amplitudes of the fundamental and second harmonic wave. As can be seen, the amplitudes of the two waves at the in- put are comparable, which leads to power efficiencies > 70% for second harmonic generation (SHG), here at 200GHz. In a second numerical experiment we have studied third harmonic generation (THG) on a bi-modal NLTL of Fig. 2(b) assuming special quantum barrier varactor diodes [12] with a symmetric capacitance voltage relationship. Fig. 4 presents the numerical results. Note that in this case of THG phase matching is achieved between frequencies f1 = 72GHz and f3 = 216GHz. As can be seen from Fig. 4, the 72GHz input signal is converted into millimeter wave power (a) InGaAs 250 nm n+ InGaAs 250 nm n InGaAs 25 nm InAlAs 25 nm undoped InGaAs 25 nm undoped InGaAs 250 nm n 500 nm 350 µm n+ InGaAs InP undoped C Co C Co (b) InAlAs 50 nm InGaAs 50 nm InGaAs 300 nm InGaAs 800 nm InP 350 µm nn+ nn+ C Co C Co (c) InGaAs 500 nm InGaAs 40 nm n+ undoped InAlAs 7.2 nm undoped InGaAs 4.3 nm undoped InAlAs 7.2 nm undoped InGaAs 40 nm undoped n+ InGaAs 500 nm InP 350 µm RTD RTD RTD RTD Fig. 2: Monolithic NLTL on InP substrate for millimeter wave generation. (a) Bi-modal NLTL with Schottky varactor diodes for SHG, (b) bi-modal NLTL with QBV for THG, and (c) RTD-NLTL for a distributed oscillator. 3.3 Millimeterwave Electronics 57 1.2 0.6 1 0.5 f1 0.8 0.4 0.6 0.3 f = 2f1 0.4 2 0.2 0.2 0.1 0 0 20 40 60 80 100 120 number of elements, k 140 160 f1 f 3 0 0 20 40 80 100 120 60 number of elements, k 140 Fig. 3: Spatial distribution of amplitudes at Fig. 4: Spatial distribution of amplitudes at frequencies f1 and f2 for the NLTL of Fig.2(a). frequencies f1 and f3=3f 1 for the NLTL of 160 Fig.2(b) at 216GHz with an efficiency of about 25%. Again, the third harmonic is available at the input of the NLTL because the propagation characteristic is that of a backward wave [5,6]. cillator and the frequency f(n, N) of the self-generated oscillations where n = 1, 2, , N defines the mode. Fig. 6 shows the results where the dots represent the results of the simulations. Clearly, a decreasing N leads to an increasing f(n,N) because the wavelength, as given by 2 x N, decreases. In Fig. 6 a comparison with analytical results is additionally carried out where f(n, N) is given by amplitude voltage, V Tunneling diode NLTL: Very recently, another type of NLTL has become known where resonant tunneling diodes are used as nonlinear elements [3]. However, such RTD-NLTLs can also be used for nonlinear active wave propagation effects leading to a travelling wave oscillator, when a transmission 1.6 line with limited length, provided by short circuits at input and out1.2 put ports, for example, is used 0.8 [7,8]. In a numerical experiment 0 100 200 300 400 we have studied the generation frequency, GHz 0.4 of millimeter waves in a RTDNLTL resonator. The results are 0 plotted in Fig.5 revealing self-gen-0.4 erated oscillation at 170 GHz after about 700 ps which is the 0 100 200 300 400 500 600 700 800 switch-on time. time, ps In a further numerical example we have studied the relationship Fig. 5: Generation of a 170 GHz signal on a tunneling diode between the length N of the osNLTL. The spectrum is shown in the inset. 58 3 RESEARCH References [1] 600 propagation in electronics, John Wiley & n=1 numeric theory 500 400 A. Scott, Active and nonlinear wave Sons, New York, 1970 [2] D. Jäger, Characteristics of travelling waves along nonlinear transmission lines for 300 monolithic integrated circuits: A review, Int. J. Electron. 58, 649-669 (1985) (invited pa- 200 per) 100 [3] 0 0 5 10 15 N 20 25 30 M.J.W. Rodwell, S.T. Allen, R.Y.Y. Yu, M.G. Case, U. Bhattacharya, M.Reddy, E. Carman, M. Kamegawa, Y. Konishi, J. Pusl, R. Pullela, Active and nonlinear wave propa- Fig. 7: Oscillation frequency vs. number N of elements gation devices in ultrafast electronics and op- of the RTD-NLTL, theory according to eq. (1) toelectronics, IEEE Proc., vol. 82, no. 7, pp. 1037-1059, 1994 (invited paper) [4] f ( n, N ) = 1 π 1 LC E. Carman, M. Case, M. Kamegawa, R. Yu, K. Giboney, and M.J.W. Rodwell, V-band and W- sin( π2 Nn ) , n = 1,2...,N band broadband, monolithic distributed frequency multipliers, in: 1992 IEEE MTT-S Digest, pp. (1) as calculated from the dispersion relation for a cascaded LC - chain neglecting losses. 819-822, 1992 [5] B. Wedding and D. Jäger, Phase-matched second harmonic generation and parametric mixing on nonlinear transmission lines, Electron. Lett. 17, 76-77 (1981) Conclusion In this paper, specific NLTLs are presented which are capable to generate millimeter waves with high conversion efficiencies. The NLTLs are compact, easily fabricated using standard InP technology, suitable for monolithic integration, and can provide high output powers. We therefore conclude that the travelling wave concept under study can provide a solution to the problem of realizing efficient millimeter wave signal sources. [6] D. Jäger, Nonlinear slow-wave propagation on periodic Schottky coplanar lines, IEEE Microwave and Millimeter-Wave Monolithic Circuits Symposium, St. Louis 1985, Symp. Dig., 15-17 (1985) [7] I. V. Ryjenkova, V. K. Mezentsev, S.L.Musher, S. K. Turitsyn, R. Hülsewede, and D. Jäger, Millimeter Wave Generation on Nonlineat Transmission Lines, Proc.1996 International Workshop on Millimeter Waves, April 11-12, Orvieto, Italy, 1996 [8] V. K. Mezentsev, S.L.Musher, I. V. Ryjenkova, S. K. Turitsyn, R. Hülsewede, D. Jäger, Travelling wave generation of millimeter waves in bi-modal 3.3 Millimeterwave Electronics NLTLs, Proc. 26th European Microwave Conference 1996, Prague [9] I. V. Ryjenkova, V. K. Mezentsev, S.L.Musher, S. K. Turitsyn, R. Hülsewede, and D. Jäger, Millimeter Wave Generation on Nonlineat Transmission Lines, Ann. des Telecomm., Special Issue (submitted). [10] R. Hülsewede, U. Effing, I. Wolff, and D. Jäger, CAD of pulse compression on nonlinear transmission lines , Proc. MIOP 95, Sindelfingen, pp. 511-515 [11] M. Dragoman, R. Kremer, and D. Jäger, Pulse generation and compression on a travelling-wave MMIC Schottky diode array, in: Ultra-Wideband, Short-Pulse Electromagnetics, H.L. Bertoni, L. Carin, and L.B. Felsen, eds., Plenum Press, New York, pp. 67-74, 1993 [12] M.A. Frerking, J.R. East, Novel heterojunction varactors, IEEE Proc., vol. 80, no. 11, pp. 18531860, 1992 3.3.3 Nonlinear RTD circuits for high-speed A/D conversion I. JÄGER I 59 range [1]. The underlying characteristic is a nonlinear N-shaped current voltage relationship even at millimeterwaves. The lack of these very interesting devices, however, is the low power conversion efficiency and the small output power levels [2]. Up to now the only solution to the latter problem which has become known is the use of a series i.e. distributed connection of several RTDs using MMIC technology [3,4]. Such a RTD nonlinear transmission line (NLTL) can further provide the basis of very interesting microwave signal processing devices as has been predicted by Crane already in 1962 [5]. In this paper, we discuss first the fundamental concept of nonlinear active wave propagation effects along monostable RTD-NLTLs utilized to generate a set of spikes from anelectrical input. The idea of such a transmission line, where losses are exactly compensated by distributed amplification, dates back to the so called ´neuristor´ [5] as a line-analog of axons in the nervous system, where the information of an input signal is converted into a number of output spikes, travelling in a stationary way for arbitrarily long distances. In a second step, we describe an electrical circuit, where the monostable RTD-NLTL is the main building block to realize a n-bit A/D con- n this report a novel nonlinear MMIC structure based upon monostable resonant tunneling diodes (RTDs) is studied. For the first time, it is shown that an input signal can be converted into a set of output spikes to be used for GHz A/D conversion. LG Introduction A huge amount of work has recently been dedicated to the study of resonant tunneling diodes (RTDs) which can provide gain and can directly be used as the key components for oscillator circuits approaching the THz frequency RTD LG RTD Fig. 1: Sketch of a nonlinear array of monostable resonant tunneling diodes in a coplanar transmission line 60 3 RESEARCH verter at GHz rates, similar to the lumped RTD A/D converter described in [7-9]. RTD-circuit The array of monostable RTDs is sketched in Fig.1. One can see a coplanar transmission line which is periodically loaded, cf.[3,4], with RTDs shunted by LG circuits -here air bridges- in order to provide monostable behavior, see [6]. The cross section of the MMIC structure in Fig. 1 has been described in [3,4]. The simulation carried out in this paper is based upon a suitable equivalent circuit, as shown in Fig.2. Each section consists of an Tequivalent representation of the transmission line. The nonlinear element in Fig.2 is determined by the RTD current voltage relationship approximated by , where an external bias current and V1,V2 > 0 have been assumed. The basic idea of the circuit in Fig. 2 is roughly the following. An input current source charges the capacitance C up to a threshold value given by J(V) of the RTD. A switching up occurs which, however, will be inverted due to the LG time constant. As a result, a spike is produced and after an RC time constant another switching occurs. Hence the spiking period is determined by the amplitude of the input current. The trans- R C J(V) V Fig. 2: Equivalent circuit of a monostable RTD-NLTL L G mission line itself ensures the generation and propagation of identical pulses - such as solitons - formed after a few diodes. Results Fig.3 shows a numerical result for an input sinusoidal wave of 25 GHz. As can be seen, the monostable RTD-NLTL produces a set of 6 puls- 1,0 0,5 0,0 -0,5 -1,0 10 20 Time40 30 50 60 70 Fig. 3: Generation of five pulses per period of sinusoidal input wave (dashed line) es per period. When the input frequency or the input amplitude are changed, the number, phase, and position of the spikes are altered in a characteristic way. Fig.4 shows an example where the width and amplitude of a rectangular input signal have been changed. As a result, the generated spiking as obvious from the regions with different shadings is a characteristic pattern for the input signal. In particular, we observe that the number of spikes per time depends linearly on the applied current amplitude providing a linear voltage-frequency conversion. Such a NLTL can be used to realise a high speed n-bit A/D converter similar to the lumped version in [7-9]. Correspondingly, we propose 3.3 Millimeterwave Electronics 61 Resonant-Tunneling Diodes, Appl.Phys.Lett., vol.58, no. 20, pp. 2291-2293, 1991 10.00 [2] R.Sun, O.Boric-Lubecke, D.-S.Pan, and T.Itoh, 9.00 Considerations 8.00 and Simulations of Subfrequency Excitation of Series Integrated Resonant Tunneling Diodes Oscillator, 7.00 IEEE Trans. Microwave Theory Techn., vol. MTT 6.00 - 43, no.10, pp. 2478-2485, 1995 5.00 [3] I.V.Ryjenkova, V. K.Mezentsev, S.L.Musher, S.K.Turitsyn, R.Hülsewede, and D.Jäger, Milli- 4.00 meter Wave Generation on Nonlineat Transmis- 3.00 2.00 2.00 sion Lines, Proc.1996 International Workshop 3.00 4.00 5.00 6.00 7.00 8.00 on Millimeter Waves, April 11-12, Orvieto, Italy, 1996 Fig. 4: Contour plot of generated number of spikes (see text) [4] [4] I.V.Ryjenkova, V.K.Mezentsev, S.L.Musher, S.K.Turitsyn, R.Hülsewede, and D.Jäger, Millimeter Wave Generation on Nonlineat Transmission Lines, Ann. Telecomm., Special Issue, to realise a common n-channel (for n bits) coplanar signal devider to provide input amplitudes by powers of 2. Hence each channel delivers a spike train to the output array establishing a Gray code as in Ref.[7]. In the present case, the LC time constant will determine the bandwidth which exceeds 100 GHz in the device under test. vol.52, No 3-4, pp. 134-139, 1997 [5] H.D.Crane, Neuristor - A Novel Device and System Concept, Proc. IRE, vol.50, pp. 20482060, 1962 [6] J.Nagumo, S.Arimoto, and S.Yoshizawa, An Active Pulse Transmission Line Simulating Nerve Axon, Proc. of the IRE, vol.50, p.2061, 1962 [7] T.-H..Kuo, H.C.Lin, R.C.Potter, D.Shupe, A Conclusion In conclusion, a novel monostable RTD-NLTL in MMIC technology is proposed which can generate a characteristic pulse pattern for a given input signal. The application of such a NLTL for an ultrafast A/D conversion is discussed in a second step. The use of RTDs in the presented circuit is expected to yield a bandwidth in excess of 100 GHz. Novel A/D Converter Using Resonant Tunneling Diodes, IEEE Journal of Solid-State Circuits, vol.26, No.2, pp.145-149, 1991 [8] S.-J.Wei, H.C.Lin, R.C.Potter, D.Shupe, Dynamic Hysteresis of the RTD Folding Circuit and ist Limitation on the A/D Converter, IEEE Transaction on Circuits and Systems II: Analog and Digital Signal Processing, vol.39, No.4, pp.247251, 1992 [9] S.-J.Wei, H.C.Lin, R.C.Potter, D.Shupe, A Self- References Latching A/D Converter Using Resonant Tunnel- [1] E.R.Brown, J.R.Söderström, and T.C.McGill, ing Diodes, IEEE Journal of Solid-State Circuits, Oscillations up to 712 GHz in InAs/AlSb vol.28, No.6, pp.697-700, 1993 62 3 RESEARCH 3.4 Optical Sensor Systems 3.4.1 MQW-Electroabsorption-Modulator for Application in a fiberoptic fieldsensor M. SCHMIDT, R. HEINZELMANN , AND A. STÖHR I n this report we present electroabsorption waveguide-modulators for an operation wavelength of 1.55µm. These devices are fabricated for an application in a fiberoptical E-field sensor system [1], [2]. In this system the task of the modulator is to convert electrical signals with frequencies up to 6 GHz into optical signals. Introduction In recent years there has been an increasing interest in electrooptical modulators. The main application of these devices is in fiberoptical communication systems for the external modulation of laserdiodes. Electroptical modulators have been realised in lithium niobate as well as in semiconductors using the Franz-Keldish-effect in bulk materials and the quantum confined starck effect in multiple quantum well structures. As MQW structures exhibit the strongest electrooptic effect they allow the use of smaller electrodes than the other mentioned modulator principles. The resulting lower capacitance has the advantage of higher cut off frequencies. Furthermore MQW modulators can be made insensitive to the polarisation of the modulated light by introducing tensile strain to the quantum wells. This avoids the need for expensive polarisation maintaining fibers. Layer structure: contact n InAlAs nid InAlAs InGaAs/InAlAs - MQW nid InAlAs n+InAlAs s. i. InP Fig. 1: Sketch of the electroabsorption waveguide modulator 3.4 Optical Sensor Systems 0 100 % 63 Field (106 V / m) 10 5 waveguide is formed by wet chemical etching with cytric acid down to the n+ layer. Afterwards the electrical contact are produced by vacuum coating. The ohmic contact on the n+ layer is realised in GeNiAu, for the Schottky contact on the n- layer we use CrAu. The design of the device structure was supported by BPM-Simulations and by calculations of the electrooptical behaviour. For the BPMSimulations we used the comercial BPM-Software BPM-Cad. The main aim of this simulation was to determine the optical confinement factor, i. e. which part of the guided modes overlaps with the absorbing MWQ layer. We calculated a confinement factor of 14 %. For the calculation of the absorption coefficient of the MQW-material in dependence of the electrical field we used a transfer matrix method. From the obtained results we calculated the optical absorption of the modulator in dependence of the applied electric field as shown in Fig. 2. 15 80 % Absorption Slope: 0,38 / V 60 % 40 % 20 % Wavelength: 1.55µm Device lenght: 500 µm 0% 0 2 4 6 Voltage [V] Fig. 2: Calculated electrooptical behaviour of a modulator device. Device structure In Fig. 1 a sketch of the device is shown. The modulators are grown by MBE in the ternary material system InGaAs/InAlAs on InP subtrates. The device structure is n + in -. The ridge 1580 PL-Measurement at 12 K Simulation at 12 K Simulation at 300 K 1560 Excitonic wavelength [nm] 1540 1520 1500 1480 1460 1440 1420 Mod 05 Mod 07 Mod 09 Mod 08 1400 Mod 13 1380 Shg 05 Shg 07 Shg 03 Shg 04 Shg 06 Mod12 Shg 08 1360 1340 6 7 8 9 Thickness of quantum-wells [nm] Fig. 3: Excitonic wavelength of the MQW material determined by photoluminescence measurements compared with calculated values. 10 Experimental results The epitaxial wafers were characterized by photoluminescence measurements. The point of interest was the spectral position of the excitonic peak of the MQW material, which indicates the position of the absorption edge. A comparison between the experimental results and the calculations is shown in Fig. 3. The optical transmission of the modulator is characterized by coupling the light of an erbium doped fiber laser into the waveguide and detecting the transmitted light on the other fac- 64 3 RESEARCH [2] Heinzelmann, A. Stöhr, M. Groß, D. Kalinowski, 0,5 T. Alder, M. Schmidt, and D. Jäger, Optically Powered Remote Optical Field Sensor Sys- Modulation [a.u.] 0,4 tem using an Electroabsorption-Modulator, 0,3 IEEE MTT-S International Microwave Sympo- λ = 1550 nm sium, Conference Proceedings, Baltimore, June 0,2 1998 0,1 0,0 0 -1 -2 -3 -4 Voltage [V] Fig. 4: Modulation of the MQW modulator versus applied voltage. et of the waveguide. The optical tranmission is measured in dependence of the applied voltage as shown in Fig. 4. Conclusions An electrooptical MQW waveguide modulator has been designed. The epitaxial layers have been grown by MBE and characterised by photoluminescence measurements. The position of the measured exciton peaks was in good agreement with the calculated values. Modulator devices have been processed from these wafers by wet chemical etching and vacuum coating of the electrical contacts. The devices have been characterized by optical transmisson measurements. References [1] Stöhr, R. Heinzelmann, T. Alder, M. Schmidt, M. Groß, and D. Jäger, Integrated Optical E-Field Sensors using TW EA-Modulators, Interational Topical Workshop on Contemporary Photonic Technologies CPT98, Technical Digest, Tokyo, January 1998 3.4.2 Photovoltaic cells for fiber optic EMC - Sensor power supply D. KALINOWSKI P hotovoltaic cells play an important role in power supply of hybrid sensors. A photovoltaic cell array is under construction to supply an active fiber optic hybrid sensor head. A prototype with first experimental results will be shown. Introduction Due to more and more restrictive laws regulating the electromagnetic compatibility (EMC) of electronic equipment the necessity of developing precise and reliable sensors to measure electromagnetic fields steadily increases. One request for such sensors is non invasiveness. Hence, our approach to reach this goal is to develop a hybrid fiber optic fiels sensor. This concept takes advantage of the fact, that optical fibers do not interfere with the electromagnetic field that is to measure, but that they are capable to transmit optical information. By this distortion of the E-field is minimized. The photovoltaic cell array (PVC) described in this article is part of this optical E-field sensor which is shown in Fig. 1. 3.4 Optical Sensor Systems 65 active region is 600 µm, i.e. the diameter of the core of the multimode fiber used. Thus, each quarter of this array, i.e. each PVC, is illuminated uniformly leading to a maximum generation of electrical power by this configuration [1]. Fig. 1: Sketch of the optically powered integrated optical field sensor Device One requirement to be matched by the PVC is a high efficient conversion of optical into electrical power. Therefore, special effort has to be laid upon the layout and the composition of the heterostructures. Since cheap and powerful laser diodes are available in the 800 to 850 nm wavelength regime and since GaAs has its optimum photovoltaic response at about 800 nm, the PVCs are designed as AlGaAs/GaAs pindiodes. Fig. 2 shows a photograph of a cell array consisting of 4 cells. The diameter of the Fig. 3: Cross section of the photovoltaic cells The GaAs and AlGaAs layers are MBE grown on semi-insulating GaAs substrate. The layer structure, illustrated in Fig. 3, consists of an 100 nm n+-AlGaAs contact layer, a 2 µm i-GaAs absorption layer and a 100 nm p+-AlGaAs window layer. A 10 nm p+-GaAs contact layer offers an aluminium protection from oxidation. The metallic contacts act as ohmic contacts. GeNiAu is used for the n-contact and PtTiPtAu for the p-contact. Fig. 2: Perspective view of the photovoltaic cell array consisting of 4 cells 66 3 RESEARCH 3.4.3 Time- and frequency-domain electro-optic field mapping of nonlinear transmission lines P. BUSSEK, TH. BRAASCH, AND G. DAVID W Fig. 4: Photovoltaic cell efficiency Results The cell array is illuminated by a laser with a emission wavelength of 800 nm. Measurements of the max. efficiency show results up to 28% depending on the optical input power (Fig. 4). Conclusion This report presents a photovoltaic cell array for power supply of our hybrid EMC - Sensor. The design and a first result are shown. References [1] M. B. Spitzer, et. al., Monolithic series-connected gallium arsenide converter development, Proc. 22nd IEEE Photovoltaic Specialists Conference, Las Vegas, USA, 1991 e report on the measurements of electric field distributions in monolithic microwave integrated circuits (MMICs) using electro-optic probing techniques. In 1996 the activities have been focused on the analysis of microwave propagation in nonlinear transmission lines (NLTLs). The measurements have been performed in frequency domain as well as in time domain as they have been done in one dimension as well as in two dimensions. As an example, in this documentation we present experimental results of periodic NLTLs demonstrating the generation of higher harmonics on these devices and the formation of shock waves. Introduction In recent years, the complexity of monolithic microwave integrated circuits (MMICs) expanded necessitating the development of measurement techniques which keep abreast of the increased demands of an appropriate characterization of these devices. So far, network-analyzers (NWA) are mostly used for onwafer microwave characterization of MMICs. This measurement technique is well established but its application is limited due to the fact that the on-wafer probes needed for this technique only allow the access to external ports. Thus, no circuit-internal measurement or local failure test of the device nore the observation of wave propagation effects is possible using NWAs. In contrast, electro-optic sampling has become a sophisticated technique to study quan- 3.4 Optical Sensor Systems titatively field distributions and wave propagation effects insight microwave and millimeter-wave devices [1-3]. This technique can be performed in frequency as in time domain enabling the detection of the amplitude and phase of a microwave signal as of the temporal evolution of this signal. The spatial resolution of this method is measured to be down to less than 0.5 µm [4]. Hence, each MMIC component can be tested and evaluated noninvasively up to millimeter-wave frequencies. By combining the direct electro-optic probing with 2D scanning of the laser beam two-dimensional field mappings of the device under test (DUT) are possible [5]. Thus, wave propagation effects can be studied which gain a growing interest by circuit designers for two reasons. On one hand, these effects influence the electrical behaviour of MMICs resulting in a limitation of their bandwidth or the 67 generation of unwanted modes [3]. On the other hand, novel types of integrated circuits such as nonlinear transmission lines (NLTLs) can be designed which make use of these effects, e.g. to generate short electrical pulses or to excite higher harmonics. This has been observed particularly in periodic NLTLs and will be shown in this report. Experimental setup The experimental setups used in this project are sketched in Fig. 1. An actively modelocked Nd:YAG laser (wavelength = 1064 nm, pulse repetition rate = 82 MHz) combined with a fibergrating pulse compressor provides short pulses of 5 ps FWHM (full width at half maximum) corresponding to a bandwidth of the setup in excess of 80 GHz. The device under test (DUT) is illuminated from the backside, i.e. the direct elec- Fig. 1: Experimental setup, (a) for the frequency domain measurements, (b) for the time domain measurements 68 3 RESEARCH tro-optic sampling is applied since the linear electro-optic effect in the sub-80 15 GHz strate itself is used for the modulation -90 of the polarization. To convert this po-100 30 GHz larization modulation into an intensity -110 modulation, polarizers and a quarterwave plate are implemented in the op-120 45 GHz 60 GHz tical pathway. The reflected intensity is -130 detected by a small area photodiode. -140 Due to the combination of a small area 0 500 1500 2500 photodiode and the confocal arrangepropagation distance (µm) ment of the setup out-of-focus-light is Fig. 2: Electro-optic signal of the fundamental microwave suppressed improving the spatial resat 15 GHz and of its higher harmonics along the center olution of the measurement system conductor of a periodic NLTL from input to output. down to less than 0.5 µm [4]. For 2D scans the probe stage is movable in For the measurements performed in time the x- and y-direction. domain some modifications of the experimental Fig 1(a) illustrates the experimental setup for setup are needed. As depicted in Fig. 1(b), the frequency domain measurements. Here, a specoptical pulses themselves generate the electritrum analyzer working as a tunable bandpass is cal signal in order to establish a phase locking used for the detection of the signal amplitude. between the probe pulses and the electrical miThe intermediate frequency is set to several crowave signal. A small part of the output beam MHz, since in this regime the high speed avaof the Nd:YAG laser is separated via a beam lanche photodiode used exhibits a maximum splitter and chopped at about 4 kHz. The photosensitivity. The microwave synthesizer, the modcurrent of a fast photodiode detecting this outelocker synthesizer of the laser system and the put signal then traverses a mechanical delay line spectrum analyzer are phase stabilized via a that periodically shifts the phase of the signal phase locked loop (PLL). For the phase meawhile the observation point is kept constant. The surements the spectrum analyzer is replaced by photodiode has to be changed by a slow Gea lock-in amplifier (not shown in Fig. 1(a)). In a diode to apply lock-in techniques at the chopsecond mode this setup is used to receive an ping frequency. optical image of the measurement region by simply detecting its front surface reflectivity. Thus, Frequency domain measurements the electro-optical signal can be normalized to The results presented here all have been the particular reflectivity of the device, and the done with periodic nonlinear transmission lines. absolute value of the voltage between the deFor a more detailed description of the examined vices top and bottom surface can be determined samples see R. Hülsewede, Investigations of [6]. 3.4 Optical Sensor Systems 69 plitudes of the higher harmonics increase indicating that they are generated along the transmission line. The obvious standing wave patterns are caused by an impedance mismatch at the end of the line and phase mismatching of the harmonics. Tw o - d i m e n sional field mappings of an NLT L are shown in figs. 3. Here, the frequency of the fundamental is 6 GHz, Fig. 3: Nonlinear transmission line; (a) metallization structure; results of 2D field and the metalmappings (b) at the fundamental at 6 GHz, (c) at the second harmonuc at 12 GHz lization strucand (d) at the third harmonic at 18 GHz. ture of the depulse compression on nonlinear transmission vice, the electro-optic signal of the fundamental, lines, in this annual report. Fig. 2 depicts the the second harmonic at 12 GHz and the third spatial distribution of the incident fundamental harmonic at 18 GHz are presented in Figs. 3(a) electrical signal at 15 GHz and the amplitudes - (d), respectively. These Figs. show the decrease of the second, third, and fourth harmonic with of the fundamental signal and the increase of the frequencies up to 60 GHz. As can be seen the harmonics while propagating along the NLTL as amplitude of the fundamental signal decreases Fig. 2 does, but additionally they reveal an unin the direction of propagation whereas the amsymmetrical distribution of the electro-optic signal 70 3 RESEARCH k=1 k=5 (a) -80 -40 0 40 time (ps) k = 10 (b) 80 -80 -40 0 time (ps) 40 80 k = 14 higher harmonics. In the time domain, this formation of a shock wave is the counterpart to the generation of harmonics in the frequency domain. The presented results validate that the electrooptic probing technique is capable of studying and demonstrating this effect as well. Conclusion In summary, electrooptic measurement tech(d) (c) niques have been used -80 -40 0 40 80 -80 -40 0 40 80 to internally investigate time (ps) time (ps) wave propagation effects along periodic nonlinear Fig. 4: Electro-optic signal of the fundamental microwave at 15 GHz and transmission lines enof its higher harmonics along the center conductor of a periodic NLTL abling circuit-designers to from input to output. get an insight into the inthat can not be detected with one-dimensional circuit electrical characteristics of complex milinescans as is the case in Fig. 2. We contribute crowave devices. The generation of harmonics this behaviour to the excitation of parasitic propand the formation of shock waves have been agation modes [3]. demonstrated showing, that this method is suitable to examine internal field distributions in Time domain measurements MMICs in both, frequency domain and time doIn time domain measurements there is a fixed main. phase relation between each particular measurement point. Thus, the evolution of a periodReferences ic signal can be observed as is elucidated in [1] K.J. Weingarten, M.J.W. Rodwell, and D. Bloom, Figs. 4(a) to 4(d) for the development of a sinuPicosecond optical sampling of GaAs integrated soidal electrical signal of 6 GHz propagating circuits, IEEE J. Quantum Electron., vol. QEalong a periodic NLTL at the 1st, the 5th, the 24, (1988), pp. 198-220 10th and the 14th diode, respectively. As can [2] G. David, S. Redlich, W. Mertin, R.M. Bertenburg, be seen, shock waves are generated with fall S. Kosslowski, F.J. Tegude, E. Kubalek, and D. times down to 5 ps due to the interaction of the Jäger (1993), Two-dimensional direct electro- 3.4 Optical Sensor Systems optic field mapping in a monolithic integrated GaAs amplifier, Proc. 23rd EuMC 1993, Madrid, 71 dyne electro-optic measurement setup and present first experimental results. Spain, 1993, pp. 497-499 [3] G. David, R. Tempel, I. Wolff, and D. Jäger, Analysis of microwave propagation effects using 2D electro-optic field mapping techniques, Optical and Quantum Electronics, Special Issue on Optical Probing of Ultrafast Devices and Integrated Circuits, vol. 28, 1996, pp. 919-931 [4] G. David, P. Bussek, U. Auer, F.J. Tegude, and D. Jäger, Electro-optic probing of RF signals in submicrometre MMIC devices, Electron. Lett., 1995, Vol. 31, No. 25, pp. 2188-2189 [5] M.G. Li, E.A. Chauchard, C.H. Lee, and H.-L.A. Hung, Two-dimensional field mapping of GaAs microstrip circuit by electrooptic sensing, Proc. OSA Int. Top. Meeting `Picosecond Electronics and Optoelectronics`, March 13-15, 1991, Salt Lake City, USA, pp. 54-58 [6] G. David, W. Schröder, D. Jäger, and I. Wolff, 2D electro-optic probing combined with field theory based multimode wave amplitude extraction: a new approach to on-wafer measurement, Symposium Digest 1995 IEEE MTT-S International Symposium, May 15 -19, 1995, Orlando, USA, pp. 1049-1052 3.4.4 Characterization of monolithic microwave integrated circuits by heterodyne electro-optic sampling TH. BRAASCH T he propagation of electric signals in the millimeter- and microwave regime along monolithic microwave integrated circuits (MMICs) can be studied by electro-optic measurement techniques. In this paper, we describe the implementation of a hetero- Introduction Using common network analyzer methodes (NWA) for the characterization of MMICs, the device under test (DUT) is measured as an integral device and no in-circuit measurements are possible [1]. The increasing working frequencies of integrated circuits up to the millimeter- and microwave region [2-4] necessitate measurement techniques that allow an insight into the device. In recent years, electro-optic sampling (EOS) has become a sophisticated technique to observe field distributions in MMICs with a spatial resolution down to less than 0.5µm [5, 6]. Quantitative characterizations have been carried out, and one- as two-dimensional measurements are possible in time-domain as in frequency domain [7, 8]. Thus, circuit-designers get a knowledge of circuit-internal parameters, which is of increasing importance since the complexity of the devices expands. Nevertheless, so far the EOS has been mostly performed with a pulsed laser. Here, to convert the microwave down to frequencies, where spectrum analyzer, lock-in amplifier and the photodiode used are able to detect the electro-optic signal, the n-th harmonic of the repetition frequency of the pulsed laser is used to interact with the electric signal. The electric bandwidth of these setups is limited by the pulse width of the laser pulses. In frequency domain, measurements on nonlinear transmission lines up to 100 GHz are reported [9, 10], and fall times down to 1.5 ps have been measured in time domain [11]. However, utilizing the n-th harmonic of the repetition frequency of the pulsed laser for the down conversion of the electric signal leads to a reduction of the signal to noise ratio of the electro-optic sig- 72 3 RESEARCH of the beat frequency between the two lasers. After passing a l/ 4 polarization control the light of the first laser is back-side coupled into the device under test where the stray field of the microwave in the substrate interacts with the laser light via the Pockels-effect. The Fig. 1: Sketch of the heterodyne electro-optic measurement setup reflected light is again coupled nal since the phase noise of the setup increases into the fiber and traverses the circulator and a with the measurement frequency. To circumvent second polarization control. Thus, the polarizathis restriction two cw lasers can be used where tion modulation due to the Pockels-effect is the second laser acts as a local oscillator. Here, converted into an intensity modulation directly corphase noise of the setup only depends on the related to the strength of the electric field at the stability of the two lasers [12]. In this paper, we particular point of measurement. The DUT is present our measurement setup and show first placed on a translational stage enabling twoexperimental results. dimensional field mappings of the field distribution. Due to the confocal arrangement of the setup Experimental setup it can also be used as an optical microscope. In We operate with two identical Er-doped fiber this mode, the front surface reflectivity of the lasers exhibiting a linewidth < 10 kHz. Both ladevice can be detected and afterwards the elecsers are continuously tunable between 1530 nm tro-optic signal can be normalized to the particuand 1560 nm leading to beat frequencies up to lar reflectivity at each particular measuring point. 4 THz. They deliver > 20 mW optical output and As a consequence, the absolute value of the show almost no mode-hops once they reached voltage between the devices top and bottom side thermal equilibrium. The degree of polarization can be determined [8]. Via a fiber coupler the is > 99%. Fig. 1 demonstrates the configuration local oscillator, i.e. the second laser, is of our setup. A small percentage of both lasers superposed to the reflected light from the DUT is coupled into a Fabry-Perot optical spectrum carrying the information of the microwave signal analyzer with a finesse >150 for the detection applied to the DUT. The intermediate frequency 3.4 Optical Sensor Systems 73 Fig. 2: (a) Sketch of a coplanar waveguide structure, (b) reflected intensity measured with the heterodyne setup at 8.5 GHz. between the second laser and one sideband of the first laser, i.e. f1 ± fm with f1 the frequency of the first laser and fm the microwave frequency, can now be adjusted by the frequency f2 of the second laser. This intermediate frequency is detected by a fast travelling-wave photodetector [3]. Hence, any microwave frequency within the tuning range of the two lasers can be converted to some MHz or GHz only affected by the inherent phase noise of the two lasers but independently of the frequency. Results A coplanar waveguide structure (CPW) was used to demonstrate the feasibility of the setup as a scanning microscope. Fig. 2 depicts the surface reflectivity of the CPW. The difference frequency between the lasers was arbitrarily set to 8.5 GHz since at this value they worked extremely stable and the detected signal of the spectrum analyzer was about 50 dB larger than the noise floor. In the next step, a microwave has now to be applied to the device and the elec- tro-optic signal has to be detected as in [6], [8] or [10] with the pulsed laser system. Conclusion In summary, to bypass the phase noise restrictions of a pulsed electro-optic measurement setup two narrow linewidth tunable cw lasers have been implemented in the configuration. Thus, the phase noise only depends on the characteristics of the lasers but is not affected by the measurement frequency. Owing to the tuning range > 30 nm of the Er-doped fieber lasers used heterodyne detections of electric signals up to 4 THz should be possible. As a first result, the surface reflectivity of a coplanar waveguide structure detected at 8.5 GHz difference frequency is presented. References [1] D.J. Bannister and M. Perkins, Tracebility for onwafer s-parameter measurements, IEE Proc. A, vol. 139, 5, 1992, pp. 232-233 74 3 RESEARCH [2] M.J.W. Rodwell, S.T. Allen, R.Y. Yu, M.G. Case, [9] R. Majidi-Ahy, B.A. Auld, and D.M. Bloom, 100 U. Bhattacharya, M. Reddy, E. Carman, M. GHz on-wafer s-parameter measurements by Kamegawa, Y. Konishi, J. Pusl, and R. Pullela, electro-optic sampling, IEEE MTT-S, 1989, pp. Active and nonlinear wave propagation devices 299-302 in ultrafast electronics and optoelectronics, Proc. IEEE, vol. 82, 7, 1994, pp. 1037-1060 [10] Th. Braasch, G. David, R. Hülsewede, U. Auer, F.-J. Tegude, and D. Jäger, Propagation of mi- [3] M. Alles, Th. Braasch, R. Heinzelmann, A. Stöhr, crowaves in MMICs studied by time- and fre- and D. Jäger, Optoelectronic devices for micro- quency-domain electro-optic field mapping, Proc. wave and millimeterwave optical links, Proc. Trends in Optics and Photonics Series (TOPS) MIKON96, Workshop Optoelectronics in Micro- of OSA 1997 Spring Topical Meeting Ultrafast wave Technology, Warsaw, Poland, 1996 (in- Electronics and Optoelectronics, 1997, Lake vited) Tahoe, USA [4] I.V. Ryjenkova, V.K. Mezentsev, S.L. Musher, [11] K.S. Giboney, S.T. Allen, M.J.W. Rodwell, and S.K. Turitsyn, R. Hülsewede, and D. Jäger, Mil- J.E. Bowers, Picosecond measurements by free- limeter wave generation on nonlinear transmis- running electro-optic sampling, Phot. Tech. Lett., sion lines, Ann. des Telecomm., Special Issue, vol. 6, 11, 1994, pp. 1353-1355 1996 (invited) [12] S. Loualiche and F. Clerot, Electro-optic micro- [5] K.J. Weingarten, M.J.W. Rodwell, and D. Bloom, wave measurements in the frequency domain, Picosecond optical sampling of GaAs integrated Appl. Phys. Lett. 61, (18), 1992, pp. 2153-2155 circuits, IEEE J. Quantum Electron., 1988, QE24, pp. 198-220 [6] G. David, P. Bussek, U. Auer, F.J. Tegude, and D. Jäger, Electro-optic probing of RF signals in submicrometre MMIC devices, Electron. Lett., 1995, Vol. 31, No. 25, pp. 2188-2189 3.4.5 Development of an experimental setup for field probe measurements on nonlinear transmission lines [7] M.G. Li, E.A. Chauchard, C.H. Lee, and H.-L.A. Hung, Two-dimensional field mapping of GaAs microstrip circuit by electrooptic sensing, OSA Proc. Picosecond Electronics and Optoelectronics, March 13-15, 1991, Salt Lake City, USA, pp. 54-58 [8] G. David, R. Tempel, I. Wolff, and D. Jäger, Analysis of microwave propagation effects using 2D electro-optic field mapping techniques, Optical and Quantum Electronics, Special Issue on Optical Probing of Ultrafast Devices and In- D. KALINOWSKI AND R. HÜLSEWEDE A n experimental setup for field probe measurements has been established. The electrical fields on nonlinear transmission lines has been measured up to 60GHz. Theoretical considerations have been done up to 200GHz. Experimental results have been compared with results using electrooptic testing. tegrated Circuits, 1996, 919-931 Introduction In recent years there has been a great progress in the development on nonlinear trans- 3.4 Optical Sensor Systems 75 mission lines making them to key structures for future microwave circuits [1]. To characterize these structures measurements at external ports are not sufficient. Different noncontacting probes give a chance to take a look at the field distribution on the transmission lines. The probes are the electro-optic probe [2], the magnetic field probe [3] and the electric field probe [4]. Such an electric field probe has been established. Its potential has been demonstrated. Experimental setup A sketch of the experimental set-up is shown in Fig. 1. The nonlinear transmission lines (NLTLs) are supplied by a microwave synthesizer and a DC-voltage source. They can be loaded with variable resistance. The probe detects the electric field above the NLTL. Its signal is evaluated by a spectrum analyzer. A computer controls the position of the probe and stores the data measured by the spectrum analyzer. By that way two-dimensional field mappings can be done. Theoretical results The probe can be described by an equivalent circuit which is shown in Fig.2 [5]. The voltage Ul depends on the electric field strength. y U probe The probe impedance is given by RV and RS which characterize the thermal and radiation losses and X A describing the open line. The relation between the indicated power at the spectrum analyser and the square of the measured field intensitity is shown in Fig. 3. It demonstrates the steady increasing probe sensitivity versus frequency. Thus the probe can be used over the whole frequency range. Measurements confirm this equivalent circuit. Fig. 4 shows the relation between the indicated signal on the spectrum analyser and the signal frequency. The experimental results are given by the dots (·), the theoretical results by the line (-). A good correspondence is given between these results. So the equivalent circuit can be used to determine the electric field strength by the spectrum analyzer signal. 0,8 x synthesizer z load bias tee dc - source 0,6 0,4 0,2 0 prober jX A Fig. 2: Equivalent circuit for the electric field pA ) Vm mixer Rs Ul sensitivity ( spectrumanalyser Rv NLTL Fig. 1: Sketch of the experimental Setup 0 40 80 120 160 frequency (GHz) Fig. 3: Sensitivity versus frequency. 200 76 3 RESEARCH rel. signal (dB) -26 -30 -34 -38 -42 theory measurement -46 -50 5 10 15 20 25 30 frequency (GHz) 35 40 Experimental results One experimental result achieved with this setup is presented in Fig. 5. A 4 µm NLTL has been examined. The sketch of this line is shown in Fig. 5c. Whereas one end has been connected with a synthesizer, the other end has been unloaded. The synthesizer has supplied the line with a 7.4 GHz microwave signal. The field distribution at this frequency is shown in Fig. 5a. The 3rd harmonic generated on the line is shown in Fig. 5b. The amplitude increases in the direction of propagation indicating the generation along the transmission line. Furthermore an asymmetrical transversal distribution is revealed. Conclusion Measurements up to 60 GHz have been done successfully. Field distributions on transmission lines with electrode widths down to 12 µm has been shown. Comparisons with theoretical results and electo-optic probing are in good agreement. References [1] M.J.W. Rodwell, et al., Active and nonlinear wave propagation devices in ultrafast electronics and optoelectronics, IEEE Proc., Vol. 82, No. 7, pp. 1037-1059, 1994 [2] P.Bussek, G. David, Quantitative analysis of two-dimensional electro-optically measured field distributions in MMIC-structures, Annual Report, Gerhard-Mercator-Universität - GH - Duisburg, Fachgebiet Optoelektronik, 1995 [3] Y. Gao, I. Wolff, A new miniature magnetic field probe for measuring three-dimensional fields in planar high-frequency circuits, IEEE Transactions on Microwave Theory and Techniques, Vol. Fig. 5:2-dim. field mapping of an NLTL (a) signal with 7.4 GHz, (b) generated 3rd harmonic at 22.2 GHz c) Sketch of the NLTL 44, No. 6, June 1996, pp. 911-918 3.5 Technologies for Optoelectronic Components and Systems [4] D. Kalinowski Entwicklung eines Feldsondenmeßplatzes zur zweidimensionalen Analyse elektromagnetischer Feldverteilungen auf 77 3.5 Technologies for Optoelectronic Components and Systems nichtlinearen Leitungen, Diploma thesis, Gerhard-Mercator-Universität Duisburg, 1996 [5] R. Geißler, et al., Taschenbuch der Hochfrequenztechnik Band 2: Komponenten, Springer- 3.5.1 Development of a measurement system for the optical characterization of full-colour-LED-displays Verlag, Berlin-Heidelberg, 1992 M. WENNING, R. BUß, AND A. STÖHR F or the optical characterization of a fullcolour-LED-display, two measurement systems have been developed. The first one is to determine the spatial distribution of a LED and the second one is to receive the spectrum of a LED or a LED-pixel. The x, y, z colour coordinates of the CIE chromaticity diagram are evaluated from the spectrum. By using the measurement system, an optimization of a full-colour LED-display was performed. Introduction Among the various types of flat panel displays (i.e., CRT, VFD, PDP, LCD, LED and EL), LED displays are widely used as information boards and as transportation terminal displays due to their excellent reliability, service life and visibility. Particularly as a result of the remarkable progress made with high-brightness blue and green LEDs, full-colour displays can now be established for outdoors. Any colour can be produced using the three primary colours red, green and blue. In this report, measurement systems are developed to characterize a full-colour LEDdisplay. Furthermore, the LUMINO XTralux ML 4 C-Pixel and an optimized Pixel has been characterized. 78 3 RESEARCH Measurement systems The spatial distribution of a LED is measured with the setup shown in Fig. 1. The LED is fixed on a LED-holder and is driven by a constant current. By rotating the swivel-arm in 1°- steps, the data of the spatial distribution is received. The Fig. 2 shows the setup to determine the spectral distribution of a LED or LEDpixel. After the spectum is measured, the x, y, z colour coordinates are determined [1,2]. The CIE (Commission Internationale de lEclairage) diagram is the standard colourimetric system. The x, y, z axis of this diagram are based on three colour-matching functions, each of which is related to the spectrum of red, green and blue. A sequence of single-wavelength computer IEEE light stop with aperture chopper Lock-in-amplifier In Ref. detector = lens + photodiode monochromator with stepper motor lens Fig. 2: Measurement setup - spectrum hole single-LED or LED-Pixel 3.5 Technologies for Optoelectronic Components and Systems colours can be expressed as a curve in the x, y, z space of the CIE diagram and the projection of the curve on the x, y plane is a horseshoe-shaped pattern (Fig. 3). Any colour can be expressed as a point inside of this horseshoe-shaped curve. 0,8 single-wavelenght colours 0,6 yellowish green red 0,4 D65 Experimental results The colour coordinates of all measured LEDs are shown in Fig. 3 which are determined from the spectrum of the LEDs. In Fig. 4 (a) every colour in the triangle region, of which the vertices indicate the three primary colours of the LUMINO -Xtralux LEDs, can be radiated by adjusting the luminous intensity of each LED. As blue 0,2 0,0 0,0 0,2 0,4 0,6 0,8 x Fig. 3: CIE diagram of all LEDs (b) (a) 0,8 0,8 single-wavelenght colours 0,2 0,0 0,0 0,6 green 0,4 D65 0,2 0,4 0,6 0,4 D65 0,2 red blue single-wavelenght colours green y y 0,6 blue 0,0 0,0 0,8 red 0,2 0,4 0,6 0,8 x x Fig. 4: (a) CIE diagram of LUMINO Xtralux-pixel and (b) of the optimized pixel. (b) (a) 0,8 0,8 single-wavelenght colours single-wavelenght colours 0,6 0,6 D65 y 0,4 0,4 20° 40° 0° 80° 40° 20° 0° 80° 90° 0,2 0,2 0,0 0,0 D65 60° 60° y y green 79 0,2 0,4 0,6 0,8 0,0 0,0 0,2 x Fig. 5: (a) Colour-shift of LUMINO-Xtralux-pixel and (b) of the optimized pixel. 0,4 x 0,6 0,8 80 3 RESEARCH (a) (b) 0,8 0,8 relative intensity 1,0 relative Intensity 1,0 green 0,6 blue 0,4 red 0,2 0,0 green 0,6 red blue 0,4 0,2 0 20 40 60 viewing angle /° 80 0,0 0 20 40 60 viewing angle /° 80 Fig. 6: (a) Spatial distribution of the LUMINO-Xtralux-pixel and (b) of the optimized pixel. seen in the diagram, it is not possible to obtain the standard white D65, which is outside the triangle. The colour coordinates of an optimised Pixel are shown in Fig. 4 (b). The LEDs are wellchosen to obtain a greater triangle region. In Fig. 5 the colour coordinate variation versus the viewing angle of these two Pixels is shown. The XTralux ML 4 C-Pixel (Fig. 5 (a)) has a large colour shift to the red primary colour, which can explained with a wider spatial distribution of the red LED (cf. Fig. 6 (a)). The small shift of the optimized Pixel was performed by using a red, green and blue LED having nearly the same spatial distribution. This was obtained by modification of the lense-form (encapsulation) of the LED. Furthermore, the LED-surface is roughend. Conclusions Within the scope of this thesis, two measurement setups have been developed. Furthermore, LEDs and the LUMINO XTralux ML 4 CPixel have been characterized. With the knowledge of colour metrics and the colour coordinates of the LEDs, an optimized Pixel has been assembled. A greater colour range and a nearly constant colour coordinate versus viewing angle of the pixel has been achieved. References [1] Heinweg Lang, Farbmetrik und Fernsehen, R.Oldenburg München Wien, ISBN 3-48620661-3, 1977 [2] DIN 5033, Farbmessung 3.5.2 Opinion poll on the evaluation of the legibility of LED-based displays R. HEDTKE AND R. BU ß n this report the legibility of LED-based displays is evaluated on the base of interviews with passengers of the public local traffic. To reach this aim a model explaining the causal connection between several external influences and the legibility is made up according to DIN 1450. Based on this model a questionnaire for the interviews I 3.5 Technologies for Optoelectronic Components and Systems 81 the legibility of those display-systems (cf. Fig. 1) are acquired. The causal connection The causal connection based on DIN 1450 is modified referring to the requirements of the questionnaire. The result is shown in Fig. 2 where the arrows describe the influences. It can be seen that Fig. 1: LED-based display used in the public local traffic. there are eight major points having an influence on the legibility: is developed. The gained data of the interThe type face and size, brightness and colour, views is evaluated using statistic methods. the distance between each letter, the distance of view, the light conditions, and personal influIntroduction ences. Today, the permanent availability of information has become of increasing importance, The used questionnaire where the transmission of visual information cerBased on the causal connection, the questainly becomes to play a more and more importionnaire shown in Fig. 3 has been developed tant role. Due to the permanent development in to proof and determine the level of influence of the area of LED-technology it has become poseach point. The first three questions are so called sible to produce so-called super-luminescence icebreaker-questions to start the conversation light emitting diodes (SLED), having a very high with the person to be interviewed. The following brightness. In the public local traffic LED-based questions are related to the legibility of the disdisplays find an increasing application. In coplayed text. Legibility, type size, distance beoperation with the company LUMINO/Krefeld, a tween each letter, type face, brightness, and producer of such displays, methods to improve colour are assessed using marks between one and five. This type of assessment has been chosen because everyone knows this marks from school. This causes TYPE FACE BRIGHTNESS COLOUR that comprehension problems are avoided. DISTANCE BETWEEN TYPE SIZE LEGIBILITY LETTERS Furthermore, tendencies like small, right, DISTANCE OF VIEW LIGHT PERSONAL INFLUENCES and large where recorded if possible. Finally, the other points Fig. 2: Model of the causal connection. 82 3 RESEARCH Fig. 3: The used questionaire 3.5 Technologies for Optoelectronic Components and Systems of influence are recorded after the main part of the interview. Interview For the interviews, locations in the following cities are selected according to the practicability of the interviews and considering the local conditions: Duisburg, Essen, Düsseldorf; Oberhausen, Leipzig, Stuttgart. At each of these locations interviews with 50 passengers have been carried out. To reach a high comparability of the gathered data, the interviews only took place on platforms for the public local traffic. Conclusion The Fachgebiet Optoelektronik at the Gerhard-Mercator-Universität Duisburg has investigated the judgement of the legibility of LED-based displays in co-operation with the company LUMINO/Krefeld, a producer of such displays. The aim of this study was the recording of subjective opinions of the users of the public local traffic. The legibility of the analysed LED-based displays has been valued by up to 90% of the interviewed passengers with well or very well. Especially the displays based on yellow LEDs in the city of Leipzig have been rated very positively. The majority of the passengers have felt the brightness to be right. As well in Leipzig, the valuation has been the same, even if the display was exposed to direct sunlight. As a rule, type size, letter distance, and type were also valued as right. Up to 80% of the passengers have felt as being informed very well by those displays. In summary, the users of the dynamic passenger-information-system are contented with the quality and information provided by these 83 systems. It should be pointed out that the LEDtechnology serves the high requirements referring to the demands in the public local traffic. Even under unfavourable conditions like direct sunshine the judgement is good or very good. References [1] DIN 1450, Beuth, Berlin, Juli 1993 [2] V. Dreier, Datenanalyse für Sozialwissenschaftler, Oldenbourg, München-Wien, 1994 [3] K. Holm (Hrsg.), Die Befragung 1, Franke, München, 1975 [4] E. Noelle, Umfragen in der Massengesellschaft, Rowohlt, Reinbek bei Hamburg, 1963 3.5.3 Evaluation of possible improvements to enhance the UV-power efficiency of a xenon flashlamp system B. NEUHAUS AND A. STÖHR T his paper will present experimental results of the characterisation of a xenon flashlamp system with a flat aluminium rear reflector fabricated by Bläsing Elektronik GmbH. The spatial distribution of the UV-radiation is determined. Studies about typical gas components and some materials of the discharge tube result in possibilities to optimize the arc lamp. Furthermore the shape of a single flash is recorded and the efficiency is measured in dependence of the pulse repetition rate. To optimize this system several rear reflectors with different geometrics are tested and the reflection index of five different materials is measured in the UV and NIR wavelength range. For all these examinations different measurment setups are worked out. 84 3 RESEARCH Introduction The UV-drying process is of great significance to the industry working with printing procedures e.g. the silk-screen printing. New UV-paints are free from solvents and they harden only by irradiation with UV-light [1]. Therefore powerful lamps with great emission in the UV-range are very important. Pulsed UV-lamps offer several advantages compared with conventional UVburners. With flashlamps the intense heat emission can be reduced; they radiate only for a short pulse duration but the hardening process is more effective because the radiation output of flashlamps is greater than of conventional lamps [2]. The UV-flash drying process will become an economical and nonpolluting alternative process in the future. The intention of the following investigations is to characterize such a flashlamp system and to optimize it based on the experimental results. Arc lamp construction and priniciple of a flashlamp system Fig.1 shows the construction of a typical arc lamp. The electrodes are set in a clear quartz tube. The type of quartz depends on the desired output spectrum. The electrodes are made of tungsten to enhance electron emission. Arc + lamps are filled with inert gas under several atmoanode spheric pressure or a mixture of gas and a deffilling gas inite amount of mercury. cathode tube The internal pressure in the tube increases during operation to 15-75 bar, depending on the lamp Fig. 1: Contype. The flashlamp system operates by sending an electric charge from a pulse generator to the gas filled lamp. The gas absorbs the energy by storing it in its atoms and subsequently it releases the energy by emitting photons, which results in a high intensity flash of light [3]. Light emssions in all directions can be guided by installing a reflector behind the lamp which collects the light and reflects it onto a surface which should be treated. Caution: these lamps produce high intensity UV radiation and ozon. Precautions are necessary during operation mode. Experimental setups Fig. 2 and Fig. 3 show the two basic experimental setups used for the measurements. In order to determine the spatial distribution of emitted radiation and to determine the UV-efficiency of the flashlamp in the UV-range the setup in Fig.2 is used. The pulse generator supplies the flashlamp with electrical impulses. The pulse repetion rate (1,56Hz, 3,12Hz, 6,25Hz, 12,5 Hz, 25 Hz) can be adjusted by a rotary switch. The light pulses, radiated by the struction of arc Fig. 2: Experimental setup to measure the lamps [2]. relative optical output of an arc flashlamp. 3.5 Technologies for Optoelectronic Components and Systems Fig. 3: Experimental setup to measure the reflection index. 85 UVL-1500/TP1 fabricated by Bläsing Elektronik GmbH. The pulse shape has a fast rise time t r(10% −90%) of 31,5ms and slower decay. The pulsewidth t FWHM is 520ms. These are typical values for flashlamps [2]. Further experiments have shown, that the flash energy as well as the pulswidth stay constant for all frequencies in the range from 1,56Hz until 25Hz. b) In order to determine the spatial distribution of the radiation from the flashlamp type UVL1500/TP1 the SiC-photodiode is led along a semicircle around the lamp as shown in Fig. 5. One can record the radiant intensity (relative) for a number of angles. The radius is r = 100cm. The values are recorded in steps of optical power (rel.) flashlamp, are detected by a special silicon carbide (SiC; spectral range: 210380nm) photodiode. Fig. 3 shows the setup to determine 0 t r(10%− 90%) = 31,5µ s the UV-reflectivity of different reflector materials. The light beam of a mercury t FWHM = 520 µs lamp is deflected by a beam splitter in definite directions. The reflector reflects the incoming light beam which is then 500 µs / div focused on a SiC photodiode. By us− 2.5ms 2.5ms 0.0s ing the second detector it is possible time t to compare the input power with the reflected power in order to determine the Fig. 4: Relative optical output power of the flashlamp reflection index. UVL-1500/TP1. For all investigations in the NIR wavelength range a monochromator one degree and they are related to the 90°and a silicon photodiode (BPW20; 375-1100nm) direction. Fig. 6 shows the results (normal are used to receive spectral values; the light curve) given in a polar coordinate system. It source is a halogene lamp. To record a single can be seen that the distribition of the radiapulse an oscilloscope is used. tion is symmetrical to the axes. Besides, it can be noticed that over an angle of a= 168° Experimental results fifty percent of the radiant power compared a) Fig. 4 shows the relative optical output powwith the power in direction of 90° is still radiater with respect to time of the flashlamp type 86 3 RESEARCH η ⋅ ( 10 −9 ) flashl amp UVL 1500-TP/1 efficiency r = 100cm 210 − 380 ηUV − 280 ηU270 V pulse repetition rate fPu ls(Hz) Si C-detector JEC-1 210-380nm Fig. 5: Principle to record the spatial distribution of the radiant. ed by the flashlamp. So the angle of radiation is very wide. Mostly such a wide angle is undesirable because one looses most of the radiant in the borderlands when no object is placed very closed to the radiant source. For plenty of applications a distribution like a club (dashed curve) is desired. It is possible to achieve such a distribution with special rear reflectors. Fig. 7: UV-efficiency in addiction to the pulse repetition rate. c) Fig. 7 represents the UV-efficiency in addiction to the pulse repetition rate from the flashlamp type UVL-1500/TP1. The values are calculated for a wavelength range from 210-380nm as well as for the range from 270280nm. The detector was about two meters away from the light source and the measurements were repeated several times for all repetition rates which are adjustable. From these values a mean value is formed for each frequency. This mean value is used for the calculation of the efficiency for flashlamps. The illustration shows that the efficiency stays nearly constant for each frequency. The con- 45 40 35 IR-reflectivity 30 aluminium UV-mirror reflector III reflector II reflector I 25 20 15 10 5 0 -5 780 800 820 840 860 880 900 920 940 960 980 1000 wavelength (nm) Fig. 6: Principle to record the spatial distri- Fig. 8: .Spectral IR-reflection index of differ- bution of the radiant. ent materials. 3.5 Technologies for Optoelectronic Components and Systems 87 groups and experimental results the reflector clusion is, that the pulse repetition rate has no coatings I and III are most suitable to optimize essential influence on the efficiency at least at the xenon flashlamp system. small frequencies below f PULS = 25Hz. d) The printing industry is interested in rear reProposals for improvements flectors with a high reflection index in the UVThe proposals to optimize the flashlamp sysrange for great UV-efficiency and a low retem can be divided in four groups: flection index in the IR-range because of the 1. gas filling undesirable heat, which is produced by infra2. tube material red radiation. Therefore five different materi3. reflector material als are tested for the reflectivity in the UV 4. reflector form (210-380nm)- and IR-(780-1000) wavelength range. These five materials and their UV-reGas filling flection index are: Four typical fillings for arc lamps are xenon - highly polished aluminium, used so far by (Xe), mercury (Hg), the mixture mercury-xenon Bläsing Elektronik GmbH; the UV-reflec(Hg(Xe)) and deuterium (D 2 ). To compare the tion index is ρUV = 90,24%. - a special UV-glass-mirror that transmits the effectiveness of these fillings in the UV-area the IR-radiation and reflects the UV-radiation; efficiency as a function of the input power of difthe UV-index results in ρUV = 97,17%. ferent arc lamps with these gases is calculated - three metallic reflectors I, II and III with difover a range from 210-380nm. In Fig. 9 this comferent coatings; The compositions of these parison is shown. To achieve these results some three coatings and their designation are spectral irradiance curves from the L.O.T.-Oriel unkown. company [2] catalog are evaluated. The xenon During the measurements it was apparent, gas lamps are plainly worse than the other that the three reflectors I, II and III are in all problamps. Increasing the lamp power does little inability diffused and diffused/directed reflecting fluence to the UV-efficiency of the xenon burnmaterials with the consequence that the absolute UV-reflection index could not be determined. D2-deuterium 210 −380 But a second measurement methηUV Xe-xenon od could prove, that all five mateHg-mer cury rials could increase the radiant Hg(Xe )-mercury/xenon power better than the aluminium η *10 reflector. Fig. 8 shows the spectral IRreflectivity of these five materials over a wavelength range from 780-1000nm. The aluminium relamp power (W) flector is worse than the other four 210 − 380 materials. In comparison with all Fig. 9: UV-efficiency ηUV of different discharge lamps. −9 88 3 RESEARCH r eflectance moplastic resin and it is thermally stable to > 350°C. The reflectance is > 95% over this range. Surface contamination only decreases the reflectance at the lower ends of the spectral range. B) metallic reflectors are very sensitive to surface contamination and to overheating. These facts can decrease the reflectance greatly as well as the permanent irradiation with UVlight. wavelength (nm) Fig. 10: Spectral reflection grade of spectralon [4] er. The mixture Hg(Xe) provides the best optical radiation power but to get the optimal power a warm up time of about 15 minutes is necessary. Tube material Quartz is undoubtedly the best material for the discharge tube and it is normally used by the industry. Quartz guaranties the mechanical and thermal durability. The type of quartz depends on the desired UV-output. There exist several special quartz types: a) UV grade quartz that transmits the output to below 200nm, and b) ozone free quartz which absorbs short wavelengths to prevent ozone generation. Above 280nm the special types do not offer advantages compared with standard quartz variants. Reflector material Literary investigations give some new information about reflector materials to optimize the radiation power. a) Labsphere Ltd. company developed a diffuse reflecting material, spectralon, with reflectivity over the range from 2502500nm shown in Fig. 10. Spectralon is a ther- Reflector form Fig.11 shows how the spatial distribution of radiation could be changed when two aluminium reflectors with different forms are used. These forms are described in Fig.12 and Fig.13. The influence on the distribution is tested when the distance between the tube and the reflector varies. The values are related to the 90°-direction and to the values of a 100 80 120 60 140 40 160 180 20 0 1,0 1,25 1,5 1,5 1,25 1,0 angle (°) without reflector R1 1,5 cm R1 4,0 cm R1 6,5 cm R2 1,5 cm R2 4,0 cm R2 6,5 cm Fig. 11: Spatial distribution of the radiation with the reflectors I/II in dependence on the distance between tube and reflector. 3.5 Technologies for Optoelectronic Components and Systems measurement without a reflector. So you are in position to determine the increase of the radiant intensity as well. The reflector R2 could increase the radiant power best, with nearly 40% compared with the values without a reflector. The distance between the tube and the Fig. 12: Cross-section reflector is imporof the reflector R1. tant as well. The reflector R1 produces the best results at a distance of d=4,0cm whereas the second reflector was best at a distance of d=1,5cm. Fig. 13: Cross-section of the reflector R2. Conclusion The analysis of the flashlamp system of Bläsing Elektronik GmbH provided the following results: The flashlamp radiates only for the duration of a pulse. The pulse repetition rate up to 25Hz has no signifcant influence on the radiation intensity. The spatial distribution of the radiation is extremly wide. Over an angle of 168° fifty percent of the radiant power compared to the power in direction of 90° is still radiated by the flashlamp. The investigations to optimize this system provided the following possibilties: The mixture mercury-xenon has the best UV-efficiency and it is suitable to use for fillings in arc lamps. The change of the reflector form and material could increase the radiant power as well. First experimental measurments point to the assumption that e.g. the combination of the reflector material III 89 with the form R2 which could both increase the intensity best, would provide a better efficiency. References [1] Erhardt D. Stiebner, Bruckmann´s Handbuch der Drucktechnik, Bruckmann, München 1992 [2] L.O.T. ORIEL catalog Vol.II, Light Sources, Monochraomators & Spectrographs, Detectors & Detection Systems, Fiber Optics, Oriel Corporation, USA, 1994 [3] Polygon flashlamps http://www.polygon1.com/ technology.html [4] Labsphere catalog; Diffuse reflectance Coatings And Materials, Labsphere, North Sutton, 1996 3.5.4 Construction of a flip chip device for bonding integrated circuits J. ERVENS AND R. BUß o bond integrated circuits in flip chip technology a heating device is required, enabling precise adjustment and soldering of the solder bumps. This device was designed, built and put into operation. Furthermore, an alternative process to electroplating [1] was tested to put solder bumps onto microelectrodes. Therefore, testing chips were constructed on which solder bumps were evaporated. In the completed device test soldering points were carried out and analyzed. T Introduction In the age of space-saving integration of semiconductor circuits the use of flip chip technology is getting more and more interesting. A special advantage is the possibility to connect silicon technique to III-V-compound semiconductors, 90 3 RESEARCH Fig 1: Self-adjustment of solder bumps [2]. which are of high importance to optoelectronic applications. Optoelectronics have many components with vertical radiation. Therefore a direct connection of the silicon substrate to the light emitting chip is very useful for the third dimension. Another point of interest is the self-adjustment of the chips by the surface tension of the melted solder bumps. As shown in Fig. 1 an alignment error of several micrometer in the adjustment can be balanced out. Description of the mode of operation of the heating device The complete equipment consists of - heating unit - adjustment unit - temperature control - temperature measuring instrument The heating unit is shown at the top of Fig. 2. It comprises the halogen radiators and the mirror reflectors and serves for fastening the sample holding device. In the device, which consists of 10 millimeter thick aluminium plates, the chips get soldered, lying on top of each other. A membrane pump sucks the top chip to a heat-resistant pane of glass as shown in Fig. 3. The bottom chip is put down on an appliance fastened to the adjustment unit which consists of a rotary table, that can be moved by hand, and an x-y-zmanipulator, see Fig. 3. The adjustment of the chips can be observed with an ocular from the top. After the adjustment is completed in x- and in y-direction the bottom chip is brought into contact with the top chip by the z- manipulator. Then the nitrogen valve is opened and a low pressure is adjusted by a manometer, to heat the chips in a nitrogen atmosphere. This disables oxidation of the surfaces of the solder bumps. The halogen radiators are started and controlled by the temperature control unit, which consists of a phase-angle control, that influences the electric power. By means of a rotary potentiometer the temperature of the radiators is also influenced. A fast-response thermocouple juts as a measuring sensor into the heating unit. This digital measuring instrument shows the predominant temperature inside. After about ten minutes the chips are soldered and after cooling down thay can be taken out off the heating unit. Fig. 2: Front view of flip chip device. 3.5 Technologies for Optoelectronic Components and Systems 91 References diminished pressure panes of glass [1] G. Sadowski, D. Zeidler: Mikrogalvanik für die Herstellung lötfähiger Bumpsysteme, me Bd 6, 1992, pp 358-361 [2] M. Wale, M. Goodwin: Flip-Chip Bonding Opti- chip 1 chip 2 sealing ring rotatable and in x-y-z-direction shiftable appliance z x y Fig. 3: Principle of chip arrangement. Construction of test chips At first the substrate is covered with a steel resist pattern mask and fastened in the deposition apparatus. Then a gold layer is evaporated, because the adhesive bond strength of gold on the substrate is significantly higher than that of soft solder. The soft solder layer is evaporated in several steps of the operation. Results of the test series The temperature in the heating unit is sufficient in order to melt the solder bumps and to connect the chips. In case of stronger mechanical demand the gold layer dissolves off the substrate, which means the soldering of the solder bumps was successful. Self-adjustment can not be recognized, because the layer thickness achieved in the evaporation process is far too thin. If a higher layer thickness can be achieved, evaporation will be applicable, but regarding todays knowledge electroplating is preferred. mizes Opto-ICs, Circuits and Devices, pp 2531, Nov. 1992 92 4 TEACHING ACTIVITIES 4.1 Lectures, excercises, and practical studies 93 4 Teaching activities 4.1 Lectures, Excercises, and practical studies Technical Electronics 3: Optoelectronics D. Jäger and A. Stöhr The course Technical Electronics 3: Optoelectronics covers the basic theory and technology of modern semiconductor photonic devices as well as applications of these devices in optoelectronic integrated circuits (OEICs). The course starts with the fundamental physical phenomenon of light-material interaction in semiconductors, such as fundamental absorption, spontaneous and stimulated emission. Subsequent lectures deal with the theory and technology of photoconductive devices, photodiodes, modulators, light emitting diodes (LEDs), and laser diodes. Special attention is given to modern quantum well waveguide laserdiodes and their applications in optical communication systems, medicine, and material processing. Ultra High Frequency Transmission Techniques: Optical Signal Transmission D. JÄGER AND R. BUß The course Ultra High Frequency Transmission Techniques: Optical Signal Transmission starts with the propagation of electromagnetic waves considering the features of optical waves at surface boundaries, such as reflection and refraction. Proceeding with the description of such fundamental physical effects like scattering, absorption and dispersion, optical wave propagation in various types of dielectric waveguides is discussed. Based on this fundamentals the design, properties and technological realization of waveguides based on III/V compound semiconductors are discussed. Another main part of this course deals with fiber optic waveguides: Wave propagation in graded index fibers as well as in stepped index fibers is derived where both advantages and disadvantages of each type are elucidated. Problems such as signal distortion in fibre optic waveguides are analyzed and solutions to avoid them are given. Following the topic of wave propagation, the most important devices for optical and optoelectronic integrated circuits (OEIC) are presented. The properties and technological realization of waveguide laser diodes, vertical cavity surface emitting laser diodes (VCSEL), modulators, and detectors are discussed. Finally, economical aspects of optical communication techniques and future prospects like fiber to the home are touched Special Areas of Optoelectronics: Lasers D. JÄGER AND A. STÖHR The first lectures within the course Lasers cover the basic principles and the mathematical description of electromagnetic waves. The course proceeds with the quantum mechanical interactions between electromagnetic waves and atomic materials resulting in the two most important requirements for light amplification by stimulated emission of radiation (laser). Special attention is then given explaining the basic concepts, the functionality, and the characteristic 94 specifications of different laser sources of importance, such as the Helium-Neon laser, the Ar-ion laser, Excimer lasers, the Ti:Sapphire laser, semiconductor laser diodes etc.. Finally, examples of laser applications in various industrial areas (medicine, communication, material processing etc.) are discussed together with future trends. Optical Signal Processing D. JÄGER AND R. BUß The course Optical Signal Processing starts with the basic theory of non-linear optical effects both in dielectric materials and in semiconductors. The causes for optical bistability are described and principles like optical switching are applied to the realization of optical memories and logic elements. Within the next section of this course, the phenomenon of opto-electronic bistability is introduced. It is shown that the integration of a light modulator and a photodetector is leading to so-called self-electro-optic effect devices (SEED), showing various forms of switching behaviour which can be controlled both optically and electrically. Finally, the main advantages of optical signal processing are pointed out while discussing applications such as optical switching networks, image processing systems, optical neural networks, optical phased array antennas, optical computing, and optical interconnects. Multimedia-Techniques D. JÄGER AND R. BUß This course elucidates Multimedia from three different points of view: The optoelectron- 4 TEACHING ACTIVITIES ic area, the informatic area and the area of data processing. Starting with optoelectronic devices and interfaces for fiber-optic networks (LAN, WAN, FDDI), multiplexing (TDM, WDM) and routing techniques in the optical domain are introduced. Problems of high capacity data storage using optical techniques and mobile connections to the internet are discussed. The second part deals with modern techniques for data compression, coding and security problems together with the discussion of pattern recognition using neural networks. Large electronic databases, techniques for data retrieval, video indexing methods and electronic data interchange are presented. The last part of this course elucidates todays computer hard- and software such as Pentium MMX technology, multimedia PCs , WWW, Internet phone, electronic mail and more. Next, various types of network protocols (ATM, FDDI, Ethernet, TCP/ IP, ...) suitable for multimedia applications are discussed. Finally applications such as teleteaching, teleworking, edutainment (Education and Entertainment), video on demand, world wide web and video conferencing are treated. Information Technology 1 + 2 D. JÄGER AND CO-WORKERS Practical studies for students with emphasis on Information Technology (E3 I/IT) Exp. 1: Optical Transmission Exp. 2: Optical Signal Processing Exp. 3: Optoelectronic Sensors Exp. 4: Optical Neural Signal Processing 4.2 Seminars and Colloquia 95 4.2 Seminars and Colloquia Seminar on Optoelectronics D. JÄGER AND CO-WORKERS M. Groß, Dimensionierung und Entwicklung eines thermooptischen Schalters im polynmeren Materialsystem, Apr. 1996 M. Alles, Tagungsberichte: IPRM 96, 22.04.-25.04., Schwäbisch Gmünd und IPR 96, 29.04.-02.05., Boston, May 1996 H. Slomka, Reinstwassererzeugung für die Optoelektronik, May 1996 R. Hülsewede, Nichtlineare Leitungsstrukturen zur Frequenzerzeugung und Frequenzvervielfachung, May 1996 R. Buß, ZEMAX: Ein Softwarepaket zur Simulation optischer Systeme, May. 1996 M. Engel, 2D-Simulation von INT-HEMT für OEIC, Jun. 1996 M. Wenning, Entwicklung einer Meßtechnik zur Bestimmung der Physikalischen und fotometrischen Eigenschaften von LED-basierten FullColor-Displays, Jun. 1996 S. Redlich, Nichtlineare Vielschichtheterostrukturen für die Microwellenphotonik, Jun. 1996 R. Hedtke, Demoskopische Untersuchung und Beurteilung der Leserlichkeit von LED-basierten Anzeigesystemen, Jun. 1996 B. Neuhaus, Aufbau einer Meßtechnik zur Charakterisierung und Optimierung der UV-Ausbeute von Blitzlampen für den Einsatz in der Druckindustrie, Jul. 1996 D. Kalinowski, Entwicklung eines Feldsondenmeßplatzes zur 2-dimensionalen Analyse elektromagnetischer Feldverteilungen auf nichtlinearen Leitungen, Jul. 1996 V. Wendrix, Herstellung und Charakterisierung eines Wanderwellen-Photodetektors, Jul. 1996 A. Kreuder, Ankopplung von Sende- und Emfpangsmodulen an eine faseroptische Übertragungsstrecke, Oct. 1996 S. Redlich, Ladungsträgertransport über Heterobarrieren - Simulationsmethoden, Oct. 1996 M. Groß, Stand des Projekts EPI-RET, Oct. 1996 A. Lüddecke, Simulation der Millimeterwellengeneration eines Wanderwellenphotodetektors, Nov. 1996 J. Evens, Aufbau einer Flip-Chip Apparatur zur Verbindung integrierter Schaltungen, Nov. 1996 P. Karioja, Overview on activities at VTT, Nov. 1996 96 V. Mezentsev, Nonlinear Problems Related To The Modern Optical Communication, Nov. 1996 M. Meininger, Entwicklung photovoltaischer Zellen zur Energieversorgung einer künstlichen Sehprothese, Dec. 1996 I. Ryjenkova, Millimeterwave propagation in nonlinear transmission lines, Dec. 1996 O. Berger, Bestimmung des HF-Ersatzschaltbildes von Photodetektoren mit Hilfe der Netzwerkanalyse, Jan. 1997 4 TEACHING ACTIVITIES T. Braasch, Bericht über die Messe Laser 1997, Jul. 1997 J. Ervens, Experimentelle Untersuchungen zum Ladungsträgertransport über eine GaAs/ AlGaAs-Barriere, Oct. 1997 R. Heinzelmann, Bericht über die OFS97 in Williamsburg, Nov. 1997 R. Hedtke,Entwicklung einer optischen Energie- und Signalübertragungsstrecke, Nov. 1997 T. Baumeister, Systemanalyse des optoelektronischen Energie- und Signalübertragungssystems im Projekt EPI-RET, Jan. 1997 C. Kampermann, Implementierung eines analytischen Modells zur Simulation der optischen Eigenschaften nichtlinearer Halbleiter-Heterostrukturen, Nov. 1997 A. Kreuder, Untersuchung der dynamischen Eigenschaften von nichtlinearen Vielschichtheterostrukturen, Jan. 1997 B. Ponellis, Simulation des optischen Konversionswirkungsgrades von WanderwellenPhotodetektoren, Dec. 1997 M. Groß, Stand des Projekts EPI-RET, Feb. 1997 O. Lotz, Entwicklung einer Seelaterne in LED-Technik in den Farben Rot und Grün, Apr. 1997 R.S. Johnson, Silicon Motherboards for Fibre-Chip Coupling, Apr. 1997 M. Schmidt, Elektronische Eigenschaften von Bor, May 1997 I. Ryjenkova, Nichtlineare Leitungen für das Millimeterwellengebiet, Jun. 1997 4.2 Seminars and Colloquia 97 Colloquium on Optoelectronics D. JÄGER AND LECTURERS WITH EMPHASIS ON OPTOELECTRONICS Prof. Dr. H.G. Schuster, Universität Kiel, Komplexe Adative Systeme, Jan. 1996 Prof. Dr. M. Dragoman, Time-frequency characterization of optical pulses, Oct. 1996 Dipl.-Phys. G. David, Universität Duisburg, Elektrooptische Feldverteilungsmessungen zur Höchstfequenz-Charakterisierung von monolithisch integrierten Mikrowellenschaltungen, Feb. 1996 Dipl.-Ing. R. Buß, Fachgebiet Optoelektronik, Duisburg, Optoelektronik in der Neurotechnologie, Oct. 1996 Dr. A.L. Ivanov, Universität Frankfurt, Switching Kinetics of a Low-Intensity ElectroOptical Element due to Intrinsic Photoconductivity, May 1996 Dr. J.-Uwe Meyer, Fraunhofer-Institut St. Ingberg, Mikrotechnologien zur Kontaktierung von biologischen Zellen und Geweben, May 1996 Dipl.-Ing. R. Heinzelmann, Universität Duisburg, Elektrooptische Wellenleitermodulatoren für optische Übertragungssysteme, May 1996 Dipl.-Ing. S. van Waasen, Universität Duisburg, 20 Gb/s Wellenleiter-pin/Wanderwellenverstärker OEIC: Jüngste Ergebnisse, Jun. 1996 Ass. Prof. Dr. A. Driessen, Univ. of Enschede, Netherlands, Advanced Micro-Systems for Optical Networks (AMON), Jun. 1996 Dr. N. Vodjdani, THOMSON CSF, Orsay Cedex, France, Integrated Optoelectronics for Optical Microwave Links and Optical Communications, Oct. 1996 Dr.-Ing. M. Martin, Hahn-Meitner-Institut, Berlin, Entwicklung von GHz Komponenten am Hahn-Meitner-Institut, Nov. 1996 Dipl.-Ing. M. Alles,Fachgebiet Optoelektronik, Duisburg, 60 GHz Wanderwellen-Photodetektoren für optische Millimeterwellenverbindungen, Dec. 1996 Dipl.-Phys. T. Braasch, Fachgebiet Optoelektronik, Duisburg, Elektrooptisches Testen zur on-wafer-Charakterisierung von MMICs , Jan. 1997 Dipl.-Ing. A. Brennemann, Fachgebiet Halbleitertechnik/-technologie, Duisburg, Neuartige Photoreceiver auf Basis einer Kombination von pin-Diode und Permeable Junction Base Transistor (PJBT), Jun. 1997 Prof. Dr. W. Sohler, Universität Paderborn, Integrierte Optik in LiNbO3: neue Entwicklungen, Jun. 1997 Prof. Dr.-Ing. R. Schwarte, Universität Siegen, Neuartiges optisches 3D-Meßsystem für die schnelle Formerfassung, Jul. 1997 98 4 TEACHING ACTIVITIES 4.2 Seminars and Colloquia 99 100 4 TEACHING ACTIVITIES 4.3 Doctoral, Diploma, and Graduate theses Doctoral theses Diploma theses Gerhard David, Höchstfrequenz-Charakterisierung von monolithisch integrierten Mikrowellenbauelementen und -schaltungen durch zweidimensionale elektrooptische Feldverteilungsmessungen Ludger Brings, Implementierung eines rechnergestützten Syntheseverfahrens zur Realisierung monolithisch integerierter periodischer Hochfrequenzleitungen Steffen Knigge, Nichtlineare Optische Eigenschaften von Vielschichtheterostrukturen Andreas Stöhr, Entwicklung und Realisierung elektrooptischer Wellenleiter-Schalter für photonische Systeme im Wellenlängenbereich um 1 µm Ralf Kremer, Optisch gesteuerte Koplanarleitungen als III-V-Halbleiter-Bauelemente für die Mikrowellen-Signalverarbeitung Stefan Zumkley, Vertikale elektrooptische Modulatoren für optische Verbindungstechnik im Gbit/s-Bereich Peter Bussek, Quantitative Auswertung von zweidimensionalen elektrooptischen Feldverteilungsmessungen zur Charakterisierung von monolithisch integrierten Mikrowellenschaltungen Thomas Alder, Herstellung und Charakterisierung von Wellenleitermodulatoren für den Wellenlängenbereich um 1,3 µm Michael Wenning, Entwicklung einer Meßtechnik zur Bestimmung der physikalischen und fotometrischen Eigenschaften von LEDbasierten Full-Color-Displays Thomas Engel, 2D-Simulation von InPHEMTs für Verstärker in Empfänger-OEICs Dirk Kalinowski, Entwicklung eines Feldsondenmeßplatzes zur zweidimensionalen Analyse elektromagnetischer Feldverteilungen auf nichtlinearen Leitungen Thomas Baumeister, Systemanalyse des optischen Energie- und Signalübertragungsmoduls für eine künstliche Sehprothese Andreas Kreuder, Untersuchung der dynamischen Eigenschaften nichtlinearer Vielschichtheterostrukturen 4.3 Doctoral, Diploma, and Gratuate theses 101 Graduate theses Mark Meininger, Entwicklung photovoltaischer Zellen zur Energieversorgung einer künstlichen Sehprothese (Retina Implant) Michael Heinsdorf, Herstellung und Charakterisierung von Wanderwellen-Photodetektoren auf InP-Substrat Oliver Lotz, Entwicklung einer Seelaterne in LED-Technik in den Farben Rot und Grün Ralph Hedtke, Demoskopische Untersuchungen zur Beurteilung der Leserlichkeit von LED-basierten Anzeigesystemen Jutta Ervens, Experimentelle Untersuchungen zum Ladungsträgertransport über eine GaAs/Alx Ga1-xAs-Heterobarriere Uwe Weimann, Entwicklung einer InfrarotDatenübertragungsstrecke auf rotierenden BildText-Systemen Claus Kampermann, Implementierung eines analytischen Modells zur Simulation der optoelektronischen Eigenschaften nichtlinearer Halbleiter-Heterostrukturen Birgit Neuhaus, Aufbau einer Meßtechnik zur Charaktierisierung und Optimierung der UVAusbeute von Blitzlampen für den Einsatz in der Druckindustrie Jutta Ervens, Aufbau einer Flip-Chip-Apparatur zur Verbindung integrierter Schaltungen André Lüdecke, Simulation der Millimeterwellengeneration eines Wanderwellen-Photodetektors Oliver Berger, Bestimmung des HF-Ersatzschaltbildes von Photodetektoren mit Hilfe der Netzwerkanalyse Bernd Ponellis, Simulation des optoelektronischen Konversionswirkungsgrades von Wanderwellen-Photodetektoren 102 5 PUBLICATIONS AND PRESENTATIONS 103 5 Publications and presentations [1] G. David, R. Tempel, I. Wolff, and D. Jäger, Analysis of microwave propagation effects using 2D electro-optic field mapping techniques, Optical and Quantum Electronics, Special Issue on Optical Probing of Ultrafast Devices and Integrated Circuits, 1996, pp. 919 - 931 [2] G. David, P. Bussek, and D. Jäger, High resolution electro-optic measurements of 2D field distributions inside MMIC devices, Proceedings of CLEO 96, Anaheim, USA, 1996, pp. 450-451 [3] G. David, R. Tempel, I. Wolff, and D. Jäger, In-circuit electro-optic field mapping for function test and characterization of MMICs, 1996 IEEE MTT-S Int. Microwave Symp., June 17-24, San Francisco, USA, 1996, pp. 1533-1536 [4] [5] R. Kremer, S. Redlich, L. Brings, and D. Jäger, A novel type of constant impedance travelling wave phase shifter for InP- based MMICs, 1996 IEEE MTT-S Int. Microwave Symp., June 17-24, San Francisco, USA, 1996 M. Alles, Th. Braasch, and D. Jäger, Highspeed coplanar Schottky travelling-wave photodetectors, Int. Conf. on Integrated Photonics Research, Conference Proceedings pp. 380-383, Boston, USA 1996 [6] G. David and D. Jäger, Analysis of in-circuit electro-optic measurements of MMICs, XXVth General Assembly of the URSI, August 28 -September 5, 1996, Lille, France [7] M. Alles, Th. Braasch, R. Heinzelmann, A. Stöhr, and D. Jäger, Optoelectronic devices for microwave and millimeterwave optical links, 11th Int. MIKON 96, Conference Proceedings, Workshop Optoelectronics in Microwave Technology, pp. 68 - 76, Warsaw, 27-30 May 1996 (invited paper) [8] M. Alles, R. Heinzelmann, R. Hülsewede, R. Kremer, S. Redlich, A. Stöhr, and D. Jäger, Wave propagation in planar structures for travelling wave semiconductor devices, Progress In Elektromagnetic Research Symposium PIERS 96, Conference Proceedings, p. 136, July 1996, Innsbruck, Austria (invited paper) [9] P. Berini, A. Stöhr, K. Wu, D. Jäger, Normal Mode Analysis and Characterization of an InGaAs/GaAs MQW Field-Induced Optical Waveguide Including Electrode Effects, IEEE/OSA J. Lightwave Technol., Vol. 14, No. 10, pp. 2422 - 2435, October 1996 104 [10] D. Jäger, M. Alles, T. Braasch, R. Heinzelmann, and A. Stöhr, Integration Technology for Microwave Photonic Devices, Interaction technology of Microwaves and Light-Waves-Systems and Devices, XXVth General Assembly of the URSI, August 28 -September 5, 1996, Lille, France (invited paper) [11] D. Jäger, R. Hülsewede, I.V. Ryjenkova, V.K. Metzentsev, S. L. Musher, Microwave Propagation on Nonlinear Transmission Lines, XXVth General Assembly of the URSI, August 28 -September 5, 1996, Lille, France [12] D. Jäger, V.K. Metzentsev, I.V. Ryjenkova, S. K. Turitsyn and R. Hülsewede, Microwave Propagation on Nonlinear Transmission Lines, Proceedings of PIERS96, Hong Kong [13] M. Alles, T. Braasch, and D. Jäger, Travelling Wave Photodetector for Optical Generation of Microwave Signals, Indium Phosphide and Related Materials IPRM 96, Proc. Part II, pp. 30 - 31, Schwäbisch Gmünd, 1996 (post deadline paper) [14] R. Hülsewede, V.K. Mezentsev, S.L. Musher, I.V. Ryjenkova, S.K. Turitsyn, and D. Jäger, Travelling wave generation of millimeter waves in bi-modal NLTLs, 26th European Microwave Conference EuMC 96, Prague 5 PUBLICATIONS AND PRESENTATIONS [15] R. Heinzelmann, A. Stöhr, Th. Alder, R. Buß, and D. Jäger, EMC measurements using electrooptic waveguide modulators, International Topical Meeting on Microwave Photonics MWP 96, Conference Proceedings, Technical Digest, December 3-5, 1996, Kyoto, Japan [16] R. Hülsewede, V.K. Mezentsev, S.L. Musher, V. Ryjenkova, S.K. Turitsyn, and D. Jäger, Millimeter Wave Generation on Nonlinear Transmission Lines, 1996 Int. Workshop on Millimeter Waves Digest, 1996, Orvieto, Italy [17] D. Jäger, Optically Controlled Microwave Devices, International Topical Meeting on Microwave Photonics MWP 96 technical digest, December 3-4, 1996, Kyoto, Japan [18] D. Jäger, V.K. Mezentsev, S.L. Musher and I.V. Ryjenkova, Millimeter wave power generation on nonlinear transmission lines, Asia Pacific Microwave Conf. APMC 96, New Delhi, India [19] I. Ryjenkova, V.K. Mezentsev, S.L. Musher, S.K. Turitsyn, R. Hülsewede and D. Jäger, Nonlinear Transmission Lines for Millimeter Wave Applications, INMMC 96, Duisburg [20] Th. Braasch, G. David, R. Hülsewede, U. Auer, F.-J. Tegude and D. Jäger, Propagation of Microwaves in MMICs Studied by Time- and Frequency-Domain Electro-Optic Field Mapping, Spring Topical Meeting Ultrafast Electronics and Optoelectronics, March 17-19, 1997, Lake Tahoe, USA 105 [21] Th. Braasch, G. David, R. Hülsewede, U. Auer, F.-J. Tegude and D. Jäger, Frequency and time domain characterization of nonlinear transmission lines using electro-optic probing techniques, MIOP 97, April 22-24, 1997, Sindelfingen, Germany [27] I. Ryjenkova, V.K. Mezentsev, S.L. Musher, S.K. Turitsyn, R. Hülsewede, and D. Jäger, Millimeter Wave Generation on Nonlinear Transmission Lines, Publication in annales des télécommunications (special issue), Vol. 52, No. 3-4, 1997, pp. 134139 [22] M. Alles, U. Auer, F.-J. Tegude and D. Jäger, Millimeterwave Photodetectors, Microwaves and Optronics, MIOP 97, April 22-24, 1997, Sindelfingen, Germany [28] M. Alles, U. Auer, F.-J. Tegude, and D. Jäger, High-speed Travelling-Wave Photodetectors for Wireless Optical Millimeter Wave Transmission,MWP 97, Sep. 3-5, 1997, Duisburg/Essen, Germany [23] M. Alles, U. Auer, F.-J. Tegude and D. Jäger, High-Speed Travelling-Wave Photodetectors for optical Millimeterwave transmission operating at 1.55 µm, Workshop Mobile Millimeter Communications MMMCom, Dresden, 12-13. Mai 1997 [24] S. Redlich, A. Kreuder, and D. Jäger, Dynamics of nonlinear electro-optical GaAs/ AlAs multilayer-heterostructures, International Conference on Low Dimensional Structures (LSDS) 97, May 19-21, 1997, Lissabon, Portugal [25] A. Stöhr, Heterostructure Semiconductor Photonic Devices and Systems, Euroconference on Advanced Heterostructures, July 1997, Grenoble, France [26] I. Ryjenkova, M. Alles and D. Jäger, Nonlinear travelling wave photodetector for millimeter wave harmonic frequency generation, Journal of Communications Special Issue, Microwave Photonics, Vol. 48, Aug. 1997, pp. 14-17 [29] Th. Braasch, G. David, R. Hülsewede, and D. Jäger, 1D- and 2D-elektro-optic field mapping to study nonlinear effects in NLTLs, MWP 97, Sep. 3-5, 1997, Duisburg/Essen, Germany [30] S. Redlich, and D. Jäger, Nichtlineare Vielschichtheterostrukturen für die Mikrowellen-Photonik, Photonik Symposium, Oct. 7-9, 1997, Schwäbisch Hall, Germany [31] S. Redlich, C. Kampermann, and D. Jäger, Vielschichtheterostrukturen: Neue Materialien für die Mikrowellen-Photonik, Photonik-Symposium, Oct. 8-10, 1997, Würzburg, Germany [32] S. Redlich, C. Kampermann, and D. Jäger, Modeling and simulation of nonlinear hybrid AlGaAs/GaAs Bragg reflectors, 10th III-V Semiconductor Device Simulation Workshop, Oct. 16-17, 1997, Torino, Italy 106 [33] A. Stöhr, R. Heinzelmann, T. Alder, W. Heinrich, T. Becks, D. Kalinowski, M. Schmidt, M. Groß, and D. Jäger, Optically Powered Integrated Optical E-Field Sensor, 12th International Conference on Optical Fiber Sensors, Conference Proceedings, Oct. 1997, Williamsburg, Virginia, USA [34] M. Groß, T. Alder, R. Buß, R. Heinzelmann, M. Meininger, and D. Jäger, Micro Photovoltaic Cell Array for Energy Transmission into the Human Eye, EPVSEC14, 1997, Barcelona, Spain, Vol. 1, pp. 1165 - 1167 [35] M. Alles, U. Auer, F.-J. Tegude, and D. Jäger, High-Speed Travelling-Wave Photodetectors for Optical Generation of Millimeterwaves, APMC 97, Dec. 2-5, 1997, Hongkong, China [36] I. Ryjenkova, D. Jäger, Nonlinear RTD Circuits for High-Speed A/D Conversion, APMC 97, Dec. 2-5, 1997, Hongkong, China 5 PUBLICATIONS AND PRESENTATIONS 107 6 Guide to the Department of Optoelectronics Travel by car - The Department of Optoelectronics, now located in the Center for Solid-State Electronics and Optoelectronics (ZHO), can easily reached by car via various highways: A3 from the north and south, A40 from the east and west. Exit at Duisburg-Kaiserberg or Duisburg-Wedau, see map for details. Travel by plane - From Düsseldorf International Airport take the city-train (S-Bahn) S1 to Duisburg main station (Hauptbahnhof, Hbf). Travel by train - From Duisburg main station (Hauptbahnhof, Hbf) it is a 20 min. walk to the Department of Optoelectronics and the ZHO. You can either go by Taxi or take the bus 933 or 936 to Universität or take the tram 901 to station Zoo/Uni. 108 Notes: