Download High Speed Op-amp Design

High Speed Op-amp Design: Compensation and
Topologies for Two and Three Stage Designs
R. Jacob Baker and Vishal Saxena
Department of Electrical and Computer Engineering
Boise State University
1910 University Dr., MEC 108
Boise, ID 83725
[email protected] and [email protected]
Abstract :
As CMOS technology continues to evolve, the supply voltages are decreasing while at
the same time the transistor threshold voltages are remaining relatively constant. Making
matters worse, the inherent gain available from the nano-CMOS transistors is dropping.
Traditional techniques for achieving high gain by vertically stacking (i.e. cascoding)
transistors becomes less useful in sub-100nm processes. Horizontal cascading (multistage) must be used in order to realize op-amps in low supply voltage processes.
This seminar discusses new design techniques for the realization of multi-stage op-amps.
Both single- and fully-differential op-amps are presented where low power, small
VDD, and high speed are important. The proposed, and experimentally verified, opamps exhibit significant improvements in speed over the traditional op-amp designs
while at the same time having smaller layout area.
Baker/Saxena
Outline
‰Introduction
‰Two-stage Op-amp Compensation
‰Multi-stage Op-amp Design
‰Multi-stage Fully-Differential Op-amps
‰Conclusion
Baker/Saxena
Op-amps and CMOS Scaling
‰ The Operational Amplifier (op-amp) is a fundamental building
block in Mixed Signal design.
9 Employed profusely in data converters, filters, sensors, drivers etc.
‰ Continued scaling in CMOS technology has been challenging
the established paradigms for op-amp design.
‰ With downscaling in channel length (L)
9 Transition frequency increases (more speed).
9 Open-loop gain reduces (lower gains).
9 Supply voltage is scaled down (lower headroom) [1].
Baker/Saxena
CMOS Scaling Trends
‰ VDD is scaling down but VTHN is almost constant.
9 Design headroom is shrinking faster.
‰ Transistor open-loop gain is dropping (~10’s in nano-CMOS)
9 Results in lower op-amp open-loop gain. But we need gain!
‰ Random offsets due to device mismatches.
[3], [4].
Baker/Saxena
Integration of Analog into Nano-CMOS?
‰ Design low-VDD op-amps.
9 Replace vertical stacking (cascoding) by horizontal cascading of gain
stages (see the next slide).
‰ Explore more effective op-amp compensation techniques.
‰ Offset tolerant designs.
‰ Also minimize power and layout area to keep up with the
digital trend.
‰ Better power supply noise rejection (PSRR).
Baker/Saxena
Cascoding vs Cascading in Op-amps
A Telescopic Two-stage Op-amp
A Cascade of low-VDD
Amplifier Blocks.
(Compensation not shown here)
VDD
VDD
VDD
VDD
VDD
VDD
VDD
2
n-1
1
vm
vp
n
vout
CL
VDDmin>4Vovn+Vovp+VTHP with
wide-swing biasing. [1]
Vbiasn
Vbiasn
Vbiasn
Stage 1
Stage 2
Stage (n-1)
Stage n
VDDmin=2Vovn+Vovp+VTHP.
‰ Even if we employ wide-swing biasing for low-voltage designs, three- or
higher stage op-amps will be indispensable in realizing large open-loop DC
gain.
Baker/Saxena
TWO-STAGE OP-AMP COMPENSATION
Baker/Saxena
Direct (or Miller) Compensation
VDD
‰ Compensation capacitor (Cc)
between the output of the gain stages
causes pole-splitting and achieves
dominant pole compensation.
‰ An RHP zero exists at
VDD
VDD
M3
M4
M7
1
220/2
750Ω
vm
M1
M2
vp
iC ff
M6TL
M6BL
iC fb
CC
10pF 2
Unlabeled NMOS are 10/2.
Unlabeled PMOS are 22/2.
M6TR
M8T
M6BR
M8B
CL
30pF
Vbias3
Vbias4
vout
100/2
100/2
9 Due to feed-forward component of
the compensation current (iC).
‰ The second pole is located at
‰ The unity-gain frequency is
x10
™All the op-amps presented have been designed in AMI C5N 0.5μm CMOS process with scale=0.3 μm and Lmin=2.
The op-amps drive a 30pF off-chip load offered by the test-setup.
Baker/Saxena
Drawbacks of Direct (Miller) Compensation
VDD
‰ The RHP zero decreases phase
margin
VDD
VDD
M3
M4
M7
1
9 Requires large CC for
compensation (10pF here for a
30pF load!).
220/2
vm
M1
M2
vp
CC
vout
2
10pF
M6TL
M6BL
Vbias3
M6TR
M8T
Vbias4
M6BR
Unlabeled NMOS are 10/2.
Unlabeled PMOS are 22/2.
Baker/Saxena
CL
30pF
100/2
100/2
M8B
‰ Slow-speed for a given load, CL.
‰ Poor PSRR
9 Supply noise feeds to the output
through CC.
‰ Large layout size.
x10
Indirect Compensation
VDD
‰ The RHP zero can be eliminated by
blocking the feed-forward compensation
current component by using
VDD
VDD
VDD
M3
vm
M4
M1
M9
1
ic
M2
MCG
vp
M7
220/2
Cc
2
A
vout
CL
30pF
M6TL
M6BL
Vbias3
M6TR
M10T
M8T
M10B
M8B
100/2
Vbias4
M6BR
Unlabeled NMOS are 10/2.
Unlabeled PMOS are 22/2.
100/2
x10
An indirect-compensated op-amp
using a common-gate stage.
Baker/Saxena
9 A common gate stage,
9 A voltage buffer,
9 Common gate “embedded” in the
cascode diff-amp, or
9 A current mirror buffer.
‰ Now, the compensation current is fedback from the output to node-1 indirectly
through a low-Z node-A.
‰ Since node-1 is not loaded by CC, this
results in higher unity-gain frequency
(fun).
Indirect Compensation in a Cascoded Op-amp
VDD
VDD
M4T
M3T
A
Vbias2
M3B
ic
M4B
VDD
M7
110/2
1
vm
M1
M2
vp
CC
2
1.5pF
Vbias3
M6TL
M6TR
M8T
M6BR
M8B
Vbias4
M6BL
CC
vout
CL
30pF
vm
vp
vout
CL
50/2
50/2
Unlabeled NMOS are 10/2.
Unlabeled PMOS are 44/2.
Indirect-compensation using
cascoded current mirror load.
Indirect-compensation using
cascoded diff-pair.
‰ Employing the common gate device “embedded” in the cascode structure
for indirect compensation avoids a separate buffer stage.
9 Lower power consumption.
9 Also voltage buffer reduces the swing which is avoided here.
Baker/Saxena
Analytical Modeling of Indirect Compensation
The compensation
current (iC) is indirectly
fed-back to node-1.
Block Diagram
vout
ic ≈
1 sCc + Rc
RC is the resistance
attached to node-A.
Small signal analytical model
Baker/Saxena
Derivation of the Small-Signal Model
Resistance roc is
assumed to be large.
The small-signal model
for a common gate
indirect compensated opamp topology is
approximated to the
simplified model seen in
the last slide.
gmc>>roc-1, RA-1,
CC>>CA
Baker/Saxena
Analytical Results for Indirect Compensation
jω
−ω un
σ
p3
p2
z1
p1
Pole-zero plot
‰ Pole p2 is much farther away from fun.
9 Can use smaller gm2=>less power!
‰ LHP zero improves phase margin.
‰ Much faster op-amp with lower
power and smaller CC.
‰ Better slew rate as CC is smaller.
Baker/Saxena
LHP zero
Indirect Compensation Using Split-Length Devices
‰ As VDD scales down, cascoding is becoming tough. Then how to realize
indirect compensation as we have no low-Z node available?
‰ Solution: Employ split-length devices to create a low-Z node.
9 Creates a pseudo-cascode stack but its really a single device.
‰ In the NMOS case, the lower device is always in triode hence node-A is a
low-Z node. Similarly for the PMOS, node-A is low-Z.
NMOS
Baker/Saxena
PMOS
Split-length 44/4(=22/2)
PMOS layout
Split-Length Current Mirror Load (SLCL) Op-amp
VDD
VDD
VDD
M4T
M3T
ic
A
M3B
220/2
M7T
M4B
220/2
1
vm
M1
M2
M7B
vp
CC
2pF
M6TL
M6BL
Vbias3
M6TR
M8T
M6BR
M8B
Vbias4
vout
2
CL
30pF
50/2
50/2
Unlabeled NMOS are 10/2.
Unlabeled PMOS are 22/2.
‰ The current mirror load devices are
split-length to create low-Z node-A.
‰ Here, fun=20MHz, PM=75° and
ts=60ns.
Baker/Saxena
Frequency Response
ts
Small step-input settling in follower
configuration
SLCL Op-amp Analysis
1
gmp
1
gmp
gm1v s
+
A
-
2
1
gmp
g m1
−
v
gmp s
Cc
Cc
vout
1
id2
−
A
id1
vs
vs
2
2
vout
1
id1
2
2
vs
CL
2
v=0
(a)
(b)
rop
1
Cc
A
2
+
gm1v s
2
+
+
vsgA
v1
R1 C1
-
gmpvsgA
-
−
1
gmp
1
gmp
CA gm2v1
gm 1
v
gmp s
2
+
+
Cc
v1
R1 C1
ic
gm2v1
-
Baker/Saxena
R2
1
gmp
vout
C2
-
ic ≈
vout
1
1
+
sCc gmp
R2
vout
C2
-
1
g m1v s
CL
‰ Here fz1=3.77fun
9 LHP zero appears at a higher
frequency than fun.
Split-Length Diff-Pair (SLDP) Op-amp
VDD
VDD
VDD
M3
M4
M7
110/2
1
vm
M1T
M2T
20/2
20/2
vp
ic
vout
A
20/2
M1B
M6TL
M6BL
2pF
20/2
CC
2
CL
30pF
M2B
Vbias3
Vbias4
M6TR
M8T
M6BR
M8B
50/2
Frequency Response
50/2
Unlabeled NMOS are 10/2.
Unlabeled PMOS are 22/2.
‰ The diff-pair devices are split-length to
create low-Z node-A.
‰ Here, fun=35MHz, PM=62°, ts=75ns.
‰ Better PSRR due to isolation of node-A
from the supply rails.
Baker/Saxena
Small step-input settling in follower
configuration
SLDP Op-amp Analysis
vout
vs
−
vout
vs
2
2
1
gmn
vs
2
1
gmn
gmn v s 1
gmn
4
id 2 = −
Cc
1
ron
A
2
+
1
gmn
+
+
vgs1
gmnvgs1
-
1
gmn
vA
CA
-
gmn v s
R1
C1
R2
gm2v1
4
vs
-
2
1
2
+
gmn v s
2
Cc
v1
R1 C1
ic
gm2v1
ic ≈
Baker/Saxena
R2
1
gmn
+
vout
C2
-
vout
1
1
+
sCc gmn
vout
C2
‰ Here fz1=0.94fun,
9 LHP zero appears slightly before
fun and flattens the magnitude
response.
9 This may degrade the phase
margin.
‰ Not as good as SLCL, but is of
great utility in multi-stage op-amp
design due to higher PSRR.
Test Chip 1: Two-stage Op-amps
Miller
SLCL
Indirect
3-Stage Indirect
SLDP
Indirect
Miller with Rz
‰ AMI C5N 0.5μm CMOS, 1.5mmX1.5mm die size.
Baker/Saxena
Test Results and Performance Comparison
Performance comparison of the op-amps for CL=30pF.
Miller with Rz (ts=250ns)
SLCL Indirect (ts=60ns)
SLDP Indirect (ts=75ns)
Baker/Saxena
‰
‰
‰
‰
10X gain bandwidth (fun).
4X faster settling time.
55% smaller layout area.
40% less power consumption.
MULTI-STAGE OP-AMP DESIGN
Baker/Saxena
Three-Stage Op-amps
‰ Higher gain can be achieved by
cascading three gain stages.
9 ~100dB in 0.5μm CMOS
‰ Results in at least a third order
system
jω
9 3 poles and two zeros.
9 RHP zero(s) degrade the phase
margin.
‰ Hard to compensate and stabilize.
‰ Large power consumption compared
to the two-stage op-amps.
σ
− ω un
Pole-zero plot
Baker/Saxena
Biasing of Multi-Stage Op-amps
‰ Diff-amps should be
employed in inner gain stages
to properly bias second and
third gain stages
Robust Biasing
9 Current in third stage is
precisely set.
9 Robust against large offsets.
9 Boosts the CMRR of the opamp (needed).
‰ Common source second stage
should be avoided.
9 Will work in feedback
configuration but will have
offsets in nano-CMOS
processes.
Fallible Biasing
Baker/Saxena
Conventional Three-Stage Topologies
Nested Miller Compensation (NMC) [6]
‰ Requires p3=2p2=4ωun for stability
(Butterworth response)
9 Huge power consumption
‰ RHP zero appears before the LHP
zero and degrades the phase
margin.
‰ Second stage is non-inverting
9 Implemented using a current
mirror.
9 Excess forward path delay (not
modeled or discussed in the
literature).
Baker/Saxena
Conventional Three-Stage Topologies contd.
Nested Gm-C Compensation (NGCC) [7]
‰ Employs feed-forward gm’s to
eliminate zeros.
9 gmf1=gm1 and gmf2=gm2
‰ Class AB output stage.
‰ Hard to implement gmf1 which
tracks gm1 for large signal swings.
9 Also wasteful of power.
‰ gmf2 is a power device and will not
always be equal to gm2.
9 Compensation breaks down.
‰ Still consumes large power.
Baker/Saxena
Conventional Three-Stage Topologies contd.
Transconductance with Capacitive
Feedback Compensation (TCFC) [14]
‰ Four poles and double LHP zeros
9 One LHP zero z1 cancels the pole p3.
9 Other LHP zero z2 enhances phase
margin.
VDD
VDD
VDD
VB1
VDD
gm2/2
VB7
gmf
VB5
VB2
CC2
VB6
3
gmt
2
VB3
VB4
1:2
Baker/Saxena
gm3
Set p2=2ωun for PM=60°.
Relatively low power.
Still design criterions are complex.
Complicated bias circuit.
9 More power.
CC1
vp
gm1
VDD
VDD
1
vm
‰
‰
‰
‰
vout
‰ Excess forward path delay.
Three-Stage Topologies: Latest in the literature
Reverse Nested Miller with Voltage Buffer
and Resistance (RNMC-VBR) [8]
‰ Employs reverse nesting of
compensation capacitors
9 Since output is only loaded by only
CC2, results in potentially higher fun.
9 Third stage is always non-inverting.
VDD
VDD
VDD
vp
gm1
20uA
3
vout
2
Cc2
CL
gm3
gmVB
1
gm2
Baker/Saxena
VDD
Cc1
RC1
vm
VDD
VDD
gmf
‰ Uses pole-zero cancellation to realize
higher phase margins.
‰ Excess forward path delay.
‰ Biasing not robust against process
variations. How do you control the
current in the output buffer?
Three-Stage Topologies: Latest in the literature contd.
Active Feedback Frequency
Compensation (AFFC) [9]
‰ Reversed nested with elimination of
RHP zero.
High-Gain Block (HGB)
Cm
Input Block
vs
-A1
+A2
1
2
-A3
-gmf
Ca
+gma
High-Speed Block (HSB)
vout
3
9 High gain block (HGB) realizes gain
by cascading stages.
9 High speed block (HSB) implements
compensation at high frequencies.
‰ Complex design criterions.
‰ Excess forward path delay. Again,
uses a non-inverting gain stage.
‰ Employs a complicated bias circuit.
9 More power consumption.
Baker/Saxena
Three-Stage Topologies: Latest in the literature contd.
‰ Various topologies have been recently reported by combining the earlier
techniques.
9 RNMC feed-forward with nulling resistor (RNMCFNR) [17].
9 Reverse active feedback frequency compensation (RAFFC) [17].
‰ Further improvements are required in
9 Eliminating excess forward path delay arising due to the compulsory noninverting stages.
9 Robust biasing against random offsets in nano-CMOS.
9 Further reduction in power and circuit complexity.
9 Better PSRR.
Baker/Saxena
Indirect Compensation in Three-Stage Op-amps
‰ Indirectly feedback the compensation
currents ic1 and ic2.
9 Reversed Nested
‰ Thus named RNIC.
‰ Employ diff-amp stages for robust
biasing and higher CMRR.
‰ Use SLDP for higher PSRR.
‰ Minimum forward path delay.
‰ No compulsion on the polarity of gain
stages.
9 Can realize any permutation of stage
polarities by just changing the sign of
the fed-back compensation current
using ‘fbr’ and ‘fbl’ nodes.
‰ Low-voltage design.
‰ Note Class A (we’ll modify after
theory is discussed).
Baker/Saxena
VDD
VDD
VDD
VDD
VDD
2
-ve
1
vm
fbl
+ve
fbr
ic1
fbl
vp
vout
Cc1
3
ic2
fbr
Cc2
Vbiasn
CL
Indirect Compensation in Three-Stage Op-amps contd.
VDD
VDD
VDD
VDD
VDD
2
-ve
1
vm
fbl
+ve
fbr
ic1
fbl
vp
vout
Cc1
3
ic2
CL
fbr
Cc2
Vbiasn
‰ Note the red arrows showing the node movements and the signs of the
compensation currents.
9 fbr and fbl are the low-Z nodes used for indirect compensation (have
resistances Rc1 and Rc2 attached to them).
‰ The CC’s are connected across two-nodes which move in opposite direction
for overall negative feedback the compensation loops.
‰ Note feedback and forward delays!
Baker/Saxena
Analysis of the Indirect Compensated 3-Stage Op-amp
‰ Plug in the indirect compensation model developed for the two-stage opamps.
i c1 ≈
v2
1 sCc1 + Rc 1
ic 2 ≈
Two LHP zeros
Baker/Saxena
vout
1 sCc 2 + Rc 2
Four non-dominant poles.
Pole-zero Cancellation
‰ Poles p4,5 are parasitic conjugated poles located far away in frequency.
9 Appear due to the loading of the nodes fbr and fbl.
‰ The small signal transfer function can be written as
‰ The quadratic expression in the denominator describing the poles p2 and p3
can be canceled by the numerator which describes the LHP zeros.
9 Results in LHP zeros z1 and z2 canceling the poles p2 and p3 resp.
‰ The resulting expression looks like a single pole system for low
frequencies. →Phase margin close to 90°.
Baker/Saxena
Pole-zero Cancellation contd.
jω
‰ Place pole-zero doublets (p2-z1 and p3z2) out of fun for clean transients.
9 i.e. fp2, fp3 > fun.
‰ Best possible pole-zero arrangement
for low power design.
‰ Results into design equations
independent of parasitics (C3≈CL
here).
‰ Rc1 and Rc2 are realized by adding poly
R’s in series with CC1 and CC2.
9 Also Rc1, Rc2≥Rc0, the impedance
attached to the low-Z nodes fbr/fbl.
‰ Robust against even 50% process
variations in R’s and C’s as long as the
pole-zero doublets stay out of fun.
Baker/Saxena
σ
−ω un
Design Equations
Pole-zero cancelled Class-A Op-amp
‰ A Here, the poly resistors are estimated as
‰ Low power, simple, robust and manufacturable topology*.
™ The presented three-stage op-amps have been designed with transient and SR performances to
be comparable to their two-stage counterparts.
Baker/Saxena
Pole-zero cancelled Class-AB Op-amp 1
‰ A dual-gain path, low-power Class-AB op-amp topology (RNIC-1).
‰ The design equation for Rc1 is modified as
Baker/Saxena
Pole-zero cancelled Class-AB Op-amp 2
VDD
M4
VDD
VDD
M8
M5
Vbiasp
M9
VDD
M10
1
M1T
20/2
fbl
vm
VDD
VDD
20/2
M2T
vp
fbr
20/2
20/2
M1B
M2B
220/2
66/2
M5
fbl
R1c
7.65K 1p
Vbiasn
M3L
Vpcas
66/2
M6
Cc1
Vncas
MFCN
MFCP
11/2
10/2
2
fbr
R2c
Cc2
4.08K 2p
M11
Vbiasn
M3R
M7L
M7R
Unlabeled NMOS are 10/2.
Unlabeled PMOS are 22/2.
‰ A single-gain path, Class-AB op-amp topology for good THD
performance.
9 Floating current source for biasing the output buffer.
9 Here, Vncas= 2VGS and Vpcas=VDD − 2VSG.
9 Note the lack of gratuitous forward delay.
Baker/Saxena
vout
3
CL
100/2
30pF
Simulation of Three-stage Op-amps
Bode Diagram
Magnitude (dB)
100
50
0
-50
-100
Phase (deg)
-150
0
-45
-90
-135
-180
-225
2
10
4
10
6
10
8
10
10
10
Frequency (Hz)
‰ Analytical model of the Class-AB (RNIC-1) topology is simulated in
MATLAB.
‰ The pole-zero plot illustrates the double pole-zero cancelation (collocation).
9 p4 and p5 are parasitic poles located at frequencies close to that of the fT limited
(or mirror) poles.
‰ Here, fun≈30MHz and PM=90° for CL=30pF.
Baker/Saxena
Simulation of Three-stage Op-amps contd.
‰ SPICE simulation of the same Class AB op-amp.
‰ CL=30pF: fun=30MHz, PM≈88°, ts=70ns, 0.84mW, SR=20V/μs.
‰ As fast as a two-stage op-amp with only 20% more power, at 50% VDD and with
the same layout area (simpler bias circuit).
‰ Operates at VDD as low as 1.25V in a 5V process (25% of VDD).
‰ SPICE simulation match with the MATLAB simulation
9 Our theory for three-stage indirect compensation is validated.
Baker/Saxena
Chip 2: Low-VDD 3-Stage Op-amps
Best
Performance
‰AMI C5N 0.5μm CMOS, 1.5mmX1.5mm die size.
Baker/Saxena
Performance Comparison
‰ Figures of Merit
9
9
9
9
FoMS=funCL/Power
FoML=SR.CL/Power
IFoMS=funCL/IDD
IFoML=SR.CL/IDD
‰ RNIC op-amp designed
for 500pF load for a fair
comparison.
‰ FoMs>2X than state-ofthe-art at VDD=3V.
‰ Comparable performance
even at lower VDD=2V.
‰ Practical, stable and
production worthy.
Baker/Saxena
Performance Comparison contd.
60000
50000
40000
FOM_S
FOM_L
30000
IFOM_S
20000
IFOM_L
10000
C
M C
C
FN
R
D
FC
FC
AF
F
AC C
B
C
F
TC
FC
R DP
N
M ZC
C
VB F
SM NR
R FF
N
M C
C
F
R NR
R
A
N
F
IC RA F
C
R - 2 ( FF
N
IC Thi C L
R
N -3 s w P
IC (T
o
R -2A hi s rk )
N
IC (T wo
-3 hi rk
A sw )
(T
hi ork
s
)
w
or
k)
G
N
N
M
N
M
C
0
Baker/Saxena
‰ Higher performance figures
than state-of-the-art.
‰ 10X faster settling.
‰ Better phase margins.
‰ Layout area same or smaller.
Flowchart for RNIC Op-amp Design
Start with the initial
specifications on fun, CL,
Av, and SR.
Split DC gain AOLDC
across A1, A2 and A3.
Select the overdrive (% of
VDD) which will set VGS, fT
and transistor gain gm*ro.
Identify gm1. Can
initially set gm2 equal to
gm1 or a to lower value.
Select Cc2 = gm1/fun
Select Cc1 and gm3 such that
the p2-z1 and p3-z2 doublet
locations are outside fun.
Move the corresponding pi-zj
doublet to a lower frequency
by changing Cci and Rci. May
have to sacrifice fun.
Calculate R1c and R2c.
Yes
Is either of R1c
and R2c negative?
Are the parasitic
poles p4,5 degrading
PM by closing on
fun?
No
No
Simulate the design for
frequency response
and transient settling.
No
More
Speed?
Yes
Yes
Increase gm2.
Increase gm1 or
decrease Cc2.
No
Does the design
meet the
specifications?
Lower
power?
Yes
Decrease gm3 or gm2.
In the worst case
scenario decrease
gm1.
No
Yes
Smaller
layout
area?
End
Yes
Reduce Cc1, Cc2 or
gm3.
No
Nothing works!
Revisit biasing.
Baker/Saxena
No
Better SR?
Yes
Increase bias current in the
first stage (i.e. ISS1) or use
smaller CC’s
N-Stage Indirect Compensation Theory
‰ The three-stage indirect compensation theory has been extended to Nstages and the closed form small signal transfer function is obtained.
Cc
n−1
Cc
n−2
Cc
2
Cc
ic
1
n−1
ic
n−2
ic
2
ic
1
Baker/Saxena
MULTI-STAGE FULLY-DIFFERENTIAL
OP-AMPS
Baker/Saxena
Fully Differential Op-amps
‰ Analog signal processing uses ‘only’
fully differential (FD) circuits.
9 Cancels switch non-linearities and even
order harmonics.
9 Double the dynamic range.
‰ Needs additional circuitry to maintain
the output common-mode level.
9 Common-mode feedback circuit
(CMFB) is employed.
vop-vom=-A(vinp-vinm)
Baker/Saxena
Three-Stage FD Op-amp Design: Problems
Block Diagram
Circuit Implementation
VDD
VDD
vom1
vop1
20/2
VCM
fbl
20/2
20/2
fbr
20/2
20/2
20/2
vp vop1
vom Cc2 R2c
vop2
vom1
Vbiasp
Second Stage
VDD
vop2
R2c Cc2
fbr
50/2
vop
50/2
VCM
VCMFB3
vop
110/2
VDD
fbl
100/2
R1c Cc1
100/2
100f
First Stage
VDD
fbl
VDD
Vbiasn
vom2
100/2
fbr
VCMFB1
110/2
VDD
VCM
VCMFB1
100f
VDD
vom2 Cc1 R1c
30K
Vbiasn
VDD
30K
vm
VDD
VDD
vom
100/2
Unlabeled NMOS are 10/2.
Unlabeled PMOS are 22/2.
Output Buffer
(Third Stage)
‰ The CMFB loop disturbs the DC biasing of the intermediate gain stages.
9 Degrades the gain, performance and may cause instability.
Baker/Saxena
Three-Stage FD Op-amp Design: Solutions
Cc2
1.
‰ Employ CMFB
1. Individually across all the stages.
2. Only across the last two stages as the
biasing of the output buffer need not be
precise.
3. Only in the third stage (output buffer).
Cc1
+
vp
-A1
1p
-
2p
+A2
vbiasp
vm
-A1
-
3p
vop
vCM
vbiasn
+A2
1m
+
-A3
-A3
2m
3m
vom
Cc1
Cc2
Cc2
2.
vp
Cc1
+
-A1
-
1p
2p
+A2
-A3
vCMFB
vm
-A1
+
-
1m
3p
vop
vp
Cc1
+
-A1
-
1p
2p
+A2
+A2
2m
-A3
3m
vom
vm
-A1
+
-
1m
+A2
2m
Cc1
Cc2
-A3
vCMFB
vCM
Cc1
Baker/Saxena
Cc2
3.
Cc2
-A3
3p
vop
vCM
3m
vom
Three-Stage FD Op-amp Design
Circuit Implementation
Block Diagram
Second Gain Stage
VDD
VDD
VDD
vom2 Cc1 R1c
VDD
VDD
VDD
vom1
VDD
vop1
fbr
fbl
vm
20/2
VCM
fbl
20/2
20/2
fbr
20/2
20/2
20/2
Vbiasn
R1c Cc1
vop2
vp
Vbiasn
Vbiasn
First Gain Stage
VDD
vom2
110/2
VDD
R2c Cc2
fbl
vop
vop
30K
fbr
50/2
50/2
VCM
VCMFB3
VCM
100/2
100/2
VCMFB3
100f
100/2
100/2
110/2
VDD
Cc2 R2c
30K
vom
Unlabeled NMOS are 10/2.
Unlabeled PMOS are 22/2.
vop2
100f
VDD
vom
Output Buffer
(Third Stage)
‰ Use CMFB only in the output (third) stage. → Manufacturable design.
9 Leaves the biasing of second and third stage alone without disturbing them.
‰ Employ diff-amp pairs in the second stage for robust biasing.
Baker/Saxena
Baker/Saxena
30K
100f
30K
100f
Three-Stage FD Op-amp: Enlarged
Chip 3: Low-VDD FD Op-amps
Best
Performance
‰AMI C5N 0.5μm CMOS, 1.5mmX1.5mm die size.
Baker/Saxena
Simulation and Performance Comparison
DC behavior
Transient response
82dB gain
‰ >2.5X figure of merit (FoM).
Baker/Saxena
ts=275ns
Flowchart for Three-Stage FD Op-amp Design
Start with the initial
specifications on fun, CL,
Av, and SR.
Design a singly-ended pole-zero cancelled
three-stage op-amp for the given
specifications.
Convert the singly-ended op-amp into a fully
differential one by mirroring it. Use a pair of
diff-pairs for the second stage for robust
biasing.
Add a CMFB circuit in the output buffer.
No
Update the value of gm3 corresponding to the
output buffer and recalculate R1C and R2C.
Simulate the design.
Does the design meet
specifications?
Yes
End
Baker/Saxena
Conclusions
‰ Indirect compensation leads to significantly faster, lower power op-amps
with smaller layout area.
‰ Indirect compensation using split-length devices facilitates low-VDD opamp design.
‰ Novel pole-zero canceled three-stage RNIC op-amps exhibit substantial
improvement over the state-of-the-art.
‰ A theory for multi-stage op-amps is presented.
‰ New methodologies for designing multi-stage FD op-amps proposed which
improve the state-of-the-art.
‰ All proposed op-amps are low voltage
9 Open new avenues for low-VDD mixed signal system design.
Baker/Saxena
Future Scope
‰ Mathematical optimization of PZC op-amps.
‰ Design of low-VDD systems in nano-CMOS process
9 Pipelined and Delta-Sigma data converters,
9 Analog filters,
9 Audio drivers, etc.
‰ Further investigation into indirect-compensated op-amps for n≥4 stages.
Baker/Saxena
References
[1]
[2]
[3]
[4]
[5]
[6]
[7]
[8]
[9]
[10]
[11]
[12]
Baker, R.J., “CMOS: Circuit Design, Layout, and Simulation,” 2nd Ed., Wiley Interscience, 2005.
Saxena, V., “Indirect Compensation Techniques for Multi-Stage Operational Amplifiers,” M.S. Thesis, ECE Dept., Boise
State University, Oct 2007.
The International Technology Roadmap for Semiconductors (ITRS), 2006 [Online]. Available:
http://www.itrs.net/Links/2006Update/2006UpdateFinal.htm
Zhao, W., Cao, Yu, "New Generation of Predictive Technology Model for sub-45nm Design Exploration" [Online].
Available: http://www.eas.asu.edu/~ptm/
Slide courtesy: bwrc.eecs.berkeley.edu/People/Faculty/jan/presentations/ASPDACJanuary05.pdf
Leung, K.N., Mok, P.K.T., "Analysis of Multistage Amplifier-Frequency Compensation," IEEE Transactions on Circuits
and Systems I, Fundamental Theory and Applications, vol. 48, no. 9, Sep 2001.
You, F., Embabi, S.H.K., Sanchez-Sinencio, E., "Multistage Amplifier Topologies with Nested Gm-C Compensation,"
IEEE Journal of Solid State Circuits, vol.32, no.12, Dec 1997.
Grasso, A.D., Marano, D., Palumbo, G., Pennisi, S., "Improved Reversed Nested Miller Frequency Compensation
Technique with Voltage Buffer and Resistor," IEEE Transactions on Circuits and Systems-II, Express Briefs, vol.54, no.5,
May 2007.
Lee, H., Mok, P.K.T., "Advances in Active-Feedback Frequency Compensation With Power Optimization and Transient
Improvement," IEEE Transactions on Circuits and Systems I, Fundamental Theory and Applications, vol.51, no.9, Sep
2004.
Eschauzier, R.G.H., Huijsing, J.H., "A 100-MHz 100-dB operational amplifier with multipath Nested Miller
compensation," IEEE Journal of Solid State Circuits, vol. 27, no. 12, pp. 1709-1716, Dec. 1992.
Leung, K. N., Mok, P. K. T., "Nested Miller compensation in low-power CMOS design," IEEE Transaction on Circuits and
Systems II, Analog and Digital Signal Processing, vol. 48, no. 4, pp. 388-394, Apr. 2001.
Leung, K. N., Mok, P. K. T., Ki, W. H., Sin, J. K. O., "Three-stage large capacitive load amplifier with damping factor
control frequency compensation," IEEE Journal of Solid State Circuits, vol. 35, no. 2, pp. 221-230, Feb. 2000.
Baker/Saxena
References contd.
[13]
[14]
[15]
[16]
[17]
[18]
Peng, X., Sansen, W., "AC boosting compensation scheme for low-power multistage amplifiers," IEEE Journal of Solid
State Circuits, vol. 39, no. 11, pp. 2074-2077, Nov. 2004.
Peng, X., Sansen, W., "Transconductances with capacitances feedback compensation for multistage amplifiers," IEEE
Journal of Solid State Circuits, vol. 40, no. 7, pp. 1515-1520, July 2005.
Ho, K.-P.,Chan, C.-F., Choy, C.-S., Pun, K.-P., "Reverse nested Miller Compensation with voltage buffer and nulling
resistor," IEEE Journal of Solid State Circuits, vol. 38, no. 7, pp. 1735-1738, Oct 2003.
Fan, X., Mishra, C., Sanchez-Sinencio, "Single Miller capacitor frequency compensation technique for low-power
multistage amplifiers," IEEE Journal of Solid State Circuits, vol. 40, no. 3, pp. 584-592, March 2005.
Grasso, A.D., Palumbo, G., Pennisi, S., "Advances in Reversed Nested Miller Compensation," IEEE Transactions on
Circuits and Systems-I, Regular Papers, vol.54, no.7, July 2007.
Shen, Meng-Hung et al., "A 1.2V Fully Differential Amplifier with Buffered Reverse Nested Miller and Feedforward
Compensation," IEEE Asian Solid-State Circuits Conference, 2006, p 171-174.
Baker/Saxena