Download Switching Control Technique of Phase-Shift- Controlled Full

IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 25, NO. 4, APRIL 2010
1001
Switching Control Technique of Phase-ShiftControlled Full-Bridge Converter to Improve
Efficiency Under Light-Load and Standby
Conditions Without Additional
Auxiliary Components
Bo-Yuan Chen, Student Member, IEEE, and Yen-Shin Lai, Senior Member, IEEE
Abstract—The main theme of this paper is to propose a switching method to reduce the loss of phase-shift-controlled full-bridge
converter under light-load and standby conditions. It will be shown
that the efficiency can be improved using the proposed switching
control technique. The presented switching control technique controls the full-bridge converter by pulsewidth-modulated (PWM)
switching mode under light-load condition and PWM switching
with burst mode under standby condition. The transition point
between phase-shift switching method and the proposed method
is investigated and confirmed by experimental results. A fieldprogrammable-gate-array-based digital-controlled experimental
system has been set up. The specifications of the converter include:
input voltage = 400 V, output voltage = 12 V, and output power =
400 W. Experimental results show that the efficiency improvement
can be up to 26% under light-load condition and the standby power
is less than 1 W. These results confirm the effectiveness of the proposed switching control technique.
Index Terms—Full-bridge converter, phase-shift full-bridge converter, pulsewidth modulation (PWM).
COSS
D
Deff
ir (t)
Icir,rm s
ID-Q5,avg
ID-Q6,avg
Iengergy,rm s
IL ,rm s
IO
Ip
Ir 1
Ir 2
NOMENCLATURE
Parasitic capacitance of MOSFET.
Duty of MOSFET control.
Effective duty cycle.
Current of resonant inductor.
RMS value of current for circulating.
Average current of body diode of Q5 .
Average current of body diode of Q6 .
RMS value of current for energy transfer.
RMS value of output inductor current.
Output current.
Peak current of resonant inductor.
Starting current of resonant inductor for
energy transfer.
End of current of resonant inductor for
circulating.
Manuscript received April 25, 2009; revised July 24, 2009. Current version
published April 14, 2010. Recommended for publication by Associate Editor
C. K. K. Tse.
The authors are with the Center for Power Electronics, National Taipei University of Technology, Taipei 106, Taiwan (e-mail: [email protected];
[email protected]).
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TPEL.2009.2033069
Lr
L1
L2
RDS( ON)
NP
NS
RL DCR
RTR,pri
RTR,sec
tDT12
tDT34
TBurst
Paux
Pburst-loss
Pcond-loss, MOSFET
Pcond-loss, diode
Pcond-loss, inductor
Pcond-loss, TR
Psw-loss, diode
Psw-loss, MOSFET
Ptotal-loss
td( ON)
td( OFF)
tf
tr
trr
vDS (t)
VD
VIN
0885-8993/$26.00 © 2010 IEEE
Inductance of resonant inductor.
Inductance of output inductor “1.”
Inductance of output inductor “2.”
Turn-ON resistance of MOSFET.
Turn number of primary winding of transformer.
Turn number of secondary winding of
transformer.
Inductor dc resistance.
primary side dc resistance of transformer.
Secondary side dc resistance of transformer.
Deadtime between Q1 and Q2 switching
control.
Deadtime between Q3 and Q4 switching
control.
Switching period of burst PWM mode
switching control.
Losses of PCB, connector, etc.
Total loss during burst PWM switching
method.
Conduction loss of MOSFET.
Conduction loss of diode.
Conduction loss of output inductor.
Conduction loss of transformer.
Switching loss of diode.
Switching loss of MOSFET.
Total loss of converter.
Turn-ON delay of MOSFET.
Turn-OFF delay of MOSFET.
Falling time of MOSFET switching control.
Rising time of MOSFET switching control.
Reverse recovery time of body diode of
MOSFET.
Voltage across Drain and Source of
MOSFET.
Forward voltage drop of body diode of
MOSFET.
Input voltage.
1002
VO
Vsec
Z0
ω0
IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 25, NO. 4, APRIL 2010
Output voltage.
Transformer secondary voltage.
Characteristic impedance of resonant
circuit.
Angular resonant frequency.
TABLE I
SUMMARY OF CURRENT SOLUTIONS
I. INTRODUCTION
ULL-BRIDGE converters with phase-shift switching control [1]–[7] have been widely employed in high-power
applications. These converters provide zero-voltage switching
(ZVS) for the power devices of full bridge. The power devices,
MOSFET, in general, can be turned on when the voltage across
these devices are nearly zero. The switching loss can, therefore,
be significantly reduced, especially for high-power applications.
The increase of resonant inductance becomes relevant under
light-load condition or/and high dc voltage in order to achieve
ZVS. The increase of resonant inductance results in more duty
cycle loss on the secondary side and significant voltage ringing
across the output rectifiers on the secondary side.
Several methods [8]–[11] have been proposed to extend ZVS
range to improve the efficiency of the phase-shift-controlled
full-bridge converters. All these topologies require auxiliary
components to increase the converter efficiency in a wide load
current range. More details of the pros and cons are summarized
in Table I and explained as follows.
1) An additional low-power auxiliary transformer is connected in series between resonant inductor and transformer, and a capacitor connected with auxiliary transformer secondary side, as shown in [8]. The resonant inductor is magnetized by the auxiliary circuit to perform
ZVS under light-load condition. However, this additional
energy will cause more circulating current that results in
more conduction loss.
2) Auxiliary inductors and capacitors are used to achieve
full-range ZVS by keeping circulating current constant,
as presented in [9]. Similarly, conduction losses become
significant due to contribution by the circulating current.
3) A small inductor and large capacitor are added to generate
a constant voltage source on secondary side, as proposed
in [10]. Although the circulating current is reduced and the
zero-current switching (ZCS) is achieved by leading leg
of full-bridge converter, this will prevent the leading leg
from being able to achieve ZVS. Moreover, ZCS cannot
be achieved for large duty cycle.
4) Voltage-doubler-type rectifier [11] is proposed. The circulating current can be significantly reduced. However,
it is difficult to achieve ZVS using magnetizing current.
Therefore, deadtime is dramatically increased.
The world is increasingly concerned about not only the total
system efficiency, but also the standby power dissipation caused
by power supplies. For a typical home containing 20 devices
that are constantly drawing standby power, standby power dissipation is responsible for 5%–10% of the total electricity loss.
Therefore, the International Energy Agency (IEA) has proposed
a “1-W plan,” the participating countries seek to lower standby
power less than 1 W in all products by 2010 [12]. Therefore,
F
some techniques, including: 1) pulse skipping [13]; 2) burst
mode [14], [15]; and 3) off-time modulation [16], [17], are applied to the power converters for reducing standby power losses.
Full-bridge converter is operated as a double forward one
in [18] under light-load condition by changing the switching method. The efficiency improvement is up to 11% under light-load condition. The results and control method under
standby conditions are not explored in [18]. Another optimization method to improve the efficiency by tuning magnetizing
inductor and commutation inductor has also been introduced
in [19].
In this paper, full-bridge converter operation is retained under light-load condition while changing the switching control
method to give efficiency improvement up to 26%. The efficiency under light-load and standby conditions can be significantly reduced without any additional auxiliary components.
The proposed switching control technique controls the fullbridge converter by pulsewidth-modulated (PWM) switching
technique under light-load condition and burst PWM switching
method under standby condition. The transition point between
conventional phase-shifted switching method and the proposed
method is investigated and confirmed by experimental results.
Experimental results will show that the efficiency improvement
can be up to 26% under light-load condition and the standby
power is less than 1 W. These results confirm the effectiveness
of the proposed switching control technique.
II. SWITCHING LOSS ANALYSIS OF PHASE-SHIFT-CONTROLLED
FULL-BRIDGE CONVERTER
For the development to follow, the basic theory of the phaseshift-controlled full-bridge converter is briefly introduced. The
analysis of losses is given to highlight the low-efficiency issue
under light-load and standby conditions.
CHEN AND LAI: SWITCHING CONTROL TECHNIQUE OF PHASE-SHIFT-CONTROLLED FULL-BRIDGE CONVERTER
Fig. 1.
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Circuit diagram of full-bridge converter with current doubler.
A. Phase-Shift-Controlled Full-Bridge Converter
Fig. 1 shows the circuit diagram of full-bridge converter with
current-doubler rectifier. The converter is controlled with phaseshift switching method; therefore, the primary-side power devices can achieve ZVS under certain load condition; operation
waveforms are shown in Fig. 2. The current-doubler topology
is widely used for high-output current applications, due to lowcurrent ripple, ease to design output inductor, and because each
inductor conducts only half of the total output current. The operation of the converter can be divided into ten intervals. Since
1–5 and 6–10 intervals are symmetrical, only 1–5 intervals are
described as follows.
1) Interval 1 (t0 –t1 ): As the Q1 drain-to-source voltage
reaches zero, contributed by resonant operation, Q1 is turned
on, and the resonant inductor current ir starts to commutate
with a finite slope
ir (t) = ir (t0 ) +
VIN
(t − t0 ).
Lr
(1)
The voltage across the transformer primary side is VIN and
voltage of secondary side is zero; therefore, no energy is transferred to the secondary side. This interval is known as duty cycle
loss and contributed by the resonant inductor. In this time interval, the current of resonant inductor commutates, thus changing
of current polarity. The effective duty is rewritten as [20]
Deff = D − ∆D.
(2)
And the duty cycle loss is expressed as
Ir 1 + Ir 2
.
∆D =
(VIN /Lr )Ts
Fig. 2.
Operational waveforms of phase-shift-controlled full-bridge converter.
increases and iL 2 decreases. The current can be expressed by
iL 1 (t) = iL 1 (t1 ) +
(NS /NP ) VIN − VO
(t − t1 )
L1
(6)
iL 2 (t) = iL 2 (t1 ) −
VO
(t − t1 ).
L2
(7)
The current on D-Q6 is
(3)
iD-Q6 (t) = iL 1 (t) + iL 2 (t).
(8)
Because of duty cycle loss, currents of inductors on secondary
side are freewheeling through the diodes in this interval. This
interval ends when ir increase to Ir 1 , where Ir 1 is
(NS /NP ) VIN − VO
NS IO
−
Deff Ts . (4)
Ir 1 =
NP
2
2L1
Since inductor L1 is much larger than Lr , the current slope of
ir is dominated by L1 . This mode ends when ir increases to Ip ,
where Ip is
(NS /NP )VIN − VO
NS IO
+
Deff Ts . (9)
IP =
NP
2
2L1
2) Interval 2 (t1 –t2 ): Energy is transferred to output in this
interval and the voltage on secondary side becomes
3) Interval 3 (t2 –t3 ): As Q4 turns OFF, the parasitic capacitors of Q3 and Q4 start to discharge and charge, respectively. In
this interval, diode D-Q5 is OFF and current is flowing through
the transformer. The primary current is
Vsec =
NS
VIN .
NP
(5)
Therefore, the current on D-Q5 decreases to zero and D-Q6
conducts the current of L1 and L2 . The inductor current iL 1
ir (t) =
NS
iL 1 (t).
NP
(10)
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 25, NO. 4, APRIL 2010
As the inductance value of the current doubler is large, current
through inductors of current doubler is regarded as a constant
current source. The leg consisting of Q3 and Q4, which turns
ON after energy transfer, can achieve ZVS-ON easily, since the
secondary inductor current is flowing through the transformer
instead of freewheeling diode, as shown in Fig. 3(a) and (b).
The deadtime required between Q3 and Q4 can, therefore, be
shortened and expressed as
tDT34 =
4COSS VIN
.
Ip
(11)
4) Interval 4 (t3 –t4 ): As the Q3 drain-to-source voltage
reaches zero, Q3 is turned on. In this interval, primary-side current ir circulates through Q1 and Q3, and no energy is transferred
to secondary side. The circulating current can be expressed as
ir (t) = ir (t3 )e−(2R D S ( ON) /L r )(t−t 3 ) .
(12)
The secondary voltage of the transformer is zero. Diodes DQ5 and D-Q6 are turned on, and the inductors L1 and L2 are in
freewheeling status. The inductor current can be expressed as
iL 1 (t) = iL 1 (t3 ) −
VO
(t − t3 )
L1
(13)
iL 2 (t) = iL 2 (t3 ) −
VO
(t − t3 ).
L2
(14)
This interval ends when ir reaches Ir 2 , which can be derived
by the first-order equation for the circuit consisting of Lr and
RDS( ON)
Ir 2 = IP e−(2R D S ( ON) /L r )(1−2D )(T s /2) .
(15)
5) Interval 5 (t4 −t5 ): When current reaches Ir 2 , Q1 turns
OFF, and resonance occurs in the resonant inductor and the COSS
of MOSFET Q1 and Q2. Because the inductors on secondary
side are in freewheeling status, no current on secondary winding
of transformer is reflected to its primary side. As shown in
Fig. 3(c) and (d), the resonant current is
ir (t) = Ir 2 cos ω0 (t − t4 )
(16)
where ω0 is the angular resonance frequency and determined by
1
ω0 = √
.
2Lr COSS
(17)
Since Q1 and Q2 power devices are turned on after circulating,
the required energy for resonance will be contributed by the
resonant inductor that should be larger than the energy stored in
COSS of MOSFET. The condition for resonance is
1
4
2
Lr Ir22 > COSS VIN
.
2
3
Fig. 3. Charging/discharging of the parasitic capacitors of MOSFETs.
(a) Constant current discharging Q3. (b) Constant current charging Q4.
(c) Resonant current discharging Q2. (d) Resonant current charging Q1.
(18)
And deadtime requires a quarter of the resonant period
√
2π 2Lr COSS
.
(19)
tDT12 =
4
The operation of t5 –t10 is similar to t0 –t5 ; therefore, the details
for t5 –t10 are not described.
B. Loss Analysis
Phase-shift-controlled full-bridge converter provides ZVSON for all four power devices on transformer primary side. How-
ever, ZVS-ON may not be achieved at the whole load range. For
example, as shown in (18), under light-load condition and/or
large input voltage, the size of resonant inductor should be large
CHEN AND LAI: SWITCHING CONTROL TECHNIQUE OF PHASE-SHIFT-CONTROLLED FULL-BRIDGE CONVERTER
enough in order to achieve resonance and ZVS-ON. The primary
current and deadtime, as shown in (18) and (19), determine
ZVS-ON. Therefore, under light-load condition, the primary
current is small, and deadtime should be sufficient to achieve
ZVS-ON. However, this will decrease the effective duty and will
increase the switching loss under heavy-load condition. Therefore, deadtime is usually calculated under a certain load condition to reduce duty loss at the cost of no ZVS-ON. As a result,
below a certain load condition, ZVS-ON cannot be achieved.
The power devices will be operated with hard switching. For
the power device Q1, which is turned on after circulation, its
drain-to-source voltage is expressed as
vDS1 (t0 ) = VIN − Z0 ir (t9 )sin(ω0 tDT12 )
equation:
Psw-loss,MOSFET,Q3 =
1005
1
vDS3 (t3 )ir (t3 ) td( ON) + tr
2
1
fs .
+ vDS3 (t8 )ir (t8 ) td( OFF) + tf
2
(27)
The voltages vDS3 (t3 ) and vDS3 (t8 ) can be derived from (11)
and (22), and expressed by
vDS3 (t3 ) = VIN −
(20)
vDS3 (t8 ) =
where Z0 is the characteristic impedance and determined as
Lr
Z0 =
.
(21)
(8/3)COSS
The drain-to-source voltage expression for Q1, when it is
turned on after circulation, is similar to that for Q2. For the
power devices Q3, which is turned on after energy transfer, the
drain-to-source voltage can be described by
vDS3 (t3 ) = VIN −
ir (t2 )tDT34
.
4COSS
(22)
The drain-to-source voltage expression for Q4, when it is
turned on after energy transfer, is similar to that for Q3. The
switching loss before achieving ZVS can be calculated by analyzing the cross area of voltage and current during turn ON/OFF
instant using MOSFET specifications. For the leading-leg power
devices, Q1 and Q2, switching loss before achieving ZVS can
be calculated by the following equation:
1
vDS1 (t0 )ir (t0 ) td( ON) + tr
Psw-loss,M OSFET,Q1 =
2
1
fs .
+ vDS1 (t5 )ir (t5 ) td( OFF) + tf
2
(23)
The voltages vDS1 (t0 ) and vDS1 (t5 ) can be derived from (19)
and (20), and are expressed as follows:
vDS1 (t0 ) = VIN − Z0 Ir 2 sin ω0 tDT12
(24)
vDS1 (t5 ) = Z0 Ir 2 sin ω0 tDT12 .
(25)
The currents ir (t0 ) and ir (t5 ) are the same, and can be calculated from (15), (16), and (19) as
ir (t0 ) = IP e−(2R D S ( ON) /L r )(1−2D )(T s /2) cos ω0 tDT12 .
(26)
For the lagging-leg power devices, Q3 and Q4, switching
loss before achieving ZVS can be calculated by the following
Ip tDT34
4COSS
Ip tDT34
.
4COSS
(28)
(29)
The currents ir (t3 ) and ir (t8 ) are the same, and can be
described by
Ns
ir (t3 ) =
iL 1 (t3 ) ≈ Ip .
(30)
Np
Another major power loss is the conduction loss, which occurs
in energy transfer interval and circulation interval. The power
dissipation on MOSFETs can be calculated from the current rms
value and equivalent resistor on the conduction route. The rms
current value for primary-side energy transfer and circulation
can be described as follows, respectively:
2
T s /2 1
ir (t2 ) − ir (t1 )
t dt
ir (t1 ) +
Ienergy,rm s =
Ts /2 0
DTs
√
1
(31)
= 2D IP I1 + (IP − I1 )2
3
2
T s /2 1
ir (t2 ) − ir (t5 )
Icir,rm s =
t dt
ir (t2 ) +
Ts /2 0
(1 − 2D)Ts /2
√
1
(32)
= 1 − 2D IP I2 + (IP − I2 )2 .
3
The conduction loss and circulation loss for a single MOSFET
on the primary side can be expressed as
2
2
Pcond-loss,MOSFET = Ienergy,rm
(33)
s + Icir,rm s RDS( ON) .
On transformer secondary side, the currents of the rectifier
diode and output inductor are shown in Fig. 4. As shown in
Fig. 4, the average current of rectifier diode and the rms value
of output inductor current can be expressed as
1
(34)
IQ5,avg = IQ6,avg = IO
2

 2
∆i
I
1
O
L
 RL DCR .
IL ,rm s = 
1+
(35)
2
3 IO /2
The conduction loss of diodes and inductors are calculated
by
Pcond-loss,diode = IQ5,avg VD 5 + IQ6,avg VD 6
Pcond-loss,inductor =
IL2 ,rm s RL DCR .
(36)
(37)
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 25, NO. 4, APRIL 2010
transition point, phase-shift switching control method is applied
to achieve higher efficiency. Based upon the proposed switching
control method, the efficiency can be increased up to 26% as
compared to conventional phase switching control method.
A. Proposed Switching Control Technique
Fig. 4.
Current waveform of secondary inductor and rectifier diode.
The switching loss of the diode is
Psw-loss,diode =
1
IO VD trr fsw .
2
The transformer losses can be expressed as
2
2
Pcond-loss,TR = Ienergy,rm
s + Icir,rm s RTR,pri
2
NP
+
Ienergy,rm s RTR, sec .
NS
(38)
(39)
Therefore, from the equations shown before, the total losses
of full-bridge converter are
Ptotal,loss
= Psw-loss,MOSFET,Q1 + Psw-loss,MOSFET,Q2 + Psw-loss,MOSFET,Q3
+ Psw-loss,MOSFET,Q4 + 4Pcond-loss,MOSFET + 2Psw-loss,diode
+ Pcond-loss,diode + 2Pcond-loss,inductor + Pcond-loss,TR .
(40)
From the aforementioned loss analysis for the primary and
secondary side, it can be realized that under standby and lightload conditions, ZVS cannot be achieved, and the switching loss
and circulating loss become significant. Therefore, this paper
presents a switching control technique to improve the standby
and light-load power dissipation.
III. PROPOSED SWITCHING CONTROL TECHNIQUE
In this section, a switching control method for phase-shiftcontrolled full-bridge converter will be presented. The proposed switching control method aims at reducing the loss under
standby and light-load conditions. As the load increases to the
Phase-shift full-bridge converter can perform ZVS operation,
thus high efficiency is obtained. However, under light-load condition, ZVS is hard to achieve, and the duty cycle is small.
Therefore, switching loss and circulating loss are significantly
increased. The proposed method is to eliminate circulating loss
using PWM switching method under standby and light-load conditions. Fig. 5 shows the proposed switching control method.
Table II summarized the proposed switching control technique.
As shown in Table II and Fig. 5, there are three switching control mode: phase-shift switching control mode under heavyload condition, PWM switching control mode under light-load
condition, and PWM switching with burst-mode control under
no-load condition.
The full-bridge is controlled by phase-shift switching control
method under heavy-load condition. The turn-ON period for
each power device is 50% of the switching period. And the
effective duty is determined by the overlapping period between
power devices of Q1 and Q4 (or Q2 and Q3 ), as shown in
Fig. 6(a).
PWM switching control method is applied to control the
full-bridge converter under light-load condition to reduce the
switching and circulating losses. Under light-load condition,
ZVS-ON cannot be achieved unless the size of resonant inductor
is increased. Fig. 6(b) shows the PWM switching method. As
shown in Fig. 6(b), power devices of Q1 and Q4 (or Q2 and Q3 )
are turned on with duty control in each switching period. Although the power devices of full-bridge converter cannot achieve
ZVS-ON using PWM switching mode, the circulating current
is eliminated and switching loss is reduced, as analyzed in
Section III-C later. As compared with PWM switching control, phase-shift switching control results in additional circulating loss. Therefore, changing the phase-shift switching control
mode to PWM switching control mode can significantly reduce
the circulating loss.
As shown in [10] and [11], burst PWM mode is used under
standby condition to further reduce the switching losses. Considering the impact factors contributed by the implementation,
e.g., offset of A/D converter, when the load decreases to less
than 0.2 A, the converter enters standby mode, and the output
voltage is controlled with a small ripple to force the compensator to control the converter with burst PWM mode; the control
waveform and PWM is shown in Fig. 7. In the PWM switching
mode with burst control, as the output voltage increases beyond
+1% of its nominal value, switching stops. In contrast, PWM
switching is resumed as output voltage is below −1% of its
nominal value.
B. Transition Between Proposed Switching Control Modes
Since there are three switching control modes for the proposed method, as shown in Fig. 5 and Table II. There are two
CHEN AND LAI: SWITCHING CONTROL TECHNIQUE OF PHASE-SHIFT-CONTROLLED FULL-BRIDGE CONVERTER
Fig. 5.
1007
Block diagram of the proposed switching control method.
TABLE II
SUMMARY OF PROPOSED SWITCHING CONTROL TECHNIQUE
Fig. 6. Switching control methods. (a) Phase-shift switching. (b) PWM
switching.
Fig. 7.
PWM switching with burst-mode control.
Fig. 8.
Transition points of the proposed switching control technique.
transition points for changing the switching control mode. Fig. 8
shows the transition points of the proposed switching control
technique. As shown in Fig. 8, the transition point of load current between phase-shift switching control and PWM switching
control mode is the load current IO Opt Eff . As the current is
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 25, NO. 4, APRIL 2010
TABLE III
SPECIFICATIONS OF THE POWER STAGE
The switching loss for leading leg and lagging leg are the
same, and can be expressed as
1
Psw-loss,MOSFET =
vDS1 (t1 )ir (t1 ) td( ON) + tr
2
1
+ vDS1 (t2 )ir (t2 ) td( OFF) + tf
fs .
2
(41)
During turn OFF, the resonant inductor energy will be transferred to the MOSFET parasitic capacitor, and therefore, the
expression for vDS1 (t1 ) and vDS1 (t2 ) are different
vDS1 (t1 ) =
Fig. 9.
VIN
+ ir (t2 )
vDS1 (t2 ) =
2
Current and VD S waveforms of PWM switching mode.
greater than this value, phase-shift switching control provides
higher efficiency. And the second transition point between PWM
switching control modes with/without burst control is the load
current IO STN-BY , in which full converter provides 0.6% of full
load.
C. Loss Analysis of Proposed Switching Control Technique
The proposed switching control technique is to improve efficiency under light-load condition by PWM switching and
PWM switching with burst mode. Fig. 9 shows the operation
waveforms for the proposed method under PWM switching
mode. As shown in Fig. 9, the circulating current is eliminated and the drain-to-source voltage during switching is significantly reduced by 50% as compared to that for phaseshift switching mode, as shown in Fig. 2. The drain-tosource voltage reduction is contributed by the special feature
of PWM switching mode. Two power devices are turned on
or turned off simultaneously, as shown in Fig. 9. Therefore,
the voltage stress, and thus, switching loss are significantly
reduced.
VIN
2
(42)
Lr
(8/3) COSS
(43)
where ir (t1 ) and ir (t2 ) can be derived from (4) and (9),
respectively.
The conduction losses of MOSFET occur only in energy
transfer interval, and can be described as
2
Pcond-loss,MOSFET = Ienergy,rm
s RDS( ON) .
(44)
The transformer losses can be expressed as
2
NP
2
Pcond-loss,TR = Ienergy,rm s RTR,pri +
Ienergy,rm s RTR,sec .
NS
(45)
The loss on transformer secondary side is similar to that
for phase-shift switching mode, and can be derived from (34)–
(38). Therefore, the total loss of PWM switching mode can be
described as
Ptotal,loss
= 4Psw-loss,MOSFET + 4Pcond-loss,MOSFET + 2Psw-loss,diode
+ Pcond-loss,diode + 2Pcond-loss,inductor + Pcond-loss,TR .
(46)
During burst PWM mode, the switching frequency is significantly lower than fs , and therefore, the total loss during burst
CHEN AND LAI: SWITCHING CONTROL TECHNIQUE OF PHASE-SHIFT-CONTROLLED FULL-BRIDGE CONVERTER
1009
Fig. 11. Experimental result of PWM and phase-shift switching control efficiency curves.
Fig. 12. Transition points between switching modes of proposed control
technique.
Fig. 13.
Efficiency curve, experimental results.
IV. SIMULATION AND EXPERIMENTAL RESULTS
Fig. 10. Simulation results of efficiency curves. (a) CO S S = 63 pF.
(b) CO S S = 100 pF. (c) CO S S = 140 pF.
PWM mode can be described as
Pburst,loss =
nPtotal,loss
Tburst
(47)
where n is the number of switching of MOSFET in one burst
cycle.
The specifications for experimental system are shown in
Table III. The input voltage is 400 V, output voltage is 12 V,
output current is 33 A, and the switching frequency is 180 kHz.
The experimental system consists of power stage and digital
controller to realize voltage-mode control.
The transition point between PWM switching mode and
phase-shift switching mode is determined by the efficiency of
converter. The losses can be estimated from (40) and (46), and
the efficiency can, therefore, be derived, as shown in Fig. 10,
for the converter specified in Table III. Fig. 10 also reflects the
effect of COSS on the transition point of two switching control
techniques. As shown in Fig. 10, the COSS indeed contributes
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 25, NO. 4, APRIL 2010
TABLE IV
STANDBY POWER DISSIPATION
Fig. 14.
Efficiency curve for 0.08%–0.6% load, experimental results.
Fig. 16. Experimental results, mode transition waveforms, Ch 1 = Q1 gate
signal 5 V/div, Ch 2 = Q4 gate signal 5 V/div, Ch 3 = Output voltage 10 V/div,
and Ch 4 = Output current 10 A/div. (a) PWM switching mode to phase-shift
switching mode. (b) Phase-shift switching mode to PWM switching mode.
Fig. 15. Experimental results, standby-mode transition waveforms, Ch 1 =
Q1 gate signal 5 V/div, Ch 2 = Q4 gate signal 5 V/div, Ch 3 = Output voltage 100 mV/div (with ac coupling), and Ch 4 = Output current 200 mA/div.
(a) Burst mode to PWM switching mode. (b) PWM switching mode to burst
mode.
to the change of resonant point, and thereby affect the transition
point. COSS is specified under a specific VDS and switching frequency [21]. Fig. 11 shows the experimental results of PWM
and phase-shift switching control efficiency curves. As shown
in Figs. 10(c) and 11, the experiemntal results agree with the
experimental results. As shown in Fig. 10, efficiency curves
for phase-shift switching control and PWM switching control
intersects as current is around 19 A as COSS = 140 pF. The
experimental results shown in Fig. 11 reveal that the optimum
transition point is around 18 A. The results derived from analytical and experimental results are quite close. The errors may
be contributed by modeling and parameter variations.
Therefore, as shown in Fig. 12, the optimal transition point is
designed at 18 A. When the load decreases less than 18 A, the
switching mode is changed from phase-shift control to PWM
switching control and a hysteresis band is added to prevent
bouncing between two switching modes. Moreover, in order to
further reduce the switching loss under standby condition, the
CHEN AND LAI: SWITCHING CONTROL TECHNIQUE OF PHASE-SHIFT-CONTROLLED FULL-BRIDGE CONVERTER
1011
mode and PWM switching mode. Fig. 16(a) is the results for the
transition from PWM switching mode to phase-shift switching
mode. Fig. 16(b) is the result for the transition from phase-shift
switching mode to PWM switching mode.
Fig. 17 shows the dynamic response to the load range between
15 and 23 A with current slew rate = 0.3 A/µs. Fig 17(a) is result
for load change from 15 to 23 A. Fig 17(b) is the result for load
change from 23 to 15 A. Both results show the output voltage
has fast dynamic response and ripple is less than ±1%.
These results demonstrate that the proposed switching control
technique can significantly reduce the losses under standby and
light-load conditions, and offer satisfied dynamic performance
even under transition between modes.
V. CONCLUSION
Fig. 17. Experimental results, dynamic response, Ch 3 = Output voltage
10 V/div, and Ch 4 = Output current 10 A/div. (a) Load change 15–23 A.
(b) Load change 23–15 A.
switching frequency for PWM switching mode is reduced by
PWM switching with burst-mode control.
Fig. 13 shows the measured efficiency curve for the proposed
and phase-shift switching modes. It is shown that the proposed
switching technique indeed provides higher efficiency under
light-load conditions as compared to phase-shift method, and
the efficiency improvement can be up to 26%. Another advantage of the proposed method is the standby power dissipation
shown in Table IV is also reduced. The standby power is reduced from 22.4 W for phase-shift switching mode to 0.624 W
for PWM switching with burst mode. And the efficiency curve
under extremely light load for 0.08%–0.6% of load is shown
in Fig. 14. The proposed switching control method obviously
provides higher efficiency as compared to that for phase-shift
switching mode.
The measured results for the transition between burst mode
and PWM switching mode are shown in Fig. 15. As shown in
Fig. 15, the output voltage ripple is less than ±1% during mode
transition. Fig. 16 shows the PWM signals, output voltage, and
output current between the transition of phase-shift switching
A new switching control technique for full-bridge converter
has been presented. The switching control mode and the related
special features for the proposed switching control technique
are summarized as follows.
1) Under heavy-load condition: Using phase-shift switching
control to achieve ZVS to reduce turn-ON loss.
2) Under light-load condition: Using PWM switching control to remove circulating loss.
3) Under standby condition: Using PWM switching with
burst-mode control to reduce both switching loss and circulating loss.
The analytical forms for the losses and transition points between modes for full-bridge converter were presented. For the
experimental system, input voltage = 400 V, output voltage = 12
V, and output power = 400 W; experimental results showed that
the efficiency improvement can be up to 26% under light-load
condition and the standby power is less than 1 W. Experimental
results also demonstrated that the proposed switching control
technique can significantly reduce the losses under standby and
light-load conditions, and offer satisfied dynamic performance
even under transition between modes.
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 25, NO. 4, APRIL 2010
Bo-Yuan Chen (S’09) received the B.S. degree in
electrical engineering in 2006 from the National
Taipei University of Technology, Taipei, Taiwan,
where he is currently working toward the Ph.D. degree in electrical engineering.
His current research interests include design
of power converters, DSP-/field-programmable-gatearray-based implementation of digital controlled
power converters, and design of electronic circuits.
Yen-Shin Lai (M’96–SM’01) received the M.S. degree from the National Taiwan University of Science
and Technology, Taipei, Taiwan, and the Ph.D. degree
from the University of Bristol, Bristol, U.K., both in
electronic engineering.
In 1987, he was as a Lecturer in the Department
of Electrical Engineering, National Taipei University
of Technology, Taipei, where he was the Chairperson
from 2003 to 2006 and has been a Full Professor since
1999, and a Distinguished Professor since 2006. His
research interests include design of control IC, circuit
design of dc/dc converter, and inverter control.
Prof. Lai is a committee member of the IEEE IAS Industrial Drives Committee and the Industrial Power Converter Committee, and an Associate Editor of the IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS and the IEEE
TRANSACTIONS ON INDUSTRY APPLICATIONS. He received several national and
international awards, including the John Hopkinson Premium for the session
1995–1996 from the Institute of Electrical Engineers, the Technical Committee
Prize Paper Award from the IEEE Industrial Application Society (IAS) Industrial
Drives Committee for 2002, the Best Presentation Award from IEEE Industrial
Electronics Society in 2004.