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UPGRADED SWITCH MODE POWER SUPPLIES FOR TRANSPORT
LINE – 1 MAGNETS IN INDUS
Alok Singh#, Manohar Koli, Mangesh Borage, Sunil Tiwari and A.C. Thakurta
Raja Ramanna Centre for Advanced Technology, Indore-452013, India
Abstract
Upgraded dc current controlled power supplies of
ratings 5 A/12 V and 12 A/14 V with specified output
current stability of ±500 ppm and ±200 ppm, respectively
are developed to energize quadrupole and dipole magnets
in Transport Line-1 (TL-1) of INDUS. They are
developed using two switch forward converter topology
operating with variable frequency PWM control
(VFPWM). This paper describes design principle,
proposed VFPWM control, peculiarities in power supply
development, salient features and results of various tests
conducted on the power supplies.
INTRODUCTION
INDUS-1 and INDUS-2 are the synchrotron radiation
sources operational at RRCAT with electron energies of
450 MeV and 2.5 GeV, respectively. The transport line-1
(TL-1), installed in Indus complex, is used to transfer
electron beam from microtron to booster synchrotron. A
dipole magnet and six quadrupole magnets are installed in
TL-1, which are energised by stable current controlled
power supplies. Existing power supplies for quadrupole
magnets in TL-1 are based on series pass scheme and are
in operation for several years. These are being upgraded
with new, compact, efficient and stable power supplies
using two switch forward converter topology operating
with variable frequency PWM control (VFPWM).
DESIGN PRINCIPLE AND FEATURES
Topology
Switch mode power supplies (SMPS) have an edge
over series pass scheme because of its reduced size,
lighter weight and better efficiency. Two-switch forward
converter topology is chosen amongst various SMPS
topologies since it is simple, rugged, tested and proven
[1]. Maximum switching frequency of 100 kHz is chosen
as a good trade-off between size of the power supply and
switching
S1
D1
LO
D3
Cin
D4
CO
Cd
Rd
OUTPUT
+
230 V
50 Hz
D2
Rshunt
S2
Figure 1: Schematic of two-switch forward converter
___________________________________________
#
[email protected]
losses. The basic circuit diagram of two switch forward
converter scheme is shown in Fig. 1. In this dc-dc
converter, IRF840 MOSFET switches S1 and S2 are turned
on and off simultaneously causing energy transfer from
source to load. The power is coupled to output through a
high frequency transformer. MUR470E freewheeling
diodes D1 and D2 resets the transformer and clamps the
voltage across switch to the dc link voltage. MBR2060
dual common cathode schottky diodes D3 and D4
functions as secondary rectifier.
Input filter (Cin) is a capacitor filter while output filter
(LO, CO, Rd and Cd) is a damped low pass filter. A zeranin
shunt is used to sense the output current.
The feedback control scheme consists of two loops: the
inner fast voltage loop that corrects input line variations,
and outer slow current loop that corrects the drift in
analog devices due to temperature. Variable frequency
PWM (VFPWM) scheme has been incorporated in control
circuit as a result of which switching frequency varies
from 20 kHz to 100 kHz simultaneously with the increase
in duty cycle from 0 to 0.5. This is done to increase the
output current setting range. Bandwidth of voltage loop
and current loop is approx. 1 kHz and 10 Hz, respectively.
Local/Remote Operation
Each power supply is capable of being operated from
local fascia panel or in remote mode from central
computer interface via a 25-pin sub-D connector provided
on the fascia plate.
VFPWM CONTROL
Wide conversion range, that is, an ability to set and
regulate the output current over a wide range, is an
important requirement of a magnet power supply. In highfrequency-switching PWM dc-dc converters, the
requirements of fast transient response and wide
conversion range conflict. While higher switching
frequency is favoured for miniaturization of the power
converter and fast transient response, the objective of
wide conversion range is defeated because the operation
at extremely low duty-cycle is limited due to minimum
on-time of the switch. The switching converters, in which
conversion ratio has the quadratic dependence on dutycycle, can greatly extend the conversion range [2]. These
techniques suffer from low conversion efficiency and
higher component ratings. In this paper a new circuit
technique, VFPWM, is suggested to improve the
conversion range. The minimum on-time of switch, which
10
d and fs
Fsmax=100 kHz
fs,max
8
Switch turn-on time, us
Dmax
fs,min
6
4
Without Frequency Variation
Fsmin=25kHz
2
Fsmin=50kHz
Dmin=0
1
0
vc/VT
Fsmin=75kHz
0
0
(a)
Vcc
0.1
0.2
0.3
0.4
0.5
0.6
Switch duty ratio
0.7
0.8
0.9
1
Figure 3: Variation of on time as a function of duty cycle
in VFPWM control.
PWM IC
Vc
PWM output
Q1
R2
R1
C1
(b)
Figure 2: (a) Variation of D and fs, in VFPWM control.
(b) Adapting PWM controller for VFPWM operation.
in turn limits the minimum output voltage from a
converter, can be increased if the switching frequency is
appropriately reduced when the converter is required to
operate with lower duty-cycles.
A wide variety of PWM controllers are commercially
available for control of dc-dc converters. The duty-cycle
of output pulses in these controllers is controlled by
analog control voltage, vc. The frequency of operation is
fixed in conventional fixed-frequency PWM (FFPWM)
control by selecting appropriate values resistor and
capacitor in the oscillator section of the controllers. In the
proposed variable-frequency PWM (VFPWM) control
method, the switching frequency is also actively varied
from a minimum (fs,min) to maximum value (fs,max) in
synchronization with variation of duty-cycle from dmin to
dmax as the analog control voltage varies from 0 to the
peak of saw-tooth carrier, VT as shown in Fig. 2(a). A
typical circuit implementation to achieve such control
characteristics is shown in Fig. 2(b). The resistance R1
and capacitor C1 decide the value of fs,min for vc=0 when
the transistor Q1 is off. As vc progressively increases, Q1
comes into conduction reducing progressively the
effective value of resistance and thereby increasing the
switching frequency. When vc=VT, Q1 is completely on
and fs,max is decided by the effective resistance of the
parallel combination of R1, R2 and capacitor C1.
Since the switching frequency is reduced when the
converter operates at lower duty-cycle, the on-time of the
switch increases. The constraint on minimum on-time of
the switch is thereby relaxed and a wider conversion
range is achieved. Illustratively, Fig. 3 shows the switch
on-time as a function of operating duty-cycle for fs,max
=100 kHz and various values of fs,min. In contrast to
another conventional method, namely constant-on-time
variable frequency control, wherein the frequency needs
to be varied from zero to maximum, the proposed
VFPWM control achieves wide conversion range with
frequency variation in limited range.
In VFPWM control, the output voltage is a linear
function of duty-cycle. Therefore the design of feedback
control loop simpler is similar to FFPWM control. In fact,
the same control circuit can be used for both FFPWM and
VFPWM with simple modifications shown in Fig. 2(b). In
FFPWM control, the switching losses in the converter
become comparable to the output power for operation
with lower duty-cycles resulting in poor conversion
efficiency. In VFPWM control, lower switching
frequency reduces switching losses improving the
conversion efficiency. On the negative side, lowfrequency operation demands a bigger output filter.
PECULARITIES IN FABRICATION
The power supplies developed for the present
application are an improved version of those presented in
[3]. In addition to the features incorporated in power
supplies in [3] like output-current-limiting, over-current
protection, two-loop based feedback control scheme and
remote operation capability, some circuit enhancements
and layout modifications have been done in these supplies
for better performance and standardization of power
supply cards for future applications.
Layout modifications in supply card include
generalization of footprint for magnetic components so as
to accommodate various ferrite core geometries (e.g.
EE36 and EE42). Similarly, generalized footprint for
power devices is provided on PCB to suit various
packages (e.g. TO-220 and TO-247). The PCB layout is
optimized to accommodate additional input filter
capacitors and bigger heatsinks with approximately 50 %
more surface area.
In addition to the layout optimization, various
enhancements in the circuit design have been carried out.
With this, it is possible now to increase the dc link filter
capacity by 50 % to reduce the 100 Hz ripple component
in the output current. Precision MFR resistors with
temperature coefficient of ±15 ppm/°C and ±0.1 %
tolerance are used in front-end control circuit to improve
output current stability. A common mode choke is added
on the output terminals to suppress common mode noise
in the output. Grounding on the PCB is further optimized
to minimize interference of high-frequency noise in the
analog control circuit. The electronics needed for
local/remote operation has been redesigned to comply
with the existing interface.
Figure 6: Conducted EMI of the power supply along
with CISPR-11 quasi-peak and average limit lines.
RESULTS AND DISCUSSIONS
Each power supply is standardized on a 6U card and
five such power supplies are housed in one 6U, 19-inch
sub-rack. Photographs of Fig. 4 show a power supply card
and a sub-rack. Various tests like open loop test, closed
loop test and local/remote operation test have been carried
out on all power supply cards which were then subjected
to 48 hour heat run test at maximum rating.
Figure 5 shows the stability curve of 5 nos. of power
supply cards for 24 hours of continuous operation after
initial warm-up time of 1 hour. Long term stability of
output current is seen to be well within the specified
limits of ±500 ppm.
To meet the required stability criteria various steps
have been taken such as using a stable shunt made of
zeranin in the feedback loop, using OP07 precision
OPAMPs in sensitive circuits and high precision MFR
resistors in reference, feedback and error amplifiers.
Figure 7: Photograph of two TL-1 quadrupole magnet
power supplies in operation.
Conducted electromagnetic interference (EMI) was
measured and pre-compliance was achieved with
CISPR-11 norms as shown in Fig. 6, which shows that the
conducted EMI spectrum of the power supply is well
below the CISPR-11 quasi peak and average limit lines.
Two numbers of TL-1 quadrupole magnet power
supplies (Fig. 7) have already been put in round the clock
operation for last 3 years and have not encountered any
failure or trip since then.
ACKNOWLEDGEMENT
Figure 4: Power supply card and sub-rack
The authors acknowledge Shri T.S. Rawat and Shri
Vinod Somkuwar for their assistance during assembly,
wiring and testing of the reported power supplies.
REFERENCES
[1]
Alok Singh, Manohar Koli, Mangesh Borage and Sunil
Tiwari, “New Power Supplies for Transport Line-2
Quadrupole Magnets in Indus”, Proc. of InPAC 2011.
[2] D. Maksimovic and S. Cuk, IEEE Transactions on Power
Electronics, 6(1), pp. 151-157, January 1991.
[3] Mangesh Borage, Trepan Singh, Manohar Koli and Sunil
Tiwari, “Modular magnet power supplies for CUTE-FEL
beamline and photocathode gun based LINAC”, Proc. of
InPAC 2009.
Figure 5: Output current stability curves of 5 power
supply units for 24 hrs of continuous operation