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UPGRADED SWITCH MODE POWER SUPPLIES FOR TRANSPORT LINE – 1 MAGNETS IN INDUS Alok Singh#, Manohar Koli, Mangesh Borage, Sunil Tiwari and A.C. Thakurta Raja Ramanna Centre for Advanced Technology, Indore-452013, India Abstract Upgraded dc current controlled power supplies of ratings 5 A/12 V and 12 A/14 V with specified output current stability of ±500 ppm and ±200 ppm, respectively are developed to energize quadrupole and dipole magnets in Transport Line-1 (TL-1) of INDUS. They are developed using two switch forward converter topology operating with variable frequency PWM control (VFPWM). This paper describes design principle, proposed VFPWM control, peculiarities in power supply development, salient features and results of various tests conducted on the power supplies. INTRODUCTION INDUS-1 and INDUS-2 are the synchrotron radiation sources operational at RRCAT with electron energies of 450 MeV and 2.5 GeV, respectively. The transport line-1 (TL-1), installed in Indus complex, is used to transfer electron beam from microtron to booster synchrotron. A dipole magnet and six quadrupole magnets are installed in TL-1, which are energised by stable current controlled power supplies. Existing power supplies for quadrupole magnets in TL-1 are based on series pass scheme and are in operation for several years. These are being upgraded with new, compact, efficient and stable power supplies using two switch forward converter topology operating with variable frequency PWM control (VFPWM). DESIGN PRINCIPLE AND FEATURES Topology Switch mode power supplies (SMPS) have an edge over series pass scheme because of its reduced size, lighter weight and better efficiency. Two-switch forward converter topology is chosen amongst various SMPS topologies since it is simple, rugged, tested and proven [1]. Maximum switching frequency of 100 kHz is chosen as a good trade-off between size of the power supply and switching S1 D1 LO D3 Cin D4 CO Cd Rd OUTPUT + 230 V 50 Hz D2 Rshunt S2 Figure 1: Schematic of two-switch forward converter ___________________________________________ # [email protected] losses. The basic circuit diagram of two switch forward converter scheme is shown in Fig. 1. In this dc-dc converter, IRF840 MOSFET switches S1 and S2 are turned on and off simultaneously causing energy transfer from source to load. The power is coupled to output through a high frequency transformer. MUR470E freewheeling diodes D1 and D2 resets the transformer and clamps the voltage across switch to the dc link voltage. MBR2060 dual common cathode schottky diodes D3 and D4 functions as secondary rectifier. Input filter (Cin) is a capacitor filter while output filter (LO, CO, Rd and Cd) is a damped low pass filter. A zeranin shunt is used to sense the output current. The feedback control scheme consists of two loops: the inner fast voltage loop that corrects input line variations, and outer slow current loop that corrects the drift in analog devices due to temperature. Variable frequency PWM (VFPWM) scheme has been incorporated in control circuit as a result of which switching frequency varies from 20 kHz to 100 kHz simultaneously with the increase in duty cycle from 0 to 0.5. This is done to increase the output current setting range. Bandwidth of voltage loop and current loop is approx. 1 kHz and 10 Hz, respectively. Local/Remote Operation Each power supply is capable of being operated from local fascia panel or in remote mode from central computer interface via a 25-pin sub-D connector provided on the fascia plate. VFPWM CONTROL Wide conversion range, that is, an ability to set and regulate the output current over a wide range, is an important requirement of a magnet power supply. In highfrequency-switching PWM dc-dc converters, the requirements of fast transient response and wide conversion range conflict. While higher switching frequency is favoured for miniaturization of the power converter and fast transient response, the objective of wide conversion range is defeated because the operation at extremely low duty-cycle is limited due to minimum on-time of the switch. The switching converters, in which conversion ratio has the quadratic dependence on dutycycle, can greatly extend the conversion range [2]. These techniques suffer from low conversion efficiency and higher component ratings. In this paper a new circuit technique, VFPWM, is suggested to improve the conversion range. The minimum on-time of switch, which 10 d and fs Fsmax=100 kHz fs,max 8 Switch turn-on time, us Dmax fs,min 6 4 Without Frequency Variation Fsmin=25kHz 2 Fsmin=50kHz Dmin=0 1 0 vc/VT Fsmin=75kHz 0 0 (a) Vcc 0.1 0.2 0.3 0.4 0.5 0.6 Switch duty ratio 0.7 0.8 0.9 1 Figure 3: Variation of on time as a function of duty cycle in VFPWM control. PWM IC Vc PWM output Q1 R2 R1 C1 (b) Figure 2: (a) Variation of D and fs, in VFPWM control. (b) Adapting PWM controller for VFPWM operation. in turn limits the minimum output voltage from a converter, can be increased if the switching frequency is appropriately reduced when the converter is required to operate with lower duty-cycles. A wide variety of PWM controllers are commercially available for control of dc-dc converters. The duty-cycle of output pulses in these controllers is controlled by analog control voltage, vc. The frequency of operation is fixed in conventional fixed-frequency PWM (FFPWM) control by selecting appropriate values resistor and capacitor in the oscillator section of the controllers. In the proposed variable-frequency PWM (VFPWM) control method, the switching frequency is also actively varied from a minimum (fs,min) to maximum value (fs,max) in synchronization with variation of duty-cycle from dmin to dmax as the analog control voltage varies from 0 to the peak of saw-tooth carrier, VT as shown in Fig. 2(a). A typical circuit implementation to achieve such control characteristics is shown in Fig. 2(b). The resistance R1 and capacitor C1 decide the value of fs,min for vc=0 when the transistor Q1 is off. As vc progressively increases, Q1 comes into conduction reducing progressively the effective value of resistance and thereby increasing the switching frequency. When vc=VT, Q1 is completely on and fs,max is decided by the effective resistance of the parallel combination of R1, R2 and capacitor C1. Since the switching frequency is reduced when the converter operates at lower duty-cycle, the on-time of the switch increases. The constraint on minimum on-time of the switch is thereby relaxed and a wider conversion range is achieved. Illustratively, Fig. 3 shows the switch on-time as a function of operating duty-cycle for fs,max =100 kHz and various values of fs,min. In contrast to another conventional method, namely constant-on-time variable frequency control, wherein the frequency needs to be varied from zero to maximum, the proposed VFPWM control achieves wide conversion range with frequency variation in limited range. In VFPWM control, the output voltage is a linear function of duty-cycle. Therefore the design of feedback control loop simpler is similar to FFPWM control. In fact, the same control circuit can be used for both FFPWM and VFPWM with simple modifications shown in Fig. 2(b). In FFPWM control, the switching losses in the converter become comparable to the output power for operation with lower duty-cycles resulting in poor conversion efficiency. In VFPWM control, lower switching frequency reduces switching losses improving the conversion efficiency. On the negative side, lowfrequency operation demands a bigger output filter. PECULARITIES IN FABRICATION The power supplies developed for the present application are an improved version of those presented in [3]. In addition to the features incorporated in power supplies in [3] like output-current-limiting, over-current protection, two-loop based feedback control scheme and remote operation capability, some circuit enhancements and layout modifications have been done in these supplies for better performance and standardization of power supply cards for future applications. Layout modifications in supply card include generalization of footprint for magnetic components so as to accommodate various ferrite core geometries (e.g. EE36 and EE42). Similarly, generalized footprint for power devices is provided on PCB to suit various packages (e.g. TO-220 and TO-247). The PCB layout is optimized to accommodate additional input filter capacitors and bigger heatsinks with approximately 50 % more surface area. In addition to the layout optimization, various enhancements in the circuit design have been carried out. With this, it is possible now to increase the dc link filter capacity by 50 % to reduce the 100 Hz ripple component in the output current. Precision MFR resistors with temperature coefficient of ±15 ppm/°C and ±0.1 % tolerance are used in front-end control circuit to improve output current stability. A common mode choke is added on the output terminals to suppress common mode noise in the output. Grounding on the PCB is further optimized to minimize interference of high-frequency noise in the analog control circuit. The electronics needed for local/remote operation has been redesigned to comply with the existing interface. Figure 6: Conducted EMI of the power supply along with CISPR-11 quasi-peak and average limit lines. RESULTS AND DISCUSSIONS Each power supply is standardized on a 6U card and five such power supplies are housed in one 6U, 19-inch sub-rack. Photographs of Fig. 4 show a power supply card and a sub-rack. Various tests like open loop test, closed loop test and local/remote operation test have been carried out on all power supply cards which were then subjected to 48 hour heat run test at maximum rating. Figure 5 shows the stability curve of 5 nos. of power supply cards for 24 hours of continuous operation after initial warm-up time of 1 hour. Long term stability of output current is seen to be well within the specified limits of ±500 ppm. To meet the required stability criteria various steps have been taken such as using a stable shunt made of zeranin in the feedback loop, using OP07 precision OPAMPs in sensitive circuits and high precision MFR resistors in reference, feedback and error amplifiers. Figure 7: Photograph of two TL-1 quadrupole magnet power supplies in operation. Conducted electromagnetic interference (EMI) was measured and pre-compliance was achieved with CISPR-11 norms as shown in Fig. 6, which shows that the conducted EMI spectrum of the power supply is well below the CISPR-11 quasi peak and average limit lines. Two numbers of TL-1 quadrupole magnet power supplies (Fig. 7) have already been put in round the clock operation for last 3 years and have not encountered any failure or trip since then. ACKNOWLEDGEMENT Figure 4: Power supply card and sub-rack The authors acknowledge Shri T.S. Rawat and Shri Vinod Somkuwar for their assistance during assembly, wiring and testing of the reported power supplies. REFERENCES [1] Alok Singh, Manohar Koli, Mangesh Borage and Sunil Tiwari, “New Power Supplies for Transport Line-2 Quadrupole Magnets in Indus”, Proc. of InPAC 2011. [2] D. Maksimovic and S. Cuk, IEEE Transactions on Power Electronics, 6(1), pp. 151-157, January 1991. [3] Mangesh Borage, Trepan Singh, Manohar Koli and Sunil Tiwari, “Modular magnet power supplies for CUTE-FEL beamline and photocathode gun based LINAC”, Proc. of InPAC 2009. Figure 5: Output current stability curves of 5 power supply units for 24 hrs of continuous operation