Download A Closed-Loop, Wideband, 100A Active Load

Application Note 133
October 2011
A Closed-Loop, Wideband, 100A Active Load
Brute Force Marries Controlled Speed
Jim Williams
Introduction
Digital systems, particularly microprocessors, furnish
transient loads in the 100A range that a voltage regulator
must service. Ideally, regulator output is invariant during
a load transient. In practice, some variation is encountered
and becomes problematic if allowable operating voltage
tolerances are exceeded. 100A load steps, characteristic
of microprocessors, exacerbate this issue, necessitating
testing the regulator and associated components under
such transient loading conditions. To meet this need, a
closed-loop, 500kHz bandwidth, linearly responding, 100A
capacity active load is described below.
pulse switches the FET via a drive stage, generating a
transient load current out of the regulator and its output
capacitors. The size, composition and location of these
capacitors has a pronounced effect on transient response
and must be quite carefully considered. Although the
electronic control facilitates high speed switching, the
architecture cannot emulate loads between the minimum
Note 1: See Reference 1, from which the immediately following material
partially derives, for details and descriptions of very wideband load
transient generators, albeit at much lower currents.
L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear
Technology Corporation. All other trademarks are the property of their respective owners.
REGULATOR UNDER TEST,
INCLUDING OUTPUT CAPACITORS
Study of this approach is prefixed by a brief review of conventional test load types and noting their shortcomings1.
+
Basic Load Transient Generator
Figure 1 diagrams a conceptual load transient generator.
The regulator under test drives DC and switched resistive
loads, which may be variable. The switched current and
output voltage are monitored, permitting comparison of
the nominally stable output voltage versus load current
under static and dynamic conditions. The switched current
is either on or off; there is no controllable linear region.
Figure 2 develops the concept by including electronic load
switch control. Operation is straightforward. The input
INPUT
PULSE
–
REGULATOR
VOLTAGE
MONITOR
RSWITCHED
LOAD
DC
LOAD
GATE
DRIVE
STAGE
0.001Ω
AN133 F02
ISWITCHED
MONITOR
Figure 2. Conceptual FET Based Load Tester Permits Input Pulse
Controlled Step Loading. As Before, Switched Current Is Either
On or Off; There Is No Controllable Linear Region
+EREGULATOR
REGULATOR
INPUT SUPPLY
REGULATOR
UNDER TEST
CURRENT
MONITOR
DC LOAD
VOLTAGE
MONITOR
RSWITCHED
LOAD
LOAD
SWITCH
ISWITCHED =
EREGULATOR
RSWITCHED LOAD
AN133 F01
Figure 1. Conceptual Regulator Load Tester Includes Switched and DC Loads and Voltage/Current Monitors. Resistor
Values Set DC and Switched Load Currents. Switched Current Is Either On or Off; There Is No Controllable Linear Region
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Application Note 133
and maximum currents. Additionally, FET switching speed
is uncontrolled, introducing wideband harmonic into the
measurement, potentially corrupting the oscilloscope
display.
Closed-Loop Load Transient Generator
Placing Q1 within a feedback loop allows true, linear control
of the load tester. Figure 3’s conceptual closed-loop load
transient generator linearly controls Q1’s gate voltage to
set instantaneous transient current at any desired point,
100A REGULATOR UNDER TEST,
INCLUDING OUTPUT CAPACITORS
+
–
REGULATOR
OUTPUT
VOLTAGE
MONITOR
+
INPUT
PULSE
(NEGATIVE)
A1
Q1
–
0.001Ω
AN133 F03
–V
BASELINE/DC LOAD SET
ISWITCHED
AND DC LOAD
MONITOR
Figure 3. Feedback Controlled Load Step Tester Allows
Continuous FET Conductivity Control. Input Accommodates
Separate DC and Pulsed Loading Instructions
INPUT AVG
DISSIPATION LIMITER
allowing simulation of nearly any load profile. Feedback
from Q1’s source to the A1 control amplifier closes a loop
around Q1, stabilizing its operating point. Q1’s current
assumes a value dependent on the instantaneous input
control voltage and the current sense resistor over a very
wide bandwidth. Note that once A1 is biased to Q1’s conduction threshold (by the “DC Load Set”), small variations
in A1’s output result in large Q1 channel current changes.
As such, large output excursions are not required from
A1; its small signal bandwidth is the fundamental speed
limitation. Within this restriction, Q1’s current waveform
is identically shaped to A1’s input control voltage, allowing linear control of load current. This versatile capability
permits a wide variety of simulated loads.
Figure 4 further develops Figure 3, adding new elements.
A gate drive stage isolates the control amplifier from Q1’s
gate capacitance, maintaining amplifier phase margin
and providing low delay, linear current gain. An X10 differential amplifier provides high resolution sensing across
the 1mΩ current shunt. The dissipation limiter, acting
on the averaged input value and Q1 temperature, shuts
down gate drive to preclude excessive FET heating and
subsequent destruction. Amplifier associated capacitors
tailor bandwidth and optimize loop response.
REGULATOR UNDER TEST,
INCLUDING OUTPUT CAPACITORS
FET TEMP
+
–
TEMP
SENSE
+
INPUT
PULSE
(NEGATIVE)
SHUTDOWN
A1
–
GATE
DRIVE
STAGE
Q1
+
0.001Ω
X10
–
AN133 F04
–V
BASELINE/DC LOAD SET
Figure 4. Developed Form of Figure 3. Differential Amplifier Provides High Resolution Sensing Across Milliohm Shunt.
Dissipation Limiter, Acting on Average Input Value and FET Temperature, Shuts Down Gate Drive, Precluding Excessive
FET Heating. Amplifier Associated Capacitors Tailor Bandwidth, Optimize Loop Response
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Application Note 133
Detailed Circuitry Discussion
–15V supply is not present by diverting Q4 bias. A1’s 1k
resistor precludes A1 damage due to 15V supply loss.
Trims optimize dynamic response, determine loop DC
baseline idle current, set dissipation limit and control gate
drive stage bias. The DC trims are self-explanatory. The
“loop compensation” and “FET response” AC trims at A1
are more subtle. They are adjusted for the best compromise between loop stability, edge rate and pulse purity.
A1’s loop compensation trim sets roll-off for maximum
Figure 5’s detailed schematic of the 100A capacity load
tester follows Figure 4’s outline. A1, responding to DC and
pulse inputs and current indicating feedback from A3, sets
Q1’s conductivity via the actively biased Q4-Q5 gate drive
stage. A2 determines stage bias under all conditions by
comparing Q5’s averaged collector voltage to a reference
and controlling Q3’s conduction, closing a loop. C1, instructed by the average input value, shuts down FET gate
drive via Q2 under harmful conditions2. SW1, sensing
heat sink temperature, contributes thermally activated
Q1 shutdown. Q6 and the Zener prevent Q1 turn-on if the
15V
15V
DISSIPATION LIMITER
POWER
LIMIT
+
100k
C1
LT®1011
1k
+
33k
13k
Q2
TP-0610L
5
BASELINE
1M CURRENT
–5V
0V
15V
200Ω
DISSIPATION
2k LIMIT
ADJUST
100k
470k
–5
0.1
GATE DRIVE
2.2k
0.01
–
10μF
Note 2: The protection scheme is patterned after techniques utilized in
high power pulse generators. See Reference 2 and 3.
Q6
2N3904
7pF TO 45pF
LOOP
COMPENSATION
Q4
2N3904
3Ω
SW1
OPENS 70°C
1N754
6.8V
10k
INPUT
TO LOAD POINT
OF SUPPLY
UNDER TEST
100k
0V TO –1V =
0A TO 100A
51Ω
FET
RESPONSE
COMPENSATION
1.5k*
1.5k*
300pF
3.01k*
HIGH
CURRENT
CRITICAL
PATH
MINIMIZE
INDUCTANCE
GAIN
100Ω
20pF
–1V
Q1
HEAT SINK
FET
CURRENT
SINK
432Ω*
–
A1
+ LT1220
50k
20pF
3Ω
CONTROL
AMPLIFIER
Q3
VN2222L
1k
4.5k*
+
A3
LT1193 F
Q5
2N3906
0.001Ω
–
–5V
1k
–15V
IQ
10mA
5V
X10 CURRENT SENSE
1Ω
SUPPLY UNDER
TEST RETURN
POINT
10k
0.02
LT1004
2.5V
FEEDBACK
CURRENT
MONITOR
1V = 100A
150k
IQ ADJUST
1k
10k
–
+
±5V REGULATORS, + = LT1121, – = LT1175
SW1 MOUNTED ON Q1 HEAT SINK
SW1 = CANTHERM F20A07005ACFA06E
* = 1% FILM RESISTOR
Q1 = INFINEON IPB015N04LG
A2
LT1006
–15V
10k
= 1N4148
AN133 F05
GATE DRIVE BIAS CONTROL
1μF
A1, A2, C1 = ±15V POWERED
BYPASSING NOT SHOWN
0.001Ω = TEPRO-TPSM3
Figure 5. Detailed Circuitry Follows Figure 4’s Concept. A1, Responding to DC and Pulsed Inputs, Sets Q1 Conductivity via
Actively Biased Gate Drive Stage. A3, Sensing Q1 Current, Closes A1’s Feedback Loop. C1, Instructed by Average Input Value,
Shuts Down Q1 Gate Drive Under Harmful Conditions. Q6 Guards Against Lost –15V Supply. SW1 Adds Thermal Limiting. Trims
Optimize Dynamic Response, Determine Loop Baseline Idle Current, Set Dissipation Limit and Control Gate Drive Stage Bias
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bandwidth while accommodating phase shift introduced
by Q1’s gate capacitance and, to a lesser extent, by A3.
The “FET response” adjustment partially compensates
Q1’s inherent nonlinear gain characteristic, improving
front and rear pulse corner fidelity3.
Circuit Testing
The circuit is initially tested using a fixture equipped with
massive, low loss, wideband bypassing shown in Figure 6.
Additionally, the importance of exceptionally low inductance
layout in the high current path cannot be overstated. Every
attempt must be made to minimize inductance in the 100A
path; such current density at high speed demands low
inductance if waveform purity is desirable. Figure 7 shows
results with the circuit properly trimmed and a minimized
inductance high current path. The 100A amplitude, high
speed waveform is exceptionally pure with just discernible
top-front and bottom-rear corner infidelities4. AC trim effects on waveform presentation are studied by deliberate
mis-adjustment. Figure 8’s overdamped response is typical
of excess A1 feedback capacitance. The current pulse is
well controlled but edge rate is slow. Figure 9’s inadequate
A1 feedback capacitance decreases transition time but
Note 3: The trimming procedure is not described here in order to maintain
text flow and focus. For detailed trimming information see Appendix B,
“Trimming Procedure”.
Note 4: See Appendices A, “Verifying Current Measurement” and C,
“Instrumentation Considerations” for guidance on obtaining Figure 7’s
performance level.
DC POWER SUPPLY
+
–
10,000μF
COMBINATION OF
OS-CON, FILM AND
CERAMIC CAPACITORS
MINIMIZE
INDUCTANCE
MINIMIZE
INDUCTANCE
Q1
FIGURE 5’s
CIRCUIT
AN133 F06
Figure 6. Fixturing Tests Figure 5’s Dynamic Response. Massive,
Broadband Bypassing Combined with Low Inductance Layout
Provides Low Loss, High Current Power to Q1
20A/DIV
HORIZ = 2μs/DIV
AN133 F07
Figure 7. Optimized Dynamic Response Trims Yield
Exceptionally Pure, 100A Q1 Current Pulse. Residual Top-Front
and Bottom-Rear Corner Infidelities Are Just Discernable
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Application Note 133
20A/DIV
20A/DIV
HORIZ = 2μs/DIV
HORIZ = 2μs/DIV
AN133 F10
AN133 F08
Figure 10. Corner Peaking Due to
Overstated FET Response Compensation
Figure 8. Overdamped Response Characteristic
of Excessive A1 Feedback Capacitor Value
20A/DIV
20A/DIV
HORIZ = 2μs/DIV
HORIZ = 500ns/DIV
AN133 F09
Figure 9. Inadequate A1 Feedback Capacitor Decreases
Transition Time But Promotes Instability. Further Capacitor
Reduction Causes Oscillation
promotes instability. Further capacitance reduction will
cause loop oscillation because loop phase shift becomes
a significant feedback lag5. Figure 10’s corner peaking is
due to overstated FET response compensation.
If the AC trims are restored to nominal values, Figure 11’s
leading edge indicates 650 nanosecond rise time, equivalent to ≈540kHz bandwidth. The trailing edge (Figure 12),
taken under the same conditions as the previous figure, is
somewhat faster at 500 nanosecond fall time.
AN133 F11
Figure 11. Optimized Dynamic Trims Allow 650 Nanosecond
Rise Time, Corresponding to ≈540kHz Bandwidth
20A/DIV
HORIZ = 500ns/DIV
AN133 F12
Figure 12. Same Conditions as Figure 11 Show
500 Nanosecond Fall Time
Note 5: Sorry, no photo available. Uncontrolled 100 Ampere amplitude
loop oscillation is too thrilling for documenting.
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Application Note 133
Layout Effects
None of the previous responses, even the mis-compensated examples, can be remotely approached if parasitic
inductance is present in the high current path. Figure 13
deliberately places only 20nH parasitic inductance in Q1’s
drain path. Figure 14a displays enormous waveshape
degradation deriving from the inductance and the loop’s
subsequent response. A monstrous error dominates
the leading edge before recovery occurs at the pulse
middle-top. Additional aberration is evident in the falling
edge turn-off. It is especially noteworthy that the photo’s
horizontal scale is 5× slower than Figure 7’s optimized
response, repeated here for emphasis as Figure 14b. The
lesson is clear. High speed 100A excursions do not tolerate inductance.
DC POWER SUPPLY
+
–
MINIMIZE
INDUCTANCE
10,000μF
COMBINATION OF
OS-CON, FILM AND
CERAMIC CAPACITORS
DELIBERATLY
INTRODUCED
20nH PARASITIC
INDUCTANCE
MINIMIZE
INDUCTANCE
Q1
FIGURE 5’s
CIRCUIT
AN133 F13
Figure 13. Deliberately Introduced Parasitic 20nH Inductance
Tests Figure 5’s Layout Sensitivity
20A/DIV
20A/DIV
HORIZ = 10μs/DIV
HORIZ = 2μs/DIV
(14a)
(14b)
AN133 F14
Figure 14. 1.5" of 0.075" (20nH) Flat Copper Braided Wire Completely Distorts (Figure 14a)
Text Figure 7’s (Shown Here as Figure 14b) Response. Note 5× Horizontal Scale Change
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Application Note 133
Regulator Testing
The active loads true linear response and high bandwidth
permit wide ranging load waveform characteristics. While
the step load pulse shown above is the commonly desired
Regulator testing is possible after the above compensation and layout issues have been addressed. Figure 15
describes a test arrangement for an LTC®3829 based,
6-phase, 120A buck regulator6. Trace A, Figure 16 shows
the 100A load pulse. Trace B’s regulator response is well
controlled on both edges.
INPUT
7V TO 14V
+
–
Note 6: This regulator is conveniently incarnated as LTC demonstration
board 1675A.
LTC3829 BASED 6-PHASE,
1.5V, 120A POWER SUPPLY
+
–
MINIMIZE
INDUCTANCE
FIGURE 5’s
CIRCUIT
AN133 F15
Figure 15. Test Arrangement for LTC3829 Based 6-Phase,
120A Buck Regulator Emphasizes Low Impedance
Connections to Figure 5’s Circuit
A = 100A/DIV
B = 0.1V/DIV
AC-COUPLED
HORIZ = 10μs/DIV
AN133 F16
Figure 16. Figure 5’s Circuit Subjects Figure 15’s Regulator
to 100A Pulsed Load (Trace A). Response (Trace B) Is Well
Controlled on Both Edges
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Application Note 133
the high speed and current involved. In Figure 18, an 80
microsecond burst of 100A peak-peak noise forms the
load. Figure 19 summarizes active load characteristics.
test, essentially any load profile is easily generated. Figure
17’s burst of 100A, 100kHz sinewaves is an example.
Response is crisp, with no untoward dynamics despite
50A/DIV
HORIZ = 10μs/DIV
AN133 F17
Figure 17. Figure 5 Supplies 100kHz, 100A Sinewave Load
Transient Profile. Circuit’s Wide Bandwidth and Linear Operation
Permit Wide Ranging Load Waveform Characteristics
50A/DIV
HORIZ = 20μs/DIV
AN133 F18
Figure 18. Active Load Circuit Sinks 100A Peak-Peak
in Response to Gated Random Noise Input
Active Load Characteristics
TEMPERATURE DRIFT
CURRENT REGULATION
vs SUPPLY
BANDWIDTH
540kHz
AT 100A
(tRISE = 650ns)
435kHz
AT 10A
(tRISE = 800ns)
COMPLIANCE VOLTAGES
FOR FULL OUTPUT CURRENT
80
100ppm/°C OF
READING + 20mA/°C
>60dB PSRR
NO
OVERSHOOT
90
1% FS
70
AMPERES
CURRENT ACCURACY
(REFERRED TO INPUT)
4%
LEADING EDGE
OVERSHOOT
100
60
50
40
30
20
10
0.95V MIN (SEE PLOT)
MAX SET BY 70°C Q1
THERMAL DISSIPATION
LIMITER
0
0
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2
MINIMUM Q1 DRAIN VOLTS
AN133 F19
Minimum Compliance Voltage vs Load Current
Figure 19. Active Load Tabular Characteristics. Current Accuracy/Regulation Errors Are Small, Bandwidth Mildly Retards
at Low Current. Compliance Voltage Is Below 1V at 100A with 4% Leading Edge Overshoot, 1.1V with No Overshoot
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Application Note 133
REFERENCES
APPENDIX A
1. Williams, Jim, “Load Transient Response Testing For
Voltage Regulators”, Linear Technology Corporation,
Application Note 104, October 2006.
Verifying Current Measurement
Theoretically, Figure 5’s Q1 source and drain current are
equal. Practically, questions center on potential effects
of residual inductances and the huge (28000pF!) gate
capacitance of the 1.5mΩ FET specified. If these or other
terms impact drain-source current equivalence at speed
A3’s indicated instantaneous current could be erroneous.
Verification is necessary and Figure A1 diagrams a method.
An added “top-side” 1mΩ shunt and its attendant X10 differential amplifier duplicate the circuit’s resident “bottomside” current sensing section. Figure A2’s results happily
eliminate concern over dynamic Q1 current differences.
The two 100A pulse outputs are identical in amplitude and
shape, promoting confidence in circuit operation.
2. Hewlett-Packard Company, “HP-214A Pulse Generator
Operating and Service Manual”, “Overload Adjust”,
Figure 5-13. See also, “Overload Relay Adjust”, pg.
5-9. Hewlett-Packard Company, 1964.
3. Hewlett-Packard Company, “HP-214B Pulse Generator
Operating and Service Manual”, “Overload Detection/
Overload Switch …”, pg. 8-29. Hewlett-Packard Company, March, 1980.
TO FIGURE 6’s
CAPACITOR BANK
AND DC SUPPLY
+
0.001Ω
F LT1193
–
Q1
TO FIGURE 5’s
CIRCUIT
4.5k*
+
LT1193 F
432Ω*
0.001Ω
–
“BOTTOM-SIDE”
CURRENT 4.5k*
432Ω*
100Ω
GAIN
100Ω
GAIN
“TOP-SIDE”
CURRENT
AN133 FA01
* = 1% FILM RESISTOR
Figure A1. Arrangement For Observing Q1’s “Top” and “Bottom” Dynamic Currents
A = 50A/DIV
B = 50A/DIV
HORIZ = 2μs/DIV
AN133 FA02
Figure A2. Q1 “Top” (Trace A) and “Bottom” (Trace B) Currents
Show Identical Characteristics Despite High Speed Operation
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Application Note 133
APPENDIX B
Trimming Procedure
Trimming Figure 5’s circuit is a 7-step procedure which
should be performed in the numerical order listed below.
Out-of-sequence adjustments are permissible assuming
the dissipation limiter circuitry is working and adjusted.
1. Set all adjustments to mid-range except A1’s feedback
capacitor, which should be at full capacity.
2. Apply no input, bias Q1’s drain from a 1V DC supply,
turn on power and trim “Base Line Current” for 0.5A
through Q1. Monitor this current with an ammeter in
Q1’s drain line.
3. Turn power off. Lift Q2’s source lead and let it float.
This disables the dissipation limit circuitry, leaving Q1
vulnerable to damage from inappropriate inputs. Follow the remainder of this step in strict accordance with
the instructions given. Turn power on, bias Q1’s drain
from a 1V supply and apply a –0.1000V DC input while
monitoring Q1’s drain current with an ammeter. Trim the
“Gain” adjustment for a 10.50A meter reading1. Make
this adjustment fairly quickly as Q1 is dissipating 10
Watts. Turn power off and reconnect Q2’s source lead2.
4. Turn power on with no input applied and Q1’s drain
unbiased. Trim the “IQ” adjustment for +10mV at A2’s
positive input measured with respect to the –15V rail.
Turn off power.
5. Bias and bypass Q1’s drain in accordance with text
Figure 6. Set the drain DC power supply for 1.5V output
and turn on power. Apply a 1kHz, –1V amplitude, 5μs
wide pulse. Slowly increase pulse width until C1 trips,
shutting down circuit output and illuminating the “Power
Limit” LED. Tripping should occur at about 12μs to15μs
pulse width. If it does not, adjust the “Dissipation Limit”
potentiometer to bring the trip point within these limits.
This sets allowable full-amplitude (100A) duty cycle at
about 1.5% .
6. Under the same operating conditions as Step 5, set
input pulse width at 10μs and adjust A1’s capacitive
trim for the fastest positive going edge obtainable at
A3’s output without introducing pulse distortion. Pulse
clarity should approach text Figure 7 with somewhat
degraded top-front and bottom-rear corner rounding.
7. Adjust “FET Response Compensation” to correct the
corner rounding noted in Step 6. Some interaction may
occur with Step 6’s adjustment. Repeat Step 6 and 7
until A3’s output waveform looks like text Figure 7.
Note 1: The tight trim targets are mandated by trimming gain at only 10%
of scale. This is certainly undesirable but less painful than trimming at
100% of scale which would force astronomical (and brief) dissipation in
Q1 and the milliohm shunt.
Note 2: It is worth mention that the primary uncertainty necessitating gain
trimming is the sense lines mechanical placement at the milliohm shunt.
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Application Note 133
APPENDIX C
Instrumentation Considerations
The pulse edge rates discussed in the text are not particularly fast, but high fidelity response requires some diligence.
In particular, the input pulse must be cleanly defined and
devoid of parasitics which would unfairly make circuit
output pulse shape look bad. The pulse generators preshoot, rise time and pulse-transition aberrations are well
out of band and filtered by A1’s 2.1MHz (tRISE = 167ns)
input RC network. These terms are not a concern; almost
all general purpose pulse generators should perform well.
A potential offender is excessive tailing after transitions.
Meaningful dynamic testing requires a rectangular pulse
shape, flat on top and bottom within 1% to 2%. The circuit’s
input band shaping filter removes the aforementioned
high speed transition related errors but will not eliminate
lengthy tailing in the pulse flats. The pulse generator should
be checked for this with a well compensated probe at the
circuit input. The oscilloscope should register the desired
flat top/bottom waveform characteristics. In making this
measurement, if high speed transition related events are
bothersome, the probe can be moved to the band limiting 300pF capacitor. This is defensible practice because
the waveform at this point determines A1’s input signal
bandwidth.
Some pulse generator output stages produce low level DC
offset when their output is nominally at its zero volt state.
The active load circuit will process such DC potentials as
legitimate signal, resulting in DC load baseline current shift.
The active loads input scale factor of 1V = 100A means
that a 10mV “zero” state error produces 1A of DC baseline
current shift. A simple way to check a pulse generator for
this error is to place it in external trigger mode and read its
output with a DVM. If offset is present it can be accounted
for, nullified with the circuit’s “baseline current” trim or
another pulse generator can be selected.
Parasitic effects due to probe grounding and instrument
interconnection should be kept in mind. At pulsed 100A
levels parasitic current is easily induced into “grounds” and
interconnections, distorting displayed waveforms. Probes
should be coaxially grounded, particularly at A3’s output
current monitor and preferably anywhere else. Additionally, it is convenient and common practice to externally
trigger the oscilloscope from the pulse generator’s trigger
output. There is nothing wrong with this, in fact it is recommended to insure a stable trigger as probes are moved
between points. This practice does, however, potentially
introduce ground loops due to multiple paths between the
pulse generator, the circuit and the oscilloscope. This can
falsely cause apparent distortion in displayed waveforms.
This effect is avoidable by using a “trigger isolator” at the
oscilloscope external trigger input. This simple coaxial
component typically consists of isolated ground and signal
paths, often coupled to a pulse transformer to provide a
galvanically isolated trigger event. Commercial examples
include the Deerfield Labs 185 and the Hewlett-Packard
11356A. Alternately, a trigger isolator is easily constructed
in a small BNC equipped enclosure as shown in Figure C1.
COAXIAL METAL
ENCLOSURE
1k
INPUT FROM
PULSE GENERATOR
TRIGGER OUTPUT
300pF
t
50Ω
T1
t
1k
INSULATED
SHELL BNC
OUTPUT TO
OSCILLOSCOPE
EXTERNAL
TRIGGER INPUT
AN133 FC01
T1 = ALMOST ANY SMALL PULSE TRANSFORMER
1k AND 300pF VALUES ARE TYPICAL
Figure C1. Trigger Isolator Floats Input BNC Ground Using Insulated Shell BNC
Connector. Capacitively Coupled Pulse Transformer Avoids Loading Input, Maintains
Isolation, Delivers Trigger to Output. T1 Secondary Resistor Terminates Ringing
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
AN133-11
Application Note 133
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Linear Technology Corporation
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