Download B. Front end electronics - GAE

B. FRONT END ELECTRONICS
I. Introduction
Authors: Eric Delagnes
At least four different techniques can be used to read the signal of the photo-detectors of such
experiments. In the three first readout scheme the readout follow the decision based on trigger
condition. The signal is compared to a threshold by a discriminator. The discriminator output is sent to
a first level trigger. This discriminator has to be as fast as possible and with the less possible time walk
to keep the timing properties of the pulses. The signal charge is obtained by integration over a time
window of minimum duration to decrease the effect of the Night Sky Background. To use the shortest
possible time window, the analogue pulse duration must be kept as short as possible. It means that the
electronics analogue bandwidth must be large enough not to widen the photo-detector pulses. The
integrated signal is read only when a trigger signal, generated using the information of several (all)
pixels of the camera, is received. For each valid photo-detector pulse, this trigger signal arrives after a
fixed time latency Tlat due to the cable length from the front-end to (and back from) the trigger
processor and to the time required for the trigger calculation.
a) The use of an analogue gated integrator and delay lines. In this case, the analogue signal must be
delayed by duration slightly longer than Tlat, before being processed by the gated integrator. This delay
can be achieved by a delay line, a cable or a transmission line integrated on the PCB. The integration
window starts when the trigger arrives (+ a few ns delay to compensate the difference in propagation
times for the different channels). The integration window duration can be digitally programmable.
Although this technique can be implemented in a very compact way and has given excellent results [7]
it suffers from limitations. First, due to the limited factor of merit of delay elements, the analogue
pulse will be widened and/or distorted after the delay. Second, the delay element (especially if it is
integrated in the PCB) has a fixed value and cannot deal with a change of trigger latency. The compact
implementation of a 100 ns delay has been proven but the practical realization of longer delays using
the same techniques seems difficult. For longer delays, the use of expensive delay lines is probably
required.
b) Another option was chosen in the ARS1 [8] chip for the ANTARES experiment, to read the simplest
kind of events. The signal is cyclically integrated over a few ns clock period defined internally on the
chip. When the signal cross the integrator discriminator threshold, this first integration is stopped and
a new one is started over a programmable window. At the end of this window, the final integration
result, obtained as the sum of the two integrations is stored in an analogue FIFO (together with a
TIMESTAMP information marking the threshold crossing). This technique permits to take into account
the part of the signal preceding the threshold crossing. An event stored inside the FIFO is read only if a
level 1 trigger is received within a given time, otherwise it is thrown. At last, ANTARES does not use this
LVL1 trigger: all the events are validated and sent to the shore. The main advantage of this solution is
that it does not require any analogue delay line and that the trigger latency can be very long and not
known a priori. The main drawbacks of this solution are:
i.
Its complexity
ii.
The FIFO (16 cells depth in ARS1) can easily saturate if the rate of “single hits” is high or if it does
not follow a Poisson statistic (afterpulses, bursted noise…).
Finally, the AMS CMOS 0.8µm technology used for the ARS1 is not yet available.
c) A third solution consists in the use of analogue memories. The operation is almost identical for the
both chip families (DRS and SAM).
The performances of the systems based on Analogue memories have already been proven for the
capture of the fast photo-detector pulses of the Atmospheric Cerenkov experiments. The first used
was the ARS0 chip [1], originally designed for the ANTARES [2] submarine neutrino telescope, which is
used successfully to read the total 4000 PMTs of the four telescopes of the HESS-1 telescope array[3].
A new chip, the Swift Analog Memory (SAM) [3] has been designed for the big telescope of HESS-2. Its
specifications have been defined to correct the defects of the ARS chip and to match the needs of
HESS-2. In parallel, the MAGIC[4] collaboration has first evaluated the performances of a readout
based on the early versions of the Domino Ring Sampler chip [5] and is currently building a new
electronics based of the new versions of the DRS for the MAGIC-2 telescope [6]. Such a solution is also
planned for the next generation of Cerenkov Telescopes.
The last versions of these chips are all based of the voltage sampling of an analogue signal in an array
of capacitor, sequenced by command signals generated by a ring of elementary delays. But they are
using slightly different techniques and exhibit different features.
The analogue signal is continuously sampled in a circular analogue memory at a rate Fs defined by the
elementary step of a delay chain. This analogue memory is made of an array of Ns sample & hold cells
using MOS switches and capacitors. The memory is said circular because when the last memory cell
has been written; the first cell is overwritten and so on. When a trigger signal arrives, the write
operation is stopped and the samples corresponding to a region of interest around the sample marked
by the signal can be read back and digitized by an extra ADC at a lower frequency.
The main interests of this solution are the following:
 no external analogue delay required.
 the sampling frequency can be very high for a very low power consumption.
 the readout frequency can be few order of magnitude smaller than the write frequency so that
the use of low cost/low power ADC and digital electronics is possible.
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 The output data after digitization is a sampled waveform so that it is possible to perform digital
treatment in it (numerical integration, optimal filtering, pulse shape study, timing extraction…)
 Signal/ noise ratio larger than 11 bits can be obtained.
 Analogue memory chips are full custom chips using low cost CMOS processes. Several channels
can be integrated on the same chip and extra functions may be added in the same chip
(multiplexing, AD conversion…)
 Can be combined with a non linear front-end
The main drawbacks are the following:
 the memory depth must be larger than the trigger latency.
 these chips are not commercially available.
The following specifications (justified) must be defined by the collaboration. A priority should be given
for each of them:











Number of cells.
Dynamic range.
Sampling frequency.
Sampling jitter.
Power consumption/channel
Readout dead time.
Number of channels/chip.
Crosstalk.
Linearity.
Price/channel
…
c) In a last solution, the analogue signal can be directly (of after shaping) digitized by a fast Flash ADC
and stored inside a FIFO. When a trigger arrives, the pulse charge is digitally calculated by performing
the sum or a linear combination of samples corresponding to a time window around the sample
associated with the trigger. This solution was used for the MAGIC-1 telescope [11].
The main advantages of this solution are:
-
It can deal with very long trigger latency (only depends on the digital FIFO length).
-
Digital treatment on the data can be performed to improve the charge resolution or to extract
exact pulse timing.
-
If the FADC is fast enough, the analysis of the pulse shape can be used to reject noise events. Its
output could also be used to generate the trigger.
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-
Can be combined with a non linear front-end
The main drawbacks of this solution are the following:
-
The FADC are still expensive. The price of the lowest cost ADC is typically 100 Euros for a 2channel 8-bit 500MSample/s ADC that can be used as a single channel 1GS/s ADC. Highest
dynamic range (up to 10bit/1GS/s) ADC are now commercially available but are much more
expensive
-
The power consumption of such a device is around 1W/channel.
-
The digital electronics required to treat and store the digital data at the output of the ADC must
used high end FPGA and memories running at high frequency. They are expensive and their
power consumption is at least of the same order as the one of the ADC.
The large power consumption of this solution makes its integration inside the camera difficult.
In MAGIC-1, the analogue signals were multiplexed and transmitted on optical fibres outside the
camera before digitization. The digitization frequency was only 300 MHz (requires less power and
cheaper) but a shaper was required to widen the pulse. This permits to limit aliasing effect and to keep
a sufficient number of samples to perform the charge calculation with a reasonable accuracy.
The performances of these different solutions depend on constraints implied by the physics cases
studied by CTA. In the following, we describe and analyze the dynamic range and linearity, the input
bandwidth, photon detector gain and its amplification, the sampling frequency and information need
to achieve the scientific goal of CTA.
II. Dynamic range and linearity
Coordinator: Bruno Khélifi
Authors: Patrick Nayman
Start concluding concerning the dynamic range issue (Bruno khelifi)
II.1.
Introduction
For the CTA electronics it is very useful to extract the photoelectron spectrum in order to calibrate easily
all the electronics chain. For this it is necessary to assess the implications on electronics.
It may be interesting to work with a gain of photomultiplier of the order of
and with a unique range. In
this case, it is necessary to assess the implications on electronics in terms of noise and dynamic range.
One can evaluate the performances of the CTA electronics by the Noise Factor method. This evaluation will
give us some information on how noisy (or how efficient) is the electronics under test.
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We can consider a very simple model composed of a PMT with its electronics. The input level will be the
photocathode of the PMT. The output level will be the ADC count.
In the case of a photomultiplier at the photoelectron level we have to evaluate the two quantities
and
.
Electronics
PM
T
ni
Photocathode
When the statistic of the photomultiplier follows a Poisson law:
incident photons hitting the photocathode and
with
is the number of
is the efficiency of the photocathode. In this case, the
variance of the photoelectron flux is given by:
(without the pedestal) of the photoelectron spectrum and
and
with
the mean value
is the corresponding variance.
The Noise Factor (F) is used to define the performances or the contributions of a system in a low noise
environment. The basic definition for the Noise Factor is:
It is possible to derive from this equation others popular equations, for example:
We easily show that
.
For the photoelectron case (
is:
.
and the number of photoelectrons per ADC count
. If we take into account the PMT statistic:
.
In the general case, where ENC is the readout noise, we have to refer this noise to the input of the PMT. The
total relative noise of the system is defined by the following quadratic sum:
where G is the total gain of the chain.
II.2.
First stage amplifier noise
In order to evaluate the first stage amplifier noise, it is possible to use the following model:
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ENI
ER S
++
-Rf
IB N
E Rf
Rs
Vs
Rg
IRG
IBI
To evaluate the input equivalent total noise we will evaluate the output contribution of each noise source
according to the following table.
is the Noise Gain.
Noise Source
Comment
Noise Gain
Noise Input Voltage
Noise Input Current Input +
Source Resistance Noise Voltage
Noise Input Current Input Rf Noise Voltage
1
Rg Noise Current
The sources
,
and
amplifier environment.
depend on the amplifier structure while
,
and
depend on the
The variance of the noise at the amplifier output is given by:
We can derive easily the equivalent input noise:
For example, with a standard low noise amplifier, we get from the manufacture the following parameters:
,
,
.
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These values lead to
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.
Now we have to take into account the bandwidth of the amplifier. For this purpose we first compute the
equivalent band noise:
where is the amplifier -3dB bandwidth. For a
,
.
The output variance related to the amplifier output is obtained by integrating the spectral density over all
the bandwidth:
and
.
With the previous data one obtains
II.3.
.
PMT Output Signal
It is interesting to know the expected signal at the input of the amplifier in function of the gain of the PMT.
Assuming a triangular shape for the signal, the output current of the PMT at the photoelectron level is given
by:
with e the electron charge, G the PMT gain and FWMH stands for the SER pulse width.
If we assume a FWMH=3ns we can compute
and
for different PMT gains.
PMT Gain
(50 ohms)
0.53µA
26.5µV
5.3µA
265µV
10.6µA
530µV
The following plot shows the pedestal and the photoelectron spectrum located respectively at
above the mean value of the pedestal.
Pedestal
5s
3s
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Photoelectron
and
The following table summarize the different parameters for a PMT gain of
BW
Noise BW
.
(RMS noise voltage)
3
9µV
300MHz
471MHz
0.4nV
3
9µV
400MHz
628MHz
0.36nV
4
6.6µV 300MHz
471MHz
0.3nV
4
6.6µV 400MHz
628MHz
0.26nV
5
5.3µV 300MHz
471MHz
0.24nV
5
5.3µV 400MHz
628MHz
0.21nV
From the above table, in all the cases, the RMS input noise must be extremely low (less than 0.4nV). Such
values for the input noise RMS are impossible to achieve with standard components.
II.4.
Constraints linked to the dynamics
Consider that it wishes to express the dynamic range as a number of photoelectron. In order to fit a correct
Gaussian curve on the photoelectron spectrum, the latter must be spread over a certain number of ADC
bins (>30). The following table gives, for different dynamic ranges expressed in term of photoelectrons, the
corresponding ADC dynamic. 2 cases are explicated: 30 ADC bin/pe and 50 ADC bin/pe.
Dynamic range
(#pe)
Dynamic with 30-bin/pe
Dynamic with 50-bin/pe
1000
30K
15 bits
50K
16 bits
2000
60K
16 bits
100K
17 bits
3000
90K
17 bits
150K
18 bits
4000
120K
17 bits
200K
18 bits
5000
150K
18 bits
250K
18 bits
6000
180K
18 bts
300K
19 bits
From the above table, a minimum dynamic range of 18-bit is needed.
II.5.
Others parameters
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In cases where using of an analogue memory, it should be noted that the over-sampling of the signal
contributes to reduce the noise. The calculation of the charge of the signal is based on the summing of
samples, of this fact the non-correlated noise is reduced by effect of averaging. More the factor of oversampling of the signal is large and more the noise will be averaged.
For example for a signal with a triangular shape (FWMH=3ns, Max=P) then the signal charge is
proportional to:
k is the oversampling factor expressed en GHz.
The RMS noise is reduced by a factor
For a 1GHz sampling frequency the noise reduction factor is 1.7 (0.76-bit) and 2.4 (1.26-bit) for a 2GHz
sampling clock.
II.6.
Conclusions
All of the discussions above show that it is impossible to manage the dynamics in a single range. Two
effects are to take into consideration: the noise of the electronic brought to the input is impossible to
achieve and the number of bits required is also a problem.
It appears advantageous to release the constraints in considering on the one hand a gain for the
photomultiplier of the order of
and on the other hand to use two overlapping ranges one for small
dynamic and another for large dynamic.
In this case, then it is possible to work with the mean value of the photoelectron located
value of pedestal.
III. Input bandwidth of the entire electronics chain
Coordinator: Razmick Mirzoyan
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above the mean
Authors: Razmick Mirzoyan
Final bandwidth of the entire chain (could be different for middle size (1st priority), large size and small
size telescopes)
1. Needs according to signal pulse shape
2. Effect on trigger
3. Background integration
…
The bandwidth of the entire signal chain, including the readout system (FADC) depends on the
aimed sensitivity for the given energy range. It is known that the Cherenkov pulses from gamma
showers have a time structure of a few ns. This time structure is narrower for lower energies.
Beyond the position of the hump in the lateral distribution of Cherenkov photons form an EAS the
time structure is becoming more extended. This is because the measuring instrument can see only
the lower part of the shower where the e-e+ have few times lower energy than in those high-up
and correspondingly they can undergo scattering under larger angles, i.e. the lower part has larger
lateral extension and the path length difference of light can become much larger. Obviously for
best results the telescope system bandwidth shall be such as to provide a possibly low distortion
of the time structure of original light pulse from air showers. In that case one can maximize the
signal to noise ratio where the noise is due to the integration of LONS. By selecting the signal time
window as short as the light pulse by itself one can minimize the contribution of LONS into the
signal, making this component unimportant. In Fig.xyz we show the time structure of gamma and
proton air showers measured by the 17m diameter MAGIC telescope above the threshold of 30
GeV. As the lower energy showers provide the highest trigger rate the shown time structure is
representative for that part of the spectrum. The PMT will produce an output pulse that is the fold
of its time response with that of the original light pulse.
IV. PMT gain/Front end amplifier/single photo-electron calibration
Coordinator: David Gascon
Authors: David Fink, David Gascon & Patrick Nayman, Eckart Lorenz, Razmick M.,
Single amplification chain (noise of the amplifier) or double amplification chain (obviously this will
assume a double readout system)?
1.
Introduction (TBC, last thing to be written…)
Comment that a sensor upgrade must be taken into account
We are discussing amplification:
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As close as possible to the PMT (from Razmick):
a. will minimize the pick-up noise (any cable connection of low amplitude signals is prone to
electromagnetic pick-up, especially to the magnetic component of the radiation)
b. provide the maximum signal/noise ratio
c. minimize the signal ringing
d. maximize the signal bandwidth
e. provide a modularity of the light sensor part of the camera. An alternative light sensor like a
SiPM (MPPC, GAPD,…) could sometime soon compete with a PMT. A camera upgrade could be
relatively easy if any given type of the used light sensor module, that has its specific amplifier
type, includes the latter in its construction.

In the front end electronics
a. Such amplifier cannot be a universal one, hence when changing the light sensor type one cannot
use it for the other type.
b. Also, it will not have the above mentioned advantages. The main advantage offered by this
solution is its inclusion in a possible ASIC chip.

A combination
a. There is no obvious advantage offered by this solution.
2.
PMT gain (?)
VALUE?
OR AT LEAST, RANGE???
a) Low gain strategy (Razmick, Eckart)
One of the main drawbacks of a PMT is due to the “fatigue” effect: the surface material of dynodes is
gradually losing its ability to emit secondary electrons when bombarded by an intense flow of accelerated
electrons inside the dynode chain of a PMT. The higher the impinging/emanated current the stronger is the
effect. Especially the last dynodes, which are operating under high currents, are strongly affected by this. In
order to recover this gain loss of the dynodes one needs (from time to time) to increase the applied interdynode (the PMT) HV.
When operating a PMT under a current of ~(1-2) µA for ~1500 h (this corresponds to operating PMTs in a
camera of an IACT during 1 year observation period) the integral charge that leaves the last dynode of a
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PMT is (5 – 10) C. Usually after flow of an integral charge of ~150 C (rare 300 C) a PMT arrives at halve of its
initial gain. This corresponds to an estimated annual gain loss of (2 – 4) %, that could be recovered by
increasing the applied HV. Usually for a big batch of PMTs (≥ 500 pieces) the absolute value of the applied
voltage for providing the same gain
for the most sensitive and the least
sensitive PMTs differs by ~500 V. Any
type of PMT has a maximum rated
applied voltage. So after increasing
the PMT HV for a few times one may
encounter serious problems because
of approaching the limit for some part
of PMTs. With lower number of
dynodes this is a less problem. The
process of ageing is different from
PMT to PMT and is not linear in time.
It is rather steep in the beginning and
tails-off after longer periods of time
(see Fig.1). In Fig.1 are shown the
results of accelerated lifetime
measurements (the PMTs were run under a constant light flux inducing an initial average anode current of
150 µA) for a few 1’ size hemi-spherical PMTs from Hamamatsu and Electron Tubes.
i.
The night sky can be considered as a rather intense DC source of light (light of night sky, hereafter
LONS). For example, in 17m MAGIC that uses pixels with an aperture of 0.1°, the LONS induces a
ph.e. rate of 130-150 Mhz when observing a dark patch in the sky, up to 800 MHz when observing
the center of the galaxy and about 1-5 GHz when observing during partial moonshine or similarly
when having a bright star in the pixel’s field of view. The PMTs in MAGIC are operated under the
gain of (2-3)x104 that typically produces anode currents of ~(08-1.4) µA (correspondingly for
extragalactic or galactic pointing). In the wide-angle-integrating AIROBICC detector the LONS had
induced a ph.e. rate of 150 GHz. The above numbers are obtained with PMTs applying the
“classical” bialkali photo-cathodes (i.e. with peak quantum efficiency (QE) of 25-28 % in the
wavelength range ~370-400 nm and an average ph.e. collection efficiency (CE) of 85 %). For the CTA
project we are going to use PMTs with 40-60 % higher photon detection efficiency, which means
that for the same gain the anode currents will be 40-60 % higher compared to PMTs with “classical”
photo-cathode. The strong ageing prevents operating PMTs under high gain.
ii.
The simple solution to overcome this problem is to lower the PMT gain and using an AC coupled
preamplifier for arriving at the overall needed channel gain. The ac coupling suppresses the steady
component induced by the LONS, but not the fast shower signals (and the fluctuations of LONS
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around its mean value). In a PMT the last dynodes only add to the gain, making it larger. The
equivalent noise of preamplifiers in the bandwidth of few 100’s of MHz is usually in the range of
4000-7000 ph.e. (correspondingly for a good or for a mediocre amplifier designs) at room
temperature. In order to provide a reasonable signal/noise ratio the gain of the PMT shall be
several times higher than the amplifier noise. For observing the single ph.e. pulse height
distribution one needs to observe the peak of the distribution but as well the valley. Usually for a
good PMT a single ph.e. from the photo cathode produces twice higher amplitude for the peak
compared to the valley. This means that the requirement for observing the valley is more stringent
than for the peak. For observing the valley one needs to provide a valley/samp ≥ 3 (sampstands for
the noise of the amplifier. Under the given gain a (good) PMT will produce on average single ph.e.
amplitudes near the peak of its amplitude distribution. So if the gain op a PMT is ≥ 6 x samp., one
shall observe single ph.e. distribution. For a higher precision one may work under the gain of 10 x
samp. Usually after flat-fielding of the channels of the imaging camera of a telescope during real
observations one finds channel-to-channel variations of the gain around the set mean value. So a
somewhat higher gain than the minimum requested one (10sinstead of 6s) will make the setup
less sensitive to the gain spread of the PMTs always providing resolved amplitude distribution for
single ph.e.s.. Assuming that we will be using an amplifier with a 1s equivalent noise of 4000
electrons, one can safely operate the PMTs under the gain of ≥ 40000.
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Fig.XY. 2- example of 6-dynode PMTs from ET operated under the gain of ~55000. One can observe the
single ph.e. peaked distribution (the upper and middle graphs).
There exist enough good preamps with a linear output of up to 2-3 Volts , which is in the range of
current FADCs or switched capacitor arrays.
iii.
Running a PMT with less number of dynodes and low gain means less power-less cooling of the
camera because simply the bleeder current could be lower. One still saves power even when adding
the preamp power ?
iv.
A PMT with only 6-7 dynodes and running on not too excessive gains applies a considerably lower
HT than that needed for operating an 8-10 dynode PMT.
v.
A PMT with 8-10 dynodes shall have larger drift than a PMT with only 6-7 dynodes.
vi.
The time profile of a PMT with only 6-7 dynodes operated at high inter-dynode voltages has a
narrower structure. With MAGIC-I we observe shortest pulses of 2.1 nsec FWHM in the FADC
readout (while the same type of PMT with 10 dynodes has a FWHM of 3.3-3.5 nsec- see also the
data sheet of ET.
vii.
Working with a lower gain means one stays more in the linear range of the gain for large pulses, see
attached docs for box and grid dynodes.
viii.
The gain stability as a function of the bias voltage drift is much less critical for a lower gain 6-7 stage
pmt than a for a higher gain 8-10 dynode PMT, in turn one needs much more voltage increase to
correct the gain by a factor 2. Also the drift after switching on PMTs is smaller because each extra
dynode can add to the drift.
ix.
Also, less dynodes means shorter size (faster) for the PMT.
For well-resolved single ph.e. amplitude response it is important operating the PMT at a relatively high
photo cathode to 1st dynode voltage of 300-350 V that can provide high yield of secondary electrons. It is
much desired to keep this potential difference the same for all the PMTs. In principle even higher potential
difference is desired but this is not practical because of possible strong increase of the afterpulsing.
In summary: less aging, less power, faster but lower amplitude pulses, less drift, higher linearity, less cooling
of the camera, less problems with cathode at HT. The main problem is to build a good preamp and to place
it as close as possible to the PMT base and to shield it very well against electromagnetic interference (EMI).
Historically, PMTs were developed at a time when a fast low noise amplifier was as big as a NIM crate and a
lot more expensive. So for required higher gain researchers were used to prefer higher number of dynodes.
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PMTs are also archaic amplifiers and lack the feedback like in any op-amp, therefore one has to live with
some drift.
A gain of ≥ 4.104 for PMTs shall provide a reasonably well-resolved single ph.e. amplitude distribution and
minimal ageing.
b) High strategy (P. Nayman, …)
3.
Single photoelectron-calibration and dynamic range (?, ATAC?)
REQUIREMENTS ON S/N
4.
Methods to achieve required dynamic range (Patrick Nayman)
The dynamic range of each pixel for the CTA cameras may be significant. If we assume a pixel dynamic
range of 1 to 6000 photoelectrons, this correspond to about 13-bit. In other hand, consider we want to
calibrate all the electronics chain at the nominal photo multiplier high voltage with the photoelectron
technique. In this case, we need a 50-bin per photoelectron to fit properly a Gaussian shape (~6-bit).
So 19-bit are necessary to achieve the dynamic range and the resolution! So it is obvious that the 19bit goal is very difficult to reach directly.
Method 1: 2-gain complete chain with a 12-bit dynamic for channel. The advantage of this method is
the simplicity to implement. It is easy to get low cost 12-bit components. The main disadvantage is that
it doubles the amount of data. There is also a small repercussion on the front-end cost because it
doubles the amplifiers and the pipeline channels.
Open question is the gain ratio (20:1 ?):

High Gain: 1 to 200 phe ?

Low gain: 100 to >6000 phe ?
Method 1-bis: 2-gain with a selection of the most appropriate channel before the analogue to digital
conversion. This method is similar to the previous one but has the advantage to limit the amount of
data by a factor of 2. This method allows, for tests purposes, to switch between method 1 and method
2. Another possibility is to choose the range at the beginning of the chain. But this solution is very
difficult to implement
-
Is it the same as the David Fink’s suggestion?
-
For precise calibration
o Do we need to see the high and low gain for the SAME input signal (100 phe for
instance)?
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
Then we really need to double the analogue memory channel
o OR Could we look consecutively to the low and high gain repeating a given input
pulse?

LED pulses fluctuate, but could it work on average? (probably it depends
on required calibration accuracy as David says...)

Then the idea to change the gain of the amplification through a switch
may work, but

With 12 bits and 2 V ADCs, don’t we need 2 gains anyway?
Method 2: amplifier with a logarithmic scheme. In theory this method is the best but in practical it is
not obvious to implement and to calibrate. This method requires developing a special design. The
calibration needs to know each individual curve.
Method 3: limited dynamic channel. This method is the simplest one but it is not possible to achieve
the resolution need for the photoelectron calibration at the nominal high voltage value.
The following table summarizes the previous possibilities.
5.
Method
Advantages
Disadvantages
1
Very easy to implement
Amount of data
1-bis
Easy to implement
2
Minimal hardware
Difficult to calibrate
3
Very easy to implement
Limited dynamic range. No pe @ nominal HV
Technical requirements (David Fink / David Gascon)
Key points driving the design have been discussed so far; using this information, specific technical
requirements for the front end amplifier of the CTA camera must be deduced.
In the first place we will look at the input signal levels. It is basically determined by the photo-sensor
gain. For the baseline photo-sensor it is not clear yet the gain setting for normal operation (see section
2), however there 2 limit cases:
1. Low gain (40K): for moon observation and to prevent ageing
2. High gain (200K): for easy single photoelectron (phe) calibration
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Table 2 shows signal range for the two extreme cases. Peak current (IP) is computed by triangular
approximation of the pulse charge Q=1/2·w·IP, where w is the signal width (W), about 3 ns. Voltage is
obtained assuming 50 Ω load.
PMT gain
40 K
100 K
Min
(1 phe)
Max
(6 Kphe)
Min
(1 phe)
Max
(6 Kphe)
Charge (Q)
6.3 fC
38 pC
15 fC
100 pC
Peak current (Ip)
4.2 µA
25 mA
10 µA
63 mA
Peak voltage (Vp)
210 µV
1.25 V
500 µV
3.2 V
Table 2. Signal range.
According to single photo-electron calibration requirements (section 3), to have good peak-to-valley
ratio the signal over noise has to be higher than a factor 101 Thus, the total integrated equivalent input
noise of the electronics must be smaller 21 µV rms for a low PMT gain or 50 µV rms for high PMT gain.
Assuming white noise and 300 MHz bandwidth (see section X), it corresponds to an equivalent series
noise of 1.2 nV/sqrt(Hz) or 3 nV/sqrt(Hz) for low and high PMT gain respectively.
In order to perform camera calibration, the transfer function, either linear or non-linear (piece-wise
linear), of the amplifier must be know with a absolute accuracy better than 3% of the signal range and
smaller than the statistical error for a given signal (sqrt(Nphe))(MC WP?). The gain temperature
coefficient must be smaller than 0.1 %/K.
Full system bandwidth should be higher than 300 MHz, therefore combined (if more than 1 amplifier in
series) amplifier bandwidth must be higher than 400 MHz for the digitizing input capacitances
specified below. Slew rate must be higher than 1.5 V/ns (ADC input), as this parameter is not additive it
applies to each amplifier in the chain.
Interface between amplifier and digitizing electronics must be considered:
1

Digitization input signal range: 1.5-2 V

Differential input

12 bit ADC are the baseline solution, therefore, as discussed in section 4, either two range of
amplification or compression is needed.
It includes some security margin, probably factor 5 to 7 is enough.
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
Input capacitance: 3 pF (for an integrated amplifier) or 5 to 10 pF (for an external amplifier).
Requirement on power conssumption???
Table 3 summarizes requirements for the front end amplifiers:
Value
Comments
Accuracy
1-3 %?
Bandwidth
400 MHz
For all the gain range. TOTAL BW of the amplification
Output range
1.5 to 2 V
Digitizer input range
Absolute gain
??
Absolute PMT gain? 105
Temp. Coeff.
< 0.1 %/K
Gain TC
Power
< 50 mW
???
Slew rate
1500 V/µs
Series noise
Low PMT gain
High PMT gain
3
Some proposed digitizers ASICs have only positive supply voltage. If the front end amplifier is integrated
in such ASIC on possible solution is to AC couple the PMT as shown in Fig 1.
CAC
IPMT
-HV
RPMT
cpar
VI
rin
+ -
cin
VR
rin
+
cin
VR
Figure 1. AC coupling between PMT and amplifiers
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If a first amplifier is placed near PMT, the same consideration applies between this amplifier and the
ASIC input amplifiers.
Related to the use of AC coupling in electronics, one may cosnider to get rid of the offset and of the 1/f
noise at an early stage. But, what is the maximum frequency for this LF pole? (to avoid overshot, etc…)
Frequency
LF pole
300 MHz
Figure 2. AC coupling LF pole
6.
Options for amplification
a. Linear amplification near (or in) PMT base (David Fink)
b. Linear amplification in the analogue memory ASIC (David Gascon)
An amplifier / signal conditioning circuit is required at the input of the front-end electronics even if a
ultra-low noise amplifier is located at the PMT base to operate at low gain. There a number of reasons:

Produce two signal ranges in case of linear amplification

Drive the input stage of the digitization (i.e. analog memory) properly: signal range, differential
signaling, input capacitance...
This circuit can be implemented through commercial components or as an custom ASIC block that can
be integrated in the digitizing chip. An integrated solution offers several advantages: almost zero cost
for mass production, speed, lower power consumption and compactness. Main drawback is the R+D
effort needed to prove the feasibility of this solution.
Collaboration has been set-up to study the integration of the amplifier on a readout chip based on
analogue memory [NECTAr]. The design of an amplifier fulfilling requirements given in section 5 needs
to exploit all the available performance of the baseline technology (CMOS 350 nm), because of the
large gain-bandwidth (GBW) product that is required (together with low noise). It will be assumed that
either the PMT is operated at high gain either an ultra-low noise amplifier located in the PMT base
provides a first amplification (x 10), thus the requirement on input referred noise is 3 nV/sqrt(Hz) for
an integrated amplifier.
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It is worth mentioning that the full design could be directly migrated to a compatible BiCMOS
technology where SiGe bipolar transistors are available. Those transistors have a large f T and would
simplify the amplifier design. However, this migration implies a cost increment, so work has been
started to develop a solution in a pure CMOS technology.
A closed-loop solution (op-amp style with global feedback) has been discarded, as reachable GBW of
the CMOS 350nm technology is between 500 MHz and 1 GHz. Solutions based on HF transconductors
have been explored, results of simulation of basic structures are shown in Table 4.
Lin. Err.
[%]
BW
[MHz]
Noise
Bias current
[mA]
Comments
Simple dif. Pair
2
1000
2.2
4.5
W/L limited by linearity
Dif. Pair with
degeneration
1
1000
2.7
4.5
Cross coupled
(XC) mismatched
3
2000
5.7
4.5
Low Gm/Ibias
XC with offset
WangGuggenbuhl
0.5
850
3.2
8
High consumption
XC with bias
offset
Szczepanski
0.5
1000
2.5
4.5
Accurate control of Gm with bias
offset voltage
Adaptative
NedungadiViswanathan
Small
range
1000
2.5
7.5
Small linear range even for high bias
current
Table 4. Performance of basic HF transconductors in CMOS 350 nm technology.
A first prototype has been submitted to the AMS MPW run in July 2009. Two linear amplifier stages
have been implemented based on:

HF transconductor with emitter degeneration

HF cross-coupled transconductor (Szczepanski)
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Figure 3. Amplifier based on Szczepanski HF cross-coupled transconductor.
c. Non-linear amplification for dynamic range compression (David Gascon)
An alternative method to cover the dynamic range is a compressor at the very front end part of the electronics.
An adequate non-linear amplification of the PMT signal would make the job. After A/D conversion the pulse
would be reconstructed using digital signal processing. In high energy physics, bilinear (Figure 4) or multilinear
functions have been studied [FermiProject] and Error! Reference source not found..
y=a1·x
y=a2·(x-bx1)+by1
ADC output
2n
by1
for x<bx1
for xbx1
a2
a1
bx1
6000 Input [phe]
Figure 4. Bi-linear transfer function
Implementation of a compressor with the required signal range (up to 1-2 V) looks like impossible in voltage
mode, because recovery time of a differential pair would be to large for a fast amplifier.
A current mode implementation is being studied. Very low input impedance current conveyor, based on a
regulated cascade common gate transistor is followed by a hard clipping current mode stage. A first prototype
of the core of the compressor has been also included in the same CMOS AMS MPW run. Figure shows the layout.
Figure 5. Layout of the first prototipe of the compressor
Figure 6 shows the result of transient simulation of the compressor. Output signals for input pulses between 30
uA and 20 mA (peak current).
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Figure 6. Output of the compressor
The transfer function, voltage and charge, is depicted in Figure 7.
Figure 7. Transfer function of the compressor: voltage (top) and charge (bottom)
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Although the prototype submitted should allow testing the principle, several points need further R+D:
-
Control of the deep saturation of the hard-limiter
-
Single ended: differential conversion will be needed
Taking into account the power consumption (about 100 mW) and the complexity (area) of the circuit it would be
more reasonable to think on implementing the compressor on a dedicated ASIC.
Then, natural choice would be a BiCMOS technology to exploit the features of SiGe bipolar (speed and low
noise). The compressor could be placed near the PMT, working also as a ultra-low noise amplifier. An amplifier
with a similar input stage exhibiting ultra-low noise (0.35 nV/sqrt(Hz)) has been recently reported [Newcomer
TWEPP09]; the amplifier is implemented in an IBM BiCMOS technology.
The advantage of a non-linear compressor comes in the cost reduction in the digitizing electronics, a factor 2, as
duplication of the analogue memory channel needed any more.
Special care is needed to understand the pros and cons of the calibration of a non-linear transfer function:
actions?
7.
8.
Special requirements for other photo-sensors (D. Fink)
-
Suppression of SiPM long tails (recharge)…
-
?
Temperature dependence
Gain temperature coefficient: 0.1 %/K or 1%/K level?
V. Sampling frequency
Coordinator: Razmick Mirzoyan
Authors: Serge V.
Mid-size : < 3 ns time slice
…
VI. Pulse shape information
Coordinator: Emiliano
Authors: …
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…
VII. Dead time of the electronics
Coordinator:
Authors: …
…
Still open question
VIII. Power consumption
…
Open question : collect numbers from HESS and MAGIC
IX.
Summary
…
X.
References.
[1] D. Lachartre and F. Feinstein, “Application Specific Integrated Circuits For Antares Offshore
Front-End Electronics”, Nucl. Instrum. Meth. A442:99-104, 2000.
[2] A deep Sea Telescope for High Energy Neutrinos, arXiv:astro_ph/9907432.
http://antares2.in2p3.fr
[3] E. Delagnes et al., “SAM: a new GHz sampling ASIC for the H.E.S.S.-II Front-End Electronics.”,
NIM A, Volume 567, Issue 1, p. 21-26. E. Delagnes et al, “A Multigigahertz Analog Memory with
Fast Read-out for the H.E.S.S.-II Front-End Electronics”, Nuclear Science Symposium Conference
Record, 2006. IEEE( San Diego)
[4] Raffaello Pegna et al, « Performance of the Domino Ring Sampler in the MAGIC experiment”,
NIMA
[5] S. Ritt, Electronics for the MEG experiment, Nucl. Instr. and Meth. A 494 (2002) 520.
[6] Stefan Ritt, “Design and Performance of the 5 GHz Waveform Digitizing Chip DRS3”, Nuclear
Science Symposium Conference Record, 2007. IEEE( Honolulu)
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[7] G. Hermann et al, “A Smart Pixel Camera for future Cherenkov Telescopes”, arXiv:astroph/0511519 v1 17 Nov 2005.
[8] F. Druillole et al, “THE ANALOG RING SAMPLER: AN ASIC FOR THE FRONT-END ELECTRONICS
OF THE ANTARES NEUTRINO TELESCOPE”, IEEE Trans.Nucl.Sci.49:1122-1129,2002
[9] H. Bartko et al., Tests of a prototype multiplexed fiber-optic ultra-fast FADC data acquisition
system for the MAGIC telescope, / NIM A 548 (2005) 464–486
[10] D. Breton, E. Delagnes, Very High Dynamic Range and High Sampling Rate VME Digitizing
Boards for Physics Experiments IEEE TNS, VOL. 52, NO. 6, DECEMBER 2005, pp 2853-2860. D.
Breton, E. Delagnes, “14-Bit and 2GS/s Low Power Digitizing Boards for Physics Experiments”,
Nuclear Science Symposium Conference Record, 2006. IEEE( SAn Diego)
[11] H. Bartko et al., “ FADC Pulse Reconstruction Using a Digital Filter for the MAGIC Telescope ”,
arXiv:astro-ph/0506459v1 20 Jun 2005.
[12] E. Delagnes et al., A Low Power Multi-Channel Single Ramp ADC With Up to 3.2 GHz Virtual
Clock, IEEE TNS Volume 54, Issue 5, Part 2, Oct. 2007 Page(s):1735 – 1742
[13] S. Kuboki, « NonLinearity Analysis of Resistor Strings A/D converters », IEEE,TCS
Vol29,n°6,june 82
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