Download High Step-up Boost Converter Integrated With a Transformer

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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 4, APRIL 2012
High Step-up Boost Converter Integrated With
a Transformer-Assisted Auxiliary Circuit Employing
Quasi-Resonant Operation
Ki-Bum Park, Member, IEEE, Gun-Woo Moon, Member, IEEE, and Myung-Joong Youn, Senior Member, IEEE
Abstract—Stacking an auxiliary step-up circuit on top of a boost
converter is one of the most attractive structures for nonisolated
high step-up applications. In this paper, in order to avoid the large
input current ripple of coupled-inductor-based circuits, an auxiliary step-up circuit is integrated via an additional transformer and
its balancing capacitor. A voltage-doubler is adopted as an auxiliary
step-up circuit, which is inherently suitable for high-voltage applications due to its simple structure and low-voltage stress. Moreover, the transformer leakage inductor and the balancing capacitor constitute a resonant tank so that the quasi-resonant operation
makes the current sinusoidal. As a result, a reduced switch turnOFF loss and reverse recovery of the diode can be expected. The proposed converter is verified with a 24 V input, 160 W – 200 V output
prototype.
Index Terms—Coupled-inductor, high step-up converter, quasiresonant voltage-doubler.
I. INTRODUCTION
OR battery-powered systems, electric vehicles, fuel cell
systems, and photovoltaic systems, where low-voltage
sources need to be converted into high voltages, the demand for
nonisolated high step-up dc–dc conversion techniques are gradually increasing [1]–[6]. A classic boost converter is widely
used due to its simple structure and its continuous input current. However, it is hard to achieve a high-voltage conversion
ratio with just a plain boost converter, since the parasitic resistance of the circuit causes a severe loss as the duty cycle is
increased which limits the step-up gain [7]. Especially in high
output voltage applications, high-voltage stress on switches and
diodes degrades the performance of devices, causing a severe
hard switching loss, a conduction loss, and a reverse recovery
problem [8]–[10]. Moreover, an increased duty cycle to obtain
a high step-up gain has a detrimental effect on the dynamic
performance of a boost converter [7]. Therefore, to relieve the
F
Manuscript received June 23, 2011; accepted September 14, 2011. Date
of current version February 20, 2012. This paper was presented in part at
the IEEE ECCE, 2010, under the title “High step-up boost converter integrated with voltage-doubler.” Recommended for publication by Associate Editor
M. Vitelli.
K.-B. Park is with the Power Electronic Systems Group, ABB Corporate Research, Segelhofstrasse 1K 5405 Baden-Daettwil, Switzerland (e-mail:
[email protected]).
G.-W. Moon and M.-J. Youn are with the KAIST, Electrical Engineering, 373-1, Kuseong-dong, Yuseong-gu, Daejeon 305-701, Republic of Korea
(e-mail: [email protected], [email protected]).
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TPEL.2011.2170223
abovementioned limitations on boost converters in high step-up
applications, it is necessary to reduce the operating duty cycle
and distribute the voltage stresses across the devices allowing
for the use of low-voltage high-performance devices.
Till now, various types of step-up techniques based on a boost
converter have been developed [11]–[29]. Cascading a boost
converter is a simple way to achieve a high step-up gain by
employing more components. However, an increased number
of powering processes degrades the efficiency and the burden of
a high-voltage stress still remains [11], [12]. Coupled-inductor
boost converters are favorable candidates due to their simple
structure. However, their input current ripple is large due to a
coupled–inductor effect and an additional voltage clamp circuit for the switch and diode is required [13]–[20]. A voltagemultiplier cell or a switch-capacitor circuit can also be useful
to raise the step-up gain in collaboration with basic topologies [21]–[24]. However, when a higher output voltage is required, the number of step-up stages is increased, which requires more components. In addition, a current snubber is also
required to suppress the excessive peak current for charging
the capacitors. In terms of isolated type converters, current-fed
converters easily offer a high step-up gain through the turn ratio
of the transformer, which makes them inherently suitable for
high step-up applications [25], [26]. As a price for utilizing a
transformer, an additional voltage snubber is needed to limit the
switch voltage spikes caused by the transformer leakage inductance. In addition, an auxiliary circuit for operation below 0.5
duty cycles is required.
Among the abovementioned high step-up circuits, a coupledinductor-based circuit seems to be one of the most suitable candidates in low-to-medium power applications due to its simple
structure. In order to remove the large input current ripple and
improve the performance of this circuit further, an alternative
structure, which is based on a boost converter integrated with a
transformer assisted auxiliary step-up circuit, is investigated in
this paper. Furthermore, a quasi-resonant operation is adopted
to reduce the switch turn-OFF loss caused by the large switch
current in a high step-up converter, without additional circuitry.
II. PROPOSED HIGH STEP-UP CONVERTER
In order to further raise the step-up gain of a boost converter, alternative structures which combines a boost converter
with an auxiliary step-up circuit in series have been developed [15], [17], [27]–[29]. Proper selection of an auxiliary module can offer advantages such as high step-up capability, design
0885-8993/$26.00 © 2011 IEEE
PARK et al.: HIGH STEP-UP BOOST CONVERTER INTEGRATED WITH A TRANSFORMER-ASSISTED AUXILIARY CIRCUIT
1975
Fig. 2. Proposed high step-up boost converter integrated with voltage-doubler
as auxiliary step-up circuit, employing quasi-resonant operation.
Fig. 1. High step-up converter employing auxiliary step-up circuit on top of
boost converter. (a) Conventional coupled-inductor-assisted auxiliary circuit.
(b) Proposed transformer-assisted auxiliary circuit.
flexibility, and distributed voltage stress across devices which
allows for low voltage high performance devices. Among them,
the coupled-inductor-assisted auxiliary step-up circuit shown in
Fig. 1(a) is promising due to its simple structure, having only
one switch and an easy high step-up capability utilizing the
turn ratio of the coupled-inductor [15], [17]. However, these
structures also suffer from some side effects that come from the
coupled-inductor as follows.
As the output voltage of the auxiliary circuit is increased,
more power is driven by the coupled-inductor to the secondary
side. Since the input current consists of the boost inductor current, ILb , and the reflected auxiliary circuit current, nIaux , the
input current ripple can get considerably larger (‘n’ is the turn
ratio of the coupled-inductor). That is, as n increases, the input current ripple increases, and more input filtering might be
required, which has a detrimental effect on the total efficiency.
Moreover, since this input current flows through the switch, a
large input current ripple could cause a large switch turn-OFF
loss.
In terms of the size of the coupled-inductor in high stepup applications, it can be rather large when compared with
a plain inductor, since it requires many secondary turns and
a large ac-current being superimposed to a dc-current flows
through the windings. That is, the coupled-inductor needs to be
designed like a flyback transformer [30]. So that the coupledinductor size is not increased too much, a smaller inductance
design is one option. However, this increases the peak inductor
current in return, which results in an even higher switch turnOFF loss. As the input current increases, these drawbacks could
impose a greater burden on the magnetic component design and
efficiency.
In order to relieve aforementioned drawbacks of coupledinductor-based circuits, a transformer-assisted auxiliary high
step-up circuit, which has a continuous input current, is introduced in this paper as shown in Fig. 1(b). In addition, a
quasi-resonant operation utilizing the transformer leakage inductance is adopted in part [31]–[34], which provides partial
soft-switching characteristics for both the switches and the
diodes. Fig. 2 shows the proposed high step-up converter, and
its main features are as follows.
A. Voltage-Doubler as an Auxiliary Step-up Circuit
Various types of rectifiers can be adopted as an auxiliary stepup circuit. Among them, a voltage-doubler is inherently suitable
for high-voltage applications due to its simple structure, which
consists of two diodes and two capacitors, and due to its lowvoltage stress on devices, which is clamped to the output voltage
of the voltage-doubler. Therefore, they are widely adopted in
many topologies as a part of the circuit [17], [19], [27], [29],
[31], [35], [37]. For the proposed converter, a voltage-doubler is
also employed as an auxiliary step-up circuit, but it is integrated
with a boost converter in a different manner that allows for the
following distinctive properties.
B. Separated Boost-Inductor and Transformer
In the proposed circuit, unlike a coupled-inductor-based circuit, the interface between the boost converter and the voltagedoubler is accomplished by an additional transformer, which
also contributes to the step-up gain by means of the turn ratio n.
Since a square voltage waveform, i.e., an ac voltage, is applied
across the switch Q, the transformer can be inserted in parallel
with Q. Then, the balancing capacitor, CR , is inserted into the
primary side of the transformer to make up for the flux-balance
of the transformer. Thereby, the voltage-doubler is coupled with
the boost converter by sharing a common switch. Therefore,
by the switching action of Q, both the boost converter and the
voltage-doubler are operated at the same time.
Compared with the large input current ripple of the coupledinductor assisted boost converters in [13]–[17], the proposed
converter maintains a continuous input current of ILb , which
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 4, APRIL 2012
could require less input filtering. In other words, the proposed
circuit decouples the large ac-current from the input side to
the transformer at the price of additional components, i.e., the
transformer and the balancing capacitor. As a result, the input
current ripple becomes continuous, and each inductor and transformer can be designed optimally compared with the coupledinductor [30].
C. Quasi-Resonant Operation
Although the large input current ripple is removed, it is transferred to the transformer and this large current ripple still flows
through the switch in addition to the boost inductor current
ILb , which could result in a high turn-OFF loss. To relieve
this, a quasi-resonant operation, which considerably reduces
the switch turn-OFF current, is adopted as follows.
In a separated transformer structure followed by the voltagedoubler, the leakage inductance of the transformer, Llkg , and
the balancing capacitor, CR , can constitute a resonant tank similar to that of the series resonant type converters in [31]–[34].
The quasi-resonant operation between Llkg and CR makes the
current on the transformer primary side and the voltage-doubler
sinusoidal during the switch-ON state. Owing to this sinusoidal
current, the switch turn-OFF current can be considerably reduced
resulting in less turn-OFF loss [36]. Moreover, since the sinusoidal current guarantees slow di/dt, the reverse recovery on the
diode of the voltage-doubler can also be alleviated.
Fig. 3.
Key waveforms in below-resonant region (TR /2 < DTS ).
III. OPERATION PRINCIPLES
The proposed converter combines the operations of a boost
converter and a voltage-doubler, with the common switching function of Q, employing pulse-width modulation (PWM).
Since the voltage-doubler utilizes the quasi-resonant operation
between Llkg and CR , its operation can be divided into two regions according to the relationship between the resonant period,
TR , in (1) and the duty cycle, D. That is, the above-resonant
(AR) region, where TR /2 > DTS , and the below-resonant (BR)
region, where TR /2 < DTS , which is similar to conventional resonant converters [31]–[34]. The detailed operation is presented
as follows.
(1)
TR = 2π Llkg CR
.
A. Below-Resonant Region (TR /2 < DTS )
The key waveform and the topological states in the BR region
are shown in Figs. 3 and 4, respectively.
Mode 1 [t0 ∼ t1 ]: Q is in the ON-state and VS is applied
to the boost inductor LB . The boost inductor current, ILb ,
flows through Q and is increased linearly. At the same time,
the voltage-doubler is operated with the common switching action of Q. The powering path from CR to the lower output of
the voltage-doubler, Vo 2 , is formed through the transformer,
Q, and Do 2 , as represented by the dotted line. The Llkg and
CR constitute a resonant tank and derive a powering current
with a sinusoidal shape. The resonant capacitor voltage, VCr , is
decreased. The switch current, IQ , comprises ILb and the transformer primary current of the voltage-doubler, Ilkg . Since Do 2 is
turned-OFF with a very slow slope of ID o2 , the reverse recovery
can be minimized. Do3 is blocked by Vo 2 + Vo 3 .
Mode 2 [t1 ∼ t2 ]: Since TR /2 is shorter than the switch oninterval DTS , the resonant operation is finished at t1 before Q
is turned-OFF. Only ILb flows through Q. Therefore, the switch
turn-OFF loss is only affected by ILb . Since no current flows
through CR , VCr keeps its value during this interval.emphasis
Mode 3 [t2 ∼ t3 ] : Q is turned-OFF at t2 , then ILb flows through
Do1 . Meanwhile, the voltage-doubler starts to conduct in the
opposite direction. That is, the resonant powering path from the
output of the boost converter, Vo 1 , to the upper output of the
voltage-doubler, Vo 3 , is formed through Do 1 , the transformer
and Do 3 , as represented by the dotted line. Therefore, by the
resonant operation between Llkg and CR , ID o3 is increased.
Since the resonant current flows through Do 1 in the opposite
direction of ILb , ID o1 is decreased accordingly. Do 2 is blocked
by Vo 2 + Vo 3 .
Mode 4 [t3 ∼ t4 ]: ID o1 reaches zero at t3 . Then all of ILb
flows through the transformer and Do 3 of the voltage-doubler.
The ILb charges CR , increasing VCr linearly. The Do 1 is blocked
by Vo 1 -VCr -Vo 3 /n, which is slowly decreased. The same amount
of change can be observed in VQ .
Mode 5 [t4 ∼ t5 ]: At t4 , VCr is increased enough to conduct
Do 1 again. In this mode, unlike mode 3, the powering path from
Vo 3 to Vo 1 , is formed through Do 3 , the transformer and Do 1 by
PARK et al.: HIGH STEP-UP BOOST CONVERTER INTEGRATED WITH A TRANSFORMER-ASSISTED AUXILIARY CIRCUIT
1977
Fig. 4. Topological states of below-resonant region. (a) Mode 1 [t0 ∼ t1 ]. (b) Mode 2 [t1 ∼ t2 ]. (c) Mode 3 [t2 ∼ t3 ]. (d) Mode 4 [t3 ∼ t4 ]. (e) Mode 5 [t4 ∼
t5 ]. (f) Mode 6 [t5 ∼ t6 ].
the resonant operation between Llkg and CR . In other words, the
flow of ILb is shifted slowly in the resonant way from ID o3 to
ID o1 . Here, the reverser recovery on Do3 can be reduced by the
slow slope of ID o3 .
Mode 6 [t5 ∼ t6 ]: ID o3 reaches zero at t5 . Then all of ILb flow
through Do 1 . Since there is no current flowing through CR , VCr
keeps its value during this mode. As D is increased, modes 5
and 6 gradually fade and disappear.
B. Above-Resonant Region (TR /2 > DTS )
Operation in the AR region is similar to that of the BR region
except for mode 2 of the BR region, where ILb flowing solely
through Q, is ignored since TR /2 is longer than DTS . The key
waveform and the topological states in the AR region are shown
in Figs. 5 and 6, respectively. Since some of the topological
states are the same as those of the BR region, only the different
topological states, the intervals t0 ∼ t1 and t2 ∼ t3 , are presented
in Fig. 6. The topological states of the intervals t1 ∼ t2 , t3 ∼ t4 ,
and t4 ∼ t5 in the AR region correspond to Fig. 4(a), 4(c), and
4(d), respectively.
It is noted that, in the AR region, since Q is turned-OFF while
still powering through the transformer, the switch current at the
turn-OFF instant of t2 comprises ILb and Ilkg . Therefore, the
turn-OFF loss can be increased compared with that in the BR
region where only ILb flows at the switch turn-OFF instant.
Fig. 5.
Key waveforms in above-resonanat region (T r /2 > DT s ).
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Fig. 6.
IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 4, APRIL 2012
Topological states of above-resonant region. (a) t0 ∼ t1 . (b) t2 ∼ t3 .
on Do 2 and Do 3 are Vo 2 +Vo 3 , i.e., nVS /(1-D). That is, the
voltage stress on the voltage-doubler is n times higher than that
of the boost converter. Since the voltage-doubler provides n/(1
+ n) of the total output voltage, the transformer handles n/(1+n)
of the total power accordingly. The voltage ripple and the peak
voltage stress of VCr are expressed as (7) and (8), respectively.
VCr
VCr
Fig. 7.
Transformer turn ratio n according to a variation of M.
IV. ANALYSIS AND CHARACTERISTICS
A. Input-Output Voltage Gain
For the sake of analysis, assuming the ripple of VCr is ignored and using a flux-balance on the boost inductor and the
transformer, the following voltage equations are obtained:
1
VS
1−D
= nVS
Vo1 =
Vo2
nD
VS
Vo3 =
1−D
1+n
VO =
VS
1−D
VCr avg = VS .
nIO TS
CR
=
rp
= VCr
p eak
ID o2
IQ
p eak
≈ Iin
p eak
(3)
≈
(4)
IQ
rm s
(6)
The Vo 1 is the same as the output voltage of a classical boost
converter and the voltage-doubler provides a voltage that is
n times higher, nVo 1 ( = Vo 2 + Vo 3 ). Fig. 7 shows the required turn ratio n according to the variation of D and the inputoutput voltage conversion ratio M. When D becomes zero, the
voltage-doubler does not operated and ILb flows through all of
the series-connected diodes, Do 1 , Do 2 , and Do 3 . That is, Vo 2
and Vo 3 become zero and VO follows VS in the same manner as
a conventional boost converter.
B. Voltage and Current Stress on the Device
In the boost converter, the voltage stresses on Q and Do 1 are
Vo 1 , i.e., VS /(1-D). In the voltage-doubler, the voltage stresses
+
VCr rp
nIO TS
= VS +
.
2
2CR
(8)
Since Co 1 , Co 2 , and Co 3 are connected in series, the average
current of ID o1 , ID o2 , and ID o3 is the same as IO . The peak
current of ID o1 is the same as the turn-OFF current of IQ , which
is especially high in the AR region due to the remaining resonant
current at the turn-OFF instant. The peak current of ID o3 is similar
to that of ILb reflected to the transformer secondary. Assuming
that DTS ≈ TR /2, the peak current stresses on Do 2 and IQ can
be expressed as in (9) and (10), respectively.
(2)
(5)
avg
(7)
≈ IO
≈
1
TS
TR
2
0
≈
avg
πTS IO
πIO
≈
TR
2D
+
nπTS IO
TR
M (π + 2D − πD) − π
2D
(9)
nπTS IO
sin (ωR t) + Iin
TR
IO
(10)
2
avg
dt
1
{M 2 −2M + 1+(16M 2 − 14M ) D−9M 2 D2 }.
8D
(11)
C. ZCS on the Diodes
The diode currents in the voltage-doubler always flow through
Llkg that provides a current snubbing effect. Therefore, the reverse recovery on Do 2 and Do 3 can always be reduced by a
slow di/dt. The steepest slopes of ID o2 and ID o3 can occur at the
switching transition in the AR region as shown in Fig. 5. Equations (12) and (13) represent the decreasing di/dt of ID o2 and
ID o3 , respectively, which are the same. In order to sufficiently
PARK et al.: HIGH STEP-UP BOOST CONVERTER INTEGRATED WITH A TRANSFORMER-ASSISTED AUXILIARY CIRCUIT
Fig. 8.
(a) Switch peak current stress and (b) switch rms current according to a function of M.
Fig. 9.
Operating duty cycle D according to variations of VS and n.
Fig. 10.
Area-product, AP , of transformer according to variation of VS .
Fig. 11.
Resonant current waveforms according to a variation of TR .
1979
reduce the reverse recovery, di/dt should be at least less than
100 A/μs [9].
dID o2
VCR (t2 ) + Vo3 /n
1
=
=
dt
nLlkg
nLlkg
1
IO
VS −
1−D
CR FS
(12)
dID o3
Vo1 + Vo2 /n − VCR (t5 )
=
dt
nLlkg
1
1
IO
VS −
=
.
nLlkg 1 − D
CR FS
(13)
On the other hand, in the BR region, where the half-period resonant operation between Llkg and CR provides an extremely slow
slope on ID o2 , zero-current-switching (ZCS) can be achieved on
Do 2 , minimizing the reverse recovery.
In the case of Do 1 , during the switch-OFF state, ID o1 is decreased to zero and then it is increased again as described in the
mode analysis in the BR region of Section II. This operation
is also caused by the resonant operation between Llkg and CR ,
therefore, it depends on TR and DTS . In both the BR and AR
regions, as TR and DTS get smaller, there is a greater chance of
reincreasing ID o1 . In this case, when Q is turned-ON, an abrupt
change in ID o1 occurs causing a reverse recovery. Unless ID o1
is increased again, a reverse recovery on Do 1 will not occur.
V. DESIGN CONSIDERATION
In general, the power level of a certain topology is mainly
determined by the component count and rating. That is, the more
switches are employed, the higher the power level. The proposed
converter utilizes only one switch, therefore, it is expected to be
suitable for low-to-medium power applications.
To illustrate the design procedure for the proposed circuit, an
18 ∼ 30 V input, 200 V output, 160 W prototype converter is
presented. The required input-output voltage gain M is varied
from 6.7 ( = 200/30), for a maximum input of 30 V, to 11.1 ( =
200/18) for a minimum input of 18 V. The nominal input voltage
is 24 V, for which the required gain M is 8.3 ( = 200/24).
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 4, APRIL 2012
TABLE I
EXPERIMENTAL PARAMETERS
Fig. 12.
Experimental waveforms at VS = 24 V with full load condition.
A. Transformer Turn Ratio and Duty Cycle
In the proposed converter design, the selection of a switch,
which is burdened by the sum of the boost inductor current and
the resonant current, is primarily considered in terms of cost and
efficiency. As presented in (11) and Fig. 8(b), for the same M,
the rms value of the switch current is slowly decreased as the
duty cycle is increased. On the other hand, the switch voltage
stress, Vo1 = VS /(1- D), is decreased with a decrease in the duty
cycle, which leads to the use of a lower-voltage switch having a
smaller on-resistance. However, a smaller duty cycle results in a
larger turn ratio n as show in Fig. 7, which increases the voltage
stress of the voltage-doubler. Therefore, a duty cycle should be
selected to accommodate as low a voltage stress on the switch as
possible while not increasing the burden of the voltage-doubler
too much.
By selecting the transformer turn ratio n = 3.5, the duty cycle
varies from 0.4 ∼ 0.6 in response to the VS change from 30 V
PARK et al.: HIGH STEP-UP BOOST CONVERTER INTEGRATED WITH A TRANSFORMER-ASSISTED AUXILIARY CIRCUIT
Fig. 13.
1981
Experimental waveforms at VS = 18 V with full load condition.
to 18 V, as can be seen in Fig. 9. The ratio of the boost converter
output, Vo 1 , to the voltage-doubler output, Vo 2 + Vo 3 , is always
1 : n + 1 regardless of the input variation, as can be seen
in (2)–(5). When n = 3.5, Vo 1 and Vo 2 + Vo 3 become about
45 V and 155 V, respectively. Therefore, 100 V devices for the
boost converter and 200 V devices for the voltage-doubler are
available.
B. Inductor and Transformer
The design of the boost inductor is the same as those of
conventional ones. Assuming the current ripple to be 15% of
the input current 6.7 A, LB is designed for 120 μH [30].
Since the balancing capacitor, CR , is inserted into the primary side and the voltage-doubler capacitors, Co 2 and Co 3 , are
located in the secondary side, no dc-current can flow through
the transformer by the charge-balance of the capacitors, even
though the voltage applied to the transformer is asymmetrical.
Therefore, the magnetizing current has no dc-offset, which is
beneficial for the transformer design [30].
Normally, the area-product AP method can be used to predict the size of the magnetic core [30]. The AP represents the
product between a cross-section area and the window area of
the magnetic core. In the case of the proposed converter, the
AP of one transformer can be obtained as in (14), where Ku
is the window utilization factor, J is the current density, and
Bm ax is the maximum flux density. Assuming Ku to be 0.3, J to
be 300 A/cm2 , Bm ax to be 0.1 T, IO to be 0.8 A, and FS to be
100 kHz, the AP of the transformer according to the function of
VS is illustrated in Fig. 10, where the dot represents the case of
n = 3.5. The AP is varied according to a change in VS and the
maximum AP is 0.73 cm4 in the case of VS = 24 V.
D (VO (1 − D) − VS ) IO
AP =
Bm ax FS Ku J
1
π2
+
.
8D 1 − D
(14)
C. Resonant Tank
Fig. 11 shows the current waveform according to TR . In the
AR region, the switch turn-OFF loss is increased. On the other
hand, in the BR region, the switch turn-OFF loss is reduced
and Do 2 achieves a zero-current-switching (ZCS) turn-OFF that
minimizes the reverse recovery of the diode. However, the current stress and the conduction loss of the devices are increased.
Therefore, TR should be designed around the midpoint, TR /2 =
DTS , to achieve ZCS of the diode while minimizing the switch
turn-OFF loss and the conduction loss. To be designed at this
point, once Llkg is obtained from the fabricated transformer,
1982
Fig. 14.
IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 4, APRIL 2012
Experimental waveforms at VS = 30 V with full load condition.
Llkg can be set as it is and CR can then be selected as in (15).
CR =
D2 TS2
.
π 2 Llkg
(15)
With respect to the reverse recovery of the diode, Llkg should
also be considered as a current snubber. The minimum Llkg
which guarantees di/dt below 100 A/μs is approximately obtained at 150 nH from (10). If the obtained Llkg from the fabricated transformer has a larger inductance than 150 nH, a severe
reverse recovery will be relieved. As Llkg is increased, it is more
beneficial to reverse recovery. However, a larger Llkg reduces
CR to maintain the same TR as presented in (15), which lead to
a larger voltage ripple in VCr in return, as noted in (7).
VI. EXPERIMENTAL RESULTS
To verify the proposed converter, a 160 W prototype is implemented. The specifications and design parameters obtained
from the design example are presented in Table I.
Fig. 12 shows the experimental waveforms at a nominal input
of 24 V under the full load condition. The duty cycle is about
0.5 and it is similar to TR /2. The resonant operation between
Llkg and CR makes Ilkg sinusoidal and only the boost inductor
current ILb flows through the switch Q at the turn-OFF instant,
resulting in a reduced switch turn-OFF loss. That is, although
the peak switch current exceeds 16 A, the turn-OFF current is
under 9 A. Moreover, both Do 2 and Do 3 achieve ZCS turnOFF, which alleviates the reverser recovery. The boost converter
output Vo 1 is about 50 V. Therefore, the voltage stresses on Q
and Do 1 are under 100 V, including the voltage spikes caused
by parasitic inductances, which allows for the use of a Schottky
diode for Do 1 . The voltage stresses on Do 2 and Do 3 are clamped
to Vo 2 +Vo 3 , at about 150 V. The input current, i.e., the boost
inductor current ILb , is continuous.
Figs. 13 and 14 show the experimental waveforms at VS =
18 V and VS = 30 V, respectively. In the case of VS = 18 V, the
duty cycle is increased to regulate VO and the circuit is operated
in the BR region of DTS > TR /2. Here, the switch turn-OFF
current is still the same as ILb . In the case of VS = 30 V, the
circuit is operated in the AR region, i.e., DTS < TR /2, and the
switch turn-OFF current is about 12 A despite the fact that ILb
is only about 7 A. In both cases, the reverse recoveries on Do 2
and Do 3 are sufficiently suppressed by the current snubbing
effect of Llkg . The voltage stress on the switch in the steadystate is still about 50 V regardless of the input variation. This
implies that the switch turn-ON loss is rarely affected by an input
change.
Fig. 15 shows the efficiency curves with respect to the variation of VS . Since the resonant tank is designed to satisfy the
condition DTS = TR /2 at VS = 24 V, the proposed circuit shows
high efficiency, over 93%, at this point along a wide load range.
In the case of VS = 18 V, where the converter is operated in
the BR region, the increased conduction loss caused by the increased input current would degrade the efficiency. On the other
hand, in the case of VS = 30 V, the operation in the AR region
increases the switch turn-OFF loss even though the conduction
loss is decreased by a low input current. Consequently, it is noted
that the efficiency of the proposed converter is affected by the
PARK et al.: HIGH STEP-UP BOOST CONVERTER INTEGRATED WITH A TRANSFORMER-ASSISTED AUXILIARY CIRCUIT
Fig. 15.
Measured efficiency.
resonant tank design. That is, the proper selection of a balancing
capacitor, CR , can improve the circuit performance at a certain
operating point, which in this experiment is the nominal 24 V
input, as expected.
VII. CONCLUSION
A coupled-inductor-assisted auxiliary step-up circuit is an attractive candidate in low-to-medium power applications due to
its simple structure and it can easily achieve a high step-up gain
by increasing the turn ratio of the coupled-inductor. However, it
has large input current ripple, which may require more input filtering, and large switch current could cause a high turn-OFF loss.
In order to remove the large input current ripple, an alternative
structure, where the auxiliary step-up circuit is integrated via
an additional transformer and its balancing capacitor, is introduced in this paper. A voltage-doubler is adopted as an auxiliary
step-up circuit, which provides a simple structure and low voltage stress. In addition, the transformer leakage inductor and the
balancing capacitor, followed by the voltage-doubler, constitute
a series resonant tank, and thereby the sinusoidal current can
considerably reduce the switch turn-OFF loss and the reverse
recovery on the diode. It is noted that other types of rectifiers
can also be integrated with a boost converter, when interfaced
by a transformer and a balancing capacitor.
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Ki-Bum Park (S’07–M’10) was born in Korea, in
1981. He received the B.S., M.S., and Ph.D. degrees
in electrical engineering from the Korea Advanced
Institute of Science and Technology (KAIST), Daejeon, Korea, in 2003, 2005, and 2010, respectively.
He is currently a Scientist with ABB Corporate
Research Center, Baden-Dättwil, Switzerland. His
research interests include power converters, multilevel inverter, server power supply, high power density adapter, battery management system, and display
driver circuit.
Dr. Park received the Second Prize Paper Award from the International
Telecommunications Energy Conference (INTELEC) 2009 and the Third Prize
Paper Award from the Energy Conversion Congress and Exposition (ECCE)Asia 2011.
IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 4, APRIL 2012
Gun-Woo Moon (S’92–M’00) received the M.S. and
Ph.D. degrees in electrical engineering from the Korea Advanced Institute of Science and Technology
(KAIST), Daejeon, in 1992 and 1996, respectively.
He is currently a Professor in the Department of
Electrical Engineering, Korea Advanced Institute of
Science and Technology (KAIST), Daejeon, Korea.
His research interests include modeling, design and
control of power converters, soft-switching power
converters, resonant inverters, distributed power systems, power-factor correction, electric drive systems,
driver circuits of plasma display panels, and flexible ac transmission systems.
Dr. Moon is a member of the Korean Institute of Power Electronics (KIPE),
Korean Institute of Electrical Engineers (KIEE), Korea Institute of Telematics
and Electronics (KITE), Korea Institute of Illumination Electronics and Industrial Equipment (KIIEIE), and Society for Information Display (SID).
Myung-Joong Youn (S’74–M’78–SM’98) was born
in Seoul, Korea, in 1946. He received the B.S. degree
in electrical engineering from Seoul National University, Seoul, in 1970, and the M.S. and Ph.D. degrees
in electrical engineering from the University of
Missouri, Columbia, in 1974 and 1978, respectively.
In 1978, he joined the Air-Craft Equipment Division, General Electric Company, Erie, PA, where he
was an Individual Contributor on Aerospace Electrical System Engineering. Since 1983, he has been a
Professor at the Korea Advanced Institute of Science
and Technology (KAIST), Daejeon, Korea. His research activities are in the areas of power electronics and control, which include the drive systems, rotating
electrical machine design, and high-performance switching regulators.
Dr. Youn is a member of the Institution of Electrical Engnieers, U.K., the
Korean Institute of Power Electronics (KIPE), the Korean Institute of Electrical Engineers (KIEE), and the Korea Institute of Telematics and Electronics
(KITE).