Download Laboratory Test Analysis of Power Semiconductors

Laboratory Tests for Energy and Automation Technique, Laboratory Test 8
1
Laboratory Test
8
Analysis of Power Semiconductors
1
Introduction
2
Description of different power semiconductors
2.1
2.2
2.3
2.4
Bipolar transistor
Gate turn-off thyristor (GTO)
Metal-oxide-semiconductor field-effect transistor (MOSFET)
Insulated-gate bipolar transistor (IGBT)
3
Calculation of losses
4
Doing the laboratory test
5
Laboratory test setup
6
Starting the laboratory test
7
Evaluation of the laboratory test
8
Questions for preparing the laboratory test
9
Literature
Appendix: Data sheets of power semiconductors
Laboratory Tests for Energy and Automation Technique, Laboratory Test 8
1
2
Introduction
The laboratory test deals with on-state and switching behaviour and with the on-state and
switching losses of power semiconductors.
In common, power semiconductors are defined as semiconductors designed for high offstate voltage (≥ 100 V) and for high current (≥ 10 A). These semiconductors are produced
since the mass production of monosilicon started (about 1960).
The semiconductors used in power electronics can be divided in the three types:
1.
2.
3.
semiconductors without control (diode, Diac)
controlled semiconductors, controlling the switching on but not the switching off
(thyristor, Triac)
fully controlled semiconductors, controlling the switching on and off (bipolar transistor,
GTO, MOSFET, IGBT)
In the laboratory experiment the fully controlled power semiconductors are tested. These are,
as defined above, bipolar transistor, gate turn-off thyristor, metal-oxide-semiconductor fieldeffect transistor and insulated-gate bipolar transistor. They are used in all application ranges
of the modern power electronics (rectifiers, frequency converters, switching-mode power
supply, servo drives, industrial drives, traction).
In this laboratory test the basic circuit for power semiconductor test is the step-down
converter with ohmic-inductive load, as shown in figure 1.
RL
LL
DF
Vd
VSt
Fig. 1 Step-down converter with ohmic-inductive load
If the ohmic-inductive load is completed with a dc voltage source, the circuit corresponds to a
dc motor in a stationary working point.
The semiconductors can also be divided by the charge carriers which are used for the
current flow.
The MOSFET is a unipolar device using only one type of charge carriers for the current flow.
In power electronics almost exclusively N-channel MOSFETS are used, so that the charge
carrier types are electrons. The MOSFET is a voltage controlled device which needs only a
small power for switching on. Because of the great mobility of the electrons and the resulting
very short switching times switching frequencies up until 100 kHz and above are possible. A
disadvantage of these devices is that a high current capability cannot be realized together
with a high off-state voltage capability. So, the MOSFETs are used in power ranges less than
10 kVA.
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The other three devices are bipolar semiconductors. This means, that both charge carrier
types, the majority carriers and the minority carriers, are used for the current flow. Based on
the physical switching behaviour the switching frequencies are limited to about 1 kHz for the
GTO, to about 5 kHz for the bipolar transistor and to about 20 kHz for the IGBT. But with
special doping of the corresponding semiconductor layer these power semiconductor devices
can be built for high power ranges (bipolar transistor, IGBT) and for very high power ranges
(GTO).
2
Description of the different power semiconductors
2.1
Bipolar transistor
The bipolar transistor is the oldest device in this laboratory test. It was developed 1948 in the
Bell Laboratories by Shockley, Bardeen and Brattain. Its circuit symbol is shown in figure 2.
C
B
E
Fig. 2 Circuit symbol of a bipolar transistor
Used as a switching device, in opposite to the use as a linear amplifying device, the
transistor has only two characteristic working points in the output characteristic diagram.
These are the “On” state and the “Off” state. It depends on the load on which working curve
the “On” or the “Off” state is reached (Figure 3).
IC
On
IB
Off
VB
VCE
Fig. 3 Working curves for different loads
The load current, which is the same as the collector current IC, is proportional to the control
current. This control current is the same as the base current IB. The connection between
these currents is the dc amplification factor B = IC/IB. Compared with small signal bipolar
transistors the power transistors have a relatively small dc amplification factor. B is in the
range of 20…100. Based on this there are very high control currents necessary for a high
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load current. To reduce the control currents mostly Darlington transistors are used in power
electronics. The principle of the Darlington circuit is shown in figure 4.
C
C
^
=
B
B
E
E
Fig. 4: Principle of the Darlington circuit
In figure 4 a transistor T1 with the dc amplification B1 is connected in common-collector circuit
with the transistor T2. With this the resulting dc amplification factor B is [3]:
B = B1 + B2 + (B1 • B2)
(1)
B ≈ B1 + B2
(2)
For the laboratory test a Darlington transistor is used. Hereby, both transistor T1 and
transistor T2 are on the same substrate. Such transistors are called monolithic integrated
Darlington transistors. Besides, in most cases a free-wheeling diode is integrated on the
substrate.
In figure 5 the construction of a monolythic integrated Darlington transistor can be seen.
B
E1
B1
E
KD1 AD1 B 2
SIO2
E2
p+
p
nn+
C
Fig. 5: Construction of a monolythic integrated Darlington transistor
In figure 5 the base layer is not scaled. The connections KD1 and AD1 are an integrated speed
up diode. This diode reduces together with the resistors R1 and R2 the switching time of the
transistor. Besides this the resistors R1 and R2 care for the tail current of the transistor T1 (Ic1
in the off state) not being amplified by the transistor T2. This reduces also the total tail current
[3]. The resistors result on one hand from doping the base layer p and the emitter region (R2)
and on the other hand through the groove of separation between T1 and T2 not being totally
drawn to the collector region. So, a small p-channel remains which is the resistor R1. Figure 6
shows the equivalent circuit of the construction in figure 5.
Laboratory Tests for Energy and Automation Technique, Laboratory Test 8
5
C
B
T1
T2
"D1"
R1
R2
E
Fig. 6: Equivalent circuit for figure 5
The bipolar transistor is a current controlled device. The base current IB controls the collector
current IC. So, with the base current the transistor is set in on or in off state. Simple switching
off leads to a long storage time of the charge carriers. This causes high losses during the
switching off. Therefore, the power transistor is controlled by a negative base current during
switching off. This current sucks the electrons out of the base region and thus reduces the
switch off time and by this the losses.
The different switching times are defined as:
td:
t r:
ts :
tf :
delay time
rise time
storage time
fall time
In data sheets frequently ton and toff are given. These are defined as:
ton = td + tr
toff = ts + tf
(turn on time)
(turn off time)
The curves of the collector current and the base current are shown in figure 7. Besides this
the characteristic parts of the switching times of a bipolar transistor are shown.
Laboratory Tests for Energy and Automation Technique, Laboratory Test 8
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iC
tr
90%
collector current
td
ts
10%
t
tf
ton
iB
90%
base current
toff
10%
t
-iB
Fig. 7: Collector current and base current for a bipolar transistor
2.2
Gate turn-off thyristor (GTO)
The GTO is a thyristor using a gate-cathode with a fine distribution. The GTO can be
switched off using a negative control current pulse. The GTO can be used for very high
power (more than 3000 kVA). Figure 8 shows the circuit symbol of a GTO.
C
G
A
Fig. 8: Circuit symbol of an GTO
The characteristic curve of the GTO is similar to the characteristic curve of a standard
thyristor. The GTO differs from a standard thyristor only by a controlled switching off. In
figure 9 the typical three regions are shown. These are the negative non-conducting zone,
the positive non-conducting zone and the conduction region.
Laboratory Tests for Energy and Automation Technique, Laboratory Test 8
IA
7
VT0
vT
VRRM
vD
VT0
VDRM
VA
Fig. 9: Characteristic curve of a GTO (same as standard thyristor)
The negative non-conducting zone is not used during the laboratory test. The device works
on the positive non-conducting zone if a positive anode-cathode voltage UAK is used but the
ignition (= switch on) has not been done. This is the so called off-state. If the GTO is ignited
by a positive current pulse into the gate the working point on the conducting voltage-current
characteristic is adjusted by the load.
As shown in figure 10 gate and cathode have a fine distributed finger structure. By igniting
the GTO in opposite to the standard thyristor the whole cathode area conducts at the same
time. With this the GTOs have smaller switch-on times and smaller rates of rise of forward
current compared with standard thyristors. The charge carriers can be sucked out of the
cathode because of the gate-cathode structure also during switching off the GTO with the
negative gate current.
C
n+
n+
n+
p
G
n+
n+
np
n+
p
n+
p
n+
p
n+
p
A
Fig. 10:
Construction of a GTO
The gate and the cathode tapes have a typical width of 100 μm to 400 μm. In spite of the
microstructure an inhomogeneous current distribution in the device cannot be avoided. This
can result in a local thermical overload in the current conducting zones which can destroy the
Laboratory Tests for Energy and Automation Technique, Laboratory Test 8
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device if there are high external voltages. Therefore, a limiting of the voltage increase as
announced in the data sheet is absolutely necessary.
The protection circuit is a RCD network and reduces the voltage increase du/dt during
switching off the GTO. The constant load current flows for a short time through the RCD
network and charges the capacitor Cs. The time constant of the RCD network is defined only
by the capacitor, because the protecting resistor is bridged by the diode Ds. The discharge
current of the capacitor is defined by the resistor Rs and it is superposed to the load current
of the GTO. The resistor Rs must be dimensioned in a way that the inrush current will not be
too much. In common, the RCD protecting circuit reduces the maximum switching frequency
and the efficiency of the power stage. Figure 11 shows the RCD network for a GTO.
L
CS
DS
RS
Fig. 11:
RCD network for a GTO
Switch on and switch off of a GTO can be explained by using an equivalent circuit based on
the two-transistor model (Shockley). In this model the GTO (and similar the thyristor) is
understood as a series-/parallel connection or a pnp- and a npn-transistor (figure 12).
a)
A
b)
P
P
N
G
P
N
c)
A
T1
N
P
G
A
P
N
G
T21
N
C
C
Fig. 12:
C
Equivalent circuit for a GTO
For the ignition of a GTO only a relative low current pulse is necessary (typical IGT ≥ 400 mA),
as in the positive non-conducting area only the junction J2 (transistor T2) is locked. For a
GTO being conductive there only charge carriers must be injected in the area P2 (base
current of T2). By the rising of the collector current IC2 and the base current of T1 the system
reaches the conductive state by themself. Simultaneous the collector current of T1 is the
base current of T2. If there is a negative control-current pulse connected with the gate (base
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current of T2) the collector current IC2 is reduced. Simultaneous this current is the base
current of T1. This feedback process continues until the current IT through the GTO becomes
zero. In the conductive state of the GTO the whole centre zone is flooded with charge
carriers. So for sucking out the charge carriers a high negative current pulse is necessary.
This current pulse is in the range of 20 % to 35 % of the thyristor current IT (= load current).
Figure 13 shows the curve of the ignition current for switching on and switching off a GTO.
tw2
gate current
gate voltage
diGdt
0,1·i
0,1 iGM
tw3
iGM iG
0,1·iGO
tw1
VGO
0,5·iGO
VGR
Vgr
iGO
diGdt
Fig. 13:
Ignition current of a GTO
The negative voltage VGK rises the blocking ability of the GTO.
2.3
Metal-oxide-semiconductor field-effect transistor (MOSFET)
The MOSFET is the only unipolar device in this laboratory test. The field-effect transistor was
described by Shockley in 1952. But it takes a long time after this to construct FET which
could fullfil the demands of practical use. The circuit diagram of a MOSFET is shown in figure
14.
D
G
S
Fig. 14:
Circuit diagram of a n-channel MOSFET
In the power electronics exclusively self-locking MOSFETs and almost exclusively n-channel
types are used. The self-locking n-channel MOSFETs are non-conducting without a positive
gate-source voltage VGS. The device keeps in the off-state until the voltage VGS reaches the
threshold voltage VGSth (BUZ 21: typ. VGSth = 3 V). With a rising control voltage VGS the
current flow through the FET rises in dependance of its transfer characteristic (figure 15).
MOSFETs are completely blocked for voltages below the threshold voltage. The drain
residual-current has a maximum of 4 mA.
Laboratory Tests for Energy and Automation Technique, Laboratory Test 8
10
25
20
15
I D/A
10
5
0
5
0
Fig. 15:
VGS/V
10
Transfer characteristic of a MOSFET (BUZ 21)
With negative values of VGS the ability of blocking cannot be rised. So, with a unipolar control
voltage the whole output characteristic ID = f(VDS) can be passed. Figure 16 shows the output
characteristic of a MOSFET.
40
10,0 V
A
VGS = 8,0 V
PD =
75 W
7,5 V
20,0 V
7,0 V
30
ID
6,5 V
20
6,0 V
5,5 V
10
5,0 V
4,5 V
4,0 V
0
0
2
4
6
8
10
V 12
VDS
Fig. 16:
Output characteristic of a MOSFET (BUZ 21)
The construction of the MOSFET used for this laboratory test (BUZ 21, SIPMOS family)
based on a drain metalization followed by a n+-substrate and a n--epitaxial layer. The
Laboratory Tests for Energy and Automation Technique, Laboratory Test 8
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thickness and the doping of this n--layer is dependend on the off-state voltage. The n+polysilicon gate laying above this n--layer is embedded in an isolating silicon dioxide (SiO2).
The source metalization (aluminium) covers the whole structure and connects the single
transistor cells in parallel. The construction is shown in figure 17.
G
S
isolation
p
Sn
n-channel
n
D
Fig. 17:
Construction of a MOSFET
Besides the MOSFET itself the equivalent circuit shows among the MOSFET itself its
parasitic elements (figure 18).
D
RD
RG
CGD
CDS
G
inverted
diode
CGS
RK
S
Fig. 18:
Equivalent circuit for a MOSFET with its parasitic elements
Because of the simplified model the capacitances in figure 18 cannot be measured in single.
Under neglecting the channel resistor RK and the n--epitaxial resistor RD the following
capacitances, which can be found in the datasheet, are defined:
Laboratory Tests for Energy and Automation Technique, Laboratory Test 8
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Ciss ≈ CGS + CGD
Coss ≈ CDS + CGD
Crss ≈ CGD
The capacitance CGD is also named Miller capacitance.
The combination of the resistors RK and RD roughly equals the drain-source resistor RDSon in
the on-state of the MOSFET:
RDson ≈ RD + RK
The diode in the equivalent circuit represents the pn junction from the p-source area to the n-epitaxial layer. This layer is the drain connection via n+-substrate. This results in an inversion
diode between drain and source. The characteristic values of this diode can be found in the
datasheet.
MOSFETs can be controlled almost without power. Energy is only needed for switching on. It
is needed primarily for the charging of the input impedance Ciss. The capacitance Ciss has to
be charged very quickly by switching on the MOSFET. The necessary currents have to be
delivered by the gate drive circuit.
As the MOSFET is a transistor the dv/dt and the di/dt are not specified. The protection of the
semiconductor is limited on the maximum values of VGsmax, VDcmax, IDmax and ϑJmax. The
voltage VDS is limited by using a Z-diode (DZ1 in figure 19). Its value has to be between the
voltage VDS delivered from the gate drive circuit and the maxium for VGS which is VGsmax = ±20
V. The gate is protected by the series connector RG. In connection with the input capacitance
RG is also protecting the MOSFET from highfrequency oscillations.
RL
LL
DF
Vd
DZ2
RG
DZ1
Fig. 19:
VSt
Protection circuits for a MOSFET
If inductive loads are switched negative voltage (Uind) has been induced. In this case high
voltages occur. For these high voltages the blocking ability of the semiconductor must be
defined. The Z-diode DZ2 protects the MOSFET against these too high voltages VDS. It is
connected between the gate and the drain connection. The value of this diode must lay
between the highest voltage at the device (UD + Uind) and the maximum value of VDsmax (BUZ
21: typ. VDsmax = 100 V). If there is a voltage higher than the Z-diode voltage during the offstate of the MOSFET, the MOSFET switches on and the high voltage decreases.
2.4
Insulated-gate bipolar transistor
The IGBT was developed at the beginning of the 1980s and it is available since the end of
the 80s. It is a combination of a bipolar transistor and a MOSFET. This also gave the name
to it. The lower output stage is realized by a bipolar transistor which is controlled by a
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MOSFET. So the control characteristic of an IGBT is similar to the control characteristic of a
MOSFET. The control electrode is named equivalent to this as gate. Figure 20 shows the
circuit diagram of an IGBT.
C
G
E
Fig. 20:
Circuit diagram of an IGBT
The construction of an IGBT is similar to the construction of a MOSFET. In case of the
MOSFET a n+-substrate and a n--substrate follow after the drain metallization. In opposite to
this the IGBT has a pure p-substrate. For this the IGBT belongs to the group of the minority
carriers. Here, by controlling the gate the charge carriers are injected from the highly doped
emitter area into the meagerly doped collector area. With this the conductivity of the
meagerly doped area rises up for several ten powers (conductivitcy modulation). The
construction of an IGBT is shown in figure 21.
emitter
gate
n+
n+
p+
nn+
p+
collector
Fig. 21:
Construction of an IGBT
The self blocked n-channel MOSFET is the driver for the pnp bipolar transistor. For this the
whole voltage loss VCE at the IGBT can be represented as the sum of the voltage losses of
the bipolar transistors pn diode and the voltage loss of the MOSFET driver. So the on-state
voltage loss of an IGBT can, in opposite to the voltage loss of a MOSFET, never be less than
the threshold voltage of a diode. That can also be seen in the output diagram of an IGBT
(figure 22).
Laboratory Tests for Energy and Automation Technique, Laboratory Test 8
300
15V
TC = 25V VGE = 20V
14
12V
250
200
10V
150
I C/A
PC = 600W
8V
100
7V
50
6V
0
0
1
2
3
4
5
6
7
8
9
10
VCE/V
Fig. 22:
Output diagram of an IGBT
The IGBT is like a MOSFET a voltage controlled device which can be controlled almost
without power. The controlling of the IGBT has the same features as the controlling of a
MOSFET.
The protection of the IGBT is limited on the values VGE and VCE. This protecting is similar to
the protection of a MOSFET. Because VCemax of the IGBT lies in the range of the bipolar
transistors a protection against high collector-emitter voltages is not necessary.
3
Calculation of the switching losses and the on-state losses
The total losses of a power semiconductor switch are composed by several parts:
switch-on losses
switch-off losses
on-state losses
reverse losses
driver losses
Pon
Poff
Pos
PR
PD
As an example the calculation of the losses of a bipolar transistor is shown. Hereby the driver
losses PD are neglected. Figure 23 shows the definition of the switching times.
Laboratory Tests for Energy and Automation Technique, Laboratory Test 8
iC
vCE
15
vCE
iC
VCE(sat)
iCR
t2
t1
Fig. 23:
t3
t
t4
Definition of the switching times
With respect to figure 23 the losses are:
t
*
VAV
P
t
t
t
1 1
1 2
1 3
1 4
= ⋅ ∫ VCE( sat ) ⋅ i C dt + ⋅ ∫ v CE ⋅ i C dt + ⋅ ∫ v CE ⋅ i CR dt + ⋅ ∫ v CE ⋅ i C dt
T t0
T t1
T t2
T t3
In the equation above are:
f = T-1:
t0 – t1:
t2 – t1:
t3 – t2:
t4 – t3 :
vCE:
iC:
VCEsat:
iCR:
switching frequency
conducting phase
switch off time
non-conducting phase
switch-on time
collector-emitter voltage
collector current
saturation voltage
reverse current
Due to the low reverse current the reverse losses PR can be neglected. They are:
t
1 3
P = ⋅ ∫ v CE ⋅ i CR dt
T t2
*
R
The simplified calculation uses linear current curves and linear voltage curves during switch
on and switch off as shown in figure 24.
With this assumption the following equation for the calculation of the losses is used:
*
=
PVAV
t ⎞ 1 ⎛
t ⎞
1
1 ⎛
⋅ (i C ⋅ VCE( sat ) ⋅ t flw ) + ⋅ ⎜ i C ⋅ v CE ⋅ off ⎟ + ⋅ ⎜ i C ⋅ v CE ⋅ on ⎟
T ⎝
2 ⎠ T ⎝
2 ⎠
T
Laboratory Tests for Energy and Automation Technique, Laboratory Test 8
16
In the equation above are:
iC:
collector current
VCEsat:
average saturation voltage
vCE:
maximum collector voltage
tflw = t1:
conduction phase
toff = t2 – t1:
switch off time
ton = t4 – t3:
switch on time
VCE
ton
tflw
toff
t
iC
90%
10%
td
Fig. 24:
tr
tflw
ts
tf
t
Simplified linear current curves and voltage curves for the bipolar transistor
The driver losses PD can be estimated as about 10 % of the collector losses P*AV. With this
the total average losses are:
*
PVAV ≈ 1.1⋅ PVAV
In the laboratory test the times above announced can be read on an oscilloscope. The
measured switch-on time ton can be compared with the theoretical value ton in the datasheet.
The switch off-time corresponds essentially to the fall time tf, so that toff ≈ tf.
The duration of the conductive phase is calculated to:
t on t off
−
2
2
t on + t f
≈ a⋅T −
2
t flw = a ⋅ T −
t flw
The factor a in the equations above is defined as duty cycle (figure 25).
Laboratory Tests for Energy and Automation Technique, Laboratory Test 8
17
iB
ton
toff
aT
(1-a)T
T
Fig. 25:
t
Definition of the duty cycle
The switch-on and switch-off times are equally divided between both switching conditions.
With the simplification shown above the on-state losses can be calculated with:
PD* =
1
⋅ VCE( sat ) ⋅ i C ⋅ t flw
T
The switching losses result in:
*
*
PS* = Pon
+ Poff
=
t + t off
1
⋅ v CE ⋅ i C ⋅ on
T
6
For the MOSFET and the IGBT the driver losses can be neglected. So the losses are:
*
PVAV ≈ PVAV
To calculate the losses of the GTO the equation above has to be used. Likewise the driver
losses can be neglected. For the theoretical comparison the on-state voltage of the GTO VF,
the switch-on time ton and the switch-off time toff have to be considered. The fundamental
frequency f = 1/T.
*
=
PVAV
t ⎞ 1 ⎛
t ⎞
1
1 ⎛
⋅ (i T ⋅ v F ⋅ t flw ) + ⋅ ⎜ i T ⋅ v AK ⋅ off ⎟ + ⋅ ⎜ i T ⋅ VAK ⋅ off ⎟
T
T ⎝
2 ⎠ T ⎝
2 ⎠
In the equation above are:
vF:
iT:
VAK:
tflw:
toff:
ton:
on-state voltage
on-state current
maximum anode-cathode voltage
conduction time
switch off time
switch on time
Laboratory Tests for Energy and Automation Technique, Laboratory Test 8
4
18
Realization of the laboratory test
The maximum switching frequencies of the power electronic devices can be found in the
respective data sheets. These values may not be exceeded during the laboratory test.
4.1
Measurements with constant duty cycle and variable frequency
The following quantities have to be measured:
1.
2.
3.
4.
frequency f
load current id using the voltage vshunt; here 1 V = 2 A
voltages at the power transistors
bipolar transistor and IGBT: vCE
MOSFET: vDS
GTO: vAK
switching times
For the bipolar transistor and the GTO the measurements have to be done in steps of 50 Hz.
For the IGBT and the MOSFET steps of 500 Hz should be used.
4.2
Measurements with variable duty cycle and constant frequency
The following quantities have to be measured:
1.
2.
3.
4.
duty cycle a
load current id using the voltage vshunt; here 1 V = 2 A
voltages at the power transistors
bipolar transistor and IGBT: vCE
MOSFET: vDS
GTO: vAK
switching times
The duty cycle should be changed in steps of 10 %.
5
Construction of the laboratory test
The next page shows the breadboard of the laboratory test. All connections in this figure are
already done. Only the measuring devices have to be connected.
There is in the left part the general supply, the fuse protection and the supply for additonal
device mounted. The bigger right part contains the parts transformer, load, power electronic
devices and puls-width-modulation generator. Also, there are the push-button switches to
switching on and off the electronics and power electronics of the laboratory test.
The laboratory test board is switched on with the push-button switch “Elektronik EIN”. The
adjusted frequency of the puls-width modulator is shown. The load circuit is set into operation
with the push-button “Leistungsteil EIN”. This is displayed by the control lamps of the general
supply.
The voltage Vd is generated using a bridge rectifier and a capacitor bank.
The ohmic-inductive load can be connected to the different power electronic devices by a
stepping switch. The appropriate control lamp of the semiconductor device is lighted.
Laboratory Tests for Energy and Automation Technique, Laboratory Test 8
m ains
fuses
RL
LL
id
DF
L1
fuses
19
vd
L2
vshunt
L3
load circuit
ele ktronics
sock ets
load circuit
PWM
PWM
PWM
bipolar
transistor
GTO
IGBT
Fig. 26:
ON
ON
OFF
OFF
duty cycle/
frequency
signal PWM
duty cycle,
frequency
Breadboard of the laboratory test (english version)
Einspeisung
Absicherung
LL
RL
DF
L1
Gesam tabsicherung
MOSFET
elektronics
emergency
OFF
PWM
id
ud
L2
L3
ushunt
Leistungsteil
PWM
BipolarTransistor
Elektronik
Steckdosen
Leistungsteil
Elektronik
EIN
EIN
AUS
AUS
PWM
PWM
PWM
GTO
IGBT
MOS-FET
Tastgrad/
Frequenz
NOT
AUS
Fig. 27:
Signal PWM
Tastgrad,Frequenz
Breadboard of the laboratory test (german version)
Important:
Before changing the semiconductor device the load circuit must be separated from the
supply with the push-button “Leistungsteil AUS”. The new power electronic device can now
be chosen and the load circuit is reconnected (“Leistungsteil EIN”). The supply of the
electronic parts must not be disconnected.
Laboratory Tests for Energy and Automation Technique, Laboratory Test 8
20
Using semiconductor specific drivers the control signal with variable frequency or variable
duty cycle is connected to the control electrodes (gates) of the power semiconductors. The
signal generator can be switched between variable frequency and constant duty cycle or
constant frequency and variable duty cycle (“Tastgrad/Frequenz”). The respective switch is
found above the LED display on the breadboard of the laboratory test. Both, duty cycle and
frequency can be adjusted with the multiturn potentiometer on the right side of the LED
display. For measurement the control signal can be tapped at the BNC jack “Signal PWM”.
The LED display shows the frequency of the control signal.
6
Starting and course of the test
1.
2.
3.
Check whether all automatic circuit breakers are switched on.
Switch on electronic supply with the push-button switch “Elektronik EIN”.
Choose the power semiconductor device with the stepping switch. The appropriate
device is displayed by a control lamp.
Switch on the load circuit with the push-button “Leistungsteil EIN”.
The on-state is displayed with the control lamps of the transformer. If the supply for
the electronic circuit is not ready the load circuit cannot set into operation.
The measurements described in the instrucion have to be done.
Change to another power semiconductor:
1.
Switch off load circuit (“Leistungsteil AUS”)
2.
Choose the power semiconductor (stepping switch)
3.
Switch on load circuit (“Leistungsteil EIN”)
Attention: Change power semiconductor only in the off state of the load circuit!
4.
5.
6.
The “NOT-AUS” button disconnects the complete laboratory test from mains. It should be
used only in emergency cases. Starting after pushing the “NOT-AUS” begins with point 1.
(see above).
7
Evaluation of the test
The losses of the several semiconductor devices have to be calculated with the measured
values. Hereby, the switching losses and the on state losses have to be separated.
Further theoretical reference calculation with the values from the datasheet have also to be
done.
The primary data of PV = function(f) and PV = function(a) have to be transferred in one
diagram for each power semiconductor. The differences between theory and practice have to
be discussed.
8
Questions for preparing the test
1.
Using the equivalent circuit please explain the internal feedback mechanism of a
GTO during switch on and during switch off!
2.
Why and how can a GTO be switched off though it is a four-layer device like a SRC?
3.
Why must a GTO, against a SCR, not unconditionally be protected against a di/dt too
high?
4.
what is the difference regarding the structure of the layers between MOSFET and
IGBT in general?
Laboratory Tests for Energy and Automation Technique, Laboratory Test 8
21
5.
Mark the working characteristics in the output characteristic IC = f(UCE) of a bipolar
transistor!
6.
The IGBT can be announced as a bipolar power transistor with MOSFET controlling.
Which consequences has this?
7.
In which parts can the total losses of a power semiconductor be divided? Hereby
which parts can be neglected?
8.
A three-phase transformer is described with 400V/42V. Calculate the output voltage
Ud of a B6 bridge with and without a smoothing capacitor. Hereby the three-phase
transformer works in Yd circuit.
9.
Calculate the losses in an ohmic load with RL = 3.2 Ω, if with Ud (see question 8) a
step-down chopper with a duty cycle of 30 % works. Neglect the losses in the
semiconductor.
10.
Show all switching times of a bipolar transistor in an appropriate iC(t) and iB(t)
diagram.
11.
Describe the function of the reducing network of a GTO.
9
Literature
[1]
Kahlen, H.:
Material of the course „Leistungselektronik I“.
University of Kaiserslautern, 4th edition 1997
[2]
Tietze, U.; Schenk, C.:
Halbleiter-Schaltungstechnik.
Springer-Verlag Berlin, Heidelberg, New York, 11th edition 1999
[3]
Macek, O.:
Schaltnetzteile, Motorsteuerungen und ihre speziellen Bauteile.
Hüthig-Verlag, 1992
[4]
Jäger, R.:
Leistungselektronik, Grundlagen und Anwendung.
VDE-Verlag, Berlin 1977
[5]
Heumann, K.:
Grundlagen der Leistungselektronik.
Verlag B.G. Teubner, 6th edition 1996