Download LTC6601-1

LTC6601-1
Low Noise, 0.5% Tolerance,
5MHz to 28MHz, Pin Configurable
Filter/ADC Driver
DESCRIPTION
FEATURES
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Pin Configurable Gain and Filter Response
Up to 28MHz
Few External Components Required
Resistors Trimmed to 0.5% Typical
Capacitors Trimmed to 0.5% Typical
Very Low Noise: 80dB S/N in 100MHz Bandwidth
Very Low Distortion (2VP-P):
1MHz: –100dBc 2nd, –123dBc 3rd
10MHz: –72dBc 2nd, –103dBc 3rd
Adjustable Output Common Mode Voltage
Rail-to-Rail Output Swing
Power Configurability and Low Power Shutdown
Tiny 0.75mm 20-Lead (4mm × 4mm) QFN Package
APPLICATIONS
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Differential Input A/D Converter Driver
Antialiasing/Reconstruction Filter
Single-Ended to Differential Conversion/Amplification
Low Voltage, Low Noise, Differential Signal
Processing
Common Mode Voltage Translation
The LTC®6601-1 is a very easy-to-use fully differential
2nd order active RC filter and driver. On-chip resistors,
capacitors, and amplifier bandwidth are trimmed to provide
consistent and repeatable filter characteristics.
The filter characteristics are pin-strap configurable. Cutoff
frequencies range from 5MHz to 28MHz. Gain is pin-strap
programmable between –17dB and +17dB.
A three-state BIAS pin is provided to adjust amplifier
power consumption. Select between high performance,
low power (50% power reduction), and standby modes
with the BIAS pin.
The LTC6601-1 is available in a compact 4mm × 4mm
16-pin leadless QFN package.
L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
Protected by U.S. Patents including 6271719.
TYPICAL APPLICATION
19MHz, 2nd Order Lowpass Filter. Gain = 6dB
Frequency Response
10
20
19
18
17
5
16
0
LTC6601-1
+
VIN
–
+
2
3V
–
15
14
3
3V
0.1μF
13
–
4
+
11
0.1μF
7
–5
–10
–15
12
5
6
VOUT
GAIN (dB)
1
8
9
10
–20
–25
–30
66011 TA01a
1
10
FREQUENCY (MHz)
100
66011 TA01b
66011f
1
LTC6601-1
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Note 1)
Total Supply Voltage (V + to V – ) ...............................5.5V
Input Voltage (Any Pin) (Note 2) ..V + + 0.3V to V – –0.3V
Input Current (VOCM, BIAS)..................................±10mA
Input Current (Pins 1, 5) (Note 2) ........................±20mA
Input Current (Pins 2, 4) (Note 2) ........................±30mA
Input Current (Pins 6, 20) (Note 2) ......................±15mA
Input Current (Pins 7, 8, 9, 10, 16, 17, 18, 19)
(Note 2)................................................................±10mA
Output Short-Circuit Duration (Note 3) ............ Indefinite
Operating Temperature Range (Note 4)....–40°C to 85°C
Specified Temperature Range (Note 5) ....–40°C to 85°C
Junction Temperature ........................................... 150°C
Storage Temperature Range...................–65°C to 150°C
C8
C7
C6
C5
IN4+
TOP VIEW
20 19 18 17 16
IN2+ 1
15 OUT–
IN1+
14 V+
2
BIAS 3
–
13 V–
21
7
8
9 10
C3
C4
6
C2
11 OUT+
C1
12 VOCM
IN4–
4
IN2– 5
IN1
UF PACKAGE
20-LEAD (4mm s 4mm) PLASTIC QFN
TJMAX = 150°C, θJA = 37°C/W, θJC = 2°C/W
EXPOSED PAD (PIN 21) IS V–, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC6601CUF-1#PBF
LTC6601IUF-1#PBF
LTC6601CUF-1#TRPBF
LTC6601IUF-1#TRPBF
66011
66011
20-Lead (4mm × 4mm) Plastic QFN
20-Lead (4mm × 4mm) Plastic QFN
0°C to 70°C
–40°C to 85°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
DC ELECTRICAL CHARACTERISTICS + The l denotes
the specifications which apply over the full operating
–
+
temperature range, otherwise specifications are at TA = 25°C. V = 3V, V = 0V, VINCM = VOCM = mid-supply, BIAS tied to V or floating,
ILOAD = 0, RBAL = 100k. The filter is configured for a gain of 1 unless otherwise noted. VS is defined as (V+ – V–). VOUTCM is defined as
(VOUT+ + VOUT–)/2. VINCM is defined as (VINP + VINM)/2. VOUTDIFF is defined as (VOUT+ – VOUT–). VINDIFF is defined as (VINP – VINM). See
Figure 1.
SYMBOL
PARAMETER
CONDITIONS
VOSDIFF (Note 6)
Amplifier Differential Offset Voltage (Input
Referred)
VS = 2.7V to 5.25V, BIAS = V+
BIAS = Floating
ΔVOSDIFF/ΔT
(Note 6)
Ampifier Differential Offset Voltage Drift
(Input Referred)
VS = 2.7V to 5.25V
RIN (Note 14)
Input Resistance, BIAS = V+
Single Ended Input Resistance, Pin 2 or Pin 4 VS = 3V
Differential Input Resistance
VS = 3V
MIN
l
l
TYP
MAX
UNITS
±0.25
±0.25
±1
±1.5
mV
mV
1
133
200
μV/°C
Ω
Ω
66011f
2
LTC6601-1
DC ELECTRICAL CHARACTERISTICS + The l denotes
the specifications which apply over the full operating
–
+
temperature range, otherwise specifications are at TA = 25°C. V = 3V, V = 0V, VINCM = VOCM = mid-supply, BIAS tied to V or floating,
ILOAD = 0, RBAL = 100k. The filter is configured for a gain of 1 unless otherwise noted. VS is defined as (V+ – V–). VOUTCM is defined as
(VOUT+ + VOUT–)/2. VINCM is defined as (VINP + VINM)/2. VOUTDIFF is defined as (VOUT+ – VOUT–). VINDIFF is defined as (VINP – VINM). See
Figure 1.
SYMBOL
PARAMETER
ΔRIN (Note 14)
Input Resistance Match, BIAS = V+
Single Ended Input Resistance, Pin 2 or Pin 4 VS = 3V
IB (Note 7)
Internal Amplifier Input Bias
CONDITIONS
VS = 2.7V to 5V
MIN
l
BIAS = V+
BIAS = Floating
l
l
BIAS = V+
BIAS = Floating
l
l
TYP
MAX
±0.25
–50
–25
UNITS
Ω
–25
–12.5
0
0
μA
μA
±1
±1
±10
±5
μA
μA
IOS (Note 7)
Internal Amplifier Input Offset
VS = 2.7V to 5V
VINCM (Note 8)
Input Signal Common Mode Range
(VINP + VINM)/2
BIAS = V+, VOCM = 1.5V
BIAS = V+, VOCM = 2.5V
VS = 3V
VS = 5V
l
l
0
0
1.7
4.7
V
V
BIAS Pin Floating, VOCM = 1.5V
BIAS Pin Floating, VOCM = 2.5V
VS = 3V
VS = 5V
l
l
0
0
1.8
4.8
V
V
Input Common Mode Rejection Ratio
(Amplifier Input Referred) ΔVINCM/ΔVOSDIFF
ΔVINCM = 2.5V
VS = 5V
74
dB
CMRRO
(Notes 9, 14)
Output Common Mode Rejection Ratio
(Amplifier Input Referred) ΔVOCM/ΔVOSDIFF
ΔVOCM = 1V
VS = 5V
70
dB
PSRR (Note 10)
Power Supply Rejection Ratio
(Amplifier Input Referred) ΔVS /ΔVOSDIFF
BIAS = V+
BIAS Pin Floating
VS = 2.7V to 5V
VS = 2.7V to 5V
l
l
66
60
95
95
dB
dB
PSRRCM (Note 10) Common Mode Power Supply Rejection Ratio
(ΔVS /ΔVOSCM)
VS = 2.7V to 5V
l
46
60
dB
1
V/V
CMRRI
(Notes 9, 14)
gcm
Common Mode Gain (ΔVOUTCM/ΔVOCM)
ΔVOCM = 2V
VS = 5V
Common Mode Gain Error = 100 • (gcm – 1)
ΔVOCM = 2V
VS = 5V
l
±0.1
±0.3
%
Output Balance (ΔVOUTCM/ΔVOUTDIFF)
Single-Ended Input
Differential Input
ΔVOUTDIFF = 2V
VS = 5V
VS = 5V
l
l
–62
–63
–40
–40
dB
dB
VOSCM
Common Mode Offset Voltage
(VOUTCM – VOCM)
VS = 2.7V to 5V
VS = 2.7V to 5V
BIAS = V+
BIAS = Floating
l
l
±5
±8
±15
±20
mV
mV
ΔVOSCM/ΔT
Common Mode Offset Voltage Drift
(VOUTCM – VOCM)
VS = 2.7V to 5V
VS = 2.7V to 5V
BIAS = V+
BIAS = Floating
l
l
5
20
VOUTCMR (Note 8)
Output Signal Common Mode Range
(Voltage Range for the VOCM Pin)
VS = 3V
VS = 5V
VS = 3V
VS = 5V
BIAS = V+
BIAS = V+
BIAS Pin Floating
BIAS Pin Floating
l
l
l
l
1.1
1.1
1.1
1.1
BAL
μV/°C
μV/°C
1.7
4
1.8
4
V
V
V
V
RINVOCM
Input Resistance, VOCM Pin
VS = 3V
l
12.5
18
23.5
kΩ
VMID
Voltage at the VOCM PIn
VS = 3V
l
1.475
1.5
1.525
V
66011f
3
LTC6601-1
DC ELECTRICAL CHARACTERISTICS + The l denotes
the specifications which apply over the full operating
–
+
temperature range, otherwise specifications are at TA = 25°C. V = 3V, V = 0V, VINCM = VOCM = mid-supply, BIAS tied to V or floating,
ILOAD = 0, RBAL = 100k. The filter is configured for a gain of 1 unless otherwise noted. VS is defined as (V+ – V–). VOUTCM is defined as
(VOUT+ + VOUT–)/2. VINCM is defined as (VINP + VINM)/2. VOUTDIFF is defined as (VOUT+ – VOUT–). VINDIFF is defined as (VINP – VINM). See
Figure 1.
SYMBOL
PARAMETER
CONDITIONS
VOUT
Output Voltage, High, Either Output Pin
(Note 11)
VS = 3V, IL = 0mA
VS = 3V, IL = –5mA
VS = 3V, IL = –20mA
VS = 5V, IL = 0mA
VS = 5V, IL = –5mA
VS = 5V, IL = –20mA
TYP
MAX
UNITS
l
l
l
l
l
l
MIN
245
285
415
350
390
550
450
525
750
625
700
1000
mV
mV
mV
mV
mV
mV
l
l
l
l
l
l
240
290
470
370
430
650
450
525
850
675
775
1100
mV
mV
mV
mV
mV
mV
l
l
l
l
l
l
120
135
195
175
200
270
225
250
350
325
360
475
mV
mV
mV
mV
mV
mV
VS = 3V, IL = 0mA, BIAS Pin Floating
VS = 3V, IL = 5mA, BIAS Pin Floating
VS = 3V, IL = 20mA, BIAS Pin Floating
VS = 5V, IL = 0mA, BIAS Pin Floating
VS = 5V, IL = 5mA, BIAS Pin Floating
VS = 5V, IL = 20mA, BIAS Pin Floating
l
l
l
l
l
l
110
120
170
150
170
225
200
225
300
270
300
400
mV
mV
mV
mV
mV
mV
VS = 3V
VS = 5V
l
l
±45
±60
l
2.7
BIAS = V+
BIAS = V+
BIAS = V+
BIAS = V+
BIAS = V+
BIAS = V+
VS = 3V, IL = 0mA, BIAS Pin Floating
VS = 3V, IL = –5mA, BIAS Pin Floating
VS = 3V, IL = –20mA, BIAS Pin Floating
VS = 5V, IL = 0mA, BIAS Pin Floating
VS = 5V, IL = –5mA, BIAS Pin Floating
VS = 5V, IL = –20mA, BIAS Pin Floating
Output Voltage, Low, Either Output Pin
(Note 11)
VS = 3V, IL = 0mA
VS = 3V, IL = 5mA
VS = 3V, IL = 20mA
VS = 5V, IL = 0mA
VS = 5V, IL = 5mA
VS = 5V, IL = 20mA
BIAS = V+
BIAS = V+
BIAS = V+
BIAS = V+
BIAS = V+
BIAS = V+
ISC
Output Short-Circuit Current,
Either Output Pin (Note 12)
VS
Supply Voltage Range
IS
Supply Current, BIAS Pin Tied to V+
VS = 2.7V
VS = 3V
VS = 5V
l
l
l
Supply Current, BIAS Pin Floating
VS = 2.7V
VS = 3V
VS = 5V
ISHDN
Supply Current, BIAS Pin Tied to V–
VBIASSD
BIAS Input Pin Range for Shutdown
VBIASLP (Note 13)
BIAS Input for Half Power Operation
±65
±90
mA
mA
5.25
V
32.9
33.1
33.9
43
43.5
45
mA
mA
mA
l
l
l
16.0
16.2
16.9
25
25.5
26.5
mA
mA
mA
VS = 2.7V
VS = 3V
VS = 5V
l
l
l
0.34
0.35
0.55
0.9
1
1.6
mA
mA
mA
VS = 2.7V to 5V
l
V–
V– + 0.4
V
l
V– + 1.0
V– + 1.5
V
V– + 2.3
V+
V
kΩ
VS = 2.7V to 5V
VBIASHP
BIAS Input for High Performance Operation
VS = 2.7V to 5V
l
RBIAS
BIAS Input Resistance
VS = 2.7V to 5V
l
100
150
200
l
V– + 1.05
V– + 1.12
V– + 1.25
VBIAS
BIAS Float Voltage
VS = 2.7V to 5V
tON
Turn-On Time
VS = 3V, VSHDN = 0.25V to 3V
400
ns
V
tOFF
Turn-Off Time
VS = 3V, VSHDN = 3V to 0.25V
400
ns
66011f
4
LTC6601-1
AC ELECTRICAL CHARACTERISTICS + The l denotes
the specifications which apply over the full operating
–
+
temperature range, otherwise specifications are at TA = 25°C. V = 3V, V = 0V, VINCM = VOCM = mid-supply, VBIAS is tied to V or
floating, unless otherwise noted. (See Figure 2 for the AC test configuration.) VS is defined as (V+ – V–). VOUTCM is defined as (VOUT+ +
VOUT–)/2. VICM is defined as (VINP + VINM)/2. VOUTDIFF is defined as (VOUT+ – VOUT–). VINDIFF is defined as (VINP – VINM).
SYMBOL
PARAMETER
CONDITIONS
GAIN
Filter Gain, See Figure 2,
BIAS Pin Tied to V+,
AC Gain Measurements Relative to 1MHz
ΔVIN = ±0.25V, fTEST = DC (Note 14)
VIN = 600mVP-P, fTEST = 1MHz
VIN = 600mVP-P, fTEST = 2MHz
VIN = 600mVP-P, fTEST = 5MHz
VIN = 600mVP-P, fTEST = 10MHz
VIN = 600mVP-P, fTEST = 14.45MHz
VIN = 600mVP-P, fTEST = 20MHz
VIN = 600mVP-P, fTEST = 50MHz
l
l
l
l
l
l
l
l
Filter Phase, See Figure 2,
BIAS Pin Tied to V+
ΔVIN = ±0.25V, fTEST = DC
VIN = 600mVP-P, fTEST = 1MHz
VIN = 600mVP-P, fTEST = 2MHz
VIN = 600mVP-P, fTEST = 5MHz
VIN = 600mVP-P, fTEST = 10MHz
VIN = 600mVP-P, fTEST = 14.45MHz
VIN = 600mVP-P, fTEST = 20MHz
VIN = 600mVP-P, fTEST = 50MHz
l
l
l
l
l
l
l
l
NOISE
Wide Band Output Noise, 14.45MHz Cutoff,
BIAS Pin Tied to V+
BW = 100MHz
BW = 20MHz
71
54
SNR
BIAS Pin Tied to V+
BW = 100MHz
BW = 20MHz
80
82.3
dB
dB
HD2, Single-Ended Input
HD3, Single-Ended Input
HD2, Differential Input
HD3, Differential Input
–70
–103
–72
–103
dBc
dBc
dBc
dBc
–120
ppm/°C
PHASE
DISTORTION VIN = 2VP-P , 10MHz, BIAS Pin Tied to V+
MIN
TYP
MAX
UNITS
–0.25
±0.05
0
0.02
0.11
–0.34
–2.35
–6.24
–21.70
0.25
dB
dB
dB
dB
dB
dB
dB
dB
–0.08
–0.01
–0.54
–2.75
–7.14
–23.70
–6.0
–12.0
–30.7
–67.6
–100.1
–127.3
0
–5.4
–10.8
–28.2
–62.6
–94.1
–122.3
–169.3
0.12
0.23
–0.14
–1.95
–5.34
–19.70
–4.8
–9.6
–25.7
–57.6
–88.1
–117.3
Deg
Deg
Deg
Deg
Deg
Deg
Deg
Deg
μVRMS
μVRMS
fO TC
Cutoff Frequency Temperature Coefficient
GAIN
Filter Gain, See Figure 2,
BIAS Pin Floating (Remaining AC
Measurements Relative to 1MHz)
ΔVIN = ±0.25V, fTEST = DC (Note 14)
VIN = 600mVP-P, fTEST = 1MHz
VIN = 600mVP-P, fTEST = 2MHz
VIN = 600mVP-P, fTEST = 5MHz
VIN = 600mVP-P, fTEST = 10MHz
VIN = 600mVP-P, fTEST = 14.45MHz
VIN = 600mVP-P, fTEST = 20MHz
VIN = 600mVP-P, fTEST = 50MHz
l
l
l
l
l
l
l
l
Filter Phase, See Figure 2,
BIAS Pin Floating
ΔVIN = ±0.25V, fTEST = DC
VIN = 600mVP-P, fTEST = 1MHz
VIN = 600mVP-P, fTEST = 2MHz
VIN = 600mVP-P, fTEST = 5MHz
VIN = 600mVP-P, fTEST = 10MHz
VIN = 600mVP-P, fTEST = 14.45MHz
VIN = 600mVP-P, fTEST = 20MHz
VIN = 600mVP-P, fTEST = 50MHz
l
l
l
l
l
l
l
l
NOISE
Output Noise, See Figure 2,
BIAS Pin Floating
BW = 100MHz
BW = 20MHz
78
58
SNR
BIAS Pin Floating
BW = 100MHz
BW = 20MHz
79
81.7
dB
dB
Distortion
VIN = 2VP-P , 10MHz, BIAS Pin Floating
HD2, Single-Ended Input
HD3, Single-Ended Input
HD2, Differential Input
HD3, Differential Input
–64
–71
–70
–72
dBc
dBc
dBc
dBc
fO TC
Cutoff Frequency Temperature Coefficient
–120
ppm/°C
PHASE
–0.25
–0.08
–0.01
–0.54
–2.90
–7.43
–23.90
–6.0
–12.4
–31.8
–70.2
–103.5
–130.1
±0.05
0
0.02
0.11
–0.34
–2.50
–6.53
–21.90
0
–5.5
–11.2
–29.3
–65.2
–97.5
–125.1
–173.6
0.25
0.12
0.23
–0.14
–2.10
–5.63
–19.90
–4.8
–10.0
–26.8
–60.2
–91.5
–120.1
dB
dB
dB
dB
dB
dB
dB
dB
Deg
Deg
Deg
Deg
Deg
Deg
Deg
Deg
μVRMS
μVRMS
66011f
5
LTC6601-1
ELECTRICAL CHARACTERISTICS
Note 1: Stresses beyond those listed under the Absolute Maximum
Ratings may cause permanent damage to the device. Exposure to any
Absolute Maximum Rating condition for extended periods may affect
device reliability and lifetime.
Note 2: All pins are protected by steering diodes to either supply. If any
pin is driven beyond the part’s supply voltage, the excess input current
(current in excess of what it takes to drive that pin to the supply rail)
should be limited to less than 10mA.
Note 3: A heat sink may be required to keep the junction temperature
below the Absolute Maximum Rating when the output is shorted
indefinitely. Long-term application of output currents in excess of the
Absolute Maximum Ratings may impair the life of the device.
Note 4: The LTC6601C/LTC6601I are guaranteed functional over the
operating temperature range –40°C to 85°C.
Note 5: The LTC6601C is guaranteed to meet specified performance from
0°C to 70°C. The LTC6601C is designed, characterized, and expected
to meet specified performance from –40°C to 85°C but is not tested or
QA sampled at these temperatures. The LTC6601I is guaranteed to meet
specified performance from –40°C to 85°C.
Note 6: Output referred voltage offset is a function of the low frequency
gain of the LTC6601. To determine output referred voltage offset, or output
voltage offset drift, multiply this specification by the noise gain (1 + GAIN).
See Applications Information for more details.
Note 7: Input bias current is defined as the average of the currents
flowing into the noninverting and inverting inputs of the internal amplifier
and is calculated from measurements made at the pins of the IC. Input
offset current is defined as the difference of the currents flowing into the
noninverting and inverting inputs of the internal amplifier and is calculated
from measurements made at the pins of the IC.
Note 8: Input common mode range is tested using the test circuit of
Figure 1 by measuring the differential DC gain with VICM = mid-supply, and
with VICM at the input common mode range limits listed in the Electrical
Characteristics table, verifying the differential gain has not deviated from
the mid-supply common mode input case by more than 1%, and the
common mode offset (VOCMOS) has not deviated from the mid-supply
common mode offset by more than ±10mV.
The voltage range for the output common mode range is tested using the
test circuit of Figure 1 by measuring the differential DC gain with VOCM =
mid-supply, and again with a voltage set on the VOCM pin at the Electrical
Characteristics table limits, checking the differential gain has not deviated
from the mid-supply common mode input case by more than 1%, and that
the common mode offset (VOCMOS) has not deviated by more than ±10mV
from the mid-supply case.
Note 9: Input CMRR is defined as the ratio of the change in the input
common mode voltage at the amplifier input to the change in differential
input referred voltage offset. Output CMRR is defined as the ratio of the
change in the voltage at the VOCM pin to the change in differential input
referred voltage offset.
Note 10: Power supply rejection (PSRR) is defined as the ratio of the
change in supply voltage to the change in differential input referred voltage
offset. Common mode power supply rejection (PSRRCM) is defined as the
ratio of the change in supply voltage to the change in the common mode
offset, VOUTCM /VOCM.
Note 11: Output swings are measured as differences between the output
and the respective power supply rail.
Note 12: Extended operation with the output shorted may cause junction
temperatures to exceed the 150°C limit and is not recommended.
Note 13: Floating the BIAS pin will reliably place the part into the halfpower mode. The pin does not have to be driven. Care should be taken,
however, to prevent external leakage currents in or out of this pin from
pulling the pin into an undesired state.
Note 14: The variable contact resistance of the high speed test equipment
limits the accuracy of this test. These parameters only show a typical
value, or conservative minimum and maximum value.
66011f
6
LTC6601-1
TYPICAL PERFORMANCE CHARACTERISTICS
High Performance Supply Current
vs Temperature and
Supply Voltage
Low Power Supply Current
vs Temperature and
Supply Voltage
18.0
0.8
35
VINCM = VOCM = MID-SUPPLY
BIAS PIN FLOATING
VINCM = VOCM = MID-SUPPLY
BIAS PIN TIED TO V+
17.5
0.6
3V
16.5
3V
2.7V
5V
0.5
33
ICC (mA)
ICC (mA)
5V
VINCM = VOCM = MID-SUPPLY
BIAS PIN TIED TO V –
0.7
5V
34
17.0
ICC (mA)
Shutdown Supply Current
vs Temperature and
Supply Voltage
2.7V
32
0.4
3V
2.7V
0.3
16.0
0.2
31
15.5
0.1
15.0
–50
–25
25
75
0
50
TEMPERATURE (°C)
100
30
–50
125
–25
25
75
0
50
TEMPERATURE (°C)
66011 G01
0
–50
125
100
–25
25
75
0
50
TEMPERATURE (°C)
66011 G03
66011 G02
Shutdown Supply Current
vs Supply Voltage and Temperature
Supply Current vs Bias Pin
Voltage and Temperature
50
125
100
Low Power Mode Supply Current
vs Supply Voltage and Temperature
1
100
VINCM = VOCM = MID-SUPPLY
VS = 3V
125°C
125°C
40
10
–40°C
25°C
30
20
ICC (mA)
ICC (mA)
ICC (mA)
0.1
1
–40°C
0.1
0.01
10
–40°C
25°C
125°C
0
0.01
VINCM = VOCM = MID-SUPPLY
BIAS PIN TIED TO V–
0.001
1
2
3
0.5
1.5
2.5
BIAS PIN VOLTAGE WITH RESPECT TO V– (V)
0
1
2
4
3
SUPPLY VOLTAGE (V)
66011 G04
125°C
VOS INPUT REFERRED (mV)
–40°C
1
0.1
25°C
0.01
VINCM = VOCM = MID-SUPPLY
BIAS PIN TIED TO V+
0
1
3
2
4
SUPPLY VOLTAGE (V)
5
66011 G07
1
2
4
3
SUPPLY VOLTAGE (V)
1.00
VS = 3V
0.75 VINCM = VOCM = MID-SUPPLY
BIAS PIN TIED TO V+
0.50 5 REPRESENTATIVE UNITS
0.25
0.00
–0.25
–0.50
–0.75
–1.00
–50
5
Low Power Mode Differential VOS
vs Temperature
1.00
10
0
66011 G06
High Performance Mode
Differential VOS vs Temperature
100
ICC (mA)
0.001
5
66011 G05
High Performance Supply Current
vs Supply Voltage and Temperature
0.001
VINCM = VOCM = MID-SUPPLY
BIAS PIN FLOATING
VOS INPUT REFERRED (mV)
0
25°C
VS = 3V
0.75 VINCM = VOCM = MID-SUPPLY
BIAS PIN FLOATING
0.50 5 REPRESENTATIVE UNITS
0.25
0.00
–0.25
–0.50
–0.75
–25
0
50
75
25
TEMPERATURE (°C)
100
125
66011 G08
–1.00
–50
–25
0
50
75
25
TEMPERATURE (°C)
100
125
66011 G09
66011f
7
LTC6601-1
TYPICAL PERFORMANCE CHARACTERISTICS
High Performance Common Mode
VOS vs Temperature
Low Power Common Mode VOS
vs Temperature
Internal Amplifier Input Bias
Current vs Temperature
15
10
–5
10
LOW POWER MODE
(BIAS PIN FLOATING)
–10
5
0
IBIAS (μA)
VOSCM (mV)
VOSCM (mV)
5
0
–15
HIGH PERFORMANCE MODE
(BIAS PIN TIED TO V+)
–20
–5
–5 V = 3V
S
VINCM = VOCM = MID-SUPPLY
BIAS PIN TIED TO V+
5 REPRESENTATIVE UNITS
–10
–50 –25
0
50
75
25
TEMPERATURE (°C)
100
VS = 3V
–10 VINCM = VOCM = MID-SUPPLY
BIAS PIN FLOATING
5 REPRESENTATIVE UNITS
–15
–50 –25
0
50
75
25
TEMPERATURE (°C)
125
–25
VS = 3V
VINCM = VOCM = MID-SUPPLY
100
FLOAT VOLTAGE (V)
RESISTANCE (Ω)
RESISTANCE/RNOMINAL (Ω/Ω)
1.15
175
1.10
1.05
125
–25
0
50
75
25
TEMPERATURE (°C)
100
1.00
–50
125
–25
0
50
75
25
TEMPERATURE (°C)
100
125
VS = 3V
VINCM = VOCM = MID-SUPPLY
RNOMINAL = 200Ω DIFFERENTIAL
1.0025 RNOMINAL = 133.3Ω SINGLE-ENDED
SEE FIGURE 1 FOR CONFIGURATION
1.0000
0.9975
0.9950
–50
SINGLE-ENDED
DIFFERENTIAL
–25
0
50
75
25
TEMPERATURE (°C)
66011 G14
66011 G13
VS = 3V
VINCM = VOCM = MID-SUPPLY
5 REPRESENTATIVE UNITS
100
125
66011 G15
High Performance Mode
Frequency Response of 12
Possible Filter Configurations
Low Frequency Gain
vs Temperature
125
1.0050
VS = 3V
VINCM = VOCM = MID-SUPPLY
VS = 3V
VINCM = VOCM = MID-SUPPLY
100
Filter Input Resistance
vs Temperature
1.20
200
100
–50
0
50
75
25
TEMPERATURE (°C)
66011 G12
BIAS Pin Float Voltage
vs Temperature
BIAS Pin Input Resistance
vs Temperature
150
–25
66011 G11
66011 G10
1.010
–30
–50
125
Low Power Mode Frequency
Response of 12 Possible Filter
Configurations
10
10
0
0
1.000
0.995
0.990
–50
GAIN (dB)
GAIN (dB)
GAIN (V/V)
1.005
–10
–20
–20
VS = 3V
VINCM = VOCM = MID-SUPPLY
BIAS PIN FLOATING
VS = 3V
VINCM = VOCM = MID-SUPPLY
BIAS PIN TIED TO V+
–25
0
50
75
25
TEMPERATURE (°C)
100
125
66011 G16
–10
–30
0.1
1
10
FREQUENCY (MHz)
100
66011 G17
–30
0.1
1
10
FREQUENCY (MHz)
100
66011 G18
66011f
8
LTC6601-1
TYPICAL PERFORMANCE CHARACTERISTICS
High Performance Mode Gain
and Phase Repeatability of 10
Random Units
Low Power Mode Gain and Phase
Repeatability of 10 Random Units
4
3
1
0
–1
MIN – AVERAGE
–0.10
–2
ϕMIN – ϕAVERAGE
–0.15
–0.20
0.1
GAIN DEVIATION (dB)
2
0
–0.05
0.20
3
2
1
MAX – AVERAGE
0
–0.05
0
–1
MIN – AVERAGE
–2
ϕMIN – ϕAVERAGE
–0.15
–4
100
1
10
FREQUENCY (MHz)
0.05
–0.10
–3
4
VS = 3V
0.15 VINCM = VOCM = MID-SUPPLY
BIAS PIN FLOATING
SEE FIGURE 1
0.10
ϕMAX – ϕAVERAGE
–3
–0.20
0.1
–4
100
1
10
FREQUENCY (MHz)
66011 G19
High Performance Mode Phase
Error of 10 Random Units
3
GAIN ERROR (dB)
VS = 3V
VICM = VOCM = MID-SUPPLY
2 BIAS PIN TIED TO V+
10 RANDOM UNITS PLOTTED
TA = 25°C
1
+SPECIFICATION
0
–1
15
VS = 3V
VICM = VOCM = MID-SUPPLY
2 BIAS PIN FLOATING
10 RANDOM UNITS PLOTTED
TA = 25°C
1
+SPECIFICATION
PHASE ERROR (DEG)
3
0
–SPECIFICATION
–1
–SPECIFICATION
–2
–2
–3
–3
1
10
FREQUENCY (MHz)
100
1
10
FREQUENCY (MHz)
VS = 5V
4
BIAS PIN
VOUTDIFF (V)
1
1.0
0
0.8
–1
0.6
–2
0.4
–3
–10
100
66011 G24
100
–5
0
1
VS = 3V
1
0
–1
0.2
VOUTDIFF
–4
10
FREQUENCY (MHz)
2
1.2
2
–SPECIFICATION
10
FREQUENCY (MHz)
1.4
3
0
1
Pulse Response
1.6
VBIAS PIN (V)
PHASE ERROR (DEG)
5
VS = 3V
VICM = VOCM = MID-SUPPLY
10 BIAS PIN FLOATING
10 RANDOM UNITS PLOTTED
TA = 25°C
5
+SPECIFICATION
–SPECIFICATION
66011 G23
Turn On and Turn Off Transient
Response
15
1
–5
66011 G22
Low Power Mode Phase Error of
10 Random Units
–15
+SPECIFICATION
0
–15
100
66011 G21
–5
VS = 3V
VICM = VOCM = MID-SUPPLY
10 BIAS PIN TIED TO V+
10 RANDOM UNITS PLOTTED
TA = 25°C
5
–10
VOUTDIFF (V)
GAIN ERROR (dB)
66011 G20
Low Power Mode Gain Error of
10 Random Units Normalized to
1MHz
High Performance Mode Gain Error
of 10 Random Units Normalized to
1MHz
PHASE DEVIATION (DEG)
VS = 3V
0.15 VINCM = VOCM = MID-SUPPLY
BIAS PIN TIED TO V+
SEE FIGURE 2
0.10
ϕMAX – ϕAVERAGE
0.05 MAX – AVERAGE
PHASE DEVIATION (DEG)
GAIN DEVIATION (dB)
0.20
2
3
4
TIME (μs)
5
6
66011 G25
0
–2
0
1
2
3
4
5
TIME (μs)
6
7
8
66011 G26
66011f
9
LTC6601-1
TYPICAL PERFORMANCE CHARACTERISTICS
Distortion vs Frequency
Differential Output Noise
100
10
10
1
0.001
0.01
SPECTRAL DENSITY,
BIAS TIED TO V+
0.1
1
FREQUENCY (MHz)
–100
HD3
–120
–130
0.1
SINGLE ENDED INPUT
DIFFERENTIAL INPUT
1
10
100
FREQUENCY (MHz)
66011 G27
0
0
GAIN
–0.5
GAIN (dB)
–1.5
0
50
75
25
TEMPERATURE (°C)
100
125
–30
–1.0
–45
–1.5
–60
–2.0 VS = 3V
VICM = VOCM = MID-SUPPLY
–2.5 BIAS PIN TIED TO V+
TEMPERATURES PLOTTED:
–45°C, –10°C, 25°C, 70°C, 95°C, 125°C
–3.0
1
10
FREQUENCY (MHz)
–75
–90
–2
–105
–3
Normalized 125Ω Resistor Trim
1000
AVERAGE = 125Ω
900 STD. DEV = 0.22Ω
FREQUENCY
500
400
600
500
400
300
200
200
100
100
0
100
700
–10
66011 G33
10
FREQUENCY (MHz)
800
300
100
1
66011 G32
AVERAGE = 100Ω
STD. DEV = 0.19Ω
600
–SPECIFICATION
10
FREQUENCY (MHz)
–1
–SPECIFICATION
700
FREQUENCY
PHASE ERROR (dB)
800
0
1
0
Normalized 100Ω Resistor Trim
900
VS = 3V
VICM = VOCM = MID-SUPPLY
10 BIAS PIN TIED TO V+
TEMPERATURES PLOTTED:
–45°C, –10°C, 25°C,
5 70°C, 95°C, 125°C
+SPECIFICATION
–15
VS = 3V
VICM = VOCM = MID-SUPPLY
2 BIAS PIN TIED TO V+
TEMPERATURES PLOTTED:
–45°C, –10°C, 25°C,
1 70°C, 95°C, 125°C
+SPECIFICATION
66011 G31
Phase Error vs Temperature
–5
–15
PHASE
66011 G30
15
100
3
PHASE (DEG)
CHANGE OF fO (%)
0
–1.0
1
10
FREQUENCY (MHz)
Gain Error Relative to 1MHz
vs Temperature
0.5
–0.5
SINGLE ENDED INPUT
DIFFERENTIAL INPUT
66011 G29
Passband Gain and Phase
vs Temperature
0.5
–25
–130
0.1
66011 G28
% Change of fO vs Temperature
–2.0
–50
–100
–110
–110
–120
1
100
10
HD2
–90
VS = 5V
VIN = 2VP-P INPUT
–70 V
ICM = VOCM
= MID-SUPPLY
–80 BIAS PIN FLOATING
FIGURE 2
–90
GAIN ERROR (dB)
SPECTRAL DENSITY,
BIAS PIN FLOATING
VS = 5V
VIN = 2VP-P INPUT
–70 VICM = VOCM = MID-SUPPLY
BIAS PIN TIED TO V+
–80 FIGURE 2
HARMONIC (dBc)
INTEGRATED NOISE,
BIAS PIN FLOATING
INTEGRATED NOISE,
BIAS TIED TO V+
HARMONIC (dBc)
VS = 3V
FIGURE 2
Distortion vs Frequency
–60
–60
INTEGRATED NOISE (μVRMS)
NOISE SPECTRAL DENSITY (nV/√Hz)
100
0
0.993
0.997
1.001
1.005
NORMALIZED RESISTANCE
1.009
66011 G34
0.99
0.994 0.998 1.002 1.006
NORMALIZED RESISTANCE
1.01
66011 G35
66011f
10
LTC6601-1
TYPICAL PERFORMANCE CHARACTERISTICS
1000
900
1000
800
700
800
AVERAGE = 400.01Ω
800 STD. DEV = 1.0Ω
AVERAGE = 200Ω
900 STD. DEV = 0.37Ω
AVERAGE = 400.01Ω
900 STD. DEV = 0.87Ω
700
600
500
400
FREQUENCY
600
FREQUENCY
FREQUENCY
700
500
400
300
300
600
500
400
300
200
200
100
100
100
0
0
200
0.99
0.994 0.998 1.002 1.006
NORMALIZED RESISTANCE
1.01
0.99
0.994 0.998 1.002 1.006
NORMALIZED RESISTANCE
0.994 0.998 1.002 1.006
NORMALIZED RESISTANCE
1200
AVERAGE = 33.3pF
900 STD. DEV = 0.09pF
1000
700
800
FREQUENCY
FREQUENCY
400
600
500
400
600
400
300
200
200
AVERAGE = 48.2pF
STD. DEV = 0.08pF
1000
800
800
1.01
Normalized 48.2pF
Capacitor Trim
1000
AVERAGE = 21.1pF
STD. DEV = 0.07pF
600
0.99
66011 G38
Normalized 33.3pF
Capacitor Trim
Normalized 21.1pF Capacitor Trim
1200
0
1.01
66011 G37
66011 G36
FREQUENCY
Normalized Feedback 400Ω
Resistor Trim
Normalized Input 400Ω
Resistor Trim
Normalized 200Ω Resistor Trim
200
100
0
0.984
0.990 0.997 1.003 1.009
NORMALIZED CAPACITANCE
0
1.015
0
0.988
0.993 0.999 1.005 1.010
NORMALIZED CAPACITANCE
66011 G39
1.016
0.992 0.995 0.998 1.001 1.004 1.007 1.010
NORMALIZED CAPACITANCE
66011 G41
66011 G40
Normalized 10.55pF
Capacitor Trim
Normalized 81.5pF
Capacitor Trim
1000
Normalized 16.1pF
Capacitor Trim
350
400
AVERAGE = 81.5pF
900 STD. DEV = 0.1pF
AVERAGE = 10.55pF
350 STD. DEV = 0.03pF
300
AVERAGE = 16.1pF
STD. DEV = 0.05pF
800
300
600
500
400
250
250
FREQUENCY
FREQUENCY
FREQUENCY
700
200
150
300
0
50
50
100
0.993 0.996 0.999 1.002 1.004 1.007 1.010
NORMALIZED CAPACITANCE
66011 G42
0
150
100
100
200
200
0.987 0.991 0.996 1.000 1.005 1.009 1.014
NORMALIZED CAPACITANCE
66011 G43
0
0.988 0.992 0.995 0.999 1.003 1.006 1.010 1.014
NORMALIZED CAPACITANCE
66011 G44
66011f
11
LTC6601-1
PIN FUNCTIONS
(Refer to the Block Diagram)
IN1+, IN2+, IN4+ (Pins 2, 1, 20): Input to a trimmed 100Ω,
200Ω, 400Ω resistor which feeds a noninverting summing
node. Can accept an input signal, be floated or tied to OUT–.
For best performance, stray capacitance should be kept as
low as possible by keeping printed circuit connections as
short and direct as possible. If necessary, strip back the
surrounding ground plane away from these pins.
BIAS (Pin 3): Input to a three-state comparator whose
three states allow the user to tailor amplifier power. The
pin impedance appears as a 150k resistor whose default
open-circuit potential is 1.15V with respect to the V– power
supply. If BIAS is driven to within 0.4V of the V– supply, the
amplifier is placed into a low power shutdown, consuming typically 350μA. When BIAS is floated, the amplifier
operates in its low power active state. Forcing the pin 2.3V
above V– places the part into the high performance active
state. See Applications Information for more detail.
IN1–, IN2–, IN4– (Pins 4, 5, 6): Input to a trimmed 100Ω,
200Ω, 400Ω resistor which feeds an inverting summing
node. Can accept an input signal, be floated or tied to
OUT+. For best performance, it is highly recommended
that stray capacitance be kept to as low as possible by
keeping printed circuit connections as short and direct
as possible, and if necessary, stripping back nearby surrounding ground plane away from these pins.
C1, C2 (Pins 7, 8): Input to a trimmed 16.1pF, 33.3pF
capacitor which feeds a noninverting summing node.
Typically, either float or tie to OUT–. If either of these
pins is tied to a low impedance source other than OUT–,
a resistance of at least 25Ω should be placed in series.
For best performance, it is highly recommended that stray
capacitance be kept to as low as possible by keeping printed
circuit connections as short and direct as possible, and
if necessary, stripping back nearby surrounding ground
plane away from these pins.
C3, C4 (Pins 9, 10): Input to a trimmed 10.55pF, 21.1pF
capacitor which feeds the amplifier inverting summing
node. Typically, either float or tie to OUT+. For best performance, it is highly recommended that stray capacitance
be kept to as low as possible by keeping printed circuit
connections as short and direct as possible, and if necessary, stripping back nearby surrounding ground plane
away from these pins.
OUT+, OUT– (Pins 11, 15): Output Pins. Besides driving
the internal feedback network, each pin can drive an additional 50Ω to ground with typical short-circuit current
limiting of ±65mA. Capacitive loading of these pins should
be minimized by resistively decoupling the outputs from
the load with at least 25Ω.
VOCM (Pin 12): Output Common Mode Reference Voltage.
The voltage on VOCM sets the output common mode voltage
level (which is defined as the average of the voltages on
the OUT+ and OUT– pins). The VOCM pin is the midpoint
of an internal resistive voltage divider between the supplies, developing a (default) mid-supply voltage potential
to maximize output signal swing. The VOCM pin can be
overdriven by an external voltage reference capable of
driving the input impedance presented by the VOCM pin.
The VOCM pin has an input resistance of approximately 18k
to a mid-supply potential. It should be bypassed with a
high quality ceramic bypass capacitor (for instance of X7R
dielectric) of at least 0.01μF, (unless using symmetrical
split supplies, then connect directly to a low impedance,
low noise ground plane) to minimize common mode noise
from being converted to differential noise by impedance
mismatches both externally and internally to the IC.
66011f
12
LTC6601-1
PIN FUNCTIONS
(Refer to the Block Diagram)
V+, V– (Pins 14, 13): Power Supply Pins. It is critical that
close attention be paid to supply bypassing. For single
supply applications (Pin 13 grounded), it is recommended
that a high quality 0.1μF surface mount ceramic bypass
capacitor (X7R dielectric for instance) be placed between
Pins 14 and 13, with direct short connections. Pin 13
should be tied directly to a low impedance ground plane
with minimal routing. For dual (split) power supplies, it
is recommended that at least two additional high quality
0.1μF ceramic capacitors are used to bypass V+ to ground
and V– to ground, again with minimal routing. For driving
large loads (< 200Ω), additional bypass capacitance may
be added for optimal performance. Keep in mind that small
geometry (e.g., 0603) surface mount ceramic capacitors
have a much lower ESL than do leaded capacitors, and
perform best in high speed applications.
C5, C6 (Pins 19, 18): Input to a trimmed 16.1pF, 33.3pF
capacitor which feeds an inverting summing node. Typically, either float or tie to OUT+. If either of these pins are
tied to a low impedance source other than OUT+, a resistance of at least 25Ω should be placed in series. For
best performance, it is highly recommended that stray
capacitance be kept to as low as possible by keeping printed
circuit connections as short and direct as possible, and if
necessary, stripping back nearby surrounding reference
plane away from these pins.
Exposed Pad (Pin 21): Always tie the underlying Exposed
Pad to V– (Pin 13). If split supplies are used, do not tie
the pad to ground. Tie it to V–.
C7, C8 (Pins 17, 16): Input to a trimmed 10.55pF, 21.1pF
capacitor which feeds the amplifier noninverting summing node. Typically, either float or tie to OUT–. For best
performance, stray capacitance should be kept as low as
possible by keeping printed circuit connections as short
and direct as possible.If necessary, strip back the surrounding ground plane away from these pins.
66011f
13
LTC6601-1
BLOCK DIAGRAM
20
19
IN4+
18
C5
17
16.1pF
400Ω
16
C8
C7
C6
33.3pF
81.5pF
400Ω
1
IN2+
200Ω
10.55pF
OUT–
15
21.1pF
2
IN1+
100Ω
V+
48.2pF
14
V – + 2.3V
860Ω
180k
3
BIAS
125Ω
60k
BIAS
V–
+
–
125Ω
180k
860Ω
36k
48.2pF
4
IN1–
100Ω
36k
VOCM
IN2–
12
21.1pF
OUT+
10.55pF
5
13
200Ω
11
400Ω
81.5pF
400Ω
IN4–
6
33.3pF
16.1pF
C1
7
C2
8
C3
9
C4
10
66011 BD
66011f
14
LTC6601-1
TEST CIRCUITS
20
19
18
17
16
LTC6601-1
1
15
2
VOUT–
+
V+
14
VINP
–
BIAS
0.1μF
+
–
3
IL
25Ω
RBAL
0.1μF
13
V–
0.1μF
12
VOCM
3nF
VOUT(CM)
–
VINM
+
4
11
5
6
7
8
9
VOUT+
RBAL
IL
25Ω
10
66011 F01
Figure 1. DC Test Circuit
LTC6601-1
5V
6 9 10 11 12
VIN
14
LT6411
15
19
18
17
16
1
1μF
13
20
15
8
VOUT–
100Ω
1μF
2
VINP
5
16
BIAS
0.1μF
+
–
1 2 3 7 17
3
V–
0.1μF
12
VOCM
3nF
4
11
5
6
7
8
9
0.1μF
13
–5V
1μF VINM
COILCRAFT
TTWB-4-B
V+
14
VOUT+
100Ω
50Ω
1μF
10
66011 F02
Figure 2. AC Test Circuit (Frequency Response Testing)
66011f
15
LTC6601-1
APPLICATIONS INFORMATION
FUNCTIONAL DESCRIPTION
Figure 3 shows the basic filter architecture. The Laplace
transfer function from VINDIFF to VOUTDIFF is given by the
following generalized equation for a 2nd order lowpass
filter:
The LTC6601 is designed to make the implementation of
high frequency fully-differential filtering functions very
easy. A very low noise amplifier is surrounded by 8 precision
matched resistors and 12 precision matched capacitors
so that a myriad of filter transfer functions limited only by
possible combinations and imagination can be configured
by hard wiring pins. The amplifier itself is a wide band, low
noise and low distortion fully-differential amplifier with accurate output phase balancing. It is optimized for driving low
voltage, single-supply, differential input, analog-to-digital
converters (ADCs). The LTC6601’s outputs are capable
of swinging rail-to-rail on supplies as low as 2.7V, which
makes the amplifier ideal for converting ground referenced,
single-ended signals into VOCM referenced differential
signals. Unlike traditional op amps which have a single
output, the LTC6601 has two outputs to process signals
differentially. This allows for two times the signal swing
in low voltage systems when compared to single-ended
output amplifiers. The balanced differential nature of the
amplifier and matched surrounding components provide
even-order harmonic distortion cancellation, and less
susceptibility to common mode noise (like power supply
noise). The LTC6601 can be used as a single-ended input
to differential output amplifier, or as a differential input to
differential output amplifier.
VOUTDIFF
=
VINDIFF
Gain
1+
s
s2
+
2πfO • Q ( 2πf
O
)2
Both Gain and Q of the filter are based on component ratios,
which match and track extremely well over temperature.
The corner frequency of the filter is a function of an RC
product. This RC product is trimmed to ±1% (typical) and
is not expected to drift by more than ±1% from nominal
over the entire temperature range –40°C to 85°C. As a
result, fully differential filters with tight magnitude, phase
tolerance and repeatability are achieved.
Although Figure 3 implies a differential input, the LTC6601
easily accepts single-ended inputs to either input, and will
faithfully replicate the signal at the output in differential
form.
The LTC6601’s output common mode voltage, defined as
the average of the two output voltages, is independent of
the input common mode voltage, and is adjusted by applying a voltage on the VOCM pin. If the pin is left open, there
is an internal resistive voltage divider, which develops a
R2
R1
R3
C2
fO =
C1
Q=
1
2π R2 • R3 • C1• C2
C2 R3
•
C1 R2
GAIN =
+ –
VIN(DIFF)
fO •
R1
R3
f3dB =
C1
Q=
)
6089 •
0.2236 • fO •
C2
(
1+ 1+ GAIN •
R3 C2
–
R2 C1
R2
R1
VOUT(DIFF)
– +
1
(3568 • Q
4
)
(
)
1788 • Q 2 + 447 + 1.287 • 105 • 2 • Q 2 1 507.6 • Q
(9.891• 10 • f
(16 • f • (8.29 • 10 • f
5
2.109 • 10 •
12
O
2
3dB
9
4
)
(
)
5.486 • 109 • fO4 + 120 • 5.526 • 109 • f3dB2 + 3.082 • 106 • fO2 3dB
2
)
+ 4.127 • 109 • fO2 6.638 • 1010 • f3dB 4
)
R2
66011 F03
Figure 3. Basic Filter Topology and Equations
66011f
16
LTC6601-1
APPLICATIONS INFORMATION
potential halfway between the V+ and V– pins. Whenever
this pin is not hard tied to a low impedance ground plane,
a high quality ceramic capacitor should be used to bypass
the VOCM pin to a low impedance ground plane (see Layout
Considerations). The LTC6601’s internal common mode
feedback path forces accurate output phase balancing to
reduce even order harmonics, and centers each individual
output about the potential set by the VOCM pin.
VOUT + + VOUT –
VOUTCM = VOCM =
2
The outputs (OUT+ and OUT–) of the LTC6601 are capable
of swinging rail-to-rail. They can source or sink up to approximately 75mA of current. Load capacitances should
be decoupled with at least 25Ω of series resistance from
each output.
The LTC6601 Electrical Characteristics table specifies an
input referred offset. This specification actually lumps voltage offsets due to offset bias currents (IOS), and amplifier
voltage offset into one specification. To refer this specification to the output, you simply multiply the specification
by the noise gain the LTC6601 is configured in:
VOSODIFF = 1 + Gain
where Gain is the closed loop gain in the particular filter
application:
Gain =
R2
R1
COMPONENT INPUT PIN PROTECTION
All of the LTC6601 pins with the exception of V+ and V– are
protected with steering diodes to either power supply. In
the event that a pin is driven beyond the supply rails, the
excess current should be limited to under 10mA to prevent
damage to the IC.
BIAS Pin
The LTC6601 has a BIAS pin (Pin 3) whose function is to
tailor both performance and power of the LTC6601. The
pin has a Thevenin equivalent impedance of approximately
150kΩ to a voltage source whose potential is 1.15V above
the V– supply. This pin has fixed logic levels relative to V–
(see the Electrical Characteristics table), and can be driven
by an external source keeping in mind its equivalent input
impedance and equivalent input voltage. If the BIAS pin is
floated, care should be taken to control external leakage
currents to this pin to under 1μA to prevent putting the
LTC6601 an undesired state.
If BIAS is tied to the positive supply, the LTC6601 differential filter will be in a fully active state configured for
highest performance (lowest noise and lowest distortion).
If the BIAS pin is floated or left unconnected, the LTC6601
filter will be in a fully active state, with amplifier currents
reduced and performance scaled back to preserve power
consumption. If the BIAS pin is tied to the most negative
supply (V–), the LTC6601 will be placed into a low power
shutdown mode with amplifier outputs disabled. In this
state, the LTC6601 draws approximately 350μA.
In low power shutdown, all internal biasing current sources
are shut off, and the output pins, OUT+ and OUT–, will each
appear as open collectors with a non-linear capacitor in
parallel and steering diodes to either supply. The turn-on
and turn-off time constant between states are on the order
of 0.4μs. Using this function to wire-OR outputs together
is not recommended.
General Design and Usage
As levels of integration have increased and correspondingly, system supply voltages decreased, there has been
a need for ADCs to process signals differentially in order
to maintain good signal-to-noise ratios. These ADCs are
typically supplied from a single supply voltage which
can be as low as 3V (2.7V min), and will have an optimal
common mode input range near mid-supply. The LTC6601
makes interfacing to these ADCs easy, by providing antialias filtering, single-ended to differential conversion and
common mode level shifting (translation). Figure 3 shows
a general application of this. The low frequency gain to
VOUTDIFF from VIN is simply:
VOUTDIFF = VOUT + – VOUT – ≈
R2
•V
R1 INDIFF
The differential output voltage (VOUT+ – VOUT–) is completely
independent of input and output common mode voltages,
or the voltage at the common mode pin. This makes the
66011f
17
LTC6601-1
APPLICATIONS INFORMATION
LTC6601 ideally suited for pre-amplification, level shifting and conversion of single-ended signals to differential
output signals for driving differential input ADCs.
INPUT IMPEDANCE
Figure 4 shows a simplified low frequency equivalent
circuit of the LTC6601. For balanced input sources (VINP
= –VINM), the low frequency input impedance is given by
the equation:
RINP = RINM = R1
The differential input impedance is simply:
RINDIFF = 2 • R1
For single-ended inputs (VINM = 0), the input impedance
actually increases over the balanced differential case due
to the fact the summing node (at the junction of R1, R2
and R3) moves in phase with VINP to bootstrap the input
impedance. Referring to Figure 4 with VINM = 0, the input
impedance looking into either input is:
R1
⎛ 1 ⎛ R2 ⎞ ⎞
⎜⎝ 1– 2 • ⎜⎝ R1+ R2 ⎟⎠ ⎟⎠
+
VINP
–
–
VINM
RINM
VOUT–
+
VOUTDIFF
–
R1
+
R2
For the general case, the upper input common mode voltage limit should be constrained to:
VOCM •
R1
R2
+ VINCM •
≤ V + – 1.4V
R1+ R2
R1+ R2
(
–
R3
The lower limit of the input common mode range is dictated by the ESD protection diodes at the input. While it
is possible for the inputs to swing below V–, the diodes
will conduct if the inputs are taken a diode drop below V–.
The upper limit of the input common mode range varies
as a function of the filter configuration (GAIN), VOCM potential, and whether or not the inputs are single-ended or
differential. While it is possible to exceed the upper limit
of the common mode range, doing so will degrade filter
linearity. Referring to Figure 4, for linear operation, the
summing junction where R1, R2 and R3 merge together
should be prevented from swinging to within 1.4V of the
V+ power supply.
)
R1
⎛ R1⎞
VINCM ≤ ⎜ 1+ ⎟ V + − 1.4V −
•V
⎝ R2 ⎠
R2 OCM
R1
R3
VINP + VINM
2
Or equivalently:
R2
RINP
The input common mode voltage is defined as the average
of the two inputs:
VINCM =
Calculating the low frequency input impedance of the
LTC6601 depends on how the inputs are driven (whether
they are driven from a single-ended or a differential
source).
RINP = RINM
Input and Output Common Mode Voltage Range
+
VOUT+
VOCM
The specifications for input common mode range (VINCMR)
are based on these constraints with R1 = R2 = 100Ω, and
VOCM = mid-supply. Substituting the numbers for a single
3V power supply, (V+ = 3V, V– = 0V) with VOCM =1.5V, and
R1 = R2 = 100Ω, into the above equation, the input common mode range (VINCMR) is between the two limits:
0V ≤ VINCM ≤ 1.7V
which is as is specified for a 3V supply.
0.1μF
66011 F04
Figure 4. Input Impedance
66011f
18
LTC6601-1
APPLICATIONS INFORMATION
Likewise, substituting the numbers for a single 5V power
supply, (V+ = 5V, V– = 0V) with VOCM = 2.5V, and R1 = R2
= 100Ω, into the above equation, the input common mode
range (VINCMR) is between the two limits:
0V ≤ VINCM ≤ 4.7V
The output common mode voltage is defined as the average of the two outputs:
VOUTCM = VOCM =
VOUT + + VOUT –
2
The VOCM pin sets this average by an internal common
mode feedback loop which internally forces VOUT+ =
–VOUT–. The output common mode range extends from
1.1 V above V– to 1V below V+. The VOCM pin sits in the
middle of a voltage divider which sets the default midsupply open circuit potential.
In single supply applications, where the LTC6601 is used
to interface to an ADC, the optimal common mode input
to the ADC is often determined by the ADC’s reference. If
the ADC makes a reference available for setting the input
common mode voltage, it can be directly tied to the VOCM
pin, but must be capable of driving the input impedance of
the VOCM pin (RVOCM). This impedance can be assumed
to be connected to a mid-supply potential. If an external
reference drives the VOCM pin, it should still be bypassed
with a high quality 0.01μF or higher capacitor to a low
impedance ground plane to filter any thermal noise and
to prevent common mode signals on this pin from being
inadvertently converted to differential signals.
Noise Considerations
When comparing the LTC6601 noise to other amplifiers,
be sure to compare similar specifications. Competing
devices often specify noise referred to the inputs of the
amplifier. The input referred voltage noise of the LTC6601-1
is 2.1nV/√Hz. This level is one of the lowest available for
amplifiers in this speed and power range.
In addition to the noise generated by the amplifier, the
surrounding feedback resistors also contribute noise. A
noise model is shown in Figure 5. The output spot noise
generated by both the amplifier and the feedback components is governed by the equation:
2
2
2
2
⎛
⎛
⎛
⎛
⎛
R2 ⎞ ⎞ ⎞
⎛ R2 ⎞ ⎞
⎛ R2 ⎞ ⎞
⎛ R2 ⎞ ⎞
2
2
2 ⎛
eno = ⎜ eni • ⎜ 1+ ⎟ ⎟ + 2 • ⎜ In • ⎜ R2 + R3 • ⎜ 1+ ⎟ ⎟ ⎟ + 2 • ⎜ enR1 • ⎜ ⎟ ⎟ + 2 ⎜ enR3 • ⎜ 1+ ⎟ ⎟ + 2 • enR2 2
⎝ R1⎠ ⎠ ⎟⎠
⎝ R1⎠ ⎠
⎝ R1⎠ ⎠
⎝ R1⎠ ⎠
⎜⎝
⎝
⎝
⎝
⎝
Substituting the equation for Johnson noise of a resistor (enR2 = 4kTR), and simplifying:
2
2
2
⎛
⎛
⎛ ⎛ R2 ⎞
⎛
R2 ⎞ ⎞ ⎞
⎛ R2 ⎞ ⎞
⎛ R2 ⎞ ⎞
2
2
2 ⎛
eno = ⎜ eni • ⎜ 1+ ⎟ ⎟ + 2 • ⎜ In • ⎜ R2 + R3 • ⎜ 1+ ⎟ ⎟ ⎟ + 8 • k • T ⎜ R2 ⎜ 1+ ⎟ + R3 ⎜ 1+ ⎟ ⎟
⎝ R1⎠ ⎠ ⎟⎠
⎝ R1⎠ ⎠
⎝ R1⎠ ⎠
⎜⎝
⎝
⎝
⎝ ⎝ R1⎠
66011f
19
LTC6601-1
APPLICATIONS INFORMATION
enR22
R2
*
enR12
In+2
R1
enR32
*
R3
*
+
*
enR32
enR12
eni2
eno2
R3
–
*
R1
In–2
*
enR22
*
R2
66011 F05
Figure 5. Differential Noise Model of the LTC6601
Table 1 lists the amplifier input referred noise for the
LTC6601-1. Tables 2 to10 list the noise referred to the input
pins of the IC for common configurations of the LTC6601-1.
To determine the spot noise at the output, simply multiply
the noise by the Gain = R2/R1. To estimate the integrated
noise at the output, multiply the noise by the gain, and the
square root of the noise bandwidth. The noise bandwidth
depends on the filter configuration. For Figure 2, the noise
bandwidth is 100MHz, or approximately 7 times the filter
bandwidth. Improvements in SNR can be made by adding
an additional RC filter at the output to band limit wide band
noise before feeding ADCs. See the section “Interfacing
the LTC6601 to ADC Converters” for more detail.
Table 1. Amplifier (Input Referred) Noise Characteristics for the
LTC6601-1
BIAS PIN PULLED TO V+
BIAS PIN FLOATING
eni
nV/√Hz
in
pA/√Hz
eni
nV/√Hz
in
pA/√Hz
2.1
3
2.6
2.1
LAYOUT CONSIDERATIONS
Because the LTC6601 is a very high speed amplifier, it is
sensitive to both stray capacitance and stray inductance.
It is critical that close attention be paid to supply bypassing. For single supply applications, it is recommended
that a high quality 0.1μF surface mount ceramic bypass
capacitor be placed between Pins 14 and 13 with direct
short connections. Pin 13 and the Exposed Pad, Pin 21,
should be tied directly to a low impedance ground plane
with minimal routing. For dual (split) power supplies, it
is recommended that an additional high quality, 0.1μF
ceramic capacitor be used to bypass pin V+ to ground
and V– to ground, again with minimal routing. For driving large differential loads (<200Ω), additional bypass
capacitance may be needed between V+ and V– for optimal performance. Note that small geometry (e.g., 0603)
surface mount ceramic capacitors have a much higher
self resonant frequency than capacitors with leads, and
perform best in high speed applications.
The VOCM pin should be bypassed to ground with a high
quality ceramic capacitor whose value exceeds 0.01μF,
with direct, short connections. In split supply applications,
the VOCM pin can be either bypassed to ground or directly
hardwired to ground. Be careful not to violate the output
common mode range specifications for the VOCM pin.
Stray parasitic capacitances to unused component pins
that set up the filter’s characteristics, should be kept to an
absolute minimum. This prevents deviations from the ideal
frequency response. An ideal layout technique would be to
remove the solder pads for the unused component pins,
and strip away the ground plane underneath these pins to
lower capacitance to an absolute minimum. Floating unused
component pins which set up the filter characteristics will
not reduce the reliability of the LTC6601.
At the output, always keep in mind the differential nature of
the LTC6601, and that it is critical that the load impedances
seen by both outputs (stray or intended), should be as balanced and symmetric as possible. This will help preserve
the natural balance of the LTC6601, which minimizes the
generation of even order harmonics and preserves the
rejection of common mode signals and noise.
66011f
20
LTC6601-1
APPLICATIONS INFORMATION
INTERFACING THE LTC6601 TO ADC CONVERTERS
The LTC6601’s rail-to-rail differential output and adjustable
output common mode voltage make the LTC6601 ideal
for interfacing to low voltage, single supply, differential
input ADCs. The sampling process of ADCs creates a
sampling transient that is caused by the switching-in
of the ADC sampling capacitor. The switching-in of this
sampling capacitor momentarily “shorts” the output of the
amplifier as charge is transferred between amplifier and
sampling capacitor. The amplifier must recover and settle
from this load transient before this acquisition period has
ended, for a valid representation of the input signal. The
LTC6601 will settle much more quickly from these periodic load impulses than it does from a 2V input step, but
it is a good idea to add an RC network after the outputs
of the LTC6601 to decouple the sampling transient of the
ADC (See Figure 6). The capacitance of the decoupling
network serves to provide the bulk of the charge during
the sampling process, while the two resistors of the filter
network are used to dampen and attenuate any transient
induced by the ADC. The ADC’s sampling bandwidth will
LTC6601-1
20
19
18
17
15
2
VOUT–
C1
R
CONTROL
VIN
–
3V
14
BIAS
The selection of the RC time constant is trial and error
for a given ADC, but the following guidelines are recommended. Choose an RC pole frequency greater than the
cutoff frequency of the LTC6601. 80MHz RC filters are
good for filtering broadband noise. Lower frequency RC
filters improve SNR at the expense of settling time. The
resistors in the decoupling network should be at least 25Ω.
Too much resistance in the decoupling network leaves
insufficient settling time and will create a voltage divider
between the dynamic input impedance of the ADC and the
decoupling resistors. Using insufficient resistance might
prevent proper dampening of the load transient caused by
the sampling process, and prolong the time required for
settling. In 16-bit applications, this will typically require
a minimum of 11 RC time constants. It is recommended
that the capacitor is chosen with low dielectric absorption
(such as a C0G multilayer ceramic capacitor).
16
1
+
often be much greater than that of the LTC6601, so having this discrete RC filter will give the additional benefit
of band limiting broadband output noise.
0.1μF
+
–
3
C2
13
AIN–
VOCM
10nF
12
4
11
5
D15
•
•
D0
AIN+
1μF
VOUT+
VCM
GND
1μF
3.3V
1μF
2.2μF
R
C1
66011 F06
6
7
8
9
10
t = R • (C1 + 2 • C2)
Figure 6. Interfacing the LTC6601 to A/D Converters
66011f
21
LTC6601-1
APPLICATIONS INFORMATION
A GALLERY OF BASIC FILTER TOPOLOGIES
Tables 2 through 10 list (sorted by Gain) a hundred possible
filter topologies that can be easily implemented with the
LTC6601. The tables also list the LTC6601-1 approximate
midband (1MHz) spot noise ein referred to the input resistor, R1 (with the BIAS pin pulled to V+). The gains for
these topologies range from 1V/V to 7V/V. The Qs listed
are within the range of 0.54 and 1.72. The fOs listed are
in the range of 6.96MHz and 22.71MHz, and the –3dB
frequencies listed range from 5.5MHz to 27.5MHz. For
all filters listed, R3 = 125Ω. Figures 7 to 10 show how to
pin-strap each filter configuration.
Table 2. Gain of 7 Filter Configurations
GAIN
V/V
dB
fO (MHz)
7.0
16.902
7.0
16.902
7.0
7.0
C2 (pF)
ein
(nV/√Hz)
48.2
97.6
3.7
48.2
114.8
3.7
400.00
48.2
130.9
3.7
400.00
58.75
130.9
3.7
C2 (pF)
ein
(nV/√Hz)
f–3dB (MHz)
Q
R1 (Ω)
R2 (Ω)
C1 (pF)
10.38
7.43
0.539
57.14
400.00
9.57
10.36
0.771
57.14
400.00
16.902
8.96
12.10
1.175
57.14
16.902
8.12
7.49
0.656
57.14
Table 3. Gain of 6 Filter Configurations
GAIN
V/V
dB
fO (MHz)
f–3dB (MHz)
Q
R1 (Ω)
R2 (Ω)
C1 (pF)
6.0
15.563
10.38
10.03
0.684
66.67
400.00
48.2
97.6
3.8
6.0
15.563
9.57
12.52
1.071
66.67
400.00
48.2
114.8
3.8
6.0
15.563
8.67
7.67
0.634
66.67
400.00
58.75
114.8
3.8
6.0
15.563
8.12
9.59
0.870
66.67
400.00
58.75
130.9
3.8
6.0
15.563
7.47
6.07
0.592
66.67
400.00
69.3
130.9
3.8
Table 4. Gain of 5 Filter Configurations
GAIN
V/V
dB
fo (MHz)
f–3dB (MHz)
Q
R1 (Ω)
R2 (Ω)
C1 (pF)
C2 (pF)
ein
nV/√Hz
5.0
13.979
11.36
9.67
0.614
80.00
400.00
48.2
81.5
4.0
5.0
13.979
10.38
12.78
0.936
80.00
400.00
48.2
97.6
4.0
5.0
13.979
9.40
7.67
0.594
80.00
400.00
58.75
97.6
4.0
5.0
13.979
8.67
10.07
0.849
80.00
400.00
58.75
114.8
4.0
5.0
13.979
8.12
11.25
1.290
80.00
400.00
58.75
130.9
4.0
5.0
13.979
7.98
6.46
0.591
80.00
400.00
69.3
114.8
4.0
5.0
13.979
7.47
8.16
0.779
80.00
400.00
69.3
130.9
4.0
5.0
13.979
6.96
5.50
0.579
80.00
400.00
79.85
130.9
4.0
66011f
22
LTC6601-1
APPLICATIONS INFORMATION
Table 5. Gain of 4 Filter Configurations
GAIN
V/V
dB
fO (MHz)
f–3dB MHz
Q
R1 (Ω)
R2 (Ω)
C1 (pF)
C2 (pF)
ein
nV/√Hz
4.0
12.041
11.36
13.05
0.834
100.00
400.00
48.2
81.5
4.2
4.0
12.041
10.38
14.80
1.480
100.00
400.00
48.2
97.6
4.2
4.0
12.041
9.40
10.47
0.799
100.00
400.00
58.75
97.6
4.2
4.0
12.041
8.67
12.00
1.284
100.00
400.00
58.75
114.8
4.2
4.0
12.041
8.65
6.76
0.575
100.00
400.00
69.3
97.6
4.2
4.0
12.041
7.98
8.84
0.794
100.00
400.00
69.3
114.8
4.2
4.0
12.041
7.43
6.09
0.596
100.00
400.00
79.85
114.8
4.2
4.0
12.041
7.47
10.00
1.141
100.00
400.00
69.3
130.9
4.2
4.0
12.041
6.96
7.57
0.775
100.00
400.00
79.85
130.9
4.2
Table 6. Gain of 3 Filter Configurations
GAIN
V/V
dB
fO (MHz)
f–3dB (MHz)
Q
R1 (Ω)
R2 (Ω)
C1 (pF)
C2 (pF)
ein
(nV/√Hz)
3.0
9.542
16.06
12.36
0.568
66.67
200.00
48.2
81.5
4.3
3.0
9.542
14.68
15.74
0.763
66.67
200.00
48.2
97.6
4.3
3.0
9.542
13.53
17.83
1.091
66.67
200.00
48.2
114.8
4.3
3.0
9.542
13.29
9.88
0.554
66.67
200.00
58.75
97.6
4.3
3.0
9.542
12.26
12.39
0.715
66.67
200.00
58.75
114.8
4.3
3.0
9.542
11.36
15.77
1.300
133.33
400.00
48.2
81.5
4.6
3.0
9.542
11.48
14.07
0.928
66.67
200.00
58.75
130.9
4.3
3.0
9.542
11.29
8.34
0.552
66.67
200.00
69.3
114.8
4.3
3.0
9.542
10.29
11.04
0.763
133.33
400.00
58.75
81.5
4.6
3.0
9.542
10.57
10.06
0.674
66.67
200.00
69.3
130.9
4.3
3.0
9.542
9.40
12.85
1.224
133.33
400.00
58.75
97.6
4.6
3.0
9.542
8.65
9.54
0.788
133.33
400.00
69.3
97.6
4.6
3.0
9.542
8.06
6.69
0.601
133.33
400.00
79.85
97.6
4.6
3.0
9.542
7.98
10.88
1.212
133.33
400.00
69.3
114.8
4.6
3.0
9.542
7.43
8.48
0.825
133.33
400.00
79.85
114.8
4.6
3.0
9.542
6.96
9.40
1.172
133.33
400.00
79.85
130.9
4.6
3.0
9.542
9.85
7.13
0.544
66.67
200.00
79.85
130.9
4.3
66011f
23
LTC6601-1
APPLICATIONS INFORMATION
Table 7. Gain of 2 Filter Configurations
GAIN
C1 (pF)
C2 (pF)
ein
(nV/√Hz)
200.00
48.2
81.5
5.0
200.00
58.75
81.5
5.0
200.00
48.2
97.6
5.0
200.00
58.75
97.6
5.0
100.00
200.00
69.3
97.6
5.0
1.200
100.00
200.00
58.75
114.8
5.0
0.835
100.00
200.00
69.3
114.8
5.0
13.97
1.197
200.00
400.00
58.75
81.5
5.5
10.51
9.76
0.660
100.00
200.00
79.85
114.8
5.0
10.57
13.97
1.102
100.00
200.00
69.3
130.9
5.0
6.021
9.47
10.52
0.796
200.00
400.00
69.3
81.5
5.5
6.021
9.85
11.17
0.819
100.00
200.00
79.85
130.9
5.0
2.0
6.021
8.82
7.55
0.616
200.00
400.00
79.85
81.5
5.5
2.0
6.021
8.65
11.91
1.254
200.00
400.00
69.3
97.6
5.5
2.0
6.021
8.06
9.48
0.864
200.00
400.00
79.85
97.6
5.5
2.0
6.021
7.43
10.40
1.341
200.00
400.00
79.85
114.8
5.5
V/V
dB
fO (MHz)
f–3dB (MHz)
Q
R1 (Ω)
R2 (Ω)
2.0
6.021
16.06
18.95
0.868
100.00
2.0
6.021
14.55
12.69
0.626
100.00
2.0
6.021
14.68
20.46
1.323
100.00
2.0
6.021
13.29
15.34
0.840
100.00
2.0
6.021
12.24
10.96
0.640
2.0
6.021
12.26
16.66
2.0
6.021
11.29
12.98
2.0
6.021
10.29
2.0
6.021
2.0
6.021
2.0
2.0
Table 8. Gain of 1.667 Filter Configurations
GAIN
V/V
dB
fO (MHz)
f–3dB MHz
Q
R1 (Ω)
R2 (Ω)
C1 (pF)
C2 (pF)
ein
nV/√Hz
1.667
4.437
19.67
19.35
0.696
80.00
133.33
48.2
81.5
5.1
1.667
4.437
17.97
22.12
0.934
80.00
133.33
48.2
97.6
5.1
1.667
4.437
16.57
23.16
1.336
80.00
133.33
48.2
114.8
5.1
1.667
4.437
16.28
15.60
0.679
80.00
133.33
58.75
97.6
5.1
1.667
4.437
15.01
17.80
0.875
80.00
133.33
58.75
114.8
5.1
1.667
4.437
14.33
18.58
1.046
80.00
133.33
58.75
126
5.1
1.667
4.437
13.82
13.19
0.676
80.00
133.33
69.3
114.8
5.1
1.667
4.437
12.94
14.77
0.826
80.00
133.33
69.3
130.9
5.1
1.667
4.437
12.06
11.32
0.666
80.00
133.33
79.85
130.9
5.1
66011f
24
LTC6601-1
APPLICATIONS INFORMATION
Table 9. Gain of 1.333 Filter Configurations
GAIN
C1 (pF)
C2 (pF)
ein
nV/√Hz
133.33
48.2
81.5
5.7
133.33
58.75
81.5
5.7
133.33
48.2
97.6
5.7
133.33
58.75
97.6
5.7
100.00
133.33
69.3
97.6
5.7
1.097
100.00
133.33
58.75
114.8
5.7
1.506
100.00
133.33
58.75
130.9
5.7
15.61
0.814
100.00
133.33
69.3
114.8
5.7
12.88
12.03
0.663
100.00
133.33
79.85
114.8
5.7
12.94
16.64
1.025
100.00
133.33
69.3
130.9
5.7
12.06
13.45
0.801
100.00
133.33
79.85
130.9
5.7
V/V
dB
fO (MHz)
f–3dB MHz
Q
R1 (Ω)
R2 (Ω)
1.333
2.499
19.67
22.73
0.841
100.00
1.333
2.499
17.82
15.77
0.633
100.00
1.333
2.499
17.97
24.34
1.185
100.00
1.333
2.499
16.28
18.44
0.818
100.00
1.333
2.499
14.99
13.58
0.646
1.333
2.499
15.01
19.82
1.333
2.499
14.06
20.12
1.333
2.499
13.82
1.333
2.499
1.333
2.499
1.333
2.499
Table 10. Gain of 1 Filter Configurations
GAIN
V/V
dB
fO (MHz)
f–3dB MHz
Q
R1 (Ω)
R2 (Ω)
C1 (pF)
C2 (pF)
ein
nV/√Hz
1.0
0.0
22.71
25.40
0.804
100.0
100.0
48.2
81.5
6.4
1.0
0.0
20.75
27.23
1.079
100.0
100.0
48.2
97.6
6.4
1.0
0.0
20.57
17.86
0.623
100.0
100.0
58.75
81.5
6.4
1.0
0.0
19.14
27.50
1.543
100.0
100.0
48.2
114.8
6.4
1.0
0.0
18.80
20.62
0.784
100.0
100.0
58.75
97.6
6.4
1.0
0.0
17.31
15.35
0.634
100.0
100.0
69.3
97.6
6.4
1.0
0.0
17.33
22.15
1.011
100.0
100.0
58.75
114.8
6.4
1.0
0.0
16.23
22.58
1.312
100.0
100.0
58.75
130.9
6.4
1.0
0.0
15.96
17.45
0.781
100.0
100.0
69.3
114.8
6.4
1.0
0.0
14.55
19.09
1.079
200.0
200.0
58.75
81.5
6.9
1.0
0.0
14.87
13.57
0.650
100.0
100.0
79.85
114.8
6.4
1.0
0.0
14.95
18.59
0.954
100.0
100.0
69.3
130.9
6.4
1.0
0.0
13.39
14.90
0.798
200.0
200.0
69.3
81.5
6.9
1.0
0.0
13.92
15.04
0.769
100.0
100.0
79.85
130.9
6.4
1.0
0.0
12.48
11.38
0.650
200.0
200.0
79.85
81.5
6.9
1.0
0.0
12.24
16.25
1.115
200.0
200.0
69.3
97.6
6.9
1.0
0.0
11.40
13.27
0.850
200.0
200.0
79.85
97.6
6.9
1.0
0.0
11.29
16.47
1.715
200.0
200.0
69.3
114.8
6.9
1.0
0.0
10.51
14.17
1.167
200.0
200.0
79.85
114.8
6.9
1.0
0.0
9.47
13.26
1.350
400.0
400.0
69.3
81.5
7.9
1.0
0.0
8.82
10.86
0.935
400.0
400.0
79.85
81.5
7.9
1.0
0.0
8.06
11.57
1.535
400.0
400.0
79.85
97.6
7.9
66011f
25
LTC6601-1
APPLICATIONS INFORMATION
20
19
18
17
16
20
LTC6601-1
1
R1
57.14W
15
11
6
7
8
9
10
20
19
18
17
16
15
11
6
7
8
9
10
20
19
18
17
16
10
20
19
18
17
16
15
+
–
11
6
7
8
9
10
20
19
18
17
16
LTC6601-1
1
15
1
+
11
8
9
+
4
–
5
15
2
R1
200Ω
7
9
5
LTC6601-1
6
8
4
–
4
7
2
R1
100Ω
2
6
1
+
5
R1
133.33Ω
11
LTC6601-1
1
4
–
5
LTC6601-1
2
16
+
4
–
5
R1
80Ω
17
15
2
R1
66.66W
4
18
1
+
2
19
LTC6601-1
–
5
11
10
6
20
19
18
17
7
8
9
10
16
LTC6601-1
1
R1
400Ω
15
+
2
4
–
5
11
6
7
8
9
10
66011 F07
Figure 7. Pin-Strap Hookup for a Particular R1
66011f
26
LTC6601-1
APPLICATIONS INFORMATION
20
19
18
17
16
20
LTC6601-1
1
15
5
11
6
7
8
9
10
20
19
18
17
16
–
11
6
7
8
9
10
20
19
18
17
16
LTC6601-1
1
15
1
+
11
7
8
9
+
R2
400Ω
4
–
5
15
2
R2
200Ω
6
R2
133Ω
5
LTC6601-1
4
16
+
4
–
2
17
15
2
R2
100Ω
4
18
1
+
2
19
LTC6601-1
–
5
11
10
6
7
8
9
66011 F08
10
Figure 8. Pin-Strap Hookup for a Particular R2
20
19
18
17
16
20
LTC6601-1
1
15
C1
48.2pF
4
–
5
11
6
7
8
9
10
20
19
18
17
16
C1
58.75pF
4
–
11
6
7
8
9
10
20
19
18
17
16
1
+
C1
69.3pF
–
5
11
7
+
5
15
6
16
LTC6601-1
1
4
17
15
2
LTC6601-1
2
18
1
+
2
19
LTC6601-1
8
9
10
15
+
2
C1
79.85pF
4
–
5
11
6
7
8
9
10
66011 F09
Figure 9. Pin-Strap Hookup for a Particular C1
66011f
27
LTC6601-1
APPLICATIONS INFORMATION
20
19
18
17
16
20
LTC6601-1
1
15
C2
81.5pF
4
–
5
11
6
7
8
9
10
20
19
18
17
16
C2
114.8pF
4
–
11
6
7
8
9
10
20
19
18
17
16
1
+
C2
97.6pF
–
5
11
7
+
5
15
6
16
LTC6601-1
1
4
17
15
2
LTC6601-1
2
18
1
+
2
19
LTC6601-1
8
9
10
15
+
2
C2
130.9pF
4
–
5
11
6
7
8
9
10
66011 F10
Figure 10. Pin-Strap Hookup for a Particular C2
66011f
28
LTC6601-1
APPLICATIONS INFORMATION
Example Filter Configurations of Basic 2nd Order
Filters
Figure 11 shows some simplified component hookups of
a selection of filters taken from Tables 7, 9 and 10. For
20
19
18
17
simplicity, VOCM pin bypass and power supply bypass
are not shown.
16
20
LTC6601-1
1
15
VOUT(DIFF)
–
11
7
19
VOUT(DIFF)
8
9
10
18
17
16
20
11
18
17
16
VOUT(DIFF)
8
9
–
5
10
11
6
GAIN = 6dB
fO = 9.85MHz
Q = 0.819
7
8
9
10
18
17
16
GAIN = 6dB
fO = 16.06MHz
Q = 0.868
19
18
17
16
20
LTC6601-1
15
1
+
15
+
2
VIN
VOUT(DIFF)
VIN
VOUT(DIFF)
4
–
5
11
7
19
LTC6601-1
1
6
19
+
4
5
4
10
VIN
–
2
9
15
2
VOUT(DIFF)
20
8
1
+
VIN
7
7
LTC6601-1
15
6
11
GAIN = 0dB
fO = 22.71MHz
Q = 0.804
1
4
–
6
LTC6601-1
2
+
5
GAIN = 0dB
fO = 13.92MHz
Q = 0.769
20
16
VIN
4
5
6
17
15
2
VIN
4
18
1
+
2
19
LTC6601-1
8
9
10
–
5
11
6
7
8
9
10
66011 F11
GAIN = 2.5dB
fO = 12.06MHz
Q = 0.801
GAIN = 2.5dB
fO = 19.67MHz
Q = 0.841
Figure 11. Basic 2nd Order Filter Configurations
66011f
29
LTC6601-1
APPLICATIONS INFORMATION
Figure 12 shows some simplified component hookups
of a selection of filters taken from Tables 4, 5, and 6. For
20
19
18
17
simplicity, VOCM pin bypass and power supply bypass
are not shown.
16
20
LTC6601-1
1
15
VOUT(DIFF)
11
VOUT(DIFF)
8
9
10
18
17
16
20
15
VOUT(DIFF)
11
17
16
VOUT(DIFF)
8
9
–
5
10
11
6
GAIN = 14dB
fO = 6.96MHz
Q = 0.579
7
8
9
10
18
17
16
GAIN = 14dB
fO = 11.36MHz
Q = 0.614
19
18
17
16
20
LTC6601-1
15
1
+
15
+
2
VIN
VOUT(DIFF)
VIN
VOUT(DIFF)
4
–
5
11
7
19
LTC6601-1
1
6
18
+
4
5
7
19
VIN
–
4
10
15
2
VIN
2
9
1
+
20
8
LTC6601-1
1
6
7
GAIN = 12dB
fO = 11.36MHz
Q = 0.834
19
4
11
6
LTC6601-1
2
–
5
GAIN = 12dB
fO = 6.96MHz
Q = 0.775
20
+
4
5
7
16
VIN
–
6
17
15
2
VIN
4
18
1
+
2
19
LTC6601-1
8
9
10
–
5
11
6
7
8
9
10
66011 F12
GAIN = 9.54dB
fO = 9.85MHz
Q = 0.544
GAIN = 9.54dB
fO = 16.06MHz
Q = 0.568
Figure 12. Basic 2nd Order Filter Configurations
66011f
30
LTC6601-1
APPLICATIONS INFORMATION
COMPLEX FILTER CONFIGURATIONS
A Modified 2nd Order Lowpass Filter Topology
The basic filter topology of Figure 3 can be modified as
shown in Figure 13. The Figure 13 circuit includes an
impedance path between the two summing nodes (the
circuit nodes common to resistors R1, R2 and R3). A
resistor and/or a capacitor connection between the summing nodes provide even more flexibility, and enhance
the filter design options (the fO and Q equations shown
in Figure 13 reduce to equations of Figure 3 if C3 is zero
and R4 is infinite).
The modified second order filter topology provides for
setting the Q value (with R4) without changing the fO
value and increasing the passband gain to greater than
one without changing the Q value (in the Q equation of
Figure 13 the value of Q does not change if the value of
the [1 + GAIN + 2(R2/R4)] denominator factor does not
change). Using R4 to set the Q value allows the option
to design the –3dB frequency (f3dB). If the Q value varies
and the fO value is constant then the f3dB frequency varies in a second order lowpass function (refer to the f3dB
equation of Figure 13).
Figures 15 to 17 show additional circuits highlighting the
use of R4 in the modified second order cicuit to set the f3dB
frequency to 7.5MHz, 10MHz and 15MHz respectively.
The design procedure for a specified f3dB frequency is
as follows:
1 Using the chosen C1, C2 and C3 values calculate the
fO value.
2. Using fO of step 1 and the specified f3dB calculate the
Q value.
3. Calculate the R4 value using the Q value of step 3.
4. Calculate the required external resistor REXT value for
the R4 value in step 3. Example, in Figure 14 the Q
value for f3dB = 5MHz is 0.54, the required R4 resistor
is 350Ω, the R4A and R4B resistors are the internal
100Ω and the REXT resistor is 150Ω [REXT = R4 – (R4A
+ R4B)].
Note: The modified second order filter topology requires
the use of at least two of the three input resistor pairs (two
of the three 400Ω, 200Ω and 100Ω pairs).
Figure 14 shows three configurations using a capacitor
(C3) and a resistor (R4) between the summing nodes.
The external 49.9Ω resistor isolates the LTC6601 outputs
from driving directly a capacitive load. The three circuits
of Figure 14 have equal fO and Q values and differ only in
the passband gain. The 150Ω R4 resistor sets a Q value
equal to 0.54 for an f3dB = 5MHz for fO = 6.954MHz.
66011f
31
LTC6601-1
APPLICATIONS INFORMATION
R2
C2
R1
C1
R3
R4A
+ –
C3A
VIN(DIFF)
REXT
R4B
VOUT(DIFF)
49.9Ω
– +
C3B
C1
R3
R1
C2
R2
66011 F13
R4 = R4A + R4B + REXT
C3 = C3A || C3B
(3568 • Q
6089 •
fO •
f3dB =
4
)
)
507.6 • Q
(9.891• 10 • f
(16 • f • (8.29 • 10 • f
5
2.109 • 10 •
0.2236 • fO •
Q=
12
O
2
3dB
9
4
)
(
)
5.486 • 109 • fO4 + 120 • 5.526 • 109 • f3dB2 + 3.082 • 106 • fO2 3dB
2
)
+ 4.127 • 109 • fO2 6.638 • 1010 • f3dB 4
)
1.25 • 10 4 • C1• Q • R2
R4 =
559 • C1•
VOUT(DIFF)
VIN(DIFF)
GAIN = –
fO =
(
1788 • Q 2 + 447 + 1.287 • 105 • 2 • Q 2 1 R2 •
=–
S2 +
VOUT(DIFF)
VIN(DIFF)
)
(
GAIN
R2 • R3 • C1• (C2 + 2 • C3)
)
R1• R2 • (2 • R3 + R4) + R3 • R4 + R2 • R3 • R4
=–
1
2••
(
C2 + 2 • C3 50 • Q • C1• (125 • GAIN + R2 + 125) C2 • R2
C1
R1• R2 • R3 • R4 • (C2 + 2 • C3)
•S+
1
R2 • R3 • C1• (C2 + 2 • C3)
R2
R1
R2 • R3 • C1• (C2 + 2 • C3)
Q=
R3 R2 •
C2
C3 C1 + 2 • C1
R2 R3 C2
1+ 1+| GAIN | + 2 • •
–
R4 R2 C1
Figure 13. Modified Filter Topology and Equations
66011f
32
LTC6601-1
APPLICATIONS INFORMATION
20
19
18
17
16
20
LTC6601-1
1
20Ω
75Ω
15
VIN
VOUT(DIFF)
+
VOUT(DIFF)
4
75Ω
5
11
7
16
20Ω
–
6
17
15
2
150Ω
4
18
1
+
2
VIN
19
LTC6601-1
8
9
–
5
11
10
6
GAIN = 1
fO = 6.954MHz
Q = 0.54
f–3dB = 5MHz
7
8
9
10
GAIN = 2.3
fO = 6.964MHz
Q = 0.54
f–3dB = 5MHz
20
19
18
17
16
LTC6601-1
75Ω
1
15
+
2
VIN
20Ω
VOUT(DIFF)
4
–
75Ω
5
11
6
7
GAIN = 3.3
fO = 6.964MHz
Q = 0.54
f–3dB = 5MHz
8
9
10
66011 F14a
Gain Magnitude vs Frequency (Gain = 1)
Passband Phase and Group Delay
10
30
0
0
PHASE
–30
PHASE (DEG)
GAIN (dB)
–60
–20
–30
–90
–120
50
GROUP DELAY
40
30
–40
GROUP DELAY (ns)
–10
20
–50
–60
100k
10
1M
10M
FREQUENCY (Hz)
100M
66011 F14b
0
100k
2M
4M
6M
FREQUENCY (Hz)
8M
0
10M
66011 F14c
Figure 14. Modified Filter Configuration Using a Capacitor and a Resistor Between Summing Nodes (f–3dB = 5MHz)
66011f
33
LTC6601-1
APPLICATIONS INFORMATION
20
19
18
17
16
20
12.4Ω
LTC6601-1
1
VIN
ZIN(DIFF) = 800
15
2
VOUT(DIFF)
–
12.4Ω
11
6
7
8
9
16
15
4
5
17
+
2
VIN
ZIN(DIFF) = 225
VOUT(DIFF)
4
18
1
+
24.9Ω
19
LTC6601-1
–
5
11
10
6
GAIN = 1
fO = 7.971MHz
Q = 0.67
f–3dB = 7.5MHz
7
8
9
10
GAIN = 3.56
fO = 7.971MHz
Q = 0.67
f–3dB = 7.5MHz
20
19
18
17
16
LTC6601-1
12.4Ω
1
VIN
ZIN(DIFF) = 175.6
15
+
2
VOUT(DIFF)
4
–
12.4Ω
5
11
6
7
8
9
10
66011 F15a
GAIN = 4.55
fO = 7.971MHz
Q = 0.67
f–3dB = 7.5MHz
Gain Magnitude vs Frequency (Gain = 1)
Passband Phase and Group Delay
10
30
0
0
PHASE
–30
PHASE (DEG)
GAIN (dB)
–60
–20
–30
–90
–120
50
40
GROUP DELAY
30
–40
GROUP DELAY (ns)
–10
20
–50
10
–60
100k
1M
10M
FREQUENCY (Hz)
100M
66011 F15b
0
100k
2M
4M
6M
FREQUENCY (Hz)
8M
0
10M
66011 F15c
Figure 15. Modified Filter Configuration Using a Resistor Between Summing Nodes (f–3dB = 7.5MHz)
66011f
34
LTC6601-1
APPLICATIONS INFORMATION
20
19
18
17
16
20
24.9Ω
LTC6601-1
1
15
4
VOUT(DIFF)
24.9Ω
11
6
7
8
9
16
+
4
–
5
17
15
2
VIN
ZIN(DIFF) = 250
VOUT(DIFF)
49.9Ω
18
1
+
2
VIN
ZIN(DIFF) = 400
19
LTC6601-1
–
5
11
10
6
GAIN = 0
fO = 11.27MHz
Q = 0.64
f–3dB = 10MHz
7
8
9
10
GAIN = 1.6
fO = 11.27MHz
Q = 0.64
f–3dB = 10MHz
20
19
18
17
16
LTC6601-1
1
15
24.9Ω
VIN
ZIN(DIFF) = 164
+
2
VOUT(DIFF)
24.9Ω
4
–
5
11
6
7
GAIN = 2.6
fO = 11.27MHz
Q = 0.64
f–3dB = 10MHz
8
9
10
66011 F16a
Gain Magnitude vs Frequency (Gain = 1)
Passband Phase and Group Delay
30
10
PHASE
0
0
–30
PHASE (DEG)
GAIN (dB)
–20
–30
–90
–120
50
40
30
GROUP DELAY
GROUP DELAY (ns)
–60
–10
20
–40
10
–50
100k
1M
10M
FREQUENCY (Hz)
100M
66011 F16b
0
100k
4M
8M
12M
FREQUENCY (Hz)
0
20M
16M
66011 F16c
Figure 16. Modified Filter Configuration Using a Resistor Between Summing Nodes (f–3dB = 10MHz)
66011f
35
LTC6601-1
APPLICATIONS INFORMATION
20
19
18
17
16
20
LTC6601-1
1
15
49.9Ω
VOUT(DIFF)
–
24.9Ω
11
6
7
8
9
16
+
4
5
17
15
2
VIN
ZIN(DIFF) = 250
VOUT(DIFF)
4
18
1
+
2
VIN
ZIN(DIFF) = 400
19
LTC6601-1
24.9Ω
–
5
11
10
6
7
8
9
10
GAIN = 1.6
fO = 16.04MHz
Q = 0.66
f–3dB = 15MHz
GAIN = 1
fO = 16.04MHz
Q = 0.56
f–3dB = 15MHz
20
19
18
17
16
LTC6601-1
1
15
24.9Ω
VIN
ZIN(DIFF) = 164
+
2
VOUT(DIFF)
24.9Ω
4
–
5
11
6
7
8
9
10
66011 F17a
GAIN = 2.6
fO = 16.04MHz
Q = 0.66
f–3dB = 15MHz
Gain Magnitude vs Frequency (Gain = 1)
Passband Phase and Group Delay
10
30
PHASE
0
0
–30
PHASE (DEG)
GAIN (dB)
–20
–90
–120
50
40
–30
30
GROUP DELAY
GROUP DELAY (ns)
–60
–10
20
–40
10
–50
100k
1M
10M
FREQUENCY (Hz)
100M
66011 F17b
0
100k
4M
8M
12M
FREQUENCY (Hz)
0
20M
16M
66011 F17c
Figure 17. Modified Filter Configuration Using a Resistor Between Summing Nodes (f–3dB = 15MHz)
66011f
36
LTC6601-1
APPLICATIONS INFORMATION
DC1251A Demonstration Board
The DC1251A demonstration circuit contains an LTC6601-1
(DC1251A-A). On a DC1251A the LTC6601-1 programming
pins can be connected through 0603 resistor jumpers. In
addition, optional surface mount capacitors and inductors
at the LTC6601 input and/or output can be installed for
additional filtering (a lowpass filter up to a 5th order can
be implemented with a DC1251A demonstration circuit).
The DC1251A has SMA connectors for the differential
input and output of the LTC6601-1. An on board 106MHz
lowpass RC filters the LTC6601-1 output.
DC12351A Top Silk Screen
66011f
37
38
J2
VIN–
J1
VIN+
C13
10μF
10V
C6
(OPT)
RZ2
C1
(OPT)
RZ1
ASSY
DC1251A-A
DC1251A-B
C14
1μF
10V
RG6
RG5
RG4
C2
(OPT)
RG3
RG2
RG1
U1
LTC6601CUF-1
LTC6601CUF-2
BOARD ASSEMBLY
(24AWG-VIA)
E2
GND
E1
V+ IN
2.7V TO 5.5V
(24AWG-VIA)
RIN2
C4 (OPT)
C3 (OPT)
RIN1
C5
(OPT)
V
+
C15
0.1μF
RQ1
R4
RC3
R5
20Ω 1% (OPT)
20Ω 1% (OPT)
RC2
C3
9
V–
V+
OUT
4
6
5
JP1
2
3
RF7
IN–
6
RF10
SHDN
LP
HP
V+
RF9
RF11
RF12
C4
10
OUT
C8
16
RF6
IN2–
C2
8
17
RF5
C7
LTC6601-1
C6
18
RF4
VOCM
C1
7
C5
19
RF3
IN1–
BIAS
IN1+
IN2+
IN3+
20
RF1
1
RF8
5
4
3
2
1
RF2
11
12
13
14
15
LTC6601-X Demonstration Circuit DC1251A
C12
0.01μF
R3
49.9Ω, 1%
R2
49.9Ω, 1%
C10
0.01μF
V+
R1
49.9Ω, 1%
C11
1000pF
C9
10pF
C8
10pF
E4
GND
(24AQG-VIA)
E3
EXT
VOCM
(24AWG-VIA)
J4
VOUT+
66011 DC
C7
10pF
J3
VOUT–
LTC6601-1
APPLICATIONS INFORMATION
66011f
LTC6601-1
PACKAGE DESCRIPTION
UF Package
20-Lead Plastic QFN (4mm × 4mm)
(Reference LTC DWG # 05-08-1710)
0.70 p0.05
4.50 p 0.05
3.10 p 0.05
2.45 p 0.05
(4 SIDES)
PACKAGE OUTLINE
0.25 p0.05
0.50 BSC
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
BOTTOM VIEW—EXPOSED PAD
4.00 p 0.10
(4 SIDES)
R = 0.115
TYP
0.75 p 0.05
PIN 1 NOTCH
R = 0.30 TYP
19 20
0.38 p 0.10
PIN 1
TOP MARK
(NOTE 6)
1
2
2.45 p 0.10
(4-SIDES)
(UF20) QFN 10-04
0.200 REF
0.00 – 0.05
0.25 p 0.05
0.50 BSC
NOTE:
1. DRAWING IS PROPOSED TO BE MADE A JEDEC PACKAGE OUTLINE MO-220
VARIATION (WGGD-1)—TO BE APPROVED
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
66011f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
39
LTC6601-1
TYPICAL APPLICATION
4th Order, 10MHz, Lowpass Filter with 12dB Gain
20
19
18
17
16
20
LTC6601-1
1
VIN
ZIN(DIFF) = 200
15
18
17
16
1
+
2
19
LTC6601-1
15
+
2
49.9Ω
VOUT(DIFF)
49.9Ω
4
4
–
5
11
6
7
8
9
–
5
10
11
6
7
8
9
10
66011 TA02a
Gain Magnitude vs Frequency
20
0
GAIN (dB)
–20
–40
–60
–80
–100
100k
1M
10M
FREQUENCY (Hz)
100M
66011 TA02b
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
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LT1993-2/LT1993-4/
LT1993-10
800MHz/900MHz/700MHz Low Distortion, Low Noise
Differential Amplifier/ADC Driver
AV = 2V/V / AV = 4V/V / AV = 10V/V, NF = 12.3dB/14.5dB/12.7dB,
OIP3 = 38dBm/40dBm/40dBm at 70MHz
LT1994
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LT6402-6/LT6402-12/
LT6402-20
300MHz Low Distortion, Low Noise Differential Amplifier/
ADC Driver
AV = 6dB/AV = 12dB/AV = 20dB, NF = 18.6dB/15dB/12.4dB,
OIP3 = 49dBm/43dBm/51dBm at 20MHz
LTC6404-1
Fully Differential Amplifier, GBW = 500MHz
Very Low Distortion, (2VP-P, 10MHz): –91dBc
LTC6404-2
Fully Differential Amplifier, GBW = 900MHz
Very Low Distortion, (2VP-P, 10MHz): –96dBc
LTC6404-4
Fully Differential Amplifier, GBW = 1700MHz
Very Low Distortion, (2VP-P, 10MHz): –101dBc
LT6600-2.5/LT6600-5/
LT6600-10/LT6600-20
Very Low Noise, Fully Differential Amplifier and Filter
2.5MHz/5MHz/10MHz/20MHz Integrated Filter, 3V Supply,
SO-8 Package
LTC6602
Dual, Matched Bandpass Filter
Programmable Gain and Bandwidth for RFID Applications
(40kHz to 1MHz)
LTC6603
Dual, Matched Lowpass Filter
Programmable Gain and Bandwidth (25kHz to 2.5MHz)
LTC6604-X
Dual, Matched Lowpass Filter
2.5MHz, 5MHz, 10MHz and 15MHz
LTC6605-X
Dual, Matched Lowpass Filter
7MHz, 10MHz and 14MHz
66011f
40 Linear Technology Corporation
LT 1108 • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
●
FAX: (408) 434-0507 ● www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2008