Download Li-Ion battery charger design for laptop computer applications

Li-Ion battery charger
design for laptop
computer applications
Song Qu
Applications Engineer
Cherry Semiconductor Corporation
Abstract
The paper discusses the practical design issues of Li-Ion battery chargers using Cherry Semiconductor CS5361/2 battery
charger controller. The controller provides all functions necessary for charging Li-Ion battery packs. The theory of operation
and design guideline is described and a reference design is presented.
Battery Current
I. Introduction
In recent years, Li-Ion rechargeable batteries are becoming more
and more popular in portable applications, due to their size,
weight and energy storage advantages. The growth in the LiIon battery market has drastically boosted the demand for high
performance battery chargers. Cherry Semiconductor CS5361/
2 battery charger controller is designed for charging Li-Ion battery packs. It can regulate both output current and output voltage with high accuracy and can provide smooth transition between current and voltage regulation. This paper will discuss
topology selection, control methods for current and voltage regulation, and compensation design. Because of its unique characteristics, the charging requirement for Li-Ion battery will be
introduced first.
II. Li-Ion Battery Charge Requirement
The differences in chemistry among various battery types result in different charge requirements. Li-Ion batteries are charged
with a constant-voltage, constant-current supply. Fig. 1 illustrates the charging voltage and current profiles of a Li-Ion battery. When the battery voltage is low, the charger operates in
constant current mode. The charger regulates the output current to a preset value. The battery voltage increases when it is
being charged. When it reaches the end-of-charge voltage (4.1
or 4.2V), the current will start to taper off. The charger enters
into constant voltage mode and regulates the output voltage to
the end-of-charge voltage. The current will continue to taper
off until it almost reaches zero. Charging can be terminated
after the current falls below a preset threshold.
The charge current is usually expressed as C-rate. The Crate describes the battery capacity in Ampere-Hours. It measures how much current is required to fully charge the battery
within one hour, assuming no losses. For a 1600 mAh battery,
Battery Voltage
Time
Figur e 1: Lithium-Ion Battery Charging Profile
1C charge current means charging the battery with 1.6A. Ideally, it will take one hour to fully charge the battery at 1C rate.
But because batteries are not 100% efficient in converting charge
current into stored charge, it takes longer than one hour to fully
charge the battery at 1C rate.
III. Selection of Charger Topology
There are two categories of battery chargers: linear chargers
and switching-regulator-based chargers. Most portable applications, such as laptop computers and cellular phones, require
high efficiency, smaller size and lighter weight. Therefore,
switching-regulator-based chargers are preferred in those applications. For battery charger applications, Buck regulator is
the most suitable one in different switching regulator topologies, because the output voltage is lower than the input voltage.
Fig. 2 shows the power stage of a typical Synchronous Buck
regulator. In Synchronous Buck topology, MOSFET Q2 replaces
the low side diode in a conventional Buck converter. This eliminates the power loss associated with diode forward voltage and
increases system efficiency. The PWM control circuitry modulates the duty cycle of the high side MOSFET Q1 to regulate
the output voltage. The switching actions of Q1 and Q2 are
International IC – China • Conference Proceedings 37
where ∆VO is the peak-to-peak output voltage ripple given by
the design specs. If the ESR value obtained from the above
equation is smaller than ESR specified in the capacitor
manufacturer’s data sheet, several capacitors should be
paralleled to get the right value.
IV. Current Regulation
Figur e 2: Power Stage of Synchronous Buck Regulator
complementary. At the beginning of each switching cycle, the
gate signal turns on the Q1 and turns off the Q2. Therefore, the
difference between input and output voltage (Vin-Vo) is applied across the inductor. The inductor current increases linearly. The energy is transferred from the input source to the
inductor. When Q1 turns off and Q2 turns on, the voltage across
the inductor becomes negative Vo and the inductor current linearly decreases. The energy stored in the inductor is transferred
to the load. A non-overlapping time is always inserted between
the turn-on signal of the Q1 and Q2 to prevent the cross conduction. The body diode of the Q2 will temporarily conduct
when both switches are turned off.
The value of the output inductor can be calculated based on
the inductor current ripple requirement:
The control method used in CS5361/2 for current regulation is
called Current-Mode Constant-Current control, also known as
I2 control for the purpose of brevity. A block diagram of I2
control for a Buck regulator is shown in Fig. 3. A constantfrequency clock signal initiates the on time of the high side
switch. There are two feedback loops. The feedback signal
ISENSE is the voltage across the sense resistor and is proportional to the inductor current. The DC portion is used by the
outer loop and is fed to Average Current Error Amplifier. The
Error Amplifier compares the feedback signal ISENSE to an
externally set reference voltage IAVG to generate a control voltage ICOMP. In CS5361/2, the ICOMP pin is open to users for
compensation design. The ripple portion of the feedback signal
is used as the ramp signal of the PWM comparator. When the
ramp signal intersects with the voltage of ICOMP pin, the output latch will be set. The controller turns off the high side switch
and turns off the low side switch. The duty cycle of the high
side switch is adjusted so that the output current (inductor current) can be regulated.
(1)
where Vo is the output voltage; is the period of one switching
cycle; ∆ IL is the peak-to-peak inductor current ripple given by
the design specifications; and D is the duty cycle of the high
side switch. Because both duty cycle and the output voltage are
changing during charge operation, the designer should determine the maximum possible product (I-D) of and Vo to calculate the inductance. The peak inductor current is given by:
(2)
where IL, peak is the peak inductor current and IO is the average
output current. The peak current should not saturate the core of
the inductor.
The output capacitor is used to filter out the inductor ripple
current. If the capacitance is large enough, most of the ripple
current will flow through the output capacitor. Under this circumstance, the inductor current ripple value and Equivalent
Series Resistance (ESR) of the output capacitor will determine
the output voltage ripple. The maximum allowable ESR is given
by:
(3)
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Figur e 3: Block Diagram of I2 Control Method
I2 control method has inherent compensation for duty cycle
in response to line voltage change. When there is a line voltage
variation, the inductor current ripple will change instantaneously,
so will the ramp signal. Therefore, the duty cycle can be adjusted on a pulse-by-pulse basis even before the Error Amplifier responds to the line variation. While the inner control loop
handles the transient response, the outer loop tightly regulates
the output current and provides easy loop compensation. A high
gain, low bandwidth error amplifier can be used to improve DC
accuracy, loop stability and noise immunity.
The I2 compensation network design is based on a smaller
signal model proposed in [1]. Fig. 4 is the block diagram of the
model. The model is accurate up to half of the switching
frequency and can predict the small signal characteristics of I2
control including sub-harmonic oscillation.
Fm is the gain of PWM comparator, which is:
(8)
where Sn is the on-time slope of the current sense waveform.
The closed current-loop control-to-inductor-current transfer function can be obtained from the block diagram.
(9)
The system is third order. The transfer function possesses
three poles and one zero. Since the denominator is very complicated, an approximate transfer function can be obtained by
ignoring some minor terms in the expression.
(10)
Figur e 4: Small Signal Model of I2 Control Method
The open loop duty-cycle-to-inductor-current transfer function of the Buck regulator can be derived using a standard PWM
switch model developed in [2]. When the Buck regulator is
operating in continuous conduction mode, the transfer function
has the following form:
(4)
Where RC is the Equivalent Series Resistance (ESR) of the
output capacitor. In the block diagram, Ri is the gain of the current sense network. He (s) represents the sampling action of
current-mode control. The sampling gain is invariant for all converters and has infinite number of poles and zeros. Actually, a
complex pair of RHP zeros provides a simplified representation of this transfer function, which is accurate up to half of the
switching frequency [3]. The transfer function is given by:
(5)
where
In Equation (10), the low frequency pole and the zero can
be cancelled out. The system becomes second order. It has only
a pair of complex poles at half of the switching frequency.
(11)
Notice that the complex pole pair is at very high frequency
(half of the switching frequency) and there is no zero in the
system, the compensation design becomes very simple. Only
one integration pole is required for the compensation network.
The pole gives the system high DC gain and makes it cross
over 0dB with a slope of -20dB/dec. After the 0dB crossover,
the system will come across the complex pole pair, which causes
the amplitude to decrease rapidly and increase the stability
margin. If a transconductrance amplifier is used as the error
amplifier, the pole can be implemented by grounding the amplifier output through a capacitor. The compensation gain is
given by:
(12)
where G is the transconductance, RO is the output impedance of
the error amplifier and Ccomp is the compensation capacitor.
Fig. 5 illustrates the effect of the compensation capacitor on
(6)
and
(7)
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frequency should be chosen well below the switching frequency.
(14)
V. Voltage Regulation
Current-mode voltage control method is used to regulate the
output voltage. The VFB pin of CS5361/2 monitors the battery
voltage. A resistor divider is used to scale down the voltage to
the reference level set at the VREF pin. CS5361/2 provides a 4.2V
reference voltage, which can obviate the need for a resistor divider network if charging a single 4.2V cell. The Voltage Error
Amplifier compares VFB and VREF. The output of the Error
Amplifier VCOMP is then compared with ramp signal, which is
generated from the inductor current ripple, to adjust the duty
cycle of the high side switch.
The small signal model of current-mode control is given in
[3]. The control-to-output transfer function with closed current
loop is given by:
Figur e 5: Bode Plot of Closed Current-Loop Control-to-Output
Transfer Function
––––––– With Compensation
–– –– –– Without Compensation
(15)
the system.
The total loop gain is
(13)
The value of the compensation capacitor can be calculated
if the crossover frequency is known. Generally, the crossover
Figur e 6: 18V/16.8V Four-Cell Battery Charger Design
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Compare the above expression with equation (11), we can
find the transfer function of current mode voltage control has
the same poles as I2 control. The difference is the zero. For I2
control, both ESR of the output capacitor and the load resistor
determine the zero. And it can be cancelled out with the low
frequency pole. But for current-mode voltage control, It is a
high frequency ESR zero. The low frequency pole cannot be
cancelled. So the system is third order. The Bode plot of the
closed loop control-to-output transfer function is illustrated in
Fig. 8. For a transconductance error amplifier, a possible compensation network is shown in Fig. 9. The compensation network has two poles and one zero. The compensation gain is
given by:
(16)
VII. Conclusion
Cherry Semiconductor’s CS5361/2 battery charger controller
was introduced. The design issues, such as topology selection,
current regulation, voltage regulation and compensation design,
for Lithium-Ion battery charger for laptop computers were discussed. A reference design and experimental waveforms were
given.
Reference:
The integrator pole will give the system high DC gain. Use
the zero to compensate the excessive phase delay caused by the
low frequency pole of the control-to-output transfer function.
The other pole should be placed around the ESR zero to assure
the amplitude decrease fast after the 0dB crossover.
VI. Reference Design
A 4-cell Lithium-Ion battery charger was built utilizing Cherry
Semiconductor’s CS5361. The schematic is illustrated in Figure 6 with all the component specifications. The input voltage
is 18V, the target charge current is 2A. The end-of-charge voltage is 16.8V. A Schottky diode is added at the input to block the
possible reverse current flowing from the battery to the input
power supply when the supply is shut off. The high side Pchannel MOSFET allows high input voltage and requires no
charge pump. Besides I≤ control, the part also features another
operation mode, which is bias mode. In the bias mode, the part
is shut off and the high side MOSFET is turned on so that the
battery can discharge to the load.
(a)
(a) Line Regulation of I2 Control
CH1: Inductor Current
CH2: Input Voltage
CH4: Switching Node Voltage
[1] S. Qu, D, Goder, “A Novel Battery Charger Solution Using I2 Control Method” Proceedings of High Frequency
Power Conversion Conference, November 9-11, 1999, Chicago, Illinois.
[2] V. Vorperian, “Simplified Analysis of PWM Converters
Using the Model of PWM Switch: Part I and II” IEEE
Transactions on Aerospace and Electronic System, March
1990, Vol.26, No.2.
[3] R. Ridley, “A New Continuous-Time Model for CurrentMode Control” Proceedings of Power Conversion and Intelligent Motion Conference, October 16-19, 1989, Long
Beach, CA.
Author’s contact details:
Song Qu
Cherry Semiconductor Corporation
2000 South County Trail, East Greenwich, RI 02818 USA
Phone: 1 401 886 3850
Fax: 1 401 886 3336
E-mail: [email protected]
(b)
(b) Current Regulation of I2 Control
CH1: Switching Node Voltage
CH3: Output Voltage
CH4: Inductor Current
Figur e 7: Experimental Waveforms
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Presentation Materials
Presentation Overview
CS5361/2 Battery Charger Controller
Li-Ion Battery Charge Requirement
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Selection of Charger Topology
Select Output Innductor
Select Output Capacitor
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Current Regulation Method
I2 Control Method
Small Signal Model of I2 Control
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Small Signal Model of I2 Control
Small Signal Model of I2 Control
Design Compensation
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Simulation Results
Design of Current Compensation
Small Signal Model of Current Control
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Design of Voltage Compensation
Reference Design
Key Operation Waveforms
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Experimental Results
Conclusion
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