Download EMI Filter Design

Input Filters for Triac-Dimmable LED Drivers by Thomas Stamm, STMicroelectronics
Abstract
EMI filters for LED lighting in the US have some special characteristics, among them the extremely
wide range of input voltage (96V to 305V) and the requirement to work with triac-based dimmers.
This paper explains in detail the design process for the filter, showing the sources for the two
components of noise and a method of dealing with the normal mode noise without causing dimmer
problems.
Ohms
C
C
Ohms
L
Mode
Normal
Ohms
C
C
Ohms
L
L
Mode
Common
Measure
Filter
EMI
Ground
to
Capacitance
Load
C
50
Capacitance
Source
LISN
Interwinding
Noise
Transformer
C
C
50
LISN
L
L
C
Measure
Filter
EMI
L
50
2
Line
LISN
Source
C
Noise
50
1
Line
LISN
L
L
Two methods are given for making the filter compatible with triac-based dimmers.
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Input Filters for Triac-Dimmable LED Drivers by Thomas Stamm, STMicroelectronics
Introduction
Driver circuits for illumination LEDs must meet several requirements, and ultimately,
Eventually the price must come down to where consumers will use LEDs for everything…
must be very inexpensive. One type, the PFC-Flyback converter, is becoming quite popular – it
provides line isolation, dimmability, and efficiency in a small space. Only one stage is required – a
single FET drives the flyback transformer, driven by a control chip such as ST’s L6562A, with
regulation of the LED current provided by a chip such as ST’s TSM1052. The circuit is described in
detail in the references.
The PFC-Flyback converter can be designed to handle a wide range of line voltage. Typically, US
customers want a range of 96V to 264V, with an occasional request for 305V. Dimmability with
common triac-based wall dimmers is sometimes required at 120V nominal line. The circuit can be
designed to handle both.
The input EMI filter for the PFC-Flyback converter must be very carefully designed if it is to operate
with triac dimmers. Any reversal of current following the triac firing will cause the triac to turn off,
leaving an unknown voltage on the dimmer’s filter capacitor. Since this voltage determines when the
triac will fire next, it must be the same for each half cycle or severe variations of conduction angle
will result, generating VERY annoying flicker.
Most EMI filters are designed for steady input voltage, smoothly varying sine waves with no
discontinuities.
This paper outlines a design method for the input EMI filter that will result in both good triac
operation and wide-range input capability with the PFC-Flyback power converter. The design process
will also work with the less efficient fixed-on-time discontinuous mode flyback converter, with minor
modifications.
Figure 1, below, shows what happens if the filter is not properly designed for use with a triac.
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Input Filters for Triac-Dimmable LED Drivers by Thomas Stamm, STMicroelectronics
Figure 1 - Triac current, turnon transient with undamped input filter
TRIAC DISCHARGING
DIMMER SNUBBER CAP
RINGING OF LINE INDUCTANCE
RINGING OF CM CHOKE AGAINST DC CAPACITOR
HOLDING CURRENT LOST HERE
GATE PULSE APPLIED
Filter Design
Two requirements dominate the design of the input filter, power factor, and conducted EMI. The
analysis below will derive the limits for values of some components and suggest processes to obtain
good values for others.
We will start with conducted EMI
Most readers will be familiar with the Line Impedance Simulation Network (LISN) which is specified
by the regulatory agencies worldwide. In all cases, the LISN terminates the high-frequency noise with
50 Ohm resistors to ground. The LISN contains other components, but these can be safely ignored.
Figure 2, Line Impedance Simulation Network (LISN) schematic
This will be replaced by a better diagram.
Carol, I have sent much better figures as separate PDF files. I can’t seem to delete the fuzzy image
below.
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Input Filters for Triac-Dimmable LED Drivers by Thomas Stamm, STMicroelectronics
Figure 3, below, shows limits for voltage and current with the LISN terminating the noise-producing
hardware.
RMS
20uA
=
RMS
1mV
=
RMS
6uA
=
RMS
0.3mV
=
4
30MHz
MHz
5
Line
Limit
500KHz
150KHz
Average
RMS
4uA
=
RMS
0.2mV
=
46dBuV
50dBuV
Line
Limit
Quasi-Peak
RMS
12uA
=
RMS
0.6mV
=
56dBuV
60dBuV
decade
per
-20dB
RMS
40uA
=
RMS
2mV
=
66dBuV
Figure 3, Regulatory Conducted EMI Limits (FCC and IEC) with Voltages and Currents Into 50 Ω
Input Filters for Triac-Dimmable LED Drivers by Thomas Stamm, STMicroelectronics
It is wise to keep the peak noise below the average limit line, both for safety margin and to avoid
time-consuming quasi-peak measurements.
EMI Filter Design
The next section deals with the filter design for the two main noise sources.
The bridge rectifier is ignored - two diodes in the bridge are always conducting in power factor
correction circuits, so the line is always connected to the noise sources.
Normal Mode and Common-Mode
Impedances are shown as boxes in Figure 4, because the elements are not pure inductors or
capacitors. Only the primary impedance is used in this analysis; self-resonance and internal
resistance are ignored.
Ohms
C
C
Ohms
L
L
Mode
Common
Measure
Filter
EMI
Ground
to
Capacitance
Load
C
50
Capacitance
Source
LISN
Interwinding
Noise
Transformer
C
C
50
LISN
L
L
C
Common-Mode Noise
Figure 4, Common-Mode Noise Circuit
This function will not be treated in detail. Common-mode characteristics affect neither the operation
of the triac-based dimmer nor power factor.
The common-mode noise is mostly due to the rapid rise of drain voltage as the FET switches off. The
usual treatment is to short-circuit the noise at its source, by placing a capacitor across the power
transformer.
Common-mode noise generally increases with increasing input voltage.
Normal Mode Noise
For triac-dimmable LED drivers this is the most critical filtering function.
Figure 5, Normal Mode Noise Circuit
5
Ohms
C
C
Ohms
L
Mode
Normal
Measure
Filter
EMI
L
50
2
Line
LISN
Source
C
Noise
50
1
Line
LISN
L
L
Input Filters for Triac-Dimmable LED Drivers by Thomas Stamm, STMicroelectronics
The normal mode noise source is the current drawn by the switching FET as it charges the flyback
inductor. A typical filter design (very general…) is shown in Figure 5 above. Note that the conducted
EMI measurement is taken across only one of the 50 Ohm resistors in the LISN. The figure shows the
total differential noise; common-mode noise is cancelled out because it is the same in both lines.
Piecemeal Analysis Technique
This analysis will treat each element of the EMI filter as if it were acting alone. As long as the section
resonances are well separated and the filter is well damped, there is little interaction. Only the
element impedances are used in the calculations. While not rigorous, the results are very usable.
1
1
Ohms
3
2
C
1
2
Ohms
D
N
2
Line
G
Choke
Common-Mode
50
R
Sink
C
Current
L
50
R
Inductance
Leakage
L3
or
INDUCTOR
Line
4
L
1
L
2
L
Figure 6, Components of the Normal-Mode EMI Circuit
Voltages appearing between the DC power rails (across the current sink) must be filtered at the
source – these are the primary source of noise in the 150KHz to 2MHz range (Figure 6).
The Normal-Mode Filter
We will next derive the filter characteristics required. For all power factor correction circuits, most of
the energy transfer occurs at the peaks of the AC sinewave. For EMI design purposes, we will treat
the input as DC at the peak level. The normal-mode noise current waveform is shown in Figure 7,
below.
Figure 7, FET Drain Current Waveform
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Input Filters for Triac-Dimmable LED Drivers by Thomas Stamm, STMicroelectronics
A SPICE FFT (Fast Fourier Transform) simulation (Figure 8) was run to find the harmonic components
of the ramp waveform. Note that the SPICE FFT function delivers peak voltage of the harmonics, not
RMS – EMI tests use RMS, 3dB below the peak values. Peak current was 1A, the load was 1 Ohm.
Risetime was set to 50% of the period, pulse duration 1% of period, and fall time at 3% of period.
The switching frequency does not affect the peak current – 75KHz was selected for the simulation.
The cursor is shown on the 10th harmonic. This waveform is representative of the current drawn
from the DC filter capacitor for a 15 Watt converter at about 96VRMS line.
Figure 8, Harmonic current series for the waveform of Figure *****
Notice that the low harmonics decay at about the same rate as the -20dB per decade decay of the
conducted EMI spec.
Table 1 - Harmonic Current Components of Figure 7 as a function of peak FET current
Fundamental
0.403
2nd
0.159
3rd
0.117
4th
0.077
5th
0.067
6th
0.050
7th
0.046
8th
0.036
9th
0.034
10th
0.027
If the resistive termination is replaced with a capacitor, the decay with frequency is -40dB / decade.
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Input Filters for Triac-Dimmable LED Drivers by Thomas Stamm, STMicroelectronics
Table 2 - Relative Harmonic Voltages with Capacitor Load (capacitor of any value), from the current
waveform of Figure 7
Fundamental
1
2nd
.197
3rd
.097
4th
.048
5th
.033
6th
.021
7th
.016
8th
.011
9th
.009
10th
.007
Figure 9 illustrates the FFT of the waveform of Figure 7, 1A peak into 0.1uF capacitor as described;
the switching frequencies are:
Green = 24KHz, Red = 50KHz, Blue = 104KHz
Note that for the same power level, lower operating frequencies generate considerably less noise
above 150 KHz.
The multi-stage filter will cause the higher harmonic amplitudes to decay even faster. So, if we can
filter the RMS component at 150 KHz down to less than 0.6mV, the spec will be met at all points.
The LISN divides the differential mode voltage in half (6dB) and the SPICE models show peak
voltages, not RMS (3 more dB) - giving us a design margin of 9dB. Use only if necessary… Design
margin is nice – don’t give it up unless you have to.
Select a value for the main filter capacitor, C1
There are two limits for the minimum value of the part:
1. The ability of the part to handle the fundamental current
2. The ability of the circuit and the part to tolerate ripple voltage
This part is generally a metalized mylar or polypropylene type. Detailed specifications for the part
used MUST include graphs of maximum voltage or current vs frequency. An agency-approved
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Input Filters for Triac-Dimmable LED Drivers by Thomas Stamm, STMicroelectronics
capacitor (“X” capacitor) is a poor choice and is unnecessary. This circuit location is “after” a “peak
voltage limiting circuit” - the capacitor does not need to be rated for line transients.
It has been found that a good compromise voltage on this capacitor is about 50V P-P at low line
where the current will be greatest. The voltage specified in capacitor catalogs is given in peak volts,
so the AC voltage rating required is 25V peak or 17.7V rms at the switching frequency.
For the dimmable filter in particular, the total shunt capacitance must be minimized. Each time the
triac fires, it must charge the shunt capacitance from zero to the instantaneous line voltage. At 90
degrees conduction angle, the step is the peak line voltage (170 Volts). Current through the
capacitors and unvarnished magnetic will cause the EMI filter components to audibly buzz at
120HzThe following analysis finds a practical minimum value for the filter capacitance across the
“DC” rails.
The current waveform of Figure 7 appears on the capacitor at the peak of the line voltage. The peak
current is:
Where Ipeak is the maximum current in Amps, Pout is the maximum average power to the LED load in
Watts,
is the efficiency of the converter (estimate 85%), and VinPEAK is the peak of the line voltage.
Find the capacitance for switching frequency ripple voltage on the capacitor:
Charge removed from C1 per cycle:
Find C1:
Substituting:
The capacitor current and voltage limits should be checked at the peak voltage for low line (96V
RMS, 136V peak) where the load current is greatest. The ramp duty cycle will be wider at the lower
voltage, placing maximum stress on the capacitor. For the purpose of this analysis, we will assume
50% duty at low line).
We will use a 15 Watt output PFC-Flyback converter as an example. Operating frequency will be set
to 50 KHz, a compromise between small filter size and small transformer size. Select the capacitor
for 50 Volts p-p due to the current ramp.
C1 = 2 * 15 / (0.85 * 50000 * 50 * 136) = 0.104µF
Use 0.1µF
The harmonic voltages are now known. We will concentrate on 150 KHz, the third harmonic.
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Input Filters for Triac-Dimmable LED Drivers by Thomas Stamm, STMicroelectronics
Note that C1’s capacitance can be split between the AC and DC sides of the bridge rectifier. It is best
to put most or all of the capacitance on the DC side to reduce the RMS current in the bridge diodes.
From Table 1, Vrms fundamental = 50V x 0.405 = 20.25V
From Table 2, Vrms 3rd Harmonic = Vrms fundamental x 0.097 =20.25 x 0.097 = 1.96Vrms
We now know the attenuation required at 150 KHz,
0.6mV/1.96V = 0.0003
Find the maximum value for the intermediate capacitor
The maximum total capacitance across the AC line and DC rails is determined by the power factor
requirement. Any capacitance at any location causes leading phase shift of the input current. (At
60Hz, the inductors in the filter have very little effect.) It does little good to have a power converter
with 0.99 input power factor and then reduce it to 0.8 with a poor input filter design.
Find the maximum value for the total capacitance:
(VARs)
Power
(VA)
Power
Angle
(Watts)
Power
Real
Phase
Apparent
Reactive
Figure 10, Real Power, Apparent Power, and Reactive Power relationship
Assume the power factor requirement is 0.9 minimum. The capacitive phase lead is cos-1 0.90, or
25.8 degrees maximum. The allowable ratio of reactive power to real power is tan 25.8 degrees, or
0.484.
The calculation is done for high line, where the phase angle will be greatest – the capacitance draws
current proportional to input voltage, the regulated power converter draws current inversely
proportional to input voltage.
Cmax = 0.484*Pout/(2πVin2fη)
= 1.30 microfarads for 15W, 85% efficiency at 132V 60Hz
For higher voltages the maximum capacitance drops rapidly:
0.325uF for 15W, 85% efficiency at 264V 60Hz
0.243uF for 15W, 85% efficiency at 305V 60Hz
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Input Filters for Triac-Dimmable LED Drivers by Thomas Stamm, STMicroelectronics
For our 15W example, if we want to work at 305V input, subtracting the 0.1uF already used at C1
gives a maximum value of about 0.143uF for C2. We will try 0.1uF. C2 must be a regulatory-approved
capacitor rated for 305V.
Outermost Stage
Because of the triac dimmer, we will try to operate without the capacitor at C3. If it is necessary to
put a part here, a resistor must be used in series to limit the current. Avoid this network if possible –
C3 must be a costly regulatory-approved part.
We will also try to operate with only a common-mode choke in the circuit. The part is needed
anyway for common-mode noise, and its leakage inductance can be exploited. The example part has
a two-section bobbin, which typically has a lower differential inductance than a 4-section bobbin. A
different choice here could save an external inductor and give better high-frequency attenuation.
For the inductors, we must consider parasitic resistance. The power dissipated in the resistance
limits efficiency. The effect is worst at low line voltage.
At 96Vrms input, the unit will draw about 15W/(0.85 * 96) = 0.183Amps RMS. Allow ¼ watt, then
R = P/I2 = 0.25/(0.183) 2 = 7.4 Ohms.
This is split between the two windings, so each has a maximum resistance of about 3.7 Ohms.
Select the CM inductor for 0.2A rms, 3.7 ohms max, in the smallest possible size.
Panasonic’s ELF14M020E has a 12mHy CM inductance, 4.3 Ohms resistance, and is rated for 0.2A.
Close enough.
Its differential inductance, as usual, is not specified. This, of course, is all we care about for
differential attenuation and triac compatibility.
To measure differential inductance, simply short one coil and measure the other. Several samples of
the Panasonic choke measured very close to 250uH.
Now find the maximum voltage on C2 (the intermediate capacitor) that will give less than 0.6mVRMS
at 150 KHz:
Figure 11, Equivalent circuit of line-side filter section
11
Input Filters for Triac-Dimmable LED Drivers by Thomas Stamm, STMicroelectronics
The voltage divider consisting of the differential inductance and the LISN can now be calculated. The
normal mode inductor’s impedance at 150 KHz is 235 ohms <90, and the LISN’s two inputs in series
are 100 ohms <0. Total impedance is (2352 + 1002)1/2 or 255 ohms. The limiting 150 KHz current of
the EMI spec is 12 microamps. The maximum RMS voltage across the 255 ohm impedance is 12e-6 *
255, or 3.06 millivolts.
The impedance of C2 (0.1uF) at 150KHz is 1/(2πfc) = 10.6 Ohms. The maximum current through this
capacitor at 150 KHz is 3.06mV/10.6 Ohms = 288uA.
So, starting with the maximum 150 KHz component of the voltage across C1, the combined
impedance of L1 and L2 must be 1.96V/288uA, or 6790 Ohms, or 7.2 mHy. Each inductor must be at
least half that value, or 3.6 millihenries. We will use 4.7mHy stock parts. These must be rated for the
peak low-line current for saturation, and the RMS line current for heating. Resistive losses are
present, and will reduce efficiency. TDK’s TSL1112RA-472JR21-PF will have a small loss of inductance
at peak line current at low line, not a problem – EMI tests are run at nominal line where peak
current is lower, and line dips to -20% are not common. The resistance of the part is 7.6 ohms
maximum, giving a nominal - line loss of about 0.3W each.
For a non-dimmable unit, C3 should not be needed.
The filter attenuation design is now complete. Filters designed by this procedure will work for nondimmable transition-mode or discontinuous mode flyback converters. Filters for triac-dimmable LED
drivers will require damping, covered in the next section.
We will now consider damping of the resonant networks. This is critical to operation with the triac
dimmer. The damping requirement will require filter modification for all but one of the damping
techniques.
We will continue with our 15 Watt output PFC-Flyback example as shown in Figure 6.
12
Input Filters for Triac-Dimmable LED Drivers by Thomas Stamm, STMicroelectronics
1
1
Ohms
3
L
3
2
C
1
C
1
2
Ohms
D
N
G
D
N
G
Choke
2
1
1
Ohms
2
2
Ohms
2
NETWORK
LINE-SIDE
NETWORK
LOAD-SIDE
Line
Choke
Common-Mode
50
R
Sink
C
Current
50
R
Inductance
Leakage
L3
or
INDUCTOR
Line
4
L
1
L
2
L
Line
2
L
Common-Mode
50
R
Sink
C
Current
L
50
R
Inductance
Leakage
L3
or
INDUCTOR
Line
4
L
1
L
Figure 12, EMI Filter, showing Line-Side and Load-Side Resonant Networks
The AC line impedance can be quite low (less than 10 ohms resistive due to shunt incandescent
lamps) in some locations and quite high (150 microhenries or so) in others. It is always a good idea to
include enough series inductance in the line-side filter section to stabilize the impedance and
resonance of this section. For the 15W example, 250uH may be enough, 500uH is safer. We will use
250uH for our calculations.
In the lab, a typical power source is a variable transformer (variac) which is usually followed by an
isolation transformer for safety. The transformer has significant leakage inductance, which increases
the source impedance of the line, softening current transients. Testing should also be done with the
lowest line impedance available.
The line-side network has a resonant circuit consisting of the differential inductance of L3, L4, the AC
line inductance, and capacitor C2, which participates due to the low network impedance. C1, L1 and
L2 can be neglected – C2’s impedance is much lower than the load side network at its resonance.
The line-side resonant frequency is
And the network impedance is
The resonant frequency is 31.8 KHz. Each half cycle is about 16 microseconds. Since it takes at least
20 microseconds of current reversal to turn off the triac, this resonance is not a concern - the initial
current spike causes sufficient carriers to be stored in the triac junctions to maintain conduction.
13
Input Filters for Triac-Dimmable LED Drivers by Thomas Stamm, STMicroelectronics
The network impedance is 50 Ohms. When the triac fires at 90 degrees, its peak current will be
170V/50 Ohms, or 3.4A. If the 100 Ohm LISN is representative of line impedance, the peak current
will be limited to Vline/100 Ohms, 1.2A. This is still sufficient to solidly turn on the triac, and ringing
will be less because of the additional resistance.
The load side network consists of C1 (the DC shunt capacitor), inductors L1 and L2, and the parallel
resonant network just examined. We will temporarily ignore the line side network and consider C2
shorted.
The resonant frequency calculates to 5190 Hz, and the network impedance to 306 Ohms. At this low
frequency the line impedance and the leakage inductance of L3 are truly low enough to ignore.
However, resonant current still flows in the network, the bulk of it through the line when it is
connected by the triac.
The load side resonant network cannot be ignored. Current in the triac cannot be allowed to reverse
for 96 microseconds. Damping is required to prevent the current reversal - circuit Q must be less
than 1.
Some damping is already present - the PFC-Flyback converter has a resistive input characteristic –
current is proportional to voltage over short periods. At 132V line the converter will have its highest
input resistance:
To get down to 306 Ohms of shunt resistance, we must parallel the PFC-Flyback load with additional
resistance. This can be added either of two places:
Across the load (isolated by a capacitor at least 3 times the value of C1), or
Across the inductors (which affects attenuation – the input inductance must be increased)
Figure 13, Adding damping to the load-side filter section
14
1
1
Ohms
3
L
3
2
C
2
C
1
C
1
2
Ohms
50
D
N
G
D
N
2
1
1
Ohms
4
DAMPER
2
Ohms
Choke
5
2
Line
2
L
R
Common-Mode
50
R
Sink
R-L
Current
R
50
R
Inductance
Leakage
L3
or
INDUCTOR
Line
4
L
1
L
Line
G
Choke
Common-Mode
4
C
R
Sink
C
Current
L
50
DAMPER
R-C
3
R
R
Inductance
Leakage
L3
or
INDUCTOR
Line
4
L
1
L
2
L
Input Filters for Triac-Dimmable LED Drivers by Thomas Stamm, STMicroelectronics
The R-C damper can be used if the high line limit is only 132V, but due to the power factor
capacitance limits we must use resistance across the inductors for higher line voltages. Attenuation
must be increased elsewhere to compensate.
We will first examine the effect of placing resistors across L1 and L2. The total resistance needed will
be 1/(1/306 -1/987) = 443 Ohms. Split this in half, placing a 220 Ohm resistor across each of L1 and
L2.
An aside…
Note that resistors may be needed here in any case, due to the self-resonance of L1 and L2. For the
TDK parts chosen, the resonance is 540KHz. This would appear in the EMI plot as a broad peak
around the resonant frequency. If damping only the parallel self-resonance, calculate the inductive
reactance at the resonant frequency and add a resistor of that value across each inductor. For the
TDK inductors in the example, use R = 15.9K, or 8K across each inductor. The effect on attenuation at
150KHz is negligible for the inductor chosen. The effect at the resonant frequency is dramatic.
Back to the subject…
Now we must correct the attenuation. The required impedance was 6790 Ohms. This is being
reduced to 440 Ohms, a factor of 15.3. Attenuation must be improved by this amount.
The original impedance of the circuit across C2 was 255 Ohms, consisting of the LISN and the
differential inductance of L3. The new impedance must be 15.3 times this, or 3901 Ohms. The
inductor we will add will dominate the impedance – the LISN does not contribute enough to matter.
L3’s new value is 4.14mHy. We can add 3.9mHy to the differential inductance of the CM choke.
Or we can add capacitance at C1 or C2 up to the power factor limit.
15
Input Filters for Triac-Dimmable LED Drivers by Thomas Stamm, STMicroelectronics
Or we can add a capacitor (and resistor) at C3 up to the power factor limit.
If high line is 264V, C1 or C2 can be increased to 0.22uF, and L3 reduced to 1.88mHy.
If high line is 305V, C1 or C2 can only be increased by 0.14uF. Not worth doing. For our 15 Watt
example it is not likely that a 96-305V line range will be possible with triac dimming at 120V, even if
the switching frequency is lowered to 20 KHz.
Next we examine the R-C network solution, usable only for 120V line. The capacitor value will be
0.33uF. The addition of the damping network will affect both the resonant frequency and the
network impedance. Both the resonance and the impedance will be about half the original values, or
2600Hz and 153 Ohms. The input resistance of the converter is in parallel with the network
(approximately), so the resistor of the R-C damper is 1/(1/153 – 1/987) = 181 Ohms.
Note that the dissipation in the resistor is very low if dimming is not present. The capacitor has a
reactance of 8040 Ohms at 60Hz. Neglecting the resistor, reactive current due to the damper
capacitor at 60Hz is only 15mA, dissipation is 40mW. The ripple current from the converter causes
additional dissipation of 0.224W.
With the dimmer at 90 degrees, however, dissipation increases dramatically. At high line, the
transient voltage is 186V, and the 0.33uF damper capacitor stores
= 5.7 millijoules.
This must be delivered by the damper resistor 120 times a second. The resistor dissipates the same
energy delivered to the capacitor, so the resistor power is
0.0057 x 120 = 0.685W, plus half the 50KHz current from the converter, or 0.797W.
Everything has its price…
References
ST’s AN510 – Circuits for Power Factor Correction with regards to Mains Filtering
(Google for this – several copies exist, but not on ST’s website)
ST’s AN1059 – Design Equations of High-Power-Factor Flyback Converters based on the L6561
http://www.st.com/stonline/books/pdf/docs/5956.pdf
ST’s AN1060 – Flyback Converters with the L6561 PFC Controller
http://www.st.com/stonline/products/literature/an/5957.pdf
ST’s AN2711 - 15 W offline TRIAC dimmable LED driv
http://www.st.com/stonline/products/literature/anp/14425.pdf
Data sheets for ST’s L6561, L6562, L6562A , L6563 Transition-Mode Power Factor Controllers
L6561 Datasheet
http://www.st.com/stonline/books/pdf/docs/5408.pdf
L6562 Datasheet
http://www.st.com/stonline/products/literature/ds/9461.pdf
L6562A Datasheet
http://www.st.com/stonline/products/literature/ds/13198.pdf
L6562AT (for outdoors) http://www.st.com/stonline/products/literature/ds/15310/l6562at.pdf
L6563S Datasheet
http://www.st.com/stonline/products/literature/ds/16116.pdf
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