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Transcript
E.ON Energy Research Center Series
High-Power DC-DC Converter
Nils Soltau, Robert U. Lenke
Rik W. De Doncker
Volume 5, Issue 5
E.ON Energy Research Center Series
High-Power DC-DC Converter
Nils Soltau, Robert U. Lenke
Rik W. De Doncker
Volume 5, Issue 5
Contents
1 Executive Summary
1
2 Introduction
3
2.1
Motivation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
3
2.2
Target Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
4
2.2.1
Collector Grids for Offshore Wind Farms . . . . . . . . . . . . . . . . . .
4
2.2.2
Collector Grids for Photovoltaic Applications . . . . . . . . . . . . . . .
8
2.2.3
Distribution and Transmission Grids . . . . . . . . . . . . . . . . . . . .
8
2.2.4
Solid-State AC Transformers . . . . . . . . . . . . . . . . . . . . . . . . 10
2.2.5
Subsea Production Facilities . . . . . . . . . . . . . . . . . . . . . . . . . 10
2.2.6
Technical Requirements . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
2.2.7
DC-DC Converter Concepts . . . . . . . . . . . . . . . . . . . . . . . . . 12
2.3
Operation Principle . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
2.4
Scope of Work
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
3 Control Techniques
3.1
15
Modeling . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
3.1.1
State-Variable Averaging and State-Space Averaging . . . . . . . . . . . 15
3.1.2
First Harmonic Approximation . . . . . . . . . . . . . . . . . . . . . . . 17
3.1.3
Verification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
3.2
Instantaneous Current Control . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
3.3
Current and Voltage Feed-Back Control . . . . . . . . . . . . . . . . . . . . . . 22
3.4
Balancing Control . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
4 Power-Electronic Switches and Soft-Switching Operation
27
4.1
Series Connection of IGCT’s . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
4.2
IGCTs under Soft-Switching Conditions . . . . . . . . . . . . . . . . . . . . . . 28
4.3
Application in a Dual-Active Bridge . . . . . . . . . . . . . . . . . . . . . . . . 30
4.4
Auxiliary Resonant-Commutated Pole . . . . . . . . . . . . . . . . . . . . . . . 31
5 Medium-Frequency Transformer
37
5.1
Review on Windings and Core Materials . . . . . . . . . . . . . . . . . . . . . . 37
5.2
Core Losses in a Dual-Active Bridge Application . . . . . . . . . . . . . . . . . 39
5.3
Design Aspects . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42
i
Contents
6 Demonstrator
47
6.1
Control Implementation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47
6.2
Converter Design and Construction . . . . . . . . . . . . . . . . . . . . . . . . . 53
6.3
Commissioning . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 59
6.4
Derating due to Single-Phase Setup . . . . . . . . . . . . . . . . . . . . . . . . . 64
7 Conclusion
67
8 Further Steps and Future Development
68
9 Bibliography
70
10 Attachments
77
10.1 List of Figures
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 77
10.2 List of Tables . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 79
10.3 Related Publications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 79
10.4 Short CV of Scientists Involved in the Project . . . . . . . . . . . . . . . . . . . 81
10.5 Project Timeline . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 83
10.6 Activities within the Scope of the Project . . . . . . . . . . . . . . . . . . . . . 83
ii
1 Executive Summary
Electrical energy distribution based on direct current (dc) have demonstrated superior performance as compared to conventional alternating current (ac) systems in low-voltage distribution
networks (e.g. data centers) and in high-voltage point-to-point transmission systems. Highpower medium-voltage dc-dc converters are key enablers to establish dc energy distribution
systems at medium-voltage level. The three-phase dual-active bridge dc-dc converter (DABC)
has been identified as a promising topology for efficient, high-power dc-dc conversion due to
its galvanic isolation, inherently low switching losses, bidirectional power control and small
filter components.
In this project, dynamic models of the DABC were developed for the first time to provide
high-bandwidth, instantaneous current control, that enable fast response of the DABC to load
variations (a full power step response without oscillations can be made within one third of a
switching period). Furthermore, a current balancing control algorithm compensates the unbalanced effects of asymmetric impedances of the high-frequency transformer in the converter’s
ac link. Based on these innovations, a controller could be developed that operates accurately
and stable under all operating conditions using three-phase transformers with standard core
design. The newly developed control mechanisms have been implemented on an industrial
control platform to demonstrate feasibility and functionality.
The great advantage of the DABC is its inherent soft-switching capability over a wide voltage operating range. Consequently, the switching losses in the power semiconductors are
substantially reduced, especially, when lossless snubbers are used. In applications with elevated voltage rating, these snubbers additionally ensure the dynamic voltage balancing of
series-connected devices. To benefit from the soft-switching operation mode over the entire
operating range, auxiliary circuitry has been investigated, designed and demonstrated. It has
been found that this circuitry offers the possibility of preventing short-circuiting the phase-legs
of the converters.
A key component of the DABC is a medium- to high-frequency transformer that connects the
input and output bridges. This transformer is operated at elevated frequency which allows
high power density and lower core losses as compared to 50 Hz transformers. Different core
materials have been characterized and the core design was optimized through finite-element
simulations and experiments. A compact 2.2 MVA single-phase prototype transformer was
built to demonstrate the potential of medium-frequency voltage conversion.
1
Executive Summary
Finally, a DABC with rated dc-link voltage of 5 kV and continuous power of 5 MW has been
designed and constructed. The commissioning confirms the applicability of thyristor-based
power-semiconductors and clarifies the advantages of the three-phase dual-active bridge converter compared to its single-phase equivalent – especially in high-power applications.
2
2 Introduction
2.1 Motivation
In order to assure a responsible and safe energy generation in the future, certain trends can be
observed worldwide. All over the world, people pursue the reduction of carbon emission and
the fortification of renewable energy sources. Many experts believe that technically speaking,
by 2030 the world’s electrical energy demand can be provided solely by renewable energy
sources [1].
These sources are mostly decentralized, and the transmission and distribution of the electrical
energy to urban areas is a major challenge [2]. The use of multi-terminal high-voltage dc
(HVDC) transmission grids promises a very flexible and efficient way to transport energy
over long distances [3–5]. Similarly, medium-voltage dc (MVDC) grids provide very flexible
energy distribution within a smaller area like industrial areas and city quarters. Particularly,
it has been shown that MVDC grids are feasible and that voltages and currents during fault
conditions are manageable [6, 7]. In many publications, the benefits of MVDC as collector
grids in wind farms [8–11] or solar power plants [12] are analyzed and demonstrated.
Besides the transmission challenges, the increasing amount of renewables in the ac grid and
the increased use of distributed generation have a huge impact on the power quality [13, 14].
To overcome the problems related to power fluctuation and the risk of voltage sags, storage
systems stabilizing the grid are essential [15, 16]. These storage systems, for example battery
energy storage systems (BESS), capacitor banks, flywheels, electrolyzers (power to gas) or gas
turbines, have a dc output or an ac output with variable frequency. Systems with variable
frequency outputs are usually connected to the ac grid via an ac-ac converter with intermediate
dc link. Most decentralized power sources and storage systems can be connected to a dc grid
more efficiently and with lower cost, because bulky 50 Hz filter components and transformers
as well as lossy PWM inverters can be avoided.
It is expected that in the near future energy management will benefit from dc grids. Multiterminal HVDC will be used to transmit bulk energy (tens of gigawatts) over large distances
and MVDC grids will be the preferred technique for energy distribution and collector grids.
DC technology enables an easy and efficient integration of storage systems. This is especially
interesting for industrial areas to overcome expensive peak loads.
One of the key-enabling components for high-voltage and medium-voltage dc grids is a dcdc converter, which is suitable for high-power applications. This dc-dc converter must be
3
Introduction
very efficient (above 99 %), easy to control and should offer redundancy if desired. Moreover,
galvanic isolation with high-voltage basic insulation level (BIL) requirements is need not only
for safety reasons but also when several converter modules operate in series and in parallel
connection. Concerning storage systems and distribution grids, a dc-dc converter must provide
bidirectional power flow as well.
Within this project, the potential and the performance of high-power dc-dc converters is evaluated and experimentally verified. Different topologies and their applicability in high-power
applications are compared. The most suitable candidate for a high-power dc-dc converter,
the three-phase dual-active bridge (DAB3), is investigated further. The key components of
the DAB3, namely the semiconductor devices and the medium-frequency transformer, are
analyzed towards their high-power handling capability. Based on the scientific findings, a
demonstrator of a DAB3 in the megawatt range is constructed.
2.2 Target Applications
A number of utility applications could potentially benefit from the availability of isolated
high-power dc-dc converters, either by improving the economics of existing solutions or by
providing extended or entirely new functionalities. This section provides an overview of these
applications as well as of the related requirements of the dc-dc conversion stages.
2.2.1 Collector Grids for Offshore Wind Farms
The capacity of offshore wind has tremendously increased over the last years. The development is depicted in Fig. 2.1. Since the nuclear disaster in Fukushima in 2011 and the
"Energiewende", public interest in offshore wind is growing even more.
Several new offshore wind farms have reached an advanced planning stage. As technology
develops and experience is being gained, the trend is to move large-scale wind farms into deeper
waters [17]. Figure 2.2 shows the distance from the shore and the water depth of wind farms,
planned for development after 2015. Furthermore, the European Wind Energy Association
(EWEA) assumes that wind farms, which are located around 100 km and more away from
the shore, need a high-voltage dc (HVDC) connection to generate energy economically [18].
Therefore, a great number of future offshore wind farms will be connected by a multi-terminal
HVDC station.
The main reason for using dc is the long distance to the onshore grid access points. The long
cables and the increased demand for reactive power compensation make an ac transmission
less efficient [8].
4
Introduction
1200
5000
1100
1000
Annual (MW)
800
700
3000
600
Cumulative (MW)
4000
900
500
2000
400
300
1000
200
100
0
1993 1994 1995 1996 1997 1998 1999 2000 2001 2002 2003 2004 2005 2006 2007 2008 2009 2010 2011 2012
Annual
0
2
5
17
0
3
0
4
51
170
276
90
90
93
Cumulative
5
7
12
29
29
32
32
36
86
256
532
622
712
804 1,123 1,496 2,073 2,956 3,829 4,995
318
373
577
883
874
1,166
Source: EWEA
Figure 2.1: Installed offshore wind capacity in Europe (1993-2012), Source: [17]
120
Distance to shore (km)
100
80
60
40
Online
20
Under
construction
Consented
0
20
10
20
30
Average Water depth (m)
40
50
Source: EWEA
Figure 2.2: Distance and depth of planned offshore wind farms (bubble size represents windfarm capacity), Source: [17]
5
Introduction
Figure 2.3: Schematic of a dc collector grid for offshore wind farms and the connection to
different dc sinks
Nowadays, the numerous wind turbines of one farm are connected via a 50 Hz ac collector grid.
At a central station the voltage is stepped up and converted to high-voltage dc, e.g. ±500 kV.
However, in classical designs at each wind turbine the 50 Hz ac voltage is generated from
dc. The conversion to dc within the wind turbine is necessary as the wind generators output
ac voltages of variable frequency. A more efficient and more reliable approach, however,
is to eliminate the ac converter and convert from low-voltage dc to HVDC using a dc-dc
converter [8], as shown Fig. 2.3. Not only would this be more efficient, but also heavy and
bulky 50 Hz components could be avoided, which is especially effective in offshore application.
An additional benefit of this dc collector field is the fact that dc storages such as battery
energy storage systems and electrolyzers can also be connected to the dc grid more efficiently.
Figure 2.4 shows different approaches to build a dc collector grid. The favorable topology
depends on the output voltage of the wind generator and the size of the wind farm [8].
However, the need for dc-dc converters, rated at different power levels, is evident. One dc-dc
converter has the same power rating as a wind generator. Depending on the generator, this
can be 1 MW–10 MW. A second dc-dc converter, which might be needed to step up the voltage
to HVDC level, corresponds to the power rating of the wind farm. This would be in the range
of 0.1 GW–10 GW.
Looking at requirements imposed on dc-dc converters, power flow is found to be almost unidirectional. With only a small power demand from the wind turbine during standby. Therefore,
power flow can be considered as highly asymmetric. The dc output voltage of the rectifier
6
Introduction
(a) Dispersed converter concept with series
connection (DCS)
(b) Centralized converter concept (CCC)
(c) Two step-up DC grid
Figure 2.4: Different topologies for dc collector grids
7
Introduction
located at the turbine usually is around 1.7 kV–2.6 kV [19]. The DCS and the two step-up
topologies, as depicted in Fig. 2.4, require a step up to MVDC level. The MVDC level is
around ±2.6 kV–±15 kV. The voltage level for HVDC is in the range of ±150 kV–±500 kV,
depending on the distance to the shore.
Galvanic isolation is favorable in nearly all dc-dc converters of the proposed collector-grid
topologies. Converters with galvanic isolation are more efficient at high voltage-conversion
ratios, needed for the MVDC-HVDC conversion. Considering the DCS topology, isolation
with respect to ground is mandatory, as a generator isolation for HVDC voltages is hardly
feasible. A dc-dc converter with galvanic isolation overcomes this issue.
2.2.2 Collector Grids for Photovoltaic Applications
Nowadays, large PV plants use ac collector grids as depicted in Fig. 2.5(a). In the shown
topologies one low-voltage (LV) inverter is applied per PV subfield. The energy of different
subfields is collected with an ac system, which suffers from high cable losses.
Similar to collector grids for offshore wind farms, the advantages of dc can be used in PV
applications as well. Each PV subfield is connected to a common MVDC collector grid through
a subfield dc-dc converter as shown in Fig. 2.5(b). From the dc collector grid one central
medium-voltage inverter feeds in the energy. Due to the savings of inverter and cable losses,
the European efficiency of a PV power plant can be improved from 96.3 % to 97.9 % [20]. An
additional boost in efficiency is expected when the dc configuration is connected to an MVDC
or HVDC grid.
Again, an efficient high-power dc-dc converter is the enabling technology.
2.2.3 Distribution and Transmission Grids
Medium and high-voltage dc distribution has been proposed for different applications in the
past [21–23]. Considering the increase of distributed generation, a medium-voltage dc infrastructure enhances the stability of the grid [24]. Furthermore, increasing urbanization might
make the higher power capability of dc systems a decisive feature for use in densely populated
areas [5]. If reliable high-temperature superconducting high-current cables become economically available, medium-voltage dc infrastructure might gradually replace high-voltage ac
(HVAC) distribution (e.g. gas insulated systems at 110 kV) as expensive high-voltage insulation becomes obsolete [25].
Additionally, it has been proposed to extend the idea of a dc grid infrastructure to the transmission level. In a scenario with a large number of HVDC-operated offshore wind farms, it
seems attractive to connect these to a common dc grid [26]. Similarly, projects like "Desertec"
8
Introduction
(a) ac collector grid
(b) dc collector grid
Figure 2.5: Collector grid topologies for a PV application
9
Introduction
integrate solar power from desert regions with the help of dc "superhighways" [27]. Moreover,
in new initiatives like Europe’s "Supergrid" the system’s dc voltage is increased further [28,
29]. It is reasonable to assume that in such a scenario there would arise a demand for dc-dc
converters, either to connect medium-voltage subgrids or to interconnect different HVDC grid
sections.
2.2.4 Solid-State AC Transformers
To coop with the increasing amount of distributed generation in current ac grid, solid-state
ac transformers (SST) are an effective solution to control the power flows in future ac grids.
McMurray, who named it electric transformer originally, patented it in 1970 [30]. The principle
of the SST is to achieve the ac voltage transformation through a high-frequency ac-link using
power electronics.
The structure of a SST is shown in Fig. 2.6. Exemplary, the figure depicts a transformation
from MVAC to HVAC.
Due to the dc-dc converter operated at high frequency, the weight and dimension of the
magnetic components is reduced. Its ability to compete with a conventional 50-Hz transformer
in terms of reliability, efficiency and power density still has to be proven. However, the SST
provides additional features as power-flow control, voltage sag compensation or fault current
limitation [31]. Additionally, it allows efficient connection of ac sources to dc grids.
Key component of the depicted SST is a high-power dc-dc converter. Alternatively, scientists
also promote a direct conversion from ac to ac using a bridge converter with a high-frequency
ac link and reverse-blocking semiconductor devices [32].
2.2.5 Subsea Production Facilities
Electrical submersible pumps are used for the extraction of oil and gas located under the
seabed [33]. To avoid the construction of an offshore platform, the facilities are installed on
the seabed. Supplying these facilities is a major challenge, as the systems are inaccessible
after the installation. Moreover, a high-voltage supply is necessary as the total consumption
can reach 100 MW [34].
Figure 2.6: Structure of a a solid-state ac transformer
10
Introduction
Using an existing HVDC line, these facilities could be energized efficiently. Furthermore, the
installation is simplified since bulky 50-Hz components are obsolete.
2.2.6 Technical Requirements
High-power dc-dc converters are one of the key technologies to establish a dc infrastructure.
In many applications, galvanic isolation is mandatory due to safety reasons. Especially at high
voltage conversion ratios, galvanic isolation prevents high circulating power and the resulting
losses.
Considering the exemplary applications stated above, other requirements for a high-power
dc-dc converter can be extracted.
High efficiency is a major requirement in all mentioned requirements. In some applications
a high power density is of particular importance. Depending on the application, the dcdc converter has to provide uni- or bidirectional power flow. Due to the high fluctuation
and dynamics of renewable energy sources, dc-dc converters have to provide highly dynamic
response characteristics as well.
The requirements for different applications are listed in Table 2.1 and also visualized in Fig. 2.7.
Similar to conventional ac transformers, one can observe a correlation between voltage level
and power rating.
Table 2.1: DC-DC converter requirements of utility-scale applications
Application
Offshore Wind Farms
turbine-mounted converters
central converters
PV Power Plant
subfield converters
central converters
Subsea Power Distribution
DC Grids
MVDC Grid Interlink
generator to MVDC Grid
storage to MVDC Grid
HVDC Grid Interlink
AC Grids
MV solid-state ac transformer (SST)
HV solid-state ac transformer (SST)
Power
Rating
Power
Flow
Voltage
Pri Sec
High Power
Density
3–10 MW
> 100 MW
Uni
Uni
MV
MV
MV
HV
×
×
0.5–5 MW
20–200 MW
10–100 MW
Uni
Uni
Uni
MV
MV
MV
MV
HV
HV
×
5–100 MW
0.5–20 MW
0.5–20 MW
> 100 MW
Bi
Uni
Bi
Bi
MV
MV
MV
HV
MV
MV
MV
HV
1–20 MW
> 40 MW
Bi
Bi
MV
HV
MV
MV
11
Introduction
Figure 2.7: Target ratings for different applications
2.2.7 DC-DC Converter Concepts
There is a huge variety of dc-dc converter concepts. They can be divided into the categories
"galvanically non-isolated" and "galvanically isolated". Whereas, the converters that are not
galvanically isolated usually have a poor efficiency when high voltage conversion ratios are
needed. Galvanically isolated converters can achieve high efficiency even at high conversion
ratios as they can step up (or step down) the voltage through the integrated transformer.
Considering medium-voltage high-power dc-dc conversion there are some potentially suitable
converter topologies. Examples are given in Table 2.2.
The focus in this work is on the three-phase dual-active bridge. It features soft-switching
operation, galvanic isolation, small filter components and low system complexity.
Table 2.2: Possible dc-dc converter topologies for medium-voltage applications
Topology
DC-DC Converter by Jovcic
Modular Multilevel DC Converter
Series-Resonant Converter
Dual Series-Resonant Converter
Single-Active Bridge
Single-Phase Dual-Active Bridge
Three-Phase Dual-Active Bridge
12
Comment
Galvanically non-isolated
Current source converter
Galvanically non-isolated
Unidirectional power flow
Unidirectional power flow
Alternative modulation schemes
Reduced current ripple
Higher transformer utilization
Example
[35]
[36]
[37]
[38]
[39]
[40]
Introduction
2.3 Operation Principle
In the following, the operation of the three-phase dual-active bridge is described [41, 42]. The
DAB, as depicted in Fig. 2.8, consists of two three-phase bridges that are connected by a threephase transformer in star connection. The transformer provides galvanic isolation and adjusts
the voltage ratio through its turns ratio. The bridges are operated at elevated frequencies, i.e.
in the kilohertz range for megawatt applications. Consequently, the mass and dimensions of
the transformer as well as the core losses are reduced compared to a 50 Hz transformer.
Both bridges are operated in fundamental frequency modulation. Therefore, a six-step voltage
waveform is applied to the primary and secondary side of the transformer. As also depicted
in Fig. 2.9, the output bridge is lagging the input by
∆t =
ϕ
ϕ
=
· Ts ,
2π · fs
2π
(2.1)
where fs is the switching frequency of the power-electronic switches and Ts the corresponding
period time. ϕ is referred to as load angle, analogous to a synchronous machine connected to
the ac grid.
Due to the voltage difference across the transformer, currents arise, leading to power flow
through the DAB3. The power flow is established through the stray inductance Lσ of the
three-phase transformer. The output power of the DAB3 is given by [42]
Up2
ϕ
2
Ps =
dϕ
−
ωLσ
3 2π
Up2
ϕ2
π
Ps =
d ϕ−
−
ωLσ
π
18
with the dynamic dc conversion ratio d =
Us0
Up .
voltage if a transformer with a turns ratio r =
π
3
for
0≤ϕ≤
for
π
2π
<ϕ≤
3
3
(2.2)
(2.3)
Us0 = r · Us is the primary-referred output
wp
ws
is applied. For the sake of simplicity and
without loss of generality, a transformer with a turn ratio of 1 is assumed in the following.
Figure 2.8: Schematic of the three-phase DAB
13
0
0
0
dc
currents
phase
currents
transformer voltages
secondary
primary
Introduction
0
Figure 2.9: Characteristic voltage and current waveforms in a DAB3
2.4 Scope of Work
This work is divided into several chapters. Firstly, the DAB3 converter is investigated in
terms of its dynamic behavior and dynamic models are derived. From the modeling work the
instantaneous current control is developed – a method to set any desired reference current
within one third of a switching period. Based on the current control, a voltage controller
is designed. Furthermore, a balancing strategy is developed that allows to compensate the
negative effect that an asymmetric transformers has on the DAB3.
In the following chapter, the preferred semiconductor switches, integrated gate-commutated
thyristors (IGCT), are investigated in a zero-voltage switching (ZVS) application. Hereby,
the focus is laid on the behavior in ZVS and on the series connection of devices in a DAB3.
Furthermore, the auxiliary resonant-commutated pole is introduced to achieve ZVS operation
in the entire working range.
The medium-frequency transformer is discussed in the chapter thereafter. Different core materials and winding configurations are discussed that are suitable for a high-power applications.
The loss effects in the transformer are investigated considering a DAB3 application. This
chapter closes with the design of a medium-frequency transformer for a DAB3.
The final chapter describes the construction of the medium-voltage high-power prototype.
The design of the power electronics as well as the transformer is discussed. Finally, measuring
results from the commissioning are presented.
14
3 Control Techniques
In this chapter the modeling of the three-phase dual-active bridge is introduced. From the
modeling work the instantaneous current control (ICC) was developed. With it any reference
current in a DAB3 can be set within one third of a switching period. Based on the ICC,
a voltage controller is presented. A balancing control is presented afterwards. It allows to
compensate the effect of asymmetrical transformers in a DAB3.
3.1 Modeling
The dynamic behavior of the DAB3 is modeled using two different approaches: a state-space
averaging approach and the first harmonic approximation. At first, both approaches are
introduced and afterwards they are compared to a circuit simulation.
3.1.1 State-Variable Averaging and State-Space Averaging
Firstly mentioned in [43, 44], state-space averaging (SSA) has become a common method to
describe the dynamic behavior of switched converters. State-variable averaging is used to
simplify the model. This has been applied also for three-phase dc-dc converters in [45].
The base for the model is the circuit diagram depicted in Fig. 3.1(a). Observing the corresponding voltage waveforms, one can identify six states [I–VI] within half a switching period.
In each state, the voltage applied to the transformer is constant. These states are the basis
for the SSA approach.
As described in more detail in [46], for each state, the system matrices A, B, C and D are
determined, so that
~x˙ = A · ~x + B · ~u
(3.1)
~y = C · ~x + D · ~u
(3.2)
with
~x = us ,
~y = us ,
~u = Up ,
C = 1,
D = 0.
(3.3)
Hereby, the load angle is assumed to be smaller than 60◦
15
Control Techniques
0
0
0
dc
currents
phase
currents
transformer voltages
secondary
primary
(a) Schematic
0
I II III IV V VI
(b) Waveforms
Figure 3.1: Schematics considered for the modeling approach
In general, the order of the system and with it the dimension of the system matrix A is according to the number of energy storage devices in the system. Neglecting the main (magnetizing)
inductance of the transformer and taking the stray inductances Lσ and the output dc capacitance Cout into account, there are four energy storage devices. A system of this order is quite
complex to invert analytically. Consequently, state-variable averaging is applied to reduce the
system order from four to one. Since the currents are fast changing compared to the output
voltage, the transformer current in each state can be represented by its mean-time average.
It turns out that the system matrices for the even states (Ae ,Be ) and the odd modes (Ao ,Bo )
are equal, respectively.
16
Control Techniques
This leads to the averaged system matrices
3
1
,
(Ao · ϕ + Ae · (π/3 − ϕ)) =
π
RL Cout
ϕ
ϕ 32 − 2π
3
B = (Bo · ϕ + Be · (π/3 − ϕ)) =
.
π
ωLσ Cout
A=
(3.4)
(3.5)
From the system matrices, one static and two dynamic equations can be derived that model
the DAB3. The static equation is given by
RL · ϕ 23 −
Us
=
Up
ωLσ
ϕ
2π
.
(3.6)
The same equation can also be derived from the general power-equation of a DAB3 (2.2). The
first dynamic equation gives the sensitivity of the output voltage with respect to disturbances
of the input voltage:
Us
ũs
ỹ Up
=
=
.
ũ ϕ̃=0 ũp
RL Cout · s + 1
(3.7)
The second small-signal transfer function from control to output gives the response of the
output voltage due to changes of the load angle ϕ:
Up RL 2 ϕ
ũs 1
=
−
.
ϕ̃ ũp =0
ωLσ 3 π RL Cout · s + 1
(3.8)
From (3.7) and (3.8) it is evident that the dynamic behavior of the considered DAB3 circuit
is solely determined by the RC output.
3.1.2 First Harmonic Approximation
Usually, first harmonic approximation (FHA) is used to describe the steady-state behavior
of a power-electronic circuit. The representation of ac quantities is simplified as only the
first-harmonic component is considered. All higher harmonic components are neglected.
Applying FHA in a DAB3 application results in sinusoidal transformer voltages and, consequently, to sinusoidal transformer currents. These three-phase quantities can be represented
by space vectors as demonstrated in Fig. 3.2. The secondary voltage is lagging the primary
voltage by the load angle ϕ and the transformer current is perpendicular to the voltage difference. This phasor diagram and the rotation of it in the αβ-plane is the central element of
the dynamic FHA model.
The entire dynamic FHA model is depicted in Fig. 3.3. Note the dc-link voltages are labeled
UpDC and UsDC unlike before. This improves the distinctness from the phasors of the ac
17
Control Techniques
Figure 3.2: Phasor diagram of a DAB3
Figure 3.3: Dynamic FHA model
voltages 1 U p and 1 U s .
The FHA model has two advantages. Firstly, it can be verified easily if the converter operates
in the soft-switching region. If the current phasor is located between both voltage phasors,
primary and secondary bridge are soft switched. This correlation is not affected by the simplification of the FHA. Since the rectangular waveforms only contains odd-numbered harmonics,
the actual zero-crossings are the same as for the first harmonic component. Secondly, the
model itself contains discrete models for the transformer and the output filter as indicated in
Fig. 3.3. These models, given as Laplace transform, are replaceable by other models.
3.1.3 Verification
Applying Matlab/Simulink [47] together with the PLECS power-electronics toolbox [48], both
models are compared to a detailed circuit simulation.
Figure 3.4 shows the results of the comparison. In (a) a step of the input voltage from 4.5 kV
to 5.5 kV is shown. The SSA model is in good agreement to the circuit simulation in terms
of dynamic behavior and in steady state. Since the it neglects higher harmonics, the FHA
model underestimates the transferred power. These higher harmonics contribute to the power
transfer in a DAB3. In (b) the steady-state value of the SSA model is different from the
circuit simulation as the operation point is different from the initial point around which the
system has been linearized. However, it is evident that the dynamic behavior matches very
well, which is predominantly determined by the output filter.
The fact that the dynamics of the current are limited by the leakage inductance in the ac
18
Control Techniques
4800
output voltage in V
output voltage in V
3800
3600
3400
Plecs
FHA
SSA
3200
3000
−0.02
0
0.02
time in s
4600
4400
4200
4000
3800
3600
0.04
(a) Input-voltage step from 4.5 kV to 5.5 kV
Plecs
FHA
SSA
0
0.01
0.02
time in s
0.03
(b) Load-angle step from 12◦ to 15◦
Figure 3.4: Comparison of models with circuit simulation
link is the motivation to set the ac currents as fast as possible. This directly leads to the
instantaneous current control.
3.2 Instantaneous Current Control
The transformer currents in a DAB3 can be represented in the αβ-plane using the Clarke
transformation [49]:
ip = ipα + jipβ =
2
ip1 + aip2 + a2 ip3 with a = e
3
j120◦
(3.9)
or
" #
ipα
ipβ
"
=
2 1
3 0
− 12
√
3
2
− 12
√
− 23
 
# ip1
 
ip2 
ip3
(3.10)
Consequently, all resulting current vectors are located on a hexagon-shaped trajectory as
illustrated in Fig. 3.5. The edge length of the hexagon is proportional to the load angle ϕ.
Illustratively, when in the time domain the currents are constant, the current vector rests in
one of the hexagon’s corners. As soon as a voltage is applied across the stray inductance and
the currents change, the current vector transitions from one corner to the following. Since this
time interval is proportional to the load angle, the edge length is as well.
When the rms value of the ac currents is changed from one reference value to another the edge
length will change. Changing the load angle abruptly shifts the hexagon away from the origin
of the αβ plane as illustrated in Fig. 3.6(a). Then the hexagon will travel back to the origin
according to the damping of the transformer.
19
Control Techniques
Figure 3.5: Applying the Clarke transformation to DAB3 currents
(a) Abrupt
(b) Method I
(c) Method II
Figure 3.6: Load-angle change without sign change
In the time domain, the shift of the hexagon results in diverging transformer currents and,
consequently, an oscillation on the dc currents. This effect is demonstrated in Fig. 3.7(a).
As also derived in [50], the oscillations can be nearly avoided applying one of two different
methods described in the following.
Using a two-step method, the load angle has to be applied at the same voltage transition in
every phase. Whether this is the rising of falling edge is not important. Consequently, the
current settles after two consecutive transitions. The two-step method of the instantaneous
current control (ICC) is illustrated in Fig. 3.6(b).
The second method (three-step method) applies three consecutive transitions with the mean
value of the old and the new load angle. Figure 3.6(c) and Fig. 3.7(b) demonstrate three-step
method in the αβ and the time domain.
As demonstrated in Fig. 3.8 both methods can be applied as well when the direction of the
powerflow is changed. Here the difference between the two methods is clearly visible. Applying
the two-step method, an overshoot results from a load-angle change with sign change (c.f.
Fig. 3.8(b)). The three-step method avoids this overshoot (c.f. Fig. 3.8(c)).
Improved ICC
Developing the Instantaneous Current Control (ICC), the influence of the winding resistance
Rt inductance has been neglected. Consequently, using the ICC it has to be ensured that the
20
20
0
phase currents in A
−20
5000
2500
0
−2500
−5000
dc current in A
dc current in A
phase currents in A
ϕ in degree
Control Techniques
4000
2000
0
−2000
−4000
0
0.02
0.04
0.06
0.08
5000
2500
0
−2500
−5000
4000
2000
0
−2000
−4000
0
0.02
0.04
0.06
0.08
time in s
time in s
(a) Abrupt
(b) Method II
Figure 3.7: Load-angle change in the time domain
(a) Abrupt
(b) Method I
(c) Method II
Figure 3.8: Load-angle change with sign change
decay time of the transformer
τ=
Lσ
Rt
(3.11)
is large compared to the switching period.
The Improved ICC (I2CC) has been developed that compensates the influence of the winding
resistance. As demonstrated in [51], the transition load angles are recalculated using the factor
κ=e
− 6f1 τ
s
.
(3.12)
Interestingly, this factor is constant over the entire operating range and only depends on the
parameters of the transformer.
Besides the correction value κ, the I2CC is analogue to the ICC. It can also be implemented
with the two methods (two-step and three-step) to achieve a commutation within one third
21
phase currents in A
phase currents in A
Control Techniques
1
0.5
0
−0.5
−1
1.5
ipDC in A
ipDC in A
1.5
1
0.5
0
−0.5
−1
1
0.5
0
1
0.5
0
0
0.2
0.4
0.6
0.8
0
time in ms
0.2
0.4
0.6
0.8
time in ms
(a) ICC
(b) I2CC
Figure 3.9: Comparison of original and improved ICC in measurement
and one half of a switching period, respectively.
Figure 3.9 shows experimental results, comparing the ICC with the I2CC. The experiment
was conducted with a lab prototype DAB3. In the setup, the primary and secondary dc-link
voltage are 50 V and the switches are operated with 10 kHz. The transformer has a stray
inductance of 250 µH and a winding resistance of 800 mΩ. Further details can be found in
[51]. The measurements demonstrate the outstanding performance of the proposed control.
3.3 Current and Voltage Feed-Back Control
The ICC sets an arbitrary reference current within one third of switching period. Consequently, when the parameters of the transformer are known, the current control can be made
very fast.
A feed-forward controller translates a new reference current directly into the corresponding
load angle. The controller applies, according to the power equation of a DAB3, the control
angle
ϕ=
2π
±
3
s
2π
3
2
−
2πωLσ ∗
i
.
Up sDC
(3.13)
Since (3.13) neglects the effect of the transformer’s magnetization inductance and the resistance of the winding, a feed-back control is mandatory to provide high precision. This can be
for example an PI regulator as shown in Fig. 3.10.
Applying the fast current regulator achieved through the ICC, a closed-loop voltage control
can be implemented in a cascaded manner as depicted in Fig. 3.11. With the cascaded control
structure, a robust voltage controller is achieved that is easy to design [52].
22
Control Techniques
Figure 3.10: Closed-loop current control
Figure 3.11: Closed-loop voltage control
To further improve the control performance, techniques like disturbance compensation or
dead-time compensation can be added [53, 54].
3.4 Balancing Control
A three-phase transformer may suffer from asymmetric impedances, in particular flat-core
laminated silicon-steel core transformers. Hence, the asymmetries might be due to the winding
arrangement, which is not symmetric or due to tolerances in production.
Especially asymmetric stray inductances can increase the voltage ripple and decrease the
utilization of the converter as demonstrated in Fig. 3.12. In this experiment, the primary and
secondary dc voltage is 60 V and the load angle ϕ = 38◦ . The switching frequency is 10 kHz.
The main inductances of the three-phase transformer are Lh1 = 2.7 mH, Lh2 = 2.1 mH and
Lh3 = 2.4 mH, respectively. The values of the asymmetric stray inductances are Lσ1 = 394 µH,
Lσ2 = 233 µH and Lσ3 = 248 µH, respectively. Figure 3.12 (a) shows that different series
impedances result in unequal phase currents. In the αβ-plane this corresponds to a deformed
hexagon (c.f. (b)). Due to the different impedances the ripples on the dc currents as well as on
the dc voltages increase (c.f. (c)–(f)). The dc-voltage ripples are calculated from the measured
dc currents. The current ripples are integrated assuming 100 µF dc-link capacitance.
As stated before the load angle is proportional to the edge length of the hexagon. Consequently
as also discussed in [55], introducing separate load angles for each phase, the deformed hexagon
can be corrected and the phase currents balanced.
Figure 3.13 shows a measurement with the same imperfect transformer as before. However,
separate load angles ϕ1 = 44.5◦ , ϕ2 = 33◦ and ϕ3 = 31◦ are applied. As demonstrated, the
dc voltage ripples are decreased and the utilization of the converter is improved.
23
Control Techniques
PSfrag
1
1
iβ in A
phase currents in A
2
0
−1
−1
−2
−0.2
−0.15
PSfrag
−0.1
0
−0.05
time in ms
0.05
−1
0.1
PSfrag
(a) Phase currents
output dc current in A
input dc current in A
1
0.5
0
0.05
−0.15 −0.1 −0.05
time in ms
0.1
1.5
1
0.5
0
0.15
(c) Input current ipDC
40
20
0
−20
−0.1
0
time in ms
(e) Input voltage ripple
0
0.05
−0.15 −0.1 −0.05
time in ms
0.1
(d) Output current isDC
output dc voltage ripple in mV
input dc voltage ripple in mV
1
2
1.5
−40
0
iα in A
(b) Phase currents represented in the
αβ-plane
2
0
0
0.1
40
20
0
−20
−40
−0.1
0
time in ms
0.1
(f ) Output voltage ripple
Figure 3.12: Influence of an asymmetric transformer on a DAB3
24
0.15
Control Techniques
PSfrag
1
1
iβ in A
phase currents in A
2
0
−1
−2
PSfrag
−1
−0.15 −0.1 −0.05
0.05
0
time in ms
0.1
−1
0.15
PSfrag
(a) Phase currents
output dc current in A
input dc current in A
1
0.5
0
0.05
−0.15 −0.1 −0.05
time in ms
0.1
1.5
1
0.5
0
0.15
(c) Input current ipDC
40
20
0
−20
−0.1
0
time in ms
(e) Input voltage ripple
0
0.05
−0.15 −0.1 −0.05
time in ms
0.1
0.15
(d) Output current isDC
output dc voltage ripple in mV
input dc voltage ripple in mV
1
2
1.5
−40
0
iα in A
(b) Phase currents represented in the
αβ-plane
2
0
0
0.1
40
20
0
−20
−40
−0.1
0
time in ms
0.1
(f ) Output voltage ripple
Figure 3.13: Transformer currents applying balancing angles
25
Control Techniques
26
4 Power-Electronic Switches and Soft-Switching Operation
The dual-active bridge requires power-electronic devices that are able to actively turn-off the
current. Insulated-gate bipolar transistors (IGBT) and integrated gate-commutated thyristors
(IGCT) are suitable for the considered multi-megawatt medium-voltage DAB3. As stated
before, the DAB3 inherently offers soft-switching capability in a certain operation range. This
encourages a closer investigation of IGCTs as they feature lower conduction losses compared
to IGBTs.
The first part of this chapter discusses how the lossless snubbers improve the application of
IGCT devices. On the one hand, the series connection of IGCTs and the dynamic voltage
sharing are investigated. On the other hand, the turn-off losses under quasi zero voltage
switching are measured. The measuring results are then included in a simulation to investigate
the switching losses in a dual-active bridge application.
4.1 Series Connection of IGCT’s
The series connection of power-electronic switches can be difficult in general. Timing delays
and device tolerances may lead to unequal voltage sharing across the switches. Consequently,
the maximal voltage-blocking capability of single devices might be exceeded.
In conventional hard-switched converters snubber circuits have to be used to ensure proper
voltage sharing if devices are connected in series. Figure 4.1 shows two different kinds: an
RC snubber (a) and an RCD snubber (b). The applied resistor Rsn limits the inrush current
if the IGCT switches on and the snubber is still charged (as it is the case in hard switched
converters). Consequently, the RCD snubber features slightly lower losses compared to the
RC snubber, since the resistor is bypassed during IGCT turn off [56]. However, the energy loss
in the snubber is still considerable. Moreover, the challenge of a low-inductive arrangement
near the GCT increases with device count.
As also discussed in [57], the lossless snubbers used in a DAB3 not only reduce the switching
losses but also ensure the proper voltage sharing of series-connected devices. The lossless
snubber is shown in Fig. 4.2 (a). Since the DAB3 is soft switched, the snubber capacitance
is not charged when the IGCT turns on. Consequently, the snubber resistor can be omitted.
Therefore, virtually no losses are generated in the snubber circuit. Figure 4.2 (b) and (c)
demonstrate the voltage sharing of two series-connected IGCTs during turn-off. The IGCTs
are of the type 5SHY55L4500 manufactured by ABB.
27
Power-Electronic Switches and Soft-Switching Operation
(a) RC snubber
(b) RCD snubber
3500
3000
2500
2000
1500
UGCT1 in V
UGCT2 in V
IGCT in A
1000
500
0
0
(a) Lossless snubber
device voltage and current
device voltage and current
Figure 4.1: Series connection of IGCTs using conventional snubber circuits for dynamic voltage balancing
50 100 150 200 250 300
3500
3000
2500
2000
1500
UGCT1 in V
UGCT2 in V
IGCT in A
1000
500
0
0
50 100 150 200 250 300
time in µs
time in µs
(b) Csn = 0.5 µF
(c) Csn = 2.5 µF
Figure 4.2: Voltage balancing applying lossless snubbers [57]
As the measurement shows, the voltage balancing is very effective even when using rather
small capacitances.
For the measurement both IGCTs are triggered synchronously. However, even when the timing
of the gating signals differs by 206 ns, [57] demonstrates that the voltage difference is below
400 V for 2.5 µF. Applying 7 µF, the voltage unbalance is below 200 V.
4.2 IGCTs under Soft-Switching Conditions
In a DAB3 that is operated in soft-switching mode, the IGCTs are turned on at zero voltage
(ZV) and zero current (ZC) at the instant the current commutates from the diode to the
IGCT. Furthermore, when a lossless snubber is applied, the turn off occurs at quasi zero
voltage (ZV), since the voltage rise is limited by the capacitor.
For the ZC turn-on of an IGCT a certain requirement has to be kept in mid. When the
current commutates from the anti-parallel diode to the GCT, the gate driver has to retrigger
the GCT. This is achieved by a gate pulse initiated automatically from the gate drive unit.
In order to succesfully retrigger the GCT internally, the anode-cathode current slope has to
28
Power-Electronic Switches and Soft-Switching Operation
vIGCT
vIGCT
3000
iIGCT
2000
2000
Csn=
1 μF
2 μF
3.5 μF
1000
p in MW
Csn=
1 μF
2 μF
3.5 μF
1000
0
0
3.0
2.66
2.33
2.0
1.66
1.33
1.0
0.66
0.33
0
iIGCT
10
EIGCT
1 μF
Csn= 2 μF
pIGCT
3.5 μF
8
6
4
2
0
5
t in μs
10
0
15
(a) IGCT optimized for conduction (ABB
5SHY 35L4512)
3.0
2.66
2.33
2.0
1.66
1.33
1.0
0.66
0.33
0
0
10
8
pIGCT
5
Csn=
1 μF
2 μF
3.5 μF
6
EIGCT
4
E in J
v in V, i in A
3000
2
t in μs
10
0
15
(b) IGCT optimized for switching (ABB
5SHY 35L4511)
Figure 4.3: Voltage and current transients during ZVS turn off [58]
be below a certain limit. For the considered IGCT devices, the data sheet gives the maximal
rate of rise of on-state current
! Up (1 + d)
di = 1000 A µs−1 >
.
dt crit
3Lσ
(4.1)
Although this requirement should be fulfilled for most DAB3 applications, an external retrigger
pulse can be applied to ensure homogeneous firing of the IGCT. During the commissioning of
the dc-dc converter, the IGCT ZC turn-on has been inconspicuous.
Due to the lossless snubber, the IGCTs turns off under ZV conditions virtually. Since data of
an IGCT under zero-voltage switching (ZVS) has not been available, detailed measurements
are carried out [58].
Figure 4.3 shows the measuring results for two different IGCTs. One IGCT (ABB 5SHY
35L4512) is optimized for low on-state voltage. This results in lower conduction losses, but
in high switching losses (c.f. (a)). The second IGCT (ABB 5SHY 35L4511) is optimized for
lower switching losses. This can be observed in Fig. 4.3(b) showing a reduced tail current
compared to the first IGCT.
Multiplying the device voltage and current gives the instantaneous power loss pIGCT . Integrat-
29
Power-Electronic Switches and Soft-Switching Operation
Eoff in J
15
0.0
1.0
2.0
3.5
μF
μF
μF
μF
20
15
Eoff in J
20
5
5
0
0
1000
2000
Ioff in A
Eoff in J
0.0
1.0
2.0
3.5
μF
μF
μF
μF
1000
0
0
3000
(a) ABB 5SHY 35L4512 - T = 25 ◦ ◦C
0
0
μF
μF
μF
μF
10
10
5
0.0
1.0
2.0
3.5
Ioff in A
3000
(c) ABB 5SHY 35L4511 - T = 25 ◦ ◦C
2000
Ioff in A
3000
(b) ABB 5SHY 35L4512 - T = 110 ◦ ◦C
5
2000
1000
0
0
0.0
1.0
2.0
3.5
μF
μF
μF
μF
1000
2000
Ioff in A
3000
(d) ABB 5SHY 35L4511 - T = 110 ◦ ◦C
Figure 4.4: Turn-off losses in presence of a lossless snubber for different IGCTs [58]
ing the instantaneous power gives the energy loss per turn-off cycle Eoff . Figure 4.4 gives the
turn-off energy for both devices, different snubber values and different case temperatures T .
As also observed in [59], already small capacitance values achieve a great reduction of the
switching losses. Further increase of the capacitance leads only to a slight decrease in the
switching losses, but causes extensively larger commutation times.
The data can be implemented into simulation models to evaluate the switching losses in softswitched converters applying IGCTs.
4.3 Application in a Dual-Active Bridge
Since the DAB3 is a soft-switched converter, it can be suggested that IGCTs are the superior switching device for this application. Compared to IGBTs, the IGCTs offer very low
conduction losses due to their thyristor structure.
Exemplary, a simulation is carried out to verify this assumption. For the sake of simplicity,
only the voltage conversion ratio d = 1 is considered. This is valid for most grid applications
30
Power-Electronic Switches and Soft-Switching Operation
Table 4.1: Simulation parameters
Primary dc voltage
Up = 5 kV
Secondary dc voltage
Us = 5 kV
Switching frequency
fs = 1 kHz
Transformer’s stray inductance
Lσ = 200 µH
IGBT
ABB 5SNA 2000K451300
IGCT
ABB 5SHY 40L4511
Diode
Infineon D1031SH
where only little voltage variation is expected. The parameters of the simulation are given in
Table 4.1.
The loss data of the StakPak IGBT and the diode is taken from the according data sheets [60,
61]. The losses of the IGCT are derived from the measurement described in the chapter 4.2.
Figure 4.5 shows the total semiconductor loss, including main switches and anti-parallel diodes.
The total conduction losses of the main switches and the anti-parallel diodes for IGBTs and
IGCTs are similar. This is due to the lower conduction losses of the IGBT’s anti-parallel diode
compared to the considered Infineon diodes for IGCT applications. These Infineon diodes are
optimized for the fast switching transients of the IGCTs in hard switching applications and
suffer from higher conduction losses. They are also applied in the demonstrator introduced
later due to availability reasons. Using anti-parallel diodes optimized for low conduction losses
could increase the converter efficiency further.
However, it is evident that the lossless snubbers decrease the switching losses and improve
the converter efficiency. Applying 1 µF to each switch increases the converter efficiency by
0.4 %–0.45 %.
The given simulation assumes that the IGCTs are always operated in soft-switching. In the
following chapter, an auxiliary resonant-commutated pole is introduced, which ensures that.
4.4 Auxiliary Resonant-Commutated Pole
The dual-active bridge loses its inherent soft-switching capability in certain operation regions.
The ZVS operating range can be determined through first harmonic approximation of the
transformer voltages and currents [42, 50]. Figure 4.6 illustrates that hard switching occurs at
light load and high dynamic voltage conversion ratios.
The auxiliary resonant-commutated pole (ARCP) has been mentioned first in 1989 as a circuit to ensure soft-switching operation in inverters [62–64]. Moreover, there are additional
advantages when it is used in a DAB3 [65].
31
120
120
100
100
losses in kW
losses in kW
Power-Electronic Switches and Soft-Switching Operation
80
60
40
20
0
1 2 3 4 5 6 7
transferred power in MW
(b) IGCT - Csn = 0 µF
120
120
Psw
Pcond
100
80
losses in kW
losses in kW
40
0
1 2 3 4 5 6 7
(a) IGBT
60
40
80
60
40
20
20
0
60
20
transferred power in MW
100
80
1 2 3 4 5 6 7
transferred power in MW
(c) IGCT - Csn = 1 µF
0
1 2 3 4 5 6 7
transferred power in MW
(d) IGCT - Csn = 2 µF
Figure 4.5: Semiconductor losses in a DAB3
To explain the operation principle of an ARCP, a single commutation from the switch Smain−
to Smain+ is considered (c.f. Fig. 4.7). Initially, the lower phase branch (Smain− ) is conducting
and the phase current ip1 is positive. The DAB3 now operates in the hard-switched operation,
since the upper switch Smain+ would turn on against the lower diode.
To achieve ZVS, the commutation process is initiated by triggering the switch Saux . A voltage
is applied to the inductance Laux leading to a linear current increase in the ARCP branch.
In sequence I, as indicated in Fig. 4.7 (b), the current iaux rises according to the inductance
value Laux . At the end of sequence I, the auxiliary current is equal to the load current ip1 .
Ultimately, the load current is completely carried by the ARCP. Additional to the auxiliary
current, a boost current iboost is injected in sequence II. It provides additional energy for the
resonant circuit to compensate ohmic losses. Turning off the main switch Smain− initiates
sequence III. According to the resonance between the auxiliary inductance and the snubber
32
Power-Electronic Switches and Soft-Switching Operation
Figure 4.6: Hard and soft switched operating areas
capacitors, the snubber capacitors reload. As soon as the anti-parallel diode of the switch
Smain+ becomes forward biased, Smain+ can turn on at zero voltage. The commutation is
completed when the auxiliary inductance is demagnetized at the end of sequence IV.
Figure 4.8 shows the integration of the ARCP in a DAB3. For the sake of clarity, only the
primary side of the DAB3 is depicted. Besides the connection of the lossless snubbers to the
main switches, an additional switch Saux and an inductance Laux are connected per phase leg.
It shall be noted that these components are considered as auxiliary devices. They are rated
for a small part of the total converter power. Different to ARCPs in inverter applications, in
a DAB3 the auxiliary current is only needed in some operation points when otherwise hard
switching would occur. Moreover, if an auxiliary current is needed, it is fairly low compared
to that in inverter applications.
Since the switch Saux is operated at ZCS, for Saux thyristors could be applied. However it has
been shown, that the RC-snubber which has to be connected in parallel to a thyristor, has a bad
impact on the system efficiency since it generates losses even if the ARCP is deactivated [65].
In Fig. 4.9 different configurations of Saux are compared. The black curve represents the losses
without using an ARCP. In this case also the lossless snubber is not applied since ZVS can
not be ensured. Applying the lossless snubber in hard-switching operation would increase
the losses and would lead to failure, ultimately. Figure 4.9 (a) shows the application of a
silicon thyristor. Besides the blocking capability of the thyristor, the reverse recovery time
of the thyristor has to be ensured since Saux is operated with the same frequency as the
main switches. In the simulation the silicon thyristor "Westcode R1127" is chosen. This
configuration effectively reduces the switching losses when the DAB3 would enter the hardswitching operation. However, it is evident that in the natural soft-switching operation range
the DAB3 without ARCP has lower losses. This is due to the mentioned RC-snubber losses
which are also present if the ARCP is not activated. Consequently, the choice whether to
implement an ARCP circuit strongly depends on the application the DAB3 is used in and the
33
Power-Electronic Switches and Soft-Switching Operation
(a) Schematic
(b) Characteristic waveforms during an exemplary commutation from Smain− to Smain+
Figure 4.7: Single phase leg of an Auxiliary Resonant-Commutated Pole (ARCP)
Figure 4.8: Integration of the ARCP in the DAB3
amount of time it operates in partial-load conditions leaving natural soft switching.
Using silicon carbide (SiC) thyristors, this issue can be overcome. Since the reverse recovery
effect is nearly not present using SiC, the RC-snubber is not necessary. The simulation has
been conducted with the commercially available "GeneSiC GA060TH65".
Due to the limited availability of SiC devices and the high price a third option is investigated,
which turns out as the most efficient solution. Using a reverse blocking IGCT as Saux (in this
case "Mitsubishi GCT GCU15CA-130"), the RC-snubber can be omitted and reverse recovery
losses are not present. However, additional sensors are needed to achieve active turn off at
zero current. If the GCT it not turned off actively at zero current, it behaves as a conventional
thyristor at turn-off and generates similar reverse recovery currents. Figure 4.9 (c) shows that
the overall switching energy is the lowest for this case. This is due to the lower conduction
losses of the GCT compared to the SiC thyristor.
During the investigation it turns out that the conduction losses of the ARCP branch are more
34
Power-Electronic Switches and Soft-Switching Operation
(a) Si thyristor
(b) SiC thyristor
(c) IGCT
Figure 4.9: Commutation energy for different configurations of Saux
critical than the switching losses. Higher conduction loss results in a higher boosting current
to compensate these losses. Consequently, the turn-off currents in the main switches increase
as well [65].
Table 4.2 gives an overview of the different switching devices and their evaluation for an
ARCP application. Consequently, the Si thyristor is superior considering availability and control issues. The IGCT that is turned off at zero current achieves great efficiency, however
it needs additional control effort. When SiC thyristors are available for higher current ratings, they might be the perfect switch for this application as they unite low losses and easy
controllability [65].
35
Power-Electronic Switches and Soft-Switching Operation
Table 4.2: Evaluation of different switching devices for an ARCP
Si thyristor
SiC thyristor
Si IGCT
X
XX
XX
availability
XX
×
X
control
XX
XX
X
losses
36
5 Medium-Frequency Transformer
The medium-frequency transformer designed for a high-power dual-active bridge is one of the
main challenges in this work. With the increasing operation frequency the total core losses
decrease, however the core loss density increases significantly. Moreover, in the considered
frequency range of around 1 kHz, only a few core materials are suitable.
In the following, different core materials and their performance in a high-power dc-dc converter
are evaluated. One of these materials is measured with the voltage waveforms of a dual-active
bridge. Finally, transformer design considerations in a DAB3 application are given.
5.1 Review on Windings and Core Materials
Nowadays, different transformers are designed for a huge variety of applications. Three key
objectives can be identified, that have a big impact on the design of the transformer: frequency,
voltage and current.
With increasing frequency the power loss density increases making the cooling more difficult.
Furthermore, skin and proximity effects increase the winding losses. With increasing voltage
the isolation effort is more complicated.
At voltages of 3 kV and above, partial discharge (PD) has to be considered [66]. Due to PD,
the isolation can be destroyed over the time. This aging effect of the isolation intensifies
with increasing frequency. Consequently, cast resin windings that are completely free of PD
should be considered in medium-voltage transformers, especially when operated at elevated
frequency.
The current rating mainly effects the design of the winding and the cross-sectional area of the
transformer wires. At increased frequency, skin and proximity effects increase the winding
resistance further. In addition, if the winding is casted, it is a major challenge to cool the
winding and liquid-cooled hollowed conductors might be a solution.
Summarizing, whenever voltage, current or frequency increase the design of a transformer
becomes more challenging. Transformers with high requirements for two of these objectives
are state of the art today: Figure 5.1 shows in which applications these transformers are
already used today. In a medium-voltage dc-dc converter, all three objectives have to be met.
To reduce the winding losses and with this the cooling effort, high-frequency litz wire should
be applied for the winding. Unfortunately, litz wires with a large diameter are commercially
37
Medium-Frequency Transformer
$elevated$
$voltage$
$elevated$
$frequency$
$elevated
$power$
$medical-use x-ray$
$HVAC transmission
Figure 5.1: Transformer requirements for different applications
$aeronautic$
$high-power$
not available, especially to construct a casted winding. (Such wires have a special mantle $dc-dc
to
converter$
perfectly bond with the resin.) Since these wire are producible, they will be available when
there is a market. By then, several wires with smaller diameter have to be parallelized or
non-isolated stranded wire might be an cost-effective alternative which also has an positive
effect on high-frequency issues [67].
The choice of the core material mainly depends on the fundamental frequency of the magnetic
flux and the power level. A trade-off between core-loss density, nominal flux density (determine
the core volume) and cost has to be found.
In medium-voltage high-power applications, the switching frequency and with it the frequency
of the magnetic flux is limited to about 1 kHz–2 kHz by the power electronic switches. Because of that, silicon steel and amorphous iron are the common materials in medium-voltage
applications.
Silicon steel is, on a quantity basis, the most commonly used core material. It is commercially
used in transformers of all power rating up the giga-watt range. The relatively large saturation
flux density (around 2 T) promises a compact core design. Silicon steel is used in 50 Hz
applications as well as in 400 Hz aircraft applications. The production is well-known and the
price is comparatively low. ThyssenKrupp Electrical Steel (TKES) provides sheets down to
0.18 mm. The German company Waasner offers silicon steel laminations with a thickness of
0.1 mm. This 0.1 mm material however is very expensive and only available as tape-wound
core. Besides traction applications at 400 Hz [68], the core material has also been proven up
to 1 kHz [69].
The saturation flux density of amorphous iron is lower compared to silicon steel. However,
the low hysteresis losses result in lower no-load losses. This might compensate the higher
purchase price. Due to the manufacturing process, the material can be produced as very thin
38
Medium-Frequency Transformer
ribbons, giving better performance at high frequencies. The smallest thickness of Hitachi’s
Metglas 2605SA1 is given with 920 nm. However, the high magnetostriction is a drawback of
the material, especially at high power levels [70].
If future power electronics allow higher operation frequencies (> 4 kHz), the use of nanocrystalline cores might be interesting. At lower frequency, the material might not be cost-effective.
Compared to amorphous iron, nanocrystalline material has no problem with magnetostriction
[70].
5.2 Core Losses in a Dual-Active Bridge Application
The waveform of the flux inside the core, corresponds to the integral of the phase voltage.
Consequently, the flux waveform in a DAB3 application is a piece-wise linear one. To investigate the impact of this piece-wise linear flux waveform on the core losses, measurements are
conducted [71].
The measured core material is the silicon steel from TKES called "PowerCore H". The sheets
with a thickness of 180 µm are measured using an Epstein frame. The test bench allows
exciting a magnetic material with an arbitrary flux waveform. Hence, the core losses applying
an sinusoidal voltage can be compared with the DAB3 application.
Exemplarily, Fig. 5.2 shows the measuring at a peak flux density of B̂ = 1 T and a frequency
of f = 1 kHz. The voltage applied to the material is depicted in Fig. 5.2 (a). The red solid line
corresponds to the DAB3 application, while the sinusoidal reference measurement is indicated
as dashed blue line. In Fig. 5.2 (b) the piece-wise linear flux waveform is shown and (c) depicts
the corresponding BH-hysteresis loop. From the hysteresis loop one can see that the specific
core losses for the DAB3 application are actually smaller than for the sinusoidal case since
the spanned area is smaller. Figure 5.2 (d) confirms this as it shows the specific core losses
for the sinusoidal and the DAB3 measurement.
In the next step it is investigated how well the improved Generalized Steinmetz Equation
(iGSE) is able to model the core losses considering the piece-wise linear flux waveform [72].
First the measurements under sinusoidal excitation are used to determine the Steinmetz parameters according to the original Steinmetz Equation (OSE)
Ps = kf α B̂ β
with
[Ps ] = W/kg.
(5.1)
Figure 5.3 (a) shows the measured data points. Based on the lines of best fit, the Steinmetz
parameters
α = 1.6155
β = 1.7021
k = 5.2 · 10−4
(5.2)
39
Medium-Frequency Transformer
1
m agn. flux density in T
phase voltage in V
Sinusoidal
DAB3
20
10
0
−10
0.5
0
−0.5
−1
−20
0
0.2
0.4
0.6
time in ms
0.8
0
1
1
(b) Magnetic flux density
2
1
W
kg
10
specific core losses in
m agn. flux density in T
(a) Transformer voltage
0.5
tim e in m s
0.5
0
−0.5
−1
−100
−50
0
50
A
m agn. field in m
1
10
0
10
100
(c) Magnetic flux density
−1
0
10
10
magn. peak flux density in T
(d) Core losses for sinusoidal and piece-wise linear waveform at f = 1 kHz
Figure 5.2: Measuring results
are determined. In the following, these parameters are used to apply the iGSE.
Equations (5.3) and (5.4) represent the well-known iGSE as published in [72].
Ps =
ki (∆B)β−α
T
with
ki =
Z
0
T
dB α
dt dt
k
(2π)α−1
R 2π
0
| cos θ|α 2β−α dθ
(5.3)
,
(5.4)
where ∆B is the peak-to-peak value of the flux density, T the period time of the flux density
and α, β and k being the Steinmetz parameters.
40
Medium-Frequency Transformer
10 kHz meas.
10 kHz OSE
7 kHz meas.
7 kHz OSE
5 kHz meas.
5 kHz OSE
1 kHz meas.
1 kHz OSE
3
2
10
DAB3 meas.
iGSE
sinus meas.
OSE
W
kg
2
10
specific core losses in
specific core losses in
W
kg
10
1
10
1
10
0
10
0
10
−1
0
10
10
magn. peak flux density in T
(a) Sinusoidal measurement
−1
0
10
10
magn. peak flux density in T
(b) Validation of the iGSE
Figure 5.3: Steinmetz parameter extraction and validation of the iGSE
As also published in [72], for piece-wise flux densities, (5.3) simplifies to:
ki (∆B)β−α X Bm+1 − Bm α
Ps =
tm+1 − tm (tm+1 − tm ) ,
T
m
(5.5)
where (tm , Bm ) are the supporting points of the piece-wise linear flux.
Figure 5.3 (b) demonstrates the comparison between the measured losses and the losses calculated by (5.5) at a frequency of 1 kHz.
While the results of the OSE for the sinusoidal excitation are apparently in good accordance
with the measuring, the iGSE slightly overestimates the losses of the piece-wise linear flux
waveform. As discussed in [71], this error is related to the variation of the permeability of
silicon steel at the evaluated frequencies [73]. Additional, eddy currents occur at the given
frequencies, which are not considered by the Steinmetz formulas [74].
However, as the relative error is sufficiently small, the iGSE delivers accurate core losses in a
DAB3 application.
41
Medium-Frequency Transformer
5.3 Design Aspects
As state before, the DAB3 requires a certain series inductance to ensure soft switching and
controllability. Especially for low-power applications, it is common practice to solely use the
stray inductance of the transformer as series inductance. Consequently, the stray inductance
of the transformer has to be designed for a specific value. This is in contrast to general design
approaches that usually minimize the stray inductance.
The stray inductance is determined by the magnetic field that is generated by one winding of
the transformer and is not linked to the other. For a simple coaxial winding configuration the
stray inductance can be calculated analytically to get an idea of the parameters influencing
the stray inductance.
For this calculation some assumptions have to be made. Firstly, the magnetic field inside the
core material is assumed zero. Secondly, the magnetic field in the winding window is assumed
perfectly homogeneous. The stray inductance corresponds to the magnetic energy stored in
the winding window Ŵm when the secondary is shorted. The magnetic field in the winding
window is depicted in Fig. 5.4 for a two layer and a three-layer configuration. The insulant is
colored gray.
The magnetic energy for the two-layer configuration is
Z x8
1
Ĥ · B̂ dV = µ0
Ĥ 2 (x)(lm hw dx)
2
V
0
"Z Z x3 2
x2
1 x − x1 2
µ0 hw lm 2
1
Ĥmax
dx
=
dx +
2
2
x
−
x
2
2
1
x1
x2
!
Z x5
Z x6
Z x4 2 1 x − x3 2
1
dx +
1 · dx +
+
+
2
2 x4 − x3
x4
x5
x3
2 #
Z x7 2
Z x8 1
1 x8 − x
+
dx +
dx
2
2 x8 − x7
x6
x7
2 a1
a1 µ0 hw lm 2
Ĥmax 2 ·
+ b + a1 +
+ (b + c)
=
2
2
3
3
µ0 hw lm 2
5
=
Ĥmax a1 + 3b + 2c
4
3
2
µ0 lm N1 2 5
=
î
a1 + 3b + 2c
4hw
3
! 1
= Lσ î2
2
Ŵm =
1
2
Z 2 !
1
1 x6 − x 2
dx
+
2
2 x6 − x5
(5.6)
where lm is the mean winding length and N1 the number of primary-side turns. Consequently,
this leads to
42
µ0 lm N12 5
Lσ =
a1 + 3b + 2c .
2hw
3
(5.7)
Medium-Frequency Transformer
hw
H/Hmax
H/Hmax
a1 b a1 b+c a1 b a1
a1 b a1 b a1 b+c a1 b a1 b a1
1
1
1/2
2/3
1/3
x1 x2 x3 x4
x5 x6 x7 x8
x
(a) 2-layer configuration
x1 x2 x3 x4 x5 x6
x7 x8 x9 x10 x11 x12
x
(b) 3-layer configuration
Figure 5.4: Magnetic field in the winding window
Analogue, the stray inductance for the three-layer configuration, according to Fig. 5.4 (b) is
µ0 lm N12 14a1 + 19b
Lσ =
+c .
hw
9
(5.8)
This shows that the stray inductance is most sensitive to the distance between the primary
and secondary winding in a coaxial configuration.
Based on (5.7) and (5.8) the transformer geometry is designed in a way that the stray inductance is Lσ = 100 µH. To verify the designs and hence the analytical formulas and to quantify
the relative error, a finite element method (FEM) simulation is carried out.
Figure 5.5 exemplarily shows the simulation of the first out of three phases. To determine
the stray inductance, the transformer’s secondary winding of the corresponding phase is short
circuited. Consequently, a magnetic field is generated between the primary and secondary
winding. From the energy stored in this field, the corresponding stray inductance is calculated.
Table 5.1 shows the results for a two-layer and a three-layer configuration. Firstly, it can
be concluded that the simulation results differ up to 30 % from the design target, which was
43
Medium-Frequency Transformer
Figure 5.5: Simulating the stray inductance in a 2-D FEM simulation
100 µH. The analytical formula is not suitable for a precise design of the stray inductance,
since some simplifications are made. Secondly, it is evident that there is an asymmetry of the
stray inductances. This is related to the use of a three-legged core. As can be seen in Fig. 5.5,
less energy is stored in the outer region of the core. Consequently, the outer phases have a
slightly lower stray inductance. This encourages the need for the balancing control introduced
in Section 3.4.
0.18 mm Silicon Steel: B̂max = 1.7 T, Jrms = 10.85 MA/m2 , µr = 11039
Layers Windings
Phase 1
Phase 2
Phase 3
Lσ,i
2
74
126.6 µH
127.5 µH
126.6 µH
Lσ,i
3
21
104.4 µH
112.1 µH
104.5 µH
Table 5.1: Simulation results: verification of the analytical designs
In a next step, a different winding configuration from the coaxial one is analyzed. Now, the
primary and secondary are stacked on top of each other as depicted in Fig. 5.6. Figure 5.7
shows the simulation results. In (a) the stray inductance Lσ2 of the second phase is located
on the centered core. The stray inductance Lσ1 of the left core is simulated in (b). Three
conclusion can be drawn from the results. Firstly, the stray inductance is much higher. In the
stacked configuration, the stray inductance is by a factor of 27 higher compared to the previous
constellation due to the bigger volume of the stray field. Secondly, the stray inductance suffers
from a higher asymmetry between the three phases. While the stray inductance of the outer
phases is Lσ1 = Lσ3 = 1.03 mH, the stray inductance of the centered phase is Lσ2 = 1.48 mH.
Finally, when the magnetic field generated by one phase returns, it penetrates the windings of
the neighboring phase. Consequently, higher losses due to the proximity effect are expected
in such a winding configuration.
44
Medium-Frequency Transformer
Figure 5.6: Alternative stacked winding configuration
(a) center leg
(b) outer leg
Figure 5.7: Simulating the stray inductance for a stacked winding configuration
45
Medium-Frequency Transformer
46
6 Demonstrator
The following chapter is about the design and the construction of the 5 MW medium-voltage
dc-dc converter. At first, the implementation of the control, including the instantaneous
current control and the balancing control, is introduced. Then the design and the construction
of the demonstrator is described. The chapter finishes with the commissioning and measuring
results.
6.1 Control Implementation
The control of a DAB3 as discussed in Chapter 3 is implemented on a control hardware
provided by ABB.
The unit with the model number "PC D247" is depicted in Fig. 6.1. Figure 6.2 shows the
system overview. The control unit features a PowerPC intended for tasks with a cycle time
in the range of 100 to 1000 µs. The build-in Xilinx Spartan3 FPGA performs quicker tasks at
time frames down to 25 ns. The PowerPC and the FPGA exchange data through a dual-ported
RAM (DPRAM).
Considering the DAB3 application, the voltage and current control is implemented on the
Power PC. The FPGA, however, performs the gate driving of the IGCTs and the implementation of the dead-time. Moreover the load angle regulation according the ICC is performed
by the FPGA. Consequently, a clear interface between Power PC and FPGA is achieved.
FPGA Implementation
Primarily, the FPGA provides a framework which ensures the communication between several
components inside the PC D247. This framework appears to the user as black box and can not
be modified. Within this black box, however, an application block is provided which allows
the implementation of custom-made routines.
Figure 6.3 shows the designed application block to control a DAB3 converter. The interface
to the PowerPC and the optical outputs are shown as well.
The "input" side interfaces to the PowerPC using the following signals.
EINSCHALTER_PORT: At a transition from low to high, the FPGA generates the switching
signals for the IGCTs. If at the same time a valid load angle is assigned, the starting
47
Demonstrator
Figure 6.1: PC D247 control hardware
sequence is according to the ICC. As soon as the signal turns to low level, the load angle
is set to 0◦ via the ICC and the switching signals are turned off afterwards.
NOTAUS_PORT: Gating signals are turned off immediately when this signal turns zero. To
restart the system, the error has to be acknowledged first by turning off EINSCHALTER_PORT.
CLK_PORT: A clock signal with 40 MHz is expected at this input port. It is provided by the
PC D247 internally.
TOTZT_PORT: This port gives the dead time tdead between to switches of one phase leg.
DRTLPERI_PORT: The switching frequency is determined by this signal. To apply the change,
EINSCHALTER_PORT has to be zero. The switching frequency can be set freely between
868 Hz and 27.77 kHz.
PHI_PORT: This signal gives the load angle ϕ. The load angle is limited to the range of −60◦
and 60◦ to comply with the ICC. If the ICC is turned off, load angles up to ±90◦ are
valid.
PHIiP_PORT with i∈ 1, 2, 3: These ports give the compensation angles ∆ϕi for the balancing
control. The load angle for the corresponding phase is
ϕi = ϕ + ∆ϕi
with
−90◦ ≤ ∆ϕ ≤ 90◦ .
(6.1)
The changes of the compensation angles are not performed according to the ICC.
VERBINDE_P_PORT: If this is on high level, the compensation angle of the first phase (PHI1P_PORT)
is also set for the second and third phase. PHI2P_PORT and PHI3P_PORT are ignored. This
48
Demonstrator
5V
X1
Primary Power Supply
FIG Module (optional)
3V3
FIG Interface
X2
Redundant Power Supply
PSUP
OM1 – OM6
PowerLink
Receiver
FPGA
CPLD
SPI1
X68
OMI
X65
MODBUS Interface
Transmitter
PowerPCTM
OM7 – OM22
Optical Application I/O
Receiver
CPLD
CPLD
PPI
Panel
5V
Transmitter
X101
Service Interface
24V
X700
Analog Outputs
X66
DAC
DAC
X61
X300
Digital Outputs
Ethernet Port 1
CPLD
X62
X400
Digital Inputs
SPI2
Ethernet Port 2
Ethernet
Switch
X63
Ethernet Port 3
X100
External Power Supply
PSUP
X63
Ethernet Port 3
X800 – X803
High Current Inputs
FADC
X804
Low Current Inputs
FADC
X900 - X901
HVD Inputs
JTAG
X102
JTAG
Figure 6.2: PC D247 system overview, Source:[75]
way it is possible to operate the DAB3 without ICC, either for comparative measuring
or to enhance the operating range to ±90◦ .
At the "output" side, the FPGA is connected to the optical modules as indicated in Fig. 6.3.
The optical modules transfer the physical switching signals to the IGCTs. Moreover three
feed-back signals are provided.
EINGSFEHLER_PORT: This signal indicates if the load angle or compensation angles at the input
are out of range.
BEGRENZUNG_PORT: When the sum of the load angle and the compensation angles is out of
range this output is high.
TRGR_PORT: This output gives a triggering signal for an oscilloscope. With it, for example the
switching signals during a load step can be verified.
Figure 6.4 shows the structure of the application block as implemented for the high-power
dc-dc converter. [76] describes in detail the implementation, which comprises 4000 lines of
49
Demonstrator
DPRAM
Output-Port 1
FPGA
Optische
Module
DAB_3ph_sst
Adresse 0, Bit 15
Adresse 0, Bit 14
Adresse 0, Bit 13
Adresse 1, Bit 10 bis 0
Adresse 2, Bit 15 bis 0
Adresse 5, Bit 15 bis 0
Adresse 6, Bit 15 bis 0
Adresse 7, Bit 15 bis 0
Adresse 8, Bit 15 bis 0
DAB_write_reg1(0)(15)
DAB_write_reg1(0)(14)
i_clk_40
DAB_write_reg1(0)(13)
DAB_write_reg1(1)(10 - 0)
DAB_write_reg1(2)
DAB_write_reg1(5)
DAB_write_reg1(6)
DAB_write_reg1(7)
DAB_write_reg1(8)
EINSCHALTER_PORT
NOTAUS_PORT
CLK_PORT
VERBINDE_P_PORT
TOTZT_PORT
HB1S_O_PORT
HB1S_U_PORT
HB2S_O_PORT
HB2S_U_PORT
HB3S_O_PORT
HB3S_U_PORT
DRTLPERI_PORT
HB1P_O_PORT
HB1P_U_PORT
PHI_PORT
HB2P_O_PORT
HB2P_U_PORT
PHI1P_PORT
HB3P_O_PORT
HB3P_U_PORT
PHI2P_PORT
PHI3P_PORT
EINGSFEHLER_PORT
BEGRENZUNG_PORT
TRGR_PORT
o_ppi_txd(13)(1)
o_ppi_txd(13)(2)
o_ppi_txd(14)(1)
o_ppi_txd(14)(2)
o_ppi_txd(15)(1)
o_ppi_txd(15)(2)
o_ppi_txd(9)(1)
o_ppi_txd(9)(2)
o_ppi_txd(10)(1)
o_ppi_txd(10)(2)
o_ppi_txd(10)(1)
o_ppi_txd(10)(2)
o_ppi_txd(14)(3)
o_ppi_txd(15)(3)
o_ppi_txd(9)(3)
OM19, V2
OM19, V4
OM20, V2
OM20, V4
OM21, V2
OM21, V4
OM15, V2
OM15, V4
OM16, V2
OM16, V4
OM17, V2
OM17, V4
OM21, V6
OM20, V6
OM15, V6
Figure 6.3: Application-block interface
VHDL code.
PowerPC Implementation
As stated, one of the main tasks of the PowerPC is the power or voltage control. However,
also the over-voltage/over-current detection, fault response and the human-machine interface
(HMI) are performed by the application.
The application which is depicted in Fig. 6.5 is implemented in Matlab/Simulink. The Matlab
toolbox "Real-Time Workshop" translates the application to C++ and transfers it to the
PC D247, where it is executed by the PowerPC.
Figure 6.5 shows the visualization of the application in Simulink. Besides some auxiliary
system programs, three interrupt service routines (ISR) and the operating panel can be identified. Each of the ISRs is performed with a different cycle time, from 250 µs to 5 ms. The ISR
with the lowest cycle time reads the analogue and digital inputs and checks if the limits are
kept. The system reacts on over current, over voltage and external emergency stop. Moreover,
pressure, temperature and conductivity of the cooling fluid are observed.
The second fastest ISR is executed every 1 ms corresponding to the switching frequency. The
routine contains e.g. the constant voltage controller and is responsible to set the load angle.
Using the ICC the cycle time of this routine can be decreased further to one third or one half
of the switching period.
50
Demonstrator
The slowest ISR reads and converts the data of the water cooling, since only low dynamics
are expected from these values.
Finally, the operating panel serves as HMI. It allows to control the converter and to read
sensor data.
51
Demonstrator
lastwinkel
start
umrechnung
totzeit
ausgabe
schaltzeitpunkt
ausgleich
winkel_s
begrenzung
vergleicher
zaehler
zaehler
schaltzeitpunkt
ausgleich
winkel_p
stop
schaltfrequenz
Figure 6.4: Overview of the modules in the FPGA
INT A
PEC
Interrupt
Sources
INT B
INT C
TsA
TsB
TsB = 1 ms
TsA = 250 us
TsC
Power Fail INT
Trigger()
01 Interrupt Control
Trigger()
A_Pa
{68}
[A_Pa]
[A_B]
[Pa_A]
{4}
Db_buf {4}
A_B
fromA
A_C
{13}
4{4}
{13}
Db_buf {13}
{4}
Db_buf {4}
fromA
B_Pa
fromPa
B_C
[B_Pa]
[A_B]
[Pa_B]
[A_C]
100 ISR A implementation
02 Start up configuration
Note: Required interrupts must
be enabled in
the interrupt control block !
{10}
200 ISR B implementation
TsC = 5 ms
Trigger()
[A_C]
4{4}
Db_buf 4{4}
[A_Pa]
C_Pa
fromA
4{4}
{68}
{68} {68}
{10}
{10}
{4}
[Pa_A]
[C_Pa]
[B_Pa]
{10}
300 ISR C implementation
[C_Pa]
4{4}
4{4}
{4}
{4}
4{4}
Trigger
400 Power Fail
Figure 6.5: Overview of the PowerPC Software
52
500 Operating Panel
[Pa_B]
Demonstrator
6.2 Converter Design and Construction
A demonstrator for a medium-voltage high-power dc-dc converter is constructed. The target
power rating of the converter is 5 MW. The nominal input voltage is Up = 5 kV. To prove
the concept and to ease the commissioning, the nominal output voltage is Us = 5 kV as well.
Consequently, the input and output dc link can be connected together to circulate the energy.
Therefore, an expensive high-power load is unnecessary and the dc power supply only needs to
compensate the losses. The power-electronic switches are "5SHY 3545L0001" IGCTs kindly
provided by ABB Switzerland. The IGCTs are applied with anti-parallel diodes from Infineon
("D1031SH45TS02"). Both, diodes and IGCTs, are rated for 2.8 kV. Consequently, two
devices per inverter arm have to be connected in series for the rated voltage of the converter.
The dynamic voltage balancing across the devices is ensured by the lossless turn-off snubbers.
The IGCTs are operated with 1000 Hz.
Firstly, the converter is operated with a single-phase transformer as single-phase DAB. Afterwards, the series connection of two IGCTs and the voltage balancing are implemented.
Finally, the converter is operated with two additional transformers as three-phase DAB. Figure 6.6 shows a picture of the demonstrator in an earlier construction phase. Six water-cooled
power-electronic building blocks (PEBB) contain the IGCT and diode stacks. The pump and
filter for the deionized water is visible at the right edge of the picture. In the outer cabinets,
the dc link capacitors with a total capacitance of Cp = Cs = 1 mF are located.
Clamping Circuit
Since the converter might enter the hard-switching operation area, a
di
dt
snubber is needed to
prevent the anti-parallel diodes from excessive reverse recovery currents. The spice simulation
depicted in Fig. 6.7 is used to design the clamping circuit.
Figure 6.6: Demonstrator in an early construction phase
53
Demonstrator
.ic V(C1) = 2500
R2
L2
2e-3
100e-6
.MODEL IGCT SW(Roff=1e4 Ron=10e-4 Voff=0.0V Von=1.0V)
L_I1
6e-6
R1
2e-3
D3
L3
D
L_CL
D1
L_S
R_S
.5e-6
1
0.3e-6
PWL(0 1 2e-6 1 7e-6 0 30e-6 0)
IGCT
D
SW2
V2
C_CL
D2
.5e-6
5e-6
C1
L1 .ic I(L1) = 1000
D
V1
1e-3
10e-3
2500
.tran 0 150e-6 0 0.1e-6
(a) Model
4000
IGCT voltage in V
Rc = 1 Ω
Rc = 0.15 Ω
3500
3000
2500
2000
0
20
40
60
80
time in µs
(b) Simulation result
Figure 6.7: Spice model to design the clamping circuit
The optimal value for the damping resistor is around 1 Ω. However, due to availability reasons
a lower resistance value has to be used. This leads to oscillations and poor damping which is
not harmful to the devices though. As evident from Fig. 6.7 the voltage stays below 4500 V
which is the repetitive peak off-state voltage of the IGCTs.
The parameters of this clamping circuit are: Lc = 5 µH, Cc = 5 µF and Rc = 0.15 Ω.
DC Link Capacitance
According to simulation, a capacitance of Cp = Cs = 1000 µF is sufficient in the considered
operating range. With it, the resulting voltage ripple in simulation is below ±1 %. Since a
neutral point is needed for the application of an ARCP or to investigate the use of a threelevel neutral point clamped (NPC) inverter, the dc link is split symmetrically. Consequently,
two capacitors, with 2000 uF each, are connected in series. The specifications of the applied
54
Demonstrator
capacitors MUECAP DKTFM 3K302277 are:
capacitance : 2270 µF
nominal input voltage : 3.3 kV
surge voltage : 4.95 kV
input resistance : 0.8 mΩ
input inductance : 245 nH
max. rms current : 255 A
Auxiliary Power Supply
The supply voltage of the IGCTs has to be galvanically isolated. This is achieved through
power electronic transformer which is fed by 24 V dc. A severe amount of power is needed for
the IGCT turn-off. The turn-off power can be calculated to
(6.2)
Poff = UGDU · QGQ (ITGQ , Tvj ) · fs .
As also depicted by Fig. 6.8, the power consumption of an IGCT rises significantly with the
operating frequency. The extrapolation of the given datasheet values for an operation at
ITGQ = 1 kA and fs = 1 kHz leads to PGin = 130 W. Considering safety margin and losses
in the isolation transformer, the auxiliary supply for a PEBB including four IGCT devices is
designed for 600 W. To energize and turn-off the converter safely in case of a power black out,
the auxiliary supply is supported with an uninterruptible power supply (UPS).
Medium-Frequency Transformer
The series inductance in the ac link has a major influence on the operation of the converter. It
determines the load-angle range, rms current in the ac link, dc current ripple, turn-off current
of the IGCTs, the soft-switching boundary at low load and the
di
dt
during the diode turn-off
and IGCT turn on.
An obvious optimization criterion is to minimize the rms current in the transformer and
consequently the apparent-power rating of the transformer [57].
Figure 6.9 show the rms current of the transformer in dependence of the series inductance at
a transferred power of 5 MW. If the dynamic voltage ratio d is large, the stray inductance is
desired to limit the current. At a operation with input and output voltage being equal (d = 1),
only a low inductance is needed in the ac link, since the current does not change when both
bridges are in the same switching state. Then a high inductance in the ac link results in a
higher reactive power. The intersection of both graphs gives the optimal inductance leading
to the lowest rms current to cover a given operation area.
For the given converter, the optimal inductance is 185 µH resulting in a minimum rms current
of 855 A.
55
Demonstrator
Figure 6.8: Power consumption of one IGCT, Source: [77]
There are two different design philosophies. Firstly, the series inductance can be implemented
as two separate inductances connected on the primary and secondary side of the transformer.
Secondly, as also common practice in low power applications, the stay inductance of the
transformer can be used as built-in filter element.
The first approach is easier to design. It might be preferred if the design of the transformer itself is already challenging due to thermal management or isolation issues. Moreover, designing
separate inductances, the inductance value can be set more accurately.
Implementing the series inductance into the transformer probably leads to lower overall copper
losses. In addition, the volume is lower compared to a transformer with additional inductors.
However, the design of the transformer is more complex and the stray inductance of the
transformer is very sensitive to the geometry and the environment. If tolerances in the stray
inductance have to be considered, the diagram in Fig. 6.9 shows that the transformer has to
be designed for higher currents.
Since the design of the transformer is already very complex due to the high power rating,
the issues of PD-free insulation and the increased stray inductance, firstly a single-phase
transformer is used for the commissioning of the high-power dc-dc converter.
In the scope of this project, a transformer for a rated power of 2.2 MVA at a frequency of
1000 Hz has been acquired. The transformer with a basic insulation level (BIL) of 12 kV
56
Demonstrator
transformer current in A
1050
Up = 4.5 kV, Us = 4.5 kV
Up = 4.5 kV, Us = 5.5 kV
1000
950
900
850
800
100
150
200
250
300
350
stray inductance in µH
Figure 6.9: Transformer current versus stray inductance
has a core made of 0.18 mm silicon steel sheets. The outer dimension of the transformer are
0.72 m × 0.65 m × 0.48 m and it weighs 607 kg. Consequently, it features a roughly 10-times
higher power density compared to a conventional 50 Hz transformer.
Auxiliary Resonant-Commutated Pole
An ARCP circuit as described in Chapter 4 has been designed and constructed for the application in the high-power medium-voltage dc-dc converter. From the discussed options, the
Si thyristor solution has been chosen for implementation since it offers fairly low losses, high
availability and low control effort. The thyristors have to offer low reverse recovery times as
they are operated at 1000 Hz switching frequency. The thyristor "R1127" from IXYS Westcode has been chosen. The reverse recovery effect is reduced applying "DD600S65K1" from
Infineon as series-connected diode. The auxiliary inductance is 8.3 µH while each main switch
is equipped with a 1.36 µF snubber capacitor.
Figure 6.10 shows pictures from the commissioning of the ARCP in the lab and in the dc-dc
converter itself. Note that the auxiliary devices are small compared to the main devices of
the dc-dc converter. The power ratings of the auxiliary devices are comparatively low. The
auxiliary inductor for example is rated for an rms current of 200 A.
In order to ensure safe operation, a measuring circuit is implemented together with the ARCP
that monitors the voltage across the IGCT devices. This circuit shall detect if the GCT is
under zero-voltage conditions to prevent it from short circuiting the snubber if that is still
charged. As a positive side effect the same circuit prevents an IGCT to fire on a short circuit,
since the voltage across the GCT will not reach 0 V.
The circuit originally proposed in [78] and explained applied to a DAB3 in [65] is connected
in parallel to the GCT. As soon as the voltage reaches 0 V an optical signal indicates that
57
Demonstrator
(a) Comissioning in the lab
(b) Testing in the high-power converter
Figure 6.10: ARCP prototype
+
(a) Schematic
(b) Implementation
Figure 6.11: ZV detection circuit
the zero-voltage condition is met. Figure 6.12 shows the verification where "ZV" is the logical
signal indicating the zero-voltage condition.
58
Demonstrator
5
vC /200V
ZV signal
4
3
2
1
0
−1
0
10
20
30
time in µs
Figure 6.12: Verification of the ZV detection circuit
6.3 Commissioning
Setup A
In the first phase of the commissioning, only one half-bridge leg on the primary and secondary
side of the converter is used. As depicted in Fig. 6.13, the primary and secondary are fed independently from a dc-power supply. While the resistances on the primary serve to symmetrize
the dc link, the secondary-side resistors are used as load.
In the first test the focus is on the quasi zero-voltage switching of the GCTs. It should be
clarified that the GCTs are able to turn on properly under zero-voltage or rather zero-current
conditions. Figure 6.14 shows the device voltage and current of GCT2 (according Fig. 6.13)
and the primary transformer current. The measurements do not show any abnormality considering the switching of the GCT.
Figure 6.13: Schematic Setup A
59
Demonstrator
40
20
current in A
current in A
ip1
iGCT1
0
−20
0
50
20
0
−20
−50
100
time in µs
300
300
200
200
uGCT2 in V
uGCT2 in V
−40
−50
ip1
iGCT1
100
0
−100
−50
0
50
100
time in µs
0
50
100
50
100
time in µs
100
0
−100
−50
(a) GCT2 turn on
0
time in µs
(b) GCT2 turn off
Figure 6.14: Measuring Setup A
Setup B
As next stage of the commissioning the dc links are connected together as shown in Fig. 6.15.
Consequently, the energy is fed in a loop and only the losses have to be compensated. This
makes an expensive high-power load obsolete. Moreover, the power transferred by the converter can be increased since it is not any more limited by the dc supply. However, it has to
be tested, if the two bridges effect each other.
While setup A has used an smaller transformer, the setup B includes the 2.2 MVA transformer.
With this measuring the transformer parameters and its behavior in a DAB application can
is analyzed.
Firstly, the measuring shows overshooting in the voltage, when the GCT turns off. This is due
to the clamping circuit and is in accordance to the spice simulation described in Section 6.2.
The absolute value of the overshoot is independent from the dc-link voltage. Accordingly, the
relative overshoot will be less at increased voltage levels.
Secondly, the transformer parameters, especially the stray inductance, can be verified through
this measuring. According to the voltage ∆U across the transformer and the slope of the
60
Demonstrator
Figure 6.15: Schematic Setup B
150
100
50
0
−50
−100
ip1
is1
100
transformer current in A
transformer voltage in V
up1
us1
50
0
−50
−100
0
0.5
1
−150
1.5
0
0.5
1
1.5
time in ms
time in ms
(a) Transformer voltages
(b) Transformer currents
Figure 6.16: Measuring Setup B
current the stray inductance calculates to
Lσ = ∆U
dip1
dt
−1
≈ 45 µH.
(6.3)
This is unfortunately much less than what was specified.
It turns out that designing the transformer with an increased stray inductance is extremely
difficult, especially in medium-voltage application when the windings are casted. Moreover,
we experienced that the stray field gets affected by the environment. After the transformer
manufacturer has mounted the cooling equipment in the transformer housing, the stray inductance decreased. This experience might motivate the use of separate inductances in the
ac link until the accurate design of the transformers stray inductance in a medium-voltage
application is manageable.
61
Demonstrator
Figure 6.17: Schematic Setup C
Setup C
To increase the power level further, two additional PEBBs are installed to operate the bridges
in an H-bridge configuration. Thus, also the high compensating currents through the connection of the mid points is obsolete. Initially, this should ensure that the dc-link is perfectly
symmetrical since otherwise the transformer might saturate.
Figure 6.17 shows the configuration with a H-bridge inverter on the primary and secondary
side. The two PEBBs of each bridge share one
di
dt
snubber. Contrary to the figure, each phase
leg applies a separate clamping diode and capacitor. Since the stray inductance between the
capacitor Cc and the IGCTs is crucial concerning over voltage, the diodes and capacitors for
each phase leg are integrated individually in each PEBB.
Two measurements at different dc-link voltages are shown in Fig. 6.18 and Fig. 6.19. At
increased voltage the stray inductance calculated from the voltage and current waveforms is
a few percent lower compared to the measuring before. However, this slight variation might
also result from sensor inaccuracy. When both bridge are in the same switching state, the
transformer current decreases due to the winding resistance.
Figure 6.19 shows the a measurement at a dc link voltage of 1000 V. For comparison, the
waveforms for 450 V are depicted in Fig. 6.18.
62
Demonstrator
1000
400
ip1
is1
transformer current in A
transformer voltage in V
up1
us1
500
0
−500
−1000
0
0.5
1
200
0
−200
−400
1.5
0.5
0
1
1.5
time in ms
time in ms
(a) Transformer voltages
(b) Transformer currents
Figure 6.18: Measuring Setup C, Up = 450 V, ϕ = 18◦
1500
300
up1
us1
500
0
−500
−1000
−1500
−0.5
ip1
is1
200
transformer current in A
transformer voltage in V
1000
100
0
−100
−200
0
0.5
time in ms
(a) Transformer voltages
1
−300
−0.5
0
0.5
1
time in ms
(b) Transformer currents
Figure 6.19: Measuring Setup C, Up = 1000 V, ϕ = 12◦
63
Demonstrator
6.4 Derating due to Single-Phase Setup
Although all components are rated for a power of 5 MW, there are reasons why the converter is
not able to achieve the rated power with the actual (single-phase) configuration. The reasons
are either related to the single-phase setup itself or due to the low stray inductance.
Transformer Saturation
The saturation of the transformer is related to the integral of the phase voltage, or rather the
volt seconds applied to the main inductance. Figure 6.20 shows the voltage waveforms applied
to a transformer in a single-phase and a three-phase DAB converter. Assuming the magnetic
material is utilized equally leads to:
Z
up1,DAB1 (t) dt Z
=
max
up1,DAB3 (t) dt (6.4)
max
Ts
4 Ts
· Up,DAB1 = ·
· Up,DAB3
2
3 6
4
Up,DAB1 = · Up,DAB3
9
(6.5)
(6.6)
Consequently, the dc-link voltage has to be reduced to 44 % since the converter is not operated
in a three-phase configuration.
Figure 6.20: Transformer voltage for a single-phase and three-phase DAB
64
Demonstrator
Current Ripple and DC Capacitor Stress
The DAB3 requires smaller dc-link capacitors compared to the DAB1 to achieve the same
voltage ripple. However, in the experiments the rms current flowing into the dc-link capacitor
has been the limiting factor. Even at reduced power levels, the dc link capacitor currents have
been elevated.
Firstly, the DAB1 in general shows a higher current ripple than the DAB3. Consequently,
the capacitor current in DAB1 IC,DAB1 is higher than the rms capacitor current in a DAB3
IC,DAB3 . Figure 6.21 depicts the increase of the capacitor rms current
IC,DAB1
IC,DAB3
(6.7)
for the primary and secondary side, respectively. It is evident that, especially when the voltage
conversion ratio is different from 1, the capacitor current in the single-phase configuration is
much higher compared to the three-phase configuration.
Secondly, the stray inductance, which turned out to be much smaller than intended, increases
the rms capacitor current further. Exemplary, the capacitor current is depicted for different
values of the series inductance at a moderate voltage conversion ratio of 5 % in Fig. 6.22. Even
at lower power levels, the rms current rises steeply if not enough series inductance is provided.
This effect is more critical in a single-phase operation than in three-phase.
Switching Losses
Although the dual-active bridge converters are soft-switched, switching losses occur. These
are mainly turn-off losses that depend on the current that is switched off.
In a single-phase configuration the current that has to be turned off is high (at the peak of
the waveform). At a conversion ratio of 1 it is nearly the dc current. In contrast to that, the
current that has to be switched off in a DAB3 is lower. For a conversion ratio of 1 it is only
the half current of the single-phase DAB.
Consequently, the switching losses in the main switches of a single-phase are substantially
increased as compared to the three-phase DABC.
This chapter shall underline the advantages of a three-phase dual-active bridge configuration.
The additional effort implementing a three-phase dual-active bridge converter, in contrast to
a single-phase converter, is comparatively low. However, the utilization of the components
can be improved significantly. Not only the utilization of the passive components but also of
the power-semiconductors enhances.
65
Demonstrator
5
4.5
2.5
3
3.5
4
4.5
2
2
2.5
4
3
3.5
3
2
4
4.5
4
3.5
3
2.5
2
2.5
3
3.5
4
2
1
0.95
1
1.05
4.5
4
3.5
3
2.5
2
2
2.5
3
3.5
1
4.5
4
3.5
3
2.5
2
2
2.5
3
3.5
2
3
4
2.5
2
2
2.5
3
3.5
4
power in MW
4
3
3.5
4
4.5
4
power in MW
5
2
2.5
3
3.5
2
2.5
3
3.5
4
5
0.95
voltage conversion ratio
1
1.05
voltage conversion ratio
(a) primary side
(b) secondary side
Figure 6.21: Comparison of the capacitor rms current in a single-phase and three-phase
configuration; switching frequency 1 kHz, primary dc-link voltage 5 kV, series
inductance 45 µH
capacitor current in A
capacitor current in A
5
400
160
200
240
600
800
1000
power in MW
80
80
40
50
120
200
80
120
150
160
200
240
100
2
1
400
50
600
800
1000
1
200
1200
2
3
160
200
240
0
20
400
600
3
80
4
800
1000
power in MW
4
120
120
5
100
150
series inductance in µH
series inductance in µH
(a) single-phase configuration
(b) three-phase configuration
200
Figure 6.22: Primary-side capacitor current as function of the series inductance; switching
frequency 1 kHz, primary dc-link voltage 5 kV, voltage conversion ratio 1.05
66
7 Conclusion
Future, flexible power distribution systems will use DC multi-terminal distribution networks.
To realize such systems, dc-dc converters , i.e. electronic transformers, are key enabling
components. In this project, a high-power dc-dc converter is designed and constructed.
Many decentralized power generation systems, such as wind farms, PV power plants or energy
distribution in city quarters offer a high potential for cost savings when using dc networks. To
open these application areas, a high-power dc-to-dc converter that is highly efficient, robust
and offers a high power density is needed. The three-phase dual-active bridge is in the focus
of this work, as this topology offers high potential of fulfilling all requirements.
The dual-active bridge offers high dynamic control performance. Using the developed instantaneous current control, the converter is able to achieve a load step within one third of a
switching period. The implemented balancing control compensates the effect of asymmetric
transformers in the ac link and, consequently, improves the power quality in the dc-link.
One major advantage of the converter is its inherent soft-switching capability in a large operating range. This and the use of lossless snubbers decrease the switching losses significantly.
Together with the applied thyristor-based power-electronic devices offering low conduction
losses this results in a very high efficiency. Moreover, the snubbers achieve the dynamic voltage balancing allowing an easy series connection of switches. To ensure zero-voltage switching
in the entire operating range, an auxiliary circuitry is investigated and implemented. This
small circuitry ensures minimal losses and makes the converter inherently safe to an IGCT
failure.
At the given power and voltage rating, the design of the transformer located in the ac link of
the converter is important. It is operated at elevated frequency to increase the power density
and to improve the efficiency. The operation in such a DAB converter poses great demands
on the transformer design. However, a 2.2 MVA single-phase prototype, that was built and
tested, demonstrates the great potential for an ten-fold power-density increase as compared
to a 50 Hz transformer.
No noticeable problems had occurred during the commissioning. This approves the suitability
of the thyristor-based power-electronic switches for the three-phase dual-active bridge. The
commissioning which has been carried out in a single-phase configuration first, clarifies the
advantages of the three-phase dual-active bridge particularly in high-power applications.
67
8 Further Steps and Future Development
Although, synthetic tests have shown that the three-phase DABC can achieve above 99 %
efficiency [57], further improvements can already be identified. In particular, the mediumfrequency high-power transformer offers great potential for future development. The optimal
core material and core design should be further developed. Amorphous iron offers lower specific
losses compared to silicon steel. However, the higher magnetostriction, causing mechanical
stress in the core, prohibits the use of such cores at the given power rating and frequency (1–
2 kHz). This effect needs to be investigated further. Moreover, the optimal material depends
on the operating frequency of the dual-active bridge converter. New semiconducting materials,
like silicon carbide or gallium nitride, and new power electronic devices offer higher operating
frequencies [79]. Consequently, they need to be considered in a joint analysis to determine the
optimal material combination.
In this project, the stray inductance of the transformer has been increased to serve as series
inductance in the ac link. Winding-core configurations should be investigated that allow
an improved precalculation of the stray inductance. At the same time, the applicability
in medium-voltage high-power transformers is of course essential. Alternatively, it should be
investigated to use separate series inductances. Depending on the needed inductance value, the
easier implementation should be confronted with the lower power density. These investigations
can be conducted for single-phase and three-phase transformers, respectively.
The open question considering the converter itself is how the auxiliary circuit affects the system
reliability. Implementing the auxiliary circuit, the number of devices is increased. However,
it offers inherent safety from a phase-leg short circuit. Based on the investigation of possible
failures, counteractions should be found. Possibly, alternative operating strategies offer, along
with damage limitation, continuous operation at decreased power levels.
Since the dual-active bridge is galvanically isolated, converters can be connected in series to
step up the voltage. The proper control strategies and voltage balancing for such a converter
pool have to be investigated. The insulation of the winding on the high-voltage side may
also pose new challenges in very-high-voltage applications. Furthermore, as more converters
are used, this offers additional degrees of freedom to ensure soft-switching and to continue
operation in case of failure.
Ultimately, the three-phase dual-active bridge is the perfect candidate to operate in multiterminal MVDC grids. Control strategies on system and converter level and fault isolation
68
Further Steps and Future Development
need to be investigated. The results can be supported by power-hardware in the loop applying
the high-power dc-dc converter to a real time simulator. Moreover, the results and experiences
on the device and converter level will also contribute to findings on system level.
Another future research topic is the integration of all required components into a small unit
with a high power density. Medium-voltage systems have to become standard products and
need to become more cost-effective for a wide use in future grids. The use of optimized
thyristor-based turn-off power semiconductors as the ICT, IETO and Dual-ICT [80–82] should
be evaluated to analyze the potentials for efficiency and power density of the dual-active bridge
for future setups. In addition, the gate drives and the devices can be further optimized for
DABC applications, thereby further enhancing performance. The overall cooling systems can
be simplified when air cooling is applied. The losses in the power devices of the soft-switched
DABC are considerably lower than in VSI PWM applications. Hence, liquid cooled systems
(having pumps and deionization units) can be avoided.
69
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10 Attachments
10.1 List of Figures
2.1
Installed offshore wind capacity in Europe (1993-2012), Source: [17] . . . . . . .
2.2
Distance and depth of planned offshore wind farms (bubble size represents windfarm capacity), Source: [17] . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
2.3
5
5
Schematic of a dc collector grid for offshore wind farms and the connection to
different dc sinks . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
6
2.4
Different topologies for dc collector grids . . . . . . . . . . . . . . . . . . . . . .
7
2.5
Collector grid topologies for a PV application . . . . . . . . . . . . . . . . . . .
9
2.6
Structure of a a solid-state ac transformer . . . . . . . . . . . . . . . . . . . . . 10
2.7
Target ratings for different applications . . . . . . . . . . . . . . . . . . . . . . . 12
2.8
Schematic of the three-phase DAB . . . . . . . . . . . . . . . . . . . . . . . . . 13
2.9
Characteristic voltage and current waveforms in a DAB3 . . . . . . . . . . . . . 14
3.1
Schematics considered for the modeling approach . . . . . . . . . . . . . . . . . 16
3.2
Phasor diagram of a DAB3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
3.3
Dynamic FHA model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
3.4
Comparison of models with circuit simulation . . . . . . . . . . . . . . . . . . . 19
3.5
Applying the Clarke transformation to DAB3 currents . . . . . . . . . . . . . . 20
3.6
Load-angle change without sign change . . . . . . . . . . . . . . . . . . . . . . . 20
3.7
Load-angle change in the time domain . . . . . . . . . . . . . . . . . . . . . . . 21
3.8
Load-angle change with sign change
3.9
Comparison of original and improved ICC in measurement . . . . . . . . . . . . 22
. . . . . . . . . . . . . . . . . . . . . . . . 21
3.10 Closed-loop current control . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
3.11 Closed-loop voltage control . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
3.12 Influence of an asymmetric transformer on a DAB3 . . . . . . . . . . . . . . . . 24
77
Attachments
3.13 Transformer currents applying balancing angles . . . . . . . . . . . . . . . . . . 25
4.1
Series connection of IGCTs using conventional snubber circuits for dynamic
voltage balancing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
4.2
Voltage balancing applying lossless snubbers [57] . . . . . . . . . . . . . . . . . 28
4.3
Voltage and current transients during ZVS turn off [58] . . . . . . . . . . . . . . 29
4.4
Turn-off losses in presence of a lossless snubber for different IGCTs [58] . . . . . 30
4.5
Semiconductor losses in a DAB3 . . . . . . . . . . . . . . . . . . . . . . . . . . 32
4.6
Hard and soft switched operating areas . . . . . . . . . . . . . . . . . . . . . . . 33
4.7
Single phase leg of an Auxiliary Resonant-Commutated Pole (ARCP) . . . . . . 34
4.8
Integration of the ARCP in the DAB3 . . . . . . . . . . . . . . . . . . . . . . . 34
4.9
Commutation energy for different configurations of Saux . . . . . . . . . . . . . 35
5.1
Transformer requirements for different applications . . . . . . . . . . . . . . . . 38
5.2
Measuring results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
5.3
Steinmetz parameter extraction and validation of the iGSE . . . . . . . . . . . 41
5.4
Magnetic field in the winding window
5.5
Simulating the stray inductance in a 2-D FEM simulation . . . . . . . . . . . . 44
5.6
Alternative stacked winding configuration . . . . . . . . . . . . . . . . . . . . . 45
5.7
Simulating the stray inductance for a stacked winding configuration . . . . . . . 45
6.1
PC D247 control hardware . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48
6.2
PC D247 system overview, Source:[75] . . . . . . . . . . . . . . . . . . . . . . . 49
6.3
Application-block interface . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50
6.4
Overview of the modules in the FPGA . . . . . . . . . . . . . . . . . . . . . . . 52
6.5
Overview of the PowerPC Software . . . . . . . . . . . . . . . . . . . . . . . . . 52
6.6
Demonstrator in an early construction phase . . . . . . . . . . . . . . . . . . . . 53
6.7
Spice model to design the clamping circuit . . . . . . . . . . . . . . . . . . . . . 54
6.8
Power consumption of one IGCT, Source: [77] . . . . . . . . . . . . . . . . . . . 56
6.9
Transformer current versus stray inductance . . . . . . . . . . . . . . . . . . . . 57
. . . . . . . . . . . . . . . . . . . . . . . 43
6.10 ARCP prototype . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 58
78
Attachments
6.11 ZV detection circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 58
6.12 Verification of the ZV detection circuit . . . . . . . . . . . . . . . . . . . . . . . 59
6.13 Schematic Setup A . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 59
6.14 Measuring Setup A . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 60
6.15 Schematic Setup B . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 61
6.16 Measuring Setup B . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 61
6.17 Schematic Setup C . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62
6.18 Measuring Setup C, Up = 450 V, ϕ = 18◦ . . . . . . . . . . . . . . . . . . . . . . 63
6.19 Measuring Setup C, Up = 1000 V, ϕ = 12◦ . . . . . . . . . . . . . . . . . . . . . 63
6.20 Transformer voltage for a single-phase and three-phase DAB . . . . . . . . . . . 64
6.21 Comparison of the capacitor rms current in a single-phase and three-phase
configuration; switching frequency 1 kHz, primary dc-link voltage 5 kV, series
inductance 45 µH . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 66
6.22 Primary-side capacitor current as function of the series inductance; switching
frequency 1 kHz, primary dc-link voltage 5 kV, voltage conversion ratio 1.05 . . 66
10.2 List of Tables
2.1
DC-DC converter requirements of utility-scale applications . . . . . . . . . . . . 11
2.2
Possible dc-dc converter topologies for medium-voltage applications . . . . . . . 12
4.1
Simulation parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
4.2
Evaluation of different switching devices for an ARCP . . . . . . . . . . . . . . 36
5.1
Simulation results: verification of the analytical designs
. . . . . . . . . . . . . 44
10.3 Related Publications
• P. Köllensperger, R. Lenke, S. Schröder, R. W. De Doncker, “Design of a Flexible Control Platform for Soft-Switching Multilevel Inverters,” IEEE Transactions on Power
Electronics, Volume: 22 , Issue: 5, Pages 1778 - 1785, Sept. 2007
• R. Lenke, F. Mura, R. W. De Doncker, “Comparison of Non-Resonant and SuperResonant Dual-Active ZVS-Operated High-Power DC-DC Converters,” 13th European
Conference on Power Electronics and Applications (EPE), Barcelona, Spain, Sept. 2009
79
Attachments
• R. Lenke, S. Rohde, F. Mura, R. W. De Doncker, “Characterization of Amorphous Iron
Distribution Transformer Core for Use in High-Power Medium-Frequency Applications,”
Energy Conversion Congress and Exhibition (ECCE), San Jose, USA , Sept. 2009
• R. Lenke, J. Hu, R. W. De Doncker, “Unified Steady-State Description of Phase-ShiftControlled ZVS-Operated Series-Resonant and Non-Resonant Single-Active-Bridge Converters,” Energy Conversion Congress and Exhibition (ECCE), San Jose, USA, Sept.
2009
• F. Mura, R. Lenke, H. van Hoek, R. W. De Doncker, “Wirkungsgradanalyse eines
Offshore-Windparks mit Mittelspannungs-Gleichstromnetz,” ETG-Kongress, Düsseldorf,
Germany, Oct. 2009
• R. Lenke, B. Szymanski, R. W. De Doncker, “Low-Frequency Modeling of Three-phase,
Four-core, Strip-wound Transformers in High-Power DC-DC converters,” Energy Conversion Congress and Exhibition (ECCE), Atlanta, USA , Sept. 2010
• M. Bragard, N. Soltau, S. Thomas, R. W. De Doncker, “The Balance of Renewable
Sources and User Demands in Grids: Power Electronics for Modular Battery Energy
Storage Systems,” IEEE Transactions on Power Electronics, Vol. 25, No. 12, pp. 30493056, 2010
• R. W. De Doncker, S. Engel, N. Soltau, “High-Power Semiconductors for Multi-Terminal
Medium-Voltage DC Systems,” Korea-Germany Joint Symposium on Power Electronics
2011 (KOSEF), München, Germany, Aug. 2011
• R. Lenke, H. van Hoek, S. Taraborrelli, R. W. De Doncker, J. San Sebastian, I. EtxeberriaOtadui, “Turn-off behavior of 4.5 kV asymmetric IGCTs under zero voltage switching
conditions,” Proceedings of the 2011-14th European Conference on Power Electronics
and Applications (EPE), Birmingham, UK, Sept. 2011
• S. Engel, N. Soltau, R. W. De Doncker, “Instantaneous Current Control for the ThreePhase Dual-Active Bridge DC-DC Converter,” IEEE Energy Conversion Congress and
Exposition, Raleigh, USA, Sept. 2012
• N. Soltau, H. Siddique, R. W. De Doncker, “Comprehensive Modeling and Control
Strategies for a Three-Phase Dual-Active Bridge,” International Conference on Renewable Energy Research and Applications (ICRERA), Nagasaki, Japan, Nov. 2012
• S. Engel, N. Soltau, H. Stagge, R. W. De Doncker, “Dynamic and Balanced Control
of Three-Phase High-Power Dual-Active Bridge DC-DC Converters in DC-Grid Applications,” IEEE Transactions on Power Electronics, Vol. 28, No. 4, pp. 1880-1889,
2012
• S. Engel, N. Soltau, H. Stagge, R. W. De Doncker, “Improved Instantaneous Current
80
Attachments
Control for the Three-Phase Dual-Active Bridge DC-DC Converter,” Energy Conversion
Congress and Exhibition Asia (ECCE Asia), Melbourne, Australia, Jun. 2013
• N. Soltau, S. Engel, R. W. De Doncker, “Compensation of Asymmetric Transformers in
High-Power DC-DC,” Energy Conversion Congress and Exhibition Asia (ECCE Asia),
Melbourne, Australia, Jun. 2013
• N. Soltau, D. Eggers, K. Hameyer, R. W. De Doncker, “Iron Losses in a MediumFrequency Transformer operated in a High-Power DC-DC Converter,” Conference on
the Computation of Electromagnetic Fields (COMPUMAG), Budapest, Hungary, Jul.
2013
• S. Engel, N. Soltau, H. Stagge, R. W. De Doncker, “Improved Instantaneous Current
Control for High-Power Three-Phase Dual-Active Bridge DC-DC Converters,” IEEE
Transactions on Power Electronics, 2013
• N. Soltau, D. Eggers, K. Hameyer, R. W. De Doncker, “Iron Losses in a MediumFrequency Transformer operated in a High-Power DC-DC Converter,” IEEE Transaction
on Magnetics, Vol. 50, No. 2, 2014
10.4 Short CV of Scientists Involved in the Project
Dipl.-Ing. Nils Soltau
• 2010: Diploma degree in Electrical Engineering and Information Technology, RWTH
Aachen University, Germany
• Since 2010: Research Associate at E.ON Energy Research Center, Institute for Power
Generation and Storage Systems (PGS), RWTH Aachen University, Germany
Dr.-Ing. Robert U. Lenke
• 2004: Diploma degree in Electrical Engineering and Information Technology, RWTH
Aachen University, Germany
• 2005: Project Engineer Power Electronics at Semikron Korea Ltd, Republic of Korea
• 2005-2007: Research Associate at Institute for Power Electronics and Electrical Drives
(ISEA), RWTH Aachen University, Germany
• 2007-2010: Research Associate at E.ON Energy Research Center, Institute for Power
Generation and Storage Systems (PGS), RWTH Aachen University, Germany
• 2010: Doctor in Electrical Engineering and Information Technology, RWTH Aachen
University, Germany
81
Attachments
• 2010-2012: R&D Engineer at Bonfiglioli Vectron GmbH, Krefeld, Germany
• since 2012: Head of Product Management Photovoltaics at Bonfiglioli Vectron GmbH,
Krefeld, Germany
Prof. Dr. ir. Dr. h. c. Rik W. De Doncker
• 1986: Doctor in Electrical Engineering with the highest distinction at the Katholieke
Universiteit Leuven, Belgium
• 1987: Fulbright-Hays Award and N.A.T.O. Research Grant at the University of Wisconsin, Madison, USA, Visiting Associate Professor at the University of Wisconsin,
Madison, USA
• 1988: General Electric (GE) Fellowship at the Interuniversity Microelectronic Center
(IMEC), Leuven, Belgium
• 1989 - 1994: Senior Scientist at the GE Corporate Research and Development Center,
Schenectady/New York, USA
• 1991 - 1993: Adjunct Professor at the Rensselaer Polytechnical Institute (RPI), Troy/New
York, USA, Department of Electric Power Engineering
• 1994 - 1996: Vice President at Silicon Power Corporation (SPCO), Malvern/Pennsylvania,
USA
• 1996: Professor and head of the Institute for Power Electronics and Electrical Drives
(ISEA) at Aachen University of Technology
• 1999: Senior Member IEEE
• 2001: Fellow IEEE
• 2002: IEEE IAS 2002 Outstanding Achievement Award
• 2002 - 2004: IEEE IAS IPCSD Department Chair
• 2004: IEEE German Chapter Award
• 2004 - 2006: President IEEE PELS (Power Electronics Society)
• 2006: Director of the E.ON Energy Research Center and Head of the E.ON Energy
Research Center Institute Power Generation and Storage Systems (PGS)
• 2007: E.ON International Research Award (HERMES project)
• 2009: Nari Hingorani Custom Power Award of the IEEE PES (Power and Energy Society)
• 2010: Honorary Doctor Degree "Doktor Honoris Causa" at TU Riga
• 2013: IEEE William E. Newell Power Electronics Field Award
82
Attachments
10.5 Project Timeline
Over the years, the project, which was officially launched in July 2007, fell behind schedule.
The process of outsourcing the construction and fabrication of the high-power medium-voltage
transformer has been underestimated in the planning stage and resulted in a project delay of
approximately 2 years. Due to the importance of this topic, the research and the development
of the three-phase medium-frequency transformers is continued in the gGmbH Project No. 37
“Medium-Frequency Transformer for Medium-Voltage DC-DC Converters”.
10.6 Activities within the Scope of the Project
The project "high-power dc-dc converter" unites different research topics. The E.ON Energy
Research Center has gained experience in medium-voltage devices, magnetics at increased
power density, control theory and plant engineering. This knowledge has been used in other
research project, such as the high-speed PGS test bench, and publications. In total 18 master
and diploma theses and 5 PhD theses were completed within the scope of the project. The
project produced 6 transaction papers, 12 conference proceedings and 2 patents.
Using the constructed demonstrator, the E.ON Energy Research Center is able to test and
characterize high-power transformers at medium-frequency. A dc-dc converter rated for the
given voltage and power level is unique worldwide and will play an essential role in future
project acquisitions.
Furthermore, the dc-dc converter can be used by the Institute for Automation of Complex
Power Systems (Prof. Monti) to validate dc control function in a hardware-in-the-loop test
environment.
83
Project Synopsis
Nils Soltau, Robert Lenke, Rik W. De Doncker
PGS - Institute for Power Generation and Storage Systems
E.ON Energy Research Center (E.ON ERC), RWTH Aachen University
Mathieustr. 10
52074 Aachen, Germany
Dipl.-Ing. Nils Soltau
Tel.: +49 241/80 49957
Fax.: +49 241/80 49949
nsoltau@eonerc.rwth-aachen.de
Dr.-Ing. Robert U. Lenke
Tel.: +49 2151/8396 327
Fax: +49 2151/8396 994
Robert.Lenke@bonfiglioli.com
Univ.-Prof. Dr. ir. Dr. h. c. Rik W. De Doncker
Tel.: +49 241/80 49940
Fax: +49 241/80 49949
post_pgs@eonerc.rwth-aachen.de
Categories E.ON ERC focus
2
2
2
2
Small Scale CHP
Energy Storage
Consumer Behavior
Energy and Buildings
Distribution Networks
Carbon Storage (CCS)
Large Power Plants
Energy Efficiency
Energy Economics Modeling
Power Electronics
Renewable Energy
Others: Medium-Size Power Plants
Type of project report:
Final Project Report
Start and end date of project:
July 2007 - March 2013
Project in planned timelines:
yes 2
no (see section 10.5)
Participating Chairs of E.ON ERC
2
84
Automation of Complex Power Systems (ACS)
Energy Efficient Buildings and Indoor Climate (EBC)
Future Energy Consumer Needs and Behavior (FCN)
Applied Geophysics and Geothermal Energy (GGE)
Power Generation and Storage Systems (PGS)
Attachments
External R&D partners
The project was kindly supported with power-electronic devices and hardware by ABB Switzerland.
Acknowledgments
This project was supported by a grant of E.ON ERC gGmbH.
85
Notes:
86
E.ON Energy Research Center Series
ISSN: 1868-7415
First Edition: Aachen, July 2013
E.ON Energy Research Center,
RWTH Aachen University
Mathieustraße 10
52074 Aachen
Germany
T +49 (0)241 80 49667
F +49 (0)241 80 49669
post_erc@eonerc.rwth-aachen.de
www.eonerc.rwth-aachen.de