Download Oscillators and Synthesizers

Oscillators and Synthesizers
Fig 10.1—Reduced to essentials, an oscillator consists of a filter and an amplifier operating
in a feedback loop.
Fig 10.2—A resonator lies at the heart of every oscillatory mechanical and electrical system.
A mechanical resonator (here, a pendulum) and an electrical resonator (here, a tuned circuit
consisting of L and C in parallel) share the same mechanism: the regular movement of
energy between two forms—potential and kinetic in the pendulum, electric and magnetic in
the tuned circuit. Both of these resonators share another trait: Any oscillations induced in
them eventually die out because of losses—in the pendulum, due to drag and friction; in the
tuned circuit, due to the resistance, radiation and inductance. Note that the curves
corresponding to the pendulum’s displacement vs velocity and the tuned circuit’s voltage vs
current, differ by one quarter of a cycle, or 90°.
Fig 10.3—Stimulating a resonance, 1880s style. Shock-exciting a gapped ring with high
voltage from a charged capacitor causes the ring to oscillate at its resonant frequency. The
result is a damped wave, each successive alternation of which is weaker than its
predecessor because of resonator losses. Repetitively stimulating the ring produces trains of
damped waves, but oscillation is not continuous.
Fig 10.4—The barebones oscillator block diagram of Fig 10.1 did not include two practical
essentials: Networks to couple power in and out of the resonator (A). Breaking the loop,
inserting a test signal and measuring the loop’s overall gain (B) allows us to determine
whether the system can oscillate, sustain oscillation or clip (C).
Fig 10.5—An oscillator with noise. Realworld amplifiers, no matter how quiet, generate some
internal noise; this allows real-world oscillators to self - start.
Fig 10.6—At A, a phasor diagram of a clean (ideal) oscillator. Noise creates a region of
uncertainty in the vector’s length and position (B). AM noise varies the vector’s length; PM
noise varies the vector’s relative angular position (C) Limiting a signal that includes AM and
PM noise strips off the AM and leaves the PM (D).
Fig 10.6—At A, a phasor diagram of a clean (ideal) oscillator. Noise creates a region of
uncertainty in the vector’s length and position (B). AM noise varies the vector’s length; PM
noise varies the vector’s relative angular position (C) Limiting a signal that includes AM and
PM noise strips off the AM and leaves the PM (D).
Fig 10.9—Setup for measuring receiver-oscillator phase noise.
Fig 10.12—The Colpitts (A), series-tuned Colpitts (B) and Hartley (C) oscillator circuits. Rules
of thumb: C3 and C4 at A and B should be equal and valued such that their Xc = 45 Ω at the
operating frequency; for C2 at A, Xc = 100 Ω. For best stability, use C0G or NP0 units for all
capacitors associated with the FETs’ gates and sources. Depending on the FET chosen, the
1-kΩ source-bias-resistor value shown may require adjustment for reliable starting
Fig 10.13—Three more oscillator examples: at A, a triode-tube Hartley; at B, a bipolar
junction transistor in a series-tuned Colpitts; at C, a dual-gate MOSFET Hartley..
Fig 10.15—G3PDM’s Vackar VFO has proved popular and successful for two decades. The
MPF102 can be used as a substitute for the 2N3819. Generally, VFOs can be adapted to
work at other frequencies (within the limits of the active device). To do so, compute an
adjustment factor: fold / fnew. Multiply the value of each frequency determining or feedback L
or C by the factor. As frequency increases, it may help to increase feedback even more than
indicated by the factor.
Fig 10.18—A modern voltage-controlled oscillator (VCO) uses the voltage-variable
characteristic of a diode PN junction for tuning
Fig 10.19—Oscillator temperature compensation has become more difficult because of the
scarcity of negative-temperature-coefficient capacitors. This circuit, by GM4ZNX, uses a
bridge containing two identical thermistors to steer a tuning diode for drift correction. The
6.2 V Zener diode used (a 1N821A or 1N829A) is a temperature-compensated part; just any
6.2-V Zener will not do.
Fig 10.20—A practical VCO. The tuning diodes are halves of a BB204 dual,
common-cathode tuning diode (capacitance per section at 3 V, 39 pF) or equivalent. The
ECG617, NTE617 and MV104 are suitable dual-diode substitutes, or use pairs of 1N5451s
(39 pF at 4 V) or MV2109s (33 pF at 4 V).
Fig 10.21—Thickness-shear vibration at a crystal’s fundamental and third overtone (A); B
shows how the modern crystals commonly used by radio amateurs consist of etched quartz
discs with electrodes deposited directly on the crystal surface.
Fig 10.22—Slight changes in a crystal cut’s orientation shift its frequencyversus-temperature
curve.
Fig 10.23—Exploring a crystal’s impedance (A) and equivalent circuit (B) through simplified
diagrams. C and D extend the investigation to include overtones; E, to spurious responses
not easily predictable by theory or controllable through manufacture. A crystal may oscillate
on any of its resonances under the right conditions.
Fig 10.24—A basic series-mode crystal oscillator. A 2N5179 can be used in this circuit if a
lower supply voltage is used; see text.
Fig 10.25—A Butler crystal oscillator.
Fig 10.26—The crystal in the series-tuned Colpitts oscillator at A operates in its series resonant mode. B shows KA2WEU’s low-noise version, which uses the crystal as a filter and
features high harmonic suppression. The circuit at C builds on the B version by adding a
common-base output amplifier and ALC loop.
Fig 10.27—A wide-range variable crystal oscillator (VXO).
Fig 10.28—Using an inductor to “tune out” C0 can increase a crystal oscillator’s pulling range.
Fig 10.29—In a phase-shift oscillator (A) based on logic gates, a chain of RC networks—
three or more—provide the feedback and phase shift necessary for oscillation, at the cost of
low Q and considerable loop loss. Many commercial Amateur Radio transceivers have used
a phase-lead oscillator similar to that shown at B as a sidetone generator. Replacing one of
the resistors in A with a crystal produces a Pierce oscillator (C), a cut-down version of which
(D) has become the most common clock oscillator configuration in digital systems
Fig 10.30—Oscillators that use transmission-line segments as resonators. Such oscillators
are more common than many of us may think, as Fig 10.31 reveals.
Fig 10.31—High device gain at UHF and resonances in circuit board traces can result in
spurious oscillations even in non-RF equipment.
Fig 10.32—Evolution of the cavity resonator.
Fig 10.37—A basic phase-locked-loop (PLL) synthesizer acts to keep the divided – down
signal from its voltage-controlled oscillator (VCO) phase-locked to the divided - down signal
from its reference oscillator. Fine tuning steps are therefore possible without the complication
of direct synthesis.
Fig 10.38—(A) Shows the mechanism of a programmable frequency divider. (B) shows the
function of a dual-modulus prescaler. The counter is reloaded with N when the count reaches
0 or 1, depending on the sequencer action.
Fig 10.39—Simple phase detectors: a mixer (A), a sampler (B) and an exclusive OR
gate (C).
Fig 10.40—Input signals very far off frequency can confuse a simple phase - detector;
a phase-frequency detector solves this problem.
Fig 10.41—A simple RC filter is a “pole.”
Fig 10.42—A common loop-amplifier/filter arrangement.
Fig 10.43—Loop-amplifier detail.
Fig 10.44—The loop’s resulting pole – zero “constellation.”