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Transcript

RFIC Design and Testing for Wireless Communications A PragaTI (TI India Technical University) Course July 18, 21, 22, 2008 Lecture 7: RF front-end design – LNA, mixer By Vishwani D. Agrawal Fa Foster Dai 200 Broun Hall, Auburn University Auburn, AL 36849-5201, USA 1 RFIC Design and Testing for Wireless Communications Topics Monday, July 21, 2008 9:00 – 10:30 Introduction – Semiconductor history, RF characteristics 11:00 – 12:30 Basic Concepts – Linearity, noise figure, dynamic range 2:00 – 3:30 RF front-end design – LNA, mixer 4:00 – 5:30 Frequency synthesizer design I (PLL) T Tuesday, d July J l 22, 22 2008 9:00 – 10:30 11:00 – 12:30 2:00 – 3:30 Frequency synthesizer design II (VCO) RFIC design for wireless communications Analog and mixed signal testing RF front-end design – LNA, mixer, FDAI, 2008 2 LNA Design Challenges (1) Amplify extremely low signals without adding much noise. (2) Amplify large signals without distortions. (3) Variable gain to compensate large input signal variation. (4) Input matching and flat gain over wide bandwidth for muti-mode transceivers. (5) Input dynamic range of a WLAN LAN from -80dBm to -20dBm Parameters for Microwave and RFIC Design Peak-to peak voltage: Vpp Vrms = Root-mean-square voltage: V pp 2 2 2 V 2 rms V pp Pwatt = = R 8R Power in Watt : PdBm Power in dBm : Refection coefficient : P tt [mW W]⎞ ⎛ Pwatt = 10 log 10 ⎜ ⎟ 1mW ⎝ ⎠ Z in − RS Γ = S11 = Z in + RS Return loss = −20 log Γ Voltage standing wave ratio (VSWR): V max 1 + Γ = VSWR = Vmin 1 − Γ RF front-end design – LNA, mixer, FDAI, 2008 4 Basic Amplifiers • • • The common emitter amplifier is often used as a drive for an LNA. The common-collector, with high input impedance and low output impedance, p , makes an excellent buffer between stages g or before the output driver. The common-base is often used as a cascode in combination with the common-emitter to form a LNA stage g with g gain to high g frequency. q y VCC VCC VCC vout vout vin vout vin vin VEE VEE Common-Emitter (LNA Driver) VEE Common-Collector (Buffer) Common-Base (Cascode) RF front-end design – LNA, mixer, FDAI, 2008 5 Common-Emitter Amplifier vi rb Cμ vπ Cπ rπ gmvπ vo ro ZL • Voltage gain • Note that rπ=βre and gm=1/re. for low frequencies, the parasitic capacitances have been ignored and rb << rπ. • Input impedance at low frequencies Avo = vo r Z = − π gmZ L ≈ L vi rb + rπ re Z in = rb + rπ RF front-end design – LNA, mixer, FDAI, 2008 6 Miller Capacitance in Common-Emitter Amplifier Cμ is replaced with two equivalent capacitors CA and CB Cμ vπ vo ⇓ vπ CA CB vo ⎛ vo C A = Cμ ⎜⎜1 − ⎝ vπ ⎛ v C B = Cμ ⎜⎜1 − π ⎝ vo • ⎞ ⎟⎟ = Cμ (1 + g m Z L ) ≈ Cμ g m Z L ⎠ ⎞ ⎛ 1 ⎞ ⎟⎟ ≈ Cμ ⎟⎟ = Cμ ⎜⎜1 + ⎠ ⎝ gmZ L ⎠ Two RC time constants or two poles: one consisting g of CA+ Cπ, and the other consisting CB. RF front-end design – LNA, mixer, FDAI, 2008 7 Miller Capacitance in Common-Emitter Amplifier • The dominant pole is one formed by CA and Cπ: f P1 = • 1 2π ⋅ [rπ (rb + Rs )]⋅ [Cπ + C A ] Recall Ft calculation, the output is loaded with a short circuit Æ removes Miller multiplication, 3db current gain bandwidth: fβ = 1 2π ⋅ rπ (Cπ + Cμ ) The unity current gain frequency: gm fT = 2π ⋅ (Cπ + Cμ ) RF front-end design – LNA, mixer, FDAI, 2008 8 Common-Base Amplifier Cμ rb vπ Cπ rπ iout gmvπ ZL iin Current gain (ignoring Cu and ro) Av ≈ α 0 RL RS ⋅ 1 1 + jωreCπ ⋅ iout 1 1 ≈ ≈ iin 1 + jωCπ re 1 + j ω 1 1 + jωRL (Cμ + CCS ) At low l frequencies, f i th the currentt gain i =1 1. The pole in this equation is usually at much higher frequency than the one in the common-emitter amplifier ωT re < rb + RS VCC Cascode LNA vo / vi ≈ − g m RC RC vout Cascode Q2 VCbias vc1 Common emitter dominant pole 1 f P1 = 2π ⋅ [rπ 1 (rb1 + Rs )]⋅ [Cπ + 2Cu ] • • • • vi Driver Q1 VEE re ≈ 1 / g m 2 Current ic1 through Q1 is about the same as the current ic2 through Q2 Å current gain ~1 Æ Gain is the same as for the common-emitter amplifier. The cascode transistor reduces the feedback of Cu1 (why?) Æ increased high frequency gain. Cascode has good isolation with reduced S12. Disadvantage: cascode transistor uses voltage headroom Æ reduced linearity; add another pole to the amp Æ -12dB/oct roll-off; add little extra pp , cascode NF = common emitter NF); ); noise ((to the 1st order approximation, common base may cause noise and oscillation. RF front-end design – LNA, mixer, FDAI, 2008 10 Bipolar Transistor Noise Model b vb rb Cμ b’ 4kTrb ibn ibf Cπ rπ c gmvb’e ro 2qIC 2qIB e 1/f noise icn Collector C Base e Shot Noise Shot Noise 2 vbn = 4kTrb Base vbn rb Collector Zπ i = 2qI B 2 bn ibn rπ Cπ gmVπ1 ro icn icn2 = 2qI c Emitter RF front-end design – LNA, mixer, FDAI, 2008 11 Min nimum Noisee Figure (dB)) Noise Figure versus Bias Current Output signal vso ≈ vsi ⋅ g m RL Base thermal noise vno ,rb ≈ 4kTrb ⋅ g m RL Collector shot noise vno , I c ≈ 2qI C RL 2 40 2.40 2.20 Base shot noise 2.00 base th thermal l noise 1.80 1.60 0 1.0 2.0 3.0 4.0 vno , I B ≈ 2qI C base shot noise β Rs g m RL collector shot noise (correlated) 5.0 6.0 Collector Current (mA) Req rb g m RS g m RS 1 NF = 1 + = 1+ + + + 2 RS RS 2 g m RS 2β 0 2β RF front-end design – LNA, mixer, FDAI, 2008 12 Noise Figure of A Two Stage LNA f t = 5GHz rb = 11Ω f in = 1GHz β 0 = 80 β =5 g m = 0.1Ω −1 rb 1 g m RS g m RS + + + NF dominatedNF = 1 + 2 R 2 g R 2 β 2β by base S m S 0 resistance NF increases at high frequency 11 1 5 5 = 1+ + + + = 1.62dB 50 10 160 50 RF front-end design – LNA, mixer, FDAI, 2008 13 Minimum Noise Figure of Common Emitter LNA rb g m RS g m RS 1 1 NF = 1 + + + + = 1 + a + b ⋅ RS 2 RS 2 g m RS 2β 0 RS 2β dNF 1 = −a 2 + b = 0 dRS RS RS ,opt = NFmin a 1 = b gm 1 + 2 g m rb f ≈ T 1 1 f + 2 β0 2rb gm β RS=50 Ohm Æ choose bias (gm) and emitter length to achieve hi noise i matching ⎛ 1 ⎞ 1 = 1 + 2 ab = 1 + (1 + 2 g m rb )⎜ + 2⎟ ⎜ β0 β ⎟ ⎝ ⎠ RF front-end design – LNA, mixer, FDAI, 2008 14 More on LNA Noise Figure NF = NFmin ( b + RS − RS ,opt RS High frequency, RS ,opt fT ≈ f Low frequency, RS ,opt ≈ • • • • ) 2 ≈ NFmin ( f 2 gm + RS − RS ,opt 2 2 f T RS ) 2 2rb 2rb NFmin ≈ 1 + Cπ ω gm gm 2rb β 0 gm NFmin ≈ 1 + 2 g m rb β0 For a given technology, NFmin is a strong function of bias current. For low operation frequency, NFmin can be reduced by increasing emitter length. For high operation frequency, NFmin is a weak function of emitter length. Increase device size does reduce rb, yet capacitance also increase. For high operation frequency, NFmin degrades as frequency increases. RF front-end design – LNA, mixer, FDAI, 2008 15 Input Matching of LNA Noise Power matching: RFin • Lb Input impedance (assuming Cu and rπ is not significant): Le g m Le j + jωLe + Z in = jωLb + rb − ωCπ Cπ • ( RS − rb )Cπ RS ≈ Le = gm ωT Power matching: g m Le + rb = RS Real part=Rs C π 1 Imaginary Le + Lb = 2 part=0 ω Cπ • Lb = Q1 1 Cπ ω 2 − (RS − rb )Cπ gm ≈ RS − rb ωT − g mω 2 ωT If Cu is considered, Cπreplaced by Cπ + CA, and therefore a larger inductor is required for matching. C A = C μ (1 + g m Z L ) ≈ C μ g m Z L RF front-end design – LNA, mixer, FDAI, 2008 16 LNA Design Steps (1) Noise matching: sizing the transistor (emitter length) and adjusting bias current to achieve minimum NF RS ,opt 1 = gm RFin Lb Q1 1 + 2 g m rb = 50Ω ⎛ 1 ⎞ 1 1 1 ⎜ + 2 NFmin = 1 + (1 + 2 g m rb ) + 2⎟ β0 ⎜ β0 ⎝ β (2) Power matching: adjusting Le such that the real part of the LNA input impedance equals to 50 Ohm. For 5AM, Le is about 0.2nH Æ Use multiple downbonds to reduce the package effect effect. Le = (3) Power matching: adding Lb such that the imaginary part of the LNA input impedance equals to zero (4) Gain/bandwidth: G i /b d idth Choosing Ch i lload d ttank k tto meett gain i and d bandwidth requirements. RS ωT Le β ⎟⎠ = 50 ωT ωT Lb ≈ g mω 2 (5) Using SPICE sim to fine tune the component values, it ti iterations t trade to t d off ff the th various i parameters t are expected. t d RF front-end design – LNA, mixer, FDAI, 2008 17 Mixing with Nonlinearity • • • Mixer is to convert a signal from one frequency to another Æ intrinsically needs a nonlinear transfer function. A diode or a transistor can be used as a nonlinear ca o ea de device. ce Two inputs at ω1 and ω2 , which are passed through a nonlinearity multiplier will produce mixing terms at ω1±ω2 Æwith other terms (harmonics, feed-through, intermodulation) that need to be filtered out. Mixers (multiplier) can be made from an amplifier with a controlled switch. VCC Rc VCC Rc vout - vout + v2 switch v1 i(v1) RF front-end design – LNA, mixer, FDAI, 2008 18 Controlled Transconductance Mixer • The currentt is Th i related l t d to t the th input i t voltage lt v2 by b th the ttransconductance d t of the input transistors Q1 and Q2. The transconductance is controlled by the current I0, which in turn is controlled by the input voltage v1. io i1 v2 + Q1 i2 2Io Q2 - v1 i0=i1-i2 v2 2Io -2Io RF front-end design – LNA, mixer, FDAI, 2008 IO i1 = 1+ e − v2 / vT IO i2 = 1+ e v2 / vT 19 Controlled Transconductance Mixer IO ⎛ io = i1 − i2 = I o ⎜ −v ⎝1+ e • • If v2<<vT io ≈ I o 2 / vT − IO 1 + ev 2 / vT v ⎞ ⎟ = I o tanh 2 2vT ⎠ v2 2vT Current source is modulated by small v1 Æ Io is replaced with Io+gmcv1, where gmc is transconductance of the current source: v2 v2 v2 = I o tanh + g mc v1 tanh io = ( I o + g mc v1 ) tanh 2vT 2vT 2vT 14243 144244 3 V2 feedthrough multiplication ( mixing ) not appear in differential output voltage Æ double balanced mixer RF front-end design – LNA, mixer, FDAI, 2008 20 Double-balanced Mixer VCC RC1 vo1 Use switching quad to eliminate the v2 feedthrough vo2 i3 Q3 v2 v1 • • • RC2 pair current: i6 Q6 Q5 Q4 2Io 2nd i5 i4 -v1 2Io v2 v2 i = i6 − i5 = I o tanh − g mc v1 tanh 2vT 2vT ' o Total differential current: v2 iob = io − i = 2 g mc v1 tanh 2vT ' o RF front-end design – LNA, mixer, FDAI, 2008 21 Double-balanced Mixer ⎛ v2 (i3 + i5 ) − (i4 + i6 ) = tanh⎜⎜ ⎝ 2vT ⎞ ⎛ v2 ⎟⎟(i1 − i2 ) = tanh⎜⎜ ⎠ ⎝ 2vT A Assuming i ⎞ v1 ⎟⎟ small signal ⎠ re + RE for V1 VCC RC1 vo1 RC2 i3 i4 Q4 Q3 vo2 i5 i6 Q6 Q5 v2 i2 i1 Q1 RE RE v1 Q2 i1 1 + e −v2 / vT i2 i5 = 1 + e v2 / vT i3 = i1 ≈ I o + i1 1 + e v2 / vT i2 i6 = 1 + e −v2 / vT i4 = v1 1 2 re + RE and 2Io i2 = I o − v1 1 2 re + RE VEE RF front-end design – LNA, mixer, FDAI, 2008 22 Double-balanced Mixer • • • Output differential voltage Conversion gain relative to v1 ⎛ v2 vo = − tanh⎜⎜ ⎝ 2vT ⎞ v1 ⎟⎟ RC ⎠ re + RE ⎛ v2 vo = − tanh⎜⎜ v1 ⎝ 2vT ⎞ RC ⎟⎟ ⎠ re + RE With RE=0, a general large-signal expression for the output: ⎛ v1 vo = −2 RC I o tanh⎜⎜ ⎝ 2vT ⎞ ⎛ v2 ⎟⎟ tanh⎜⎜ ⎠ ⎝ 2vT ⎞ ⎟⎟ ⎠ RF front-end design – LNA, mixer, FDAI, 2008 23 LO Level at Upper Quad Transistors • • • • • Th diff The differential i l pair i needs d an iinput voltage l swing i off about b 4 to 5 vT for f the h transistors i to be hard-switched one way or the other. LO input to the mixer should be at least 100mV peak for complete switching. At 50Ω, 100mV peak is -10 dBm. w1 -10 to 0 dBm (100~300 (100 300 mVpp = 200~600mVpp 200 600mVpp diff) is a reasonable compromise between noise figure, gain and required LO power. This is also the reasonable level for all switching circuits If the LO voltage is too large, large current has to be moved into and out of the bases of the transistors during transition Æ lead to spikes in the signals and reduce the switching speed Æ cause an increase in LO feed feed-through. through Large LO also pushes switching transistor into saturation Æ loose switching speed and inject mixer noise into substrate. a) Vc b) Rcc VLO+ Rc Vin- Vin+ VLO- Q3 V Q4 d Vd Ibias RF front-end design – LNA, mixer, FDAI, 2008 24 Slide 24 w1 weishen, 7/16/2003 Mixer Noise Noise Contributions N 0tot NF = 10 log N 0( source) 11 = 10 log = 8.65dB 1.5 Output N Noise Power (Relativve values) total 10 top transistors 8 6 bottom t transistors it 4 2 0 -0.1 0 0.1 source resistance VLO (instantaneous) • • • • • • • Top transistor contributes significant noise during transition and contributes ignorable noise when fully switched, either in cutoff or saturation (without gain). Gain from RF input is maximum when top transistors are fully switched (cascode) (cascode). Æ need sharp transition buffer for LO Æ large LO such that minimal time is spent around 0V. Mixer noise figure can be approximately analyzed using a lowly swept dc voltage at the LO input or with an actual LO signal. With a slowly swept dc voltage, the mixer becomes equivalent to a cascode amplifier and the LO input served as a gain-controlling signal. For mixer, any noise (or signal) is mixed to two output frequencies, thus reducing the output level Æ mixer having less gain than the equivalent differential pair. However, both RF and image is mixed to IF Æ doubling noise power at output. RF front-end design – LNA, mixer, FDAI, 2008 25 Mixer with Simultaneous Noise and Power Match • Use inductor degeneration and inductor input achieving simultaneous noise and power matching similar to that of a typical LNA. VCC RL RL LO reject j t IF- IF+ Z0 LE = 2πfT LO Q1 RF VBB Q2 LE LE VBB Filter harmonics • • ωg m Z 0 IIP3 ≈ 2πfT Single-to-differential Noise matching: sizing LE, and RF transistor, and operating the RF transistors at the current required for minimum NF. quad switching g transistors are sized for maximum fT, (typically ( yp y about five to The q ten times smaller than the RF transistors.) RF front-end design – LNA, mixer, FDAI, 2008 26 Mixer Design Issues Sizing Transistors • • The RF differential pair is basically an LNA stage, and the transistors and associated passives can be optimized using the LNA design techniques. The switching gq quad transistors are sized so that they y operate p close to their peak fT at the bias current that is optimal for the differential pair transistors are biased at their minimum noise current, then the switching transistors end up being about one-eighth the size. Increasing Gain • Voltage gain without matching and assuming full switching of the upper quad: RC vo = vin π re + RE 2 • • To increase the gain Æ increase the load resistance RC, to reduce degeneration resistance RE, or to increase the bias current IB. Make sure that increasing output voltage swing will not cause the switching transistors to become saturated.Æ Enough g headroom. RF front-end design – LNA, mixer, FDAI, 2008 27 Mixer Design Issues Increasing IP3 • Identify which part of the circuit is compressing. Compression can be due to overdriving of the lower differential pair, clipping at the output, or the LO bias voltage being too low, causing clipping at the collectors of the bottom pair. Adjust j the bias and voltage g swing g to avoid clipping. pp g differential p 1. If the compression is due to the bottom differential pair (RF input), then linearity can be improved by increasing RE or by increasing bias current. 2. Compression caused by clipping at the output is typically due to the quad going g into saturation. Saturation can be avoided by y reducing g the transistors g load resistance or adjust the quad transistor bias. Too large LO will also cause saturation. 3. If compression is caused by clipping at the collector of the RF input differential pair, then increasing the LO bias voltage will improve linearity; h however, this hi may result l iin clipping li i at the h output. • • • Improving Noise Figure • • • • NF will be largely determined by the choice of topology. topology Use the simultaneous matched design technique. To minimize noise, the emitter degeneration resistor should be kept as small as possible. Use inductor as degeneration to achieve low noise. Make top transistors switching fast. RF front-end design – LNA, mixer, FDAI, 2008 28 Mixer Design Issues M t hi Matching, Bias Bi Resistors, R i t and d Gain G i • • Use resistive matching to achieve broad band. For a resistively degenerated mixer, the RF input impedance will be fairly high; for example, with RE=100 Ω, Zin can be b off the th order d off a Kilo Kil Oh Ohm Æ easier i ffor LNA output t t stage t to t drive d i the th mixer. At the output, if matched, the load resistor Ro is equal to the collector resistor Rc. Furthermore, to convert from voltage gain Av to power gain Po/Pi, one must consider the output resistance Ri and load resistance Ro=Rc as follows: vo2 2 ⎛ ⎞ / 2 R R R Po Ro vo2 Ri 2 C ⎟⎟ i = 2 = 2 = Av2 i ≈ ⎜⎜ Ro ⎝ π RE ⎠ RC Pi v i Ro vi Ri RF front-end design – LNA, mixer, FDAI, 2008 29