Download Simplification of millimeter-wave radio-over

Simplification of millimeter-wave radio-overfiber system employing heterodyning of
uncorrelated optical carriers and selfhomodyning of RF signal at the receiver
A.H.M. Razibul Islam,1,2,* Masuduzzaman Bakaul,1,2,4 Ampalavanapillai Nirmalathas,1
and Graham E. Town3
2
1
Department of Electrical & Electronic Engineering, University of Melbourne, Parkville, Vic-3010, Australia
NICTA (National ICT Australia Ltd.), Department of Electrical & Electronic Engineering, University of Melbourne,
Parkville, Vic-3010, Australia
3
Department of Electronic Engineering, Macquarie University, North Ryde, NSW-2109, Australia
4
[email protected]
*
[email protected]
Abstract: A simplified millimeter-wave (mm-wave) radio-over-fiber (RoF)
system employing a combination of optical heterodyning in signal
generation and radio frequency (RF) self-homodyning in data recovery
process is proposed and demonstrated. Three variants of the system are
considered in which two independent uncorrelated lasers with a frequency
offset equal to the desired mm-wave carrier frequency are used to generate
the transmitted signal. Uncorrelated phase noise in the resulting mm-wave
signal after photodetection was overcome by using RF self-homodyning in
the data recovery process. Theoretical analyses followed by experimental
results and simulated characterizations confirm the system’s performance.
A key advantage of the system is that it avoids the need for high-speed
electro-optic and electronic devices operating at the RF carrier frequency at
both the central station and base stations.
©2012 Optical Society of America
OCIS codes: (060.2360) Fiber optic links and subsystems; (060.5625) Radio frequency
photonics; (060.2840) Heterodyne; (060.2920) Homodyning; (060.4080) Modulation;
(300.3700) Linewidth.
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1. Introduction
Applications like high-definition TV (HDTV), video-on-demand (VoD), internet video and
peer-to-peer communications will be the dominant component of the total internet traffic in
the near future [1]. A large portion of these bandwidth-intensive applications is expected to be
supported by current wireless-internet networks [2]. Such expectations can only be met by
enabling broadband wireless access (BWA) at speeds no less than 50 to 100 Mbps per user
[3]. Existing wireless access schemes, which mainly operate at lower microwave frequencies
(<5 GHz), cannot offer such speed due to spectral congestions. Millimeter-wave (mm-wave)
frequencies with aggregated channel bandwidths up to 10 GHz are therefore being actively
investigated to offer larger transmission bandwidths [4].
Mm-wave radio-over-fiber (RoF) is a key enabler in offering BWA in which base stations
(BSs) are connected to the central station (CS) via a fiber feeder network enabling centralized
control and monitoring [5,6]. For simplicity, BSs are housed with devices for effective
handover only. However, as atmospheric attenuation at mm-wave frequencies can be
extremely high, the coverage of each BS is reduced to few 10’s to few 100’s meters [6]. This
means that a large number of BSs will be required to provide extensive geographical
coverage. Consequently, to make the BWA systems commercially viable, cost-effective and
simplified sub-system designs, such as for CSs and BSs, are essential.
Various approaches for the generation and distribution of mm-wave RoF signals have
been proposed and can be categorized into two main groups. In the first group, an optical
heterodyne technique is employed where the difference between two optical frequencies
mixed in a photodetector (PD) generates the desired the mm-wave carrier. Several techniques
have been demonstrated to generate phase-correlated optical carriers. These include optical
phase locking or injection locking of two laser diodes [6,7], frequency-locked lasers [8,9],
sub-harmonically-locked lasers [10] and so on. Multimode lasers, in which the modes are not
phase locked but are related in frequency, have also been proposed as sources for microwave
systems [11,12].
The second approach employs coherent spectral lines from a single light source to
generate optical double sideband with carrier (ODSB+C), optical single sideband with carrier
(OSSB+C) or optical carrier-suppressed double sideband (OCS-DSB) signal. Such an
approach requires high-speed opto-electronic and RF devices including modulators, PDs,
local oscillators (LOs), mixers and filters [13–15]. As most of the devices are yet to mature,
they are still expensive.
In most lasers, the intensity and phase noise mainly occurs at relatively low frequencies
(i.e. kHz to MHz), with linewidths well below the bit rate of the data to be transmitted.
Consequently, heterodyned lasers do not necessarily need to be correlated in phase or
frequency to be used in mm-wave RoF systems [16–18]. RoF systems employing such
uncorrelated lasers can overcome the relevant phase-noise effects by applying incoherent
demodulation, such as RF self-homodyning.
We recently proposed and demonstrated such a system [19,20] generating amplitude-shiftkey (ASK) modulated mm-wave signal using two uncorrelated, off-the-shelf, distributed
feedback (DFB) lasers; from which data was recovered using phase-insensitive RF selfhomodyning. This paper extends the previous work by incorporating theoretical analyses and
by demonstrating the proposed concept for an additional system configuration. In also
includes comparison of results for different configurations through experiments, simulations
and numerical analyses.
The new system configuration has the added advantage of reusing the same light-source
for both up and downlinks [13]. Moreover, the extension includes analysis and impact of laser
noise properties, such as, linewidth and relative intensity noise (RIN) on the recovery of data
using RF self-homodyning.
Scheme A
Scheme B
f1
Central Station
f1
f2
f1
Central Station
f1
f2
MZM
f2
OC
f2
MZM
OC
Data
PD
PD
Base Station
Data
Base Station
(b)
(a)
Scheme C
PD
MZM
f1
( f1
0
− f2
)
0
( f1
− f2
)
0 2 ( f1 − f 2
)
0
OC
f1
Data
f2
Central Station
LPF
f2
Base Station
PD
(c)
Amp.
Data
Self-homodyning
(d)
Fig. 1. (a) Millimeter-wave generation schemes exploiting heterodyning of uncorrelated optical
carriers, where (a): data is imposed on two independent optical carriers located at CS and
separated by 35.75 GHz, similar to a OCS-DSB modulation format (Scheme A), (b): data is
imposed on one optical carrier separated by 35.75 GHz from a second independent optical
carrier both located at CS, similar to an OSSB + C modulation format (Scheme B), (c): data is
imposed on one optical carrier located at CS and coupled with an optical LO (separated by
35.75 GHz) at the BS to enable local optical heterodyning prior to photodetection (Scheme C);
(d): describes RF self-homodyning for phase-insensitive recovery of data at the receiver
irrespective of heterodyning schemes.
The paper is organized as follows: Section 2 describes the system configurations and
theoretical analyses of three heterodyning schemes noted as Scheme A, Scheme B and
Scheme C including phase noise generation and signal to noise ratio (SNR) calculations.
Section 3 presents the experimental results for each of the schemes. Section 4 characterizes
the proposed schemes by simulation and quantifies the effects of variations in laser linewidths
and RIN. It also includes a comparison of the performance of RF self-homodyning over a
conventional mixer and LO based receiver for the recovery of data. Finally, Section 5 presents
the summary of this study.
The paper is organized as follows: Section 2 describes the system configurations and
theoretical analyses of three heterodyning schemes noted as Scheme A, Scheme B and
Scheme C including phase noise generation and signal to noise ratio (SNR) calculations.
Section 3 presents the experimental results for each of the schemes. Section 4 characterizes
the proposed schemes by simulation and quantifies the effects of variations in laser linewidths
and RIN. It also includes a comparison of the performance of RF self-homodyning over a
conventional mixer and LO based receiver for the recovery of data. Finally, Section 5 presents
the summary of this study.
2. System configurations and theory of operation
Figures 1(a) to 1(c) presents three transmitter configurations of the proposed RoF system in
various generation schemes named as Scheme A, B and C showing the spectra of respective
optical mm-wave signal in insets. Figure 1(d) presents RF self-homodyning scheme for phaseinsensitive recovery of data at the receiver. The electric fields of the lasers in our system can
be represented by:
E1 exp j ( 2π f1t + φ1 )
(1)
E2 exp j ( 2π f 2 t + φ2 )
(2)
Here, E1 and E2 are the peak amplitudes of the electric fields, f1 and f 2 are the optical
frequencies of two lasers such that ( f1 − f 2 ) is the desired mm-wave frequency and φ1 and φ2
are the phase variations of each of the two independent lasers. For simplicity in our analysis,
we assume E1 = E2 = 1 . The modulating signal is sinusoidal and is represented as
m cos 2π f m t ; where m is the modulation index of the single-drive Mach-Zehnder modulator
V 
(MZM) and m =  m  , Vm and Vπ represent drive-voltage and the half-wave voltage of the
 Vπ 
modulator respectively and f m is the frequency of the modulating mm-wave signal.
2.1 Scheme A
Scheme A presents a transmitter configuration where both of the optical carriers are
modulated, as shown in inset of Fig. 1(a). The modulated optical carriers after Taylor series
expansion, can be expressed as follows:
A1 ( t )
= exp j ( 2π f1t + φ1 ) 1 + jm cos ( 2π f m t ) 
 m

(3)
= exp j ( 2π f1t + φ1 ) 1 + ( exp j 2π f m t + exp− j 2π f m t ) 
 2

m


= exp j ( 2π f1t + φ1 ) + {exp j ( 2π f1t + φ1 + 2π f m t ) + exp j ( 2π f1t + φ1 − 2π f m t )}
2


A2 ( t )
= exp j ( 2π f 2 t + φ2 ) 1 + jm cos ( 2π f m t ) 
 m

(4)
= exp j ( 2π f 2 t + φ2 ) 1 + ( exp j 2π f m t + exp− j 2π f m t ) 
 2

m


= exp j ( 2π f 2 t + φ2 ) + {exp j ( 2π f 2 t + φ2 + 2π f m t ) + exp j ( 2π f 2 t + φ2 − 2π f m t )}
2


These expressions of the modulated optical carriers will change after transmission over
single-mode fiber (SMF) and can be rewritten as:
m


exp j ( 2π f1t + φ1 + ϕ0 ) + 2 exp j {2π ( f1 + f m ) t + φ1 + ϕ1 }
Af1 ( t ) = 

 + m exp j 2π ( f − f ) t + φ + ϕ

{
}
1
m
1
2
 2

(5)
m


exp j ( 2π f 2 t + φ2 + ϕ0′ ) + 2 exp j {2π ( f 2 + f m ) t + φ2 + ϕ1′}
A f2 ( t ) = 

 + m exp j 2π ( f − f ) t + φ + ϕ ′

{
}
2
m
2
2
 2

(6)
where, ϕ0 , ϕ1 , ϕ 2 and ϕ0′ , ϕ1′ , ϕ 2′ represent the phase delays of the spectral components
while propagating through the SMF. The complex conjugates of Eqs. (5) and (6) are:
m


exp− j ( 2π f1t + φ1 + ϕ0 ) + exp − j {2π ( f1 + f m ) t + φ1 + ϕ1 }

2
A* f1 ( t ) = 

 + m exp − j 2π ( f − f ) t + φ + ϕ

{ 1 m
1
2}
 2

(7)
m


exp− j ( 2π f 2 t + φ2 + ϕ0′ ) + 2 exp − j {2π ( f 2 + f m ) t + φ2 + ϕ1′}

(t ) = 
 + m exp − j 2π ( f − f ) t + φ + ϕ ′

{ 2 m
2
2}
 2

(8)
A* f 2
We assume that both of the modulated optical signals are collinearly polarized. Therefore,
after remote heterodyning at the PD, the resulted RF signal can be expressed as:
ip (t )
(
) (
)
= ℜ ×  Af1 ( t ) + Af2 ( t ) × A*f1 ( t ) + A*f2 ( t ) 
m
m
m

2
 2 + m + 2 cos {2π f m t + (ϕ1 − ϕ0 )} + 2 cos {2π f m t + (ϕ0 − ϕ 2 )} + 2 cos {2π f m t + (ϕ0′ − ϕ 2′ )}

m2
m2
 m
 + 2 cos {2π f m t + (ϕ1′ − ϕ0′ )} + 4 cos {2π .2 f m t + (ϕ1 − ϕ 2 )} + 4 cos {2π .2 f m t + (ϕ1′ − ϕ2′ )}

m2

′
′
 + cos {2π ( f1 − f 2 ) t + (φ1 − φ2 ) + (ϕ0 − ϕ 0 )} + 4 cos {2π ( f1 − f 2 ) t + (φ1 − φ2 ) + (ϕ1 − ϕ1 )}

2
m
m
cos {2π ( f1 − f 2 ) t + (φ1 − φ2 ) + (ϕ2 − ϕ 2′ )} + cos 2π ( f1 − f 2 ) − f m  t + (φ1 − φ2 ) + (ϕ0 − ϕ1′ )
= ℜ ×  +
4
2

 + m cos 2π ( f − f ) − f  t + (φ − φ ) + (ϕ − ϕ ′ ) + m cos 2π ( f − f ) + f  t + (φ − φ ) + (ϕ − ϕ ′ )
m
1
2
2
0
m
1
2
0
2
 1 2
 1 2
 2
2

2
m
m
 + cos 2π ( f − f ) + f  t + (φ − φ ) + (ϕ − ϕ ′ ) +
cos 2π ( f1 − f 2 ) + 2 f m  t + (φ1 − φ2 ) + (ϕ1 − ϕ 2′ )
m
1
2
1
0
 1 2
 2
4

2
 + m cos 2π ( f − f ) − 2 f  t + (φ − φ ) + (ϕ − ϕ ′ )
m
1
2
2
1
 1 2
 4
{
}
{
}
{
{
}
{
{
}
}
}








 (9)











where ℜ is the responsivity of the PD.
Due to the heterodyning detection process, the last nine terms in Eq. (9) clearly show the
desired mm-wave frequency at ( f1 − f 2 ) together with its first and second-order sidebands
and the uncorrelated phase (φ1 − φ2 ) noise, which can be easily separated by using a suitable
bandpass filter. Therefore, after the bandpass filtering, Eq. (9) can be written as,
ip (t )

m2
cos {2π ( f1 − f 2 ) t + (φ1 − φ2 ) + (ϕ1 − ϕ1′ )}
cos {2π ( f1 − f 2 ) t + (φ1 − φ2 ) + (ϕ0 − ϕ0′ )} +
4

2
 m
m
 + 4 cos {2π ( f1 − f 2 ) t + (φ1 − φ2 ) + (ϕ 2 − ϕ 2′ )} + 2 cos 2π ( f1 − f 2 ) − f m  t + (φ1 − φ2 ) + (ϕ0 − ϕ1′ )

m
 m
= ℜ ×  + cos 2π ( f1 − f 2 ) − f m  t + (φ1 − φ2 ) + (ϕ 2 − ϕ0′ ) + cos 2π ( f1 − f 2 ) + f m  t + (φ1 − φ2 ) + (ϕ0 − ϕ2′ )
2
2

2
 + m cos 2π  f − f + f  t + φ − φ + ϕ − ϕ ′ + m cos 2π  f − f + 2 f  t + φ − φ + ϕ − ϕ ′
( 1 2) ( 1 2)
0)
m
( 1 2 ) m  ( 1 2 ) ( 1
( 1 2 )
 2
4

2
m
+
cos 2π ( f1 − f 2 ) − 2 f m  t + (φ1 − φ2 ) + (ϕ2 − ϕ1′ )
 4
{
}
{
}
{
{
}
{
{
}
}
}


 (10)











For simplicity, if we assume that the PD’s responsivity ℜ = 1 , and fiber dispersion is
negligible (i.e. ≈ 0 ), Eq. (10) can now be written as,

m2
cos {2π ( f1 − f 2 ) t + (φ1 − φ2 )}
 cos {2π ( f1 − f 2 ) t + (φ1 − φ2 )} +
4

2
 m
m
 + 4 cos {2π ( f1 − f 2 ) t + (φ1 − φ2 ) +} + 2 cos 2π ( f1 − f 2 ) − f m  t + (φ1 − φ2 )

m
m
i p ( t ) =  + cos 2π ( f1 − f 2 ) − f m  t + (φ1 − φ2 ) + cos 2π ( f1 − f 2 ) + f m  t + (φ1 − φ2 )
2
2

2
 + m cos 2π  f − f + f  t + φ − φ + m cos 2π  f − f + 2 f  t + φ − φ
(
)
(
)
( 1 2)
1
2
m
1
2
m


( 1 2 )
 2
4

2
 + m cos 2π ( f − f ) − 2 f  t + (φ − φ )
m
1
2
 1 2
 4
{
}
{
}
{
{
}
{
{
}
}
}





 (11)








Now, the modulation index values can be generalized as M where, M will vary for
0 ≤ m ≤ 1 . Hence, Eq. (11) can be simplified as follows:
M cos  2π {( f1 − f 2 ) + nf m } t + (φ1 − φ2 ) 
(12)
Here, n is the order of sidebands generated within the range of −2 to + 2 in Eq. (11). Now,
in order to apply RF self-homodyning in data recovery process, Eq. (12) is required to be
multiplied by itself. To enable such multiplication, let us assume the multiplying terms as:
cos C = M cos  2π {( f1 − f 2 ) + Af m } t + (φ1 − φ2 ) 
cos D = M cos  2π {( f1 − f 2 ) + Bf m } t + (φ1 − φ2 )  ,
where A and B are the respective n values in the range of −2 to +2 for both cos C and
cos D . After multiplication we get,
cos C × cos D
(13)
M
=
cos [ 2π ( A − B) f m t ] + cos  2π {2 ( f1 − f 2 ) + ( A + B ) f m } t + 2 (φ1 − φ2 ) 
2
Now, let us assume, A = 1 for cos C and B = 0 for cos D and M = 1 .Therefore, Eq. (13)
can be written as,
{
}
{
}
0.5 cos ( 2π f m t ) + cos  2π {2 ( f1 − f 2 ) + f m } t + 2 (φ1 − φ2 ) 
(14)
The first term in Eq. (14) shows the baseband data generated through the RF selfhomodyning, which can be recovered by using a suitable low-pass filter (LPF) [20–23].
2.2 Scheme B
Scheme B presents a transmitter configuration where, instead of modulating both of the
optical carriers, only one carrier is modulated and the remaining carrier is transmitted as an
optical LO alongside the modulated carrier, as shown in inset of Fig. 1(b). Hence, for scheme
B, the electric fields of the modulated and the unmodulated optical carriers can be expressed
as follows:
m


A1 ( t ) = exp j ( 2π f1t + φ1 ) + {exp j ( 2π f1t + φ1 + 2π f m t ) + exp j ( 2π f1t + φ1 − 2π f m t )} (15)
2


A2 ( t ) = exp j ( 2π f 2 t + φ2 )
After transmission over the SMF, Eqs. (5) and (6) can be rewritten as
(16)
m


exp j ( 2π f1t + φ1 + ϕ0 ) + 2 exp j {2π ( f1 + f m ) t + φ1 + ϕ1 }
Af1 ( t ) = 

 + m exp j 2π ( f − f ) t + φ + ϕ

{ 1 m
1
2}
 2

(17)
Af2 ( t ) = exp j ( 2π f 2 t + φ2 + ϕ0′ )
(18)
Similar to Eq. (9), the analytic expression after the PD can be written as


m2 m
m
2
+
+ cos {2π f m t + (ϕ1 − ϕ0 )} + cos {2π f m t + (ϕ0 − ϕ 2 )}

2
2
2


2
 m

 + 4 cos {2π .2 f m t + (ϕ1 − ϕ 2 )}



i p ( t ) = ℜ ×  + cos {2π ( f1 − f 2 ) t + (φ1 − φ2 ) + (ϕ0 − ϕ0′ )}
 (19)


 + m cos 2π ( f − f ) − f  t + (φ − φ ) + (ϕ − ϕ ′ )

m
1
2
2
0
 1 2
 2



 + m cos 2π ( f − f ) + f  t + (φ − φ ) + (ϕ − ϕ ′ )

1
2
1
0
m
 1 2
 2

The last three terms in Eq. (19) clearly show the desired modulated mm-wave frequency at
( f1 − f 2 ) together with the relevant uncorrelated phase (φ1 − φ2 ) noise, as explained earlier.
As before, phase-insensitive baseband data can be recovered by repeating steps in Eqs. (10)–
(14).
2.3 Scheme C
Similar to Scheme B, Scheme C is also comprised of one modulated and one unmodulated
optical carrier. The unmodulated carrier, instead of being sent from the CS, is added at the BS
before photodetection, as shown in inset of Fig. 1(c). Therefore, in Scheme C, the
unmodulated optical carrier is free from any fiber impairments, such as dispersion. This
modifies Eq. (18) as,
{
}
{
}
ALO ( t ) = exp j [ 2π f 2 t + φ2 ]
The analytic expression after PD in this case can be written as
(20)


m2 m
m
2
+
+ cos {2π f m t + (ϕ1 − ϕ0 )} + cos {2π f m t + (ϕ0 − ϕ 2 )}

2
2
2


2
 m

 + 4 cos {2π .2 f m t + (ϕ1 − ϕ 2 )}



i p ( t ) = ℜ ×  + cos {2π ( f1 − f 2 ) t + (φ1 − φ2 ) + ϕ0 }
 (21)


 + m cos 2π ( f − f ) − f  t + (φ − φ ) + ϕ

m
1
2
2
 1 2
 2



 + m cos 2π ( f − f ) + f  t + (φ − φ ) + ϕ

m
1
2
1
 1 2
 2

The rest of the baseband recovery process will remain similar as explained in Eqs. (10)–
(14).
2.4 Phase noise due to unlocked heterodyning
Equations (9), (19) and (21) in Sections 2.1-2.3 clearly show the presence of uncorrelated
phase noise in the generated mm-wave signals, the effects of which need to be further
explored and quantified. In order to do this, for simplicity, let us assume that the generated
{
}
{
}
mm-wave intermediate frequency (IF) signal is unmodulated. Therefore, generated output
after the PD can be simplified as follows:
i p ( t ) = ℜ ( P1 + P2 ) + 2ℜ P1 .P2 cos [ 2π f IF t + φIF ]
(22)
-40
5.2 MHz
SSB phase noise (dBc/Hz)
-50
-70
-60
-75
-70
-80
-80
-85
0
2
4
6
8
10
6
x 10
-90
-100
0
2
4
6
8
Frequency offset from carrier (Hz)
10
7
x 10
Fig. 2. SSB phase noise at mm-wave IF due to uncorrelated lasers.
where, P1 and P2 represent the optical power and E1 = 2 P1 , E2 = 2 P2 , f IF = ( f1 − f 2 ) and
φIF = (φ1 − φ2 ) . The power spectral density (PSD) of Eq. (22) can then be written as,
S I ( f ) = ℜ 2 P1 P2
1 / π∆υ IF
 2( f − f IF ) 
1+ 

 ∆υ IF 
(23)
2
where, ∆υ IF is the total full width half-maximum (FWHM) linewidth at the mm-wave IF
and ∆υ IF = ∆υ1 + ∆υ2 . Here, ∆υ1 and ∆υ2 are the linewidths of the unlocked DFB lasers, as
explained before. We also assume that the PSD shown in Eq. (22) is Lorentzian laser
lineshape.
Now, the RF power of the generated IF signal can be expressed as
PIF = I p ( t ) . RL =
2
( 2ℜ
2
1
) .R
2
P1 . P2
L
(24)
where, . denotes the average and RL is the terminal impedance at the PD output.
Now, the ratio of the Eqs. (23) and (24) is the single side-band (SSB) phase noise in
dBc/Hz, which also can be termed as inverse carrier to noise ratio (CNR) [24]. Therefore,
Eqs. (23) and (24) are used to plot the SSB phase noise of the generated mm-wave IF signal at
offset frequencies up to 100 MHz, where ∆υ IF is 5.2 MHz, as shown in Fig. 2. Due to higher
linewidth contributions (5.2 MHz), SSB phase-noise is calculated to be approximately
−72dBc/Hz at 1 KHz offset. The plot used experimental specifications (described later) of the
DFB lasers and the PD for a better clarity, as summarized in Table 1.
Table 1. Specifications used for the SSB plot
(5MHz + 200 KHz) = 5.2 MHz
Total Linewidth, ∆υ IF
Responsivity,
ℜ
Terminal impedance, RL
0.6 A/W
50 Ohm
2.5 SNR due to unlocked heterodyning
In order to understand the significance of such unlocked systems on the overall SNR
performance, Eq. (24) can be simplified as follows:
(
)
2
1
1
2ℜ P1 .P2 .RL = I av2 .RL
(25)
2
2
Here, I av = 2ℜP1 is the average photo-current and is found to be 5.5 mA ( RL = 50 Ohm)
in this case.
Now, the noise PSD of the proposed link consists of noise sources such as, thermal noise,
the shot noise, RIN, amplified spontaneous emission (ASE) noise from the erbium doped fiber
amplifier (EDFA) and the converted laser phase to intensity noise. The noise from EDFA is
regarded as the additional RIN to the system [25]. Same is true for the converted phase to
intensity noise [26]. Therefore, the total noise PSD after the PD can be expressed as [27],
PIF =
1
(26)
( pshot + pRIN total )
4
Here, pth = KT is the PSD of the thermal noise; K is the Boltzmann constant and T is the
pN = pth +
temperature in Kelvin. pshot = 2qI av RL is the PSD of the shot noise where, q is the electron
charge. Finally, pRIN total = ( RIN Lasers + RIN ASE + RIN phase ) .I av2 .RL is the PSD of the total RIN of
the system, where, RIN Lasers is the RIN from lasers, RIN ASE is the RIN due to ASE noise of
EDFA and RIN phase is the phase to intensity converted noise. Hence, the total noise PSD is
1


PN = Bn  pth + ( pshot + pRIN total ) 
(27)
4


Here, Bn is the noise bandwidth. Now, assuming that, after the PD, the system goes
through a set of low-noise amplifier (LNA) and medium power amplifier (MPA), cascaded
noise figure of amplifiers according to Friis formula becomes,
 NF − 1 
NFAmp = NFLNA +  MPA 
(28)
 GLNA 
Here, NFAmp is the total noise figure of the amplifiers, NFLNA is the noise figure of the
LNA, NFMPA is the noise figure of the MPA and GLNA is the gain of the LNA.
Now, considering a directional antenna at 35.75 GHz mm-wave frequencies with
transmitting antenna gain of GTx = 15 dB and receiving antenna gain of GRx = 20 dB , Friis
Table 2. Specifications used for SNR calculations
Thermal noise after PD
Shot noise after PD
Total RIN after PD
Total gain of amplifiers
Noise figure of the amplifiers
Path loss for 10 m wireless link
Implementation loss
Thermal noise at the receiver
1.6218 × 10-10 watt
0.88 × 10−9 watt
0.4783 × 10-7 watt
37 dB
2.4 dB
83 dB
6 dB
5.5 × 10 −11 watt
free-space path loss formula for the line-of-sight link used in our calculation can be expressed
in dB as
 4.π .d . f RF 
PL = 20 log10 
(29)

c


Here, PL is the path loss due to free space, d is the wireless link distance, f RF is the
mm-wave RF carrier frequency and finally, c is the speed of light in vacuum.
Therefore, the SNR equation for the system using Eqs. (25)–(29) becomes,
1 2
I av .RL .GLNA .GMPA .GTx .GRx
SNR = 2
PN .NFAmp .PL.IL.KTBn .NFRx
(30)
Here, GMPA is the gain of the MPA, IL is the implementation loss of the antenna, KTBn is
the thermal noise at the RF receiver and NFRx is the noise figure at the receiver antenna.
Using the parameters as shown in Table 2, the SNR of the system is calculated to be 29 dB
for a wireless link distance of 10 meters for an indoor environment. Multipath-effects were
not considered for this calculation as at mm-wave frequencies, multipath scenarios do not
affect the SNR performance of the system much as far as the directional antennas are
concerned [28]. However, due to the phase noise effects generated after the PD due to
unlocked lasers, there will be SNR degradations which can be expressed
as SNRunlocked lasers = −10 log (σ θ2 ) , where, σ θ is the phase noise in rms radian/s due to unlocked
lasers and is calculated as 0.628 rms radian/s from the CNR calculations as detailed in section
2.4. SNR degradation is calculated as 4.04 dB in this case and hence, the total SNR of the
system is approximately 25 dB.
3. Experimental demonstrations
3.1 Scheme A
Figure 3 shows the setup to demonstrate the proposed concept experimentally in Scheme A.
Two free running DFB lasers operating at 1549.214 nm and 1549.50 nm are combined with a
3 dB optical coupler (OC) that gives a frequency offset of 35.75 GHz. Polarization controllers
(PC) are used for each of the light-sources before combining. Linewidths of the lasers are 5
MHz and 200 KHz respectively. RIN of both the lasers is ≤-145 dBc/Hz. An ASK formatted
155 Mbps data with a Pseudorandom bit sequence (PRBS) pattern of 231 − 1 and an amplitude
of 2 V p − p is used to drive the low-speed MZM. VΠ of the MZM was 4.5 volts. The DSB
modulated optical signal is then amplified by an EDFA followed by an optical bandpass filter
(OBPF) of 2 nm bandwidth to minimize the out-of-band ASE noise. The signal is then
transported over 26.2 km SMF to the BS. A U2T high-speed 50 GHz PIN PD with
responsivity of 0.6 A/W is used to detect the modulated optical carriers that generates 35.75
GHz mm-wave signal modulated with 155 Mbps data and a replica of the data at baseband.
The mm-wave signal is then amplified using Miteq LNA with a noise figure of 2.3 dB and
DBS Microwave MPA to provide around + 30 dB amplification. The need for band-pass
electrical filter is avoided by amplifying the PD output with band-limited amplifiers working
within 26.5 to 40 GHz that automatically suppresses the baseband replica. In order to
demonstrate the back-to-back (BTB) wireless transmission performance of the proposed
technique, the amplified signal is then divided by an Anritsu V240C 3 dB power divider
operating within DC to 65 GHz prior to self-homodyning using a Miteq mixer. An input
power of + 15 dBm is launched to the LO port of the mixer. A variable phase shifter working
within 26 to 40 GHz is used at the RF port of the mixer to enable phase-matched
multiplication. Finally, a LPF of 545 MHz is used to recover the desired baseband data.
The relevant optical and RF spectra recovered from the experiment are shown in the insets
(i) to (iv) of Fig. 3. The spectrum of modulated optical carriers after MZM is shown in inset
(i).
DFB LD 1
Amp.
P=+15 dBm
PC
IF
Mixer
MZM
OC
SMF
Att
EDFA OBPF
PC
DFB LD 2
LO
(iii)
(ii)
(i)
BERT
RF
PD
Amp.
Phase Shifter
Base Station
PRBS Data
(iv)
LPF
RF Self-homodyning
Optical Heterodyning
Customer Unit
-20
-10
(i)
-20
-30
-40
(ii)
-30
mm-wave carrier
-40
baseband
-50
-60
-70
-80
1549.0
1549.5
Wavelength (nm)
0
1550.0
mm-wave carrier
(iii)
10
(iv)
RF Power (dBm)
-20
-40
-60
-20
-40
RF Power (dBm)
1548.5
RF Power (dBm)
RF Power (dBm)
Optical Power (dBm)
Central Station
20
30
Frequency (GHz)
40
0
-20
-40
-60
-80
0
-60
200
400
600
Frequency (MHz)
800
-80
-80
0
10
20
30
Frequency (GHz)
40
0
10
20
30
Frequency (GHz)
40
Fig. 3. Experimental set up of Scheme A (top); Insets (i)-(iv) show optical and RF spectra at
different points of the setup in Scheme A.
Insets (ii) and (iii) show the respective RF spectra after PD and after band-limited amplifiers,
whilst inset (iv) shows the recovered baseband data after LPF.
Figure 4 shows the bit-error-rate (BER) curves and eye diagrams for both BTB and after
transmission over 26.2 km SMF. The error free (at a BER of 10−9) recovery of data with a
receiver sensitivity of −16.75 dBm confirms the functionality of the proposed concept
experimentally. The incurred 0.2 dB power penalty is negligible and can be attributed to
experimental errors.
-3
BTB Exp.
After SMF Exp.
-4
Log10(BER)
-5
-6
-7
-8
-9
i) BTB Exp.
ii) After SMF Exp.
-10
-18
-17
-16
-15
Received Optical Power (dBm)
Fig. 4. BER and eye diagrams for Scheme A.
-14
DFB LD 1
PRBS Data
Amp.
PC
MZM
(i)
P=+15 dBm
LO
(iii)
(ii)
SMF Att
OBPF
Phase Shifter
RF Self-homodyning
Customer Unit
(i)
RF Power (dBm)
-20
-30
-40
1549.0
1549.5
Wavelength (nm)
-60
-80
10
20
30
Frequency (GHz)
-60
-80
0
-40
0
baseband
1550.0
mm-wave carrier
(iii)
mm-wave carrier
-40
10
-20
(iv)
-40
RF Power (dBm)
1548.5
(ii)
-20
RF Power (dBm)
Optical Power (dBm)
Central Station
RF Power (dBm)
BERT
Amp.
Base Station
DFB LD 2
-20
(iv)
LPF
RF
PD
EDFA
PC
-10
IF
Mixer
OC
20
30
Frequency (GHz)
40
0
-20
-40
-60
-60
0
200
400
600
Frequency (MHz)
800
-80
40
0
10
20
Frequency (GHz)
30
40
Fig. 5. Experimental set up of Scheme B (top); Insets (i)-(iv) show optical and RF spectra at
different points of the setup in Scheme B.
3.2 Scheme B
Figure 5 shows the setup to demonstrate the proposed concept experimentally in Scheme B
with self-explanatory insets. Compared to the setup in Scheme A, the MZM is now moved
and placed immediately after the DFB laser operating at 1549.50 nm, keeping the rest of the
setup unchanged. Therefore, the modulated optical carrier in Scheme B is combined with an
unmodulated optical carrier operating at 1549.214 nm, as shown in Fig. 5. The related power
levels of the lasers are controlled to manage equal powers after coupling. As the remaining
part of the setup is common to both Schemes A and B, the duplication of the description of
the setup is therefore excluded.
-3
BTB Exp.
After SMF Exp.
Log10(BER)
-4
-5
-6
-7
-8
-9
i) BTB Exp.
ii) After SMF Exp.
-10
-18
-16
-14
-12
-10
-8
Received Optical Power (dBm)
Fig. 6. BER and eye diagrams for Scheme B.
-6
PC
(i)
SMF
PRBS Data
DFB LD 1
(iii)
(ii)
MZM
EDFA
OC
(iv)
Att
OBPF
(v)
Self
homodyning
BERT
LPF
Amp.
PD
Customer Unit
Central Station
PC
DFB LD 2
Base Station
Optical Power (dBm)
0
(i)
-10
-20
-30
-40
-20
1549.0
1549.5
Wavelength (nm)
mm-wave carrier
-40
baseband
-80
10
20
30
Frequency (GHz)
RF Power (dBm)
0
(v)
-20
-40
40
-30
1549.0
1549.5
Wavelength (nm)
1550.0
mm-wave carrier
-20
-40
-60
0
10
20
30
Frequency (GHz)
40
0
-20
-40
-60
-80
0
-60
0
-20
1548.5
(iv)
(iii)
-60
(ii)
-10
1550.0
RF Power (dBm)
RF Power (dBm)
1548.5
RF Power (dBm)
Optical Power (dBm)
0
10
200
400
600
Frequency (MHz)
800
20
30
Frequency (GHz)
40
Fig. 7. Experimental set up of Scheme C (top); Insets (i)-(v) show optical and RF spectra at
different points of the setup in Scheme C.
Insets (i) to (iv) of Fig. 5 show the relevant optical and RF spectra at different points of
thesetup, as explained earlier with Scheme A. Similar to Fig. 4, Fig. 6 shows the BER curves
and eye diagrams for Scheme B for BTB and after transmission over 26.2 km SMF. An error
free (at a BER of 10−9) recovery of data with a receiver sensitivity of −15.9 dBm confirms the
functionality of the concept in Scheme B. The incurred power penalty is 0.3 dB which can be
attributed to experimental errors.
3.3 Scheme C
Figure 7 shows the setup to demonstrate the proposed concept experimentally in Scheme C
with self-explanatory insets. Compared to the setups in Scheme A or B, one of the lightsources is moved from the CS to BS to couple the unmodulated optical carrier with the
modulated optical carrier just before photodetection. Like before, the power levels of the
lasers are controlled to manage equal powers after coupling. Insets (i) to (v) of Fig. 7 also
show the relevant optical and RF spectra at different points of the setup, as explained earlier
in Schemes A and B. Eye-diagrams and BER curves for the recovered data are presented in
Fig. 8 for the BTB configuration and after transmission over 26.2 km SMF conditions. At the
end of the SMF link, error-free data (at a BER of 10−9) was recovered at a receiver sensitivity
BTB Exp.
After SMF Exp.
-3
Log10(BER)
-4
-5
-6
-7
-8
i) BTB Exp.
-9
-18
ii) After SMF Exp.
-16
-14
-12
Received Optical Power (dBm)
-10
Fig. 8. BER and eye diagrams for Scheme C.
155 Mbps
ASK Data
Amp
OBPF
MZM
LD 1
EDFA
LPF
SMF
PD
Data
RF Selfhomodyning
LD 2
Fig. 9. Simulation model of Scheme A for system characterization.
of −15.9 dBm. Compared to the BTB condition, a negligible 0.3 dB power penalty is observed
that again can be attributed to experimental errors.
As shown earlier in Fig. 1, Scheme A described above can be considered as a perfect
presentation of OCS-DSB modulation format, whereas, Scheme B and Scheme C both
represent OSSB + C modulation format. Both of the modulation formats are spectrally
efficient and dispersion tolerant [29,30]. While both Scheme B and Scheme C have the
potential to establish wavelength reuse in uplink path as described in [13,14], Scheme C
simplifies the system by avoiding the required optical filtering arrangements. ASK
modulation in mm-wave generation and distribution is chosen as it is relatively simple to
implement and offers better CNR compared to any multilevel modulation formats, such as
quadrature amplitude modulation [4,31].
4. Simulations and characterization
In order to characterize the effects of various system parameters, the proposed concept is
simulated by using VPI Transmission maker 7.6TM, as shown in Fig. 9. Among three
heterodyning schemes, the simulation considers only Scheme A for simplicity and expects
-3
BTB Sim.
After SMF Sim.
BTB Exp.
After SMF Exp.
-4
Log10(BER)
-5
-6
-7
-8
-9
BTB Sim.
After SMF Sim.
-10
-18
-17
-16
-15
Received Optical Power (dBm)
Fig. 10. BER and eye diagrams for Scheme A.
-14
3.0
Power Penalty (dB)
2.5
2.0
1.5
1.0
0.5
0.0
-145
-140
-135
-130
-125
-120
Relative Intensity Noise (dBc/Hz)
-115
Fig. 11. Simulated power penalty versus relative intensity noise (RIN) curve.
0
Freq. spacing=32.25 GHz
Freq. spacing=34.5 GHz
Freq. spacing=35.GHz
Freq. spacing=35.75 GHz
Freq. spacing=36.5 GHz
Freq. spacing=38.25 GHz
Log10(BER)
-2
-4
-6
-8
-10
-17.6
-17.4
-17.2
-17.0
Received Optical Power (dBm)
-16.8
Fig. 12. BER curves due to frequency drifts introduced by variation in frequency separation of
both the lasers.
minimum or no difference in outcome, if other schemes are considered. The various
simulation parameters are chosen in such a way that the simulated results closely follow the
experiments. Figure 10 shows simulated as well as experimental BER curves together. They
are almost parallel on top of each other. The difference is negligible (less than 0.2 dB) and can
be ignored.
Figure 11 shows the power penalties of the system as a function of RIN of the lightsources. It shows that almost no penalty is observed if RINs are increased from −145 to −130
dBc/Hz. The situation however changes significantly if RINs are increased beyond −130
dBc/Hz; a power penalty of 1dB is observed at a RIN of −120 dBc/Hz. Therefore, RIN needs
to be monitored while deploying the system practically, although thankfully most of the
current commercially available DFB lasers exhibit a typical RIN of ≤-130 dBc/Hz [32].
BER performance of the system for frequency drifts is measured and shown in Fig. 12. It
shows that for a drift of +/− 2.5 GHz, the BER plots sit almost on top of each other, and
200 KHz
5 MHz
10 MHz
20 MHz
30 MHz
40 MHz
50 MHz
-3
Log10(BER)
-4
-5
-6
-7
-8
-9
-17.8
-17.6
-17.4
-17.2
-17.0
Received Optical Power (dBm)
Fig. 13. BER curves at various laser linewidths.
-16.8
Power Penalty (dB)
Power Penalty (dB)
1.0
0.8
0.6
0.4
Self-homodyning
LO-based
1.0
0.8
0.6
0.4
0.2
0.0
0
25
50
Linewidth (KHz)
10
15
0.2
0.0
0
5
20 25 30 35
Linewidth (MHz)
40
45
50
Fig. 14. Power penalty vs. laser linewidths for both LO-based and self-homodyned data
recovery techniques.
-3
Noise Variance 0
Noise Variance 0.1
Noise Variance 0.2
Noise Variance 0.3
-4
Log10(BER)
-5
-6
-7
-8
-9
Variance 0.2
-10
-18
-17
-16
-15
-14
Received Optical Power (dBm)
Variance 0.3
-13
-12
Fig. 15. BER curves at various noise variances in a wireless channel.
therefore, robust to the change. Similar robustness can be expected even at higher drifts, as
RF self-homodyning is simply a self-multiplication process and has little to do with any
specific frequency band.
Typically, higher linewidths of light-sources result in increased phase-noise in any
optically heterodyned system [32,33]. To quantify this effect for the proposed system,
linewidth of one light-source is increased from 200 KHz to 50 MHz, keeping second source
unchanged to 5 MHz. Respective changes in BER performance of the system is shown in Fig.
13. Almost no power penalty is observed for such huge variation of linewidths, as opposed to
conventional mixer and LO based receivers. This again can be attributed to the phaseinsensitiveness of RF self-homodyned data recovery process, as explained in Section 2.
To further comment on the significance of this data recovery mechanism in the proposed
system, the performance of RF self-homodyning is compared with conventional mixer and
LO based technique such as coherent demodulation [34,35], as shown in Fig. 14. It shows that
with a laser linewidth of up to 40 KHz, both of the data recovery mechanisms are able to
recover error-free data at a BER of 10−9. However, this changes significantly with
conventional mixer and LO based technique if the linewidths are increased above 40 KHz and
corrupts completely with linewidths as little as 50 KHz. Unlike mixer and LO based
technique, RF self-homodyning performs consistently against the various linewidths and
shows robustness to linewidths as high as 50 MHz, as shown in Fig. 14.
To get a better insight about the performance of the proposed method in wireless
environments, we simulated scheme A with additive white Gaussian noise (AWGN) channel
at the receiver. Figure 15 shows the BER performance of the system at different level of noise
variances from 0 to 0.3. It is found that at a noise variance of 0.3, error free BER at 10−9 is not
achieved due to increased noise power, which dominates the data recovery process. However,
the system performs well within a noise variance of 0.2, as shown in Fig. 15.
5. Conclusion
The demonstrated mm-wave RoF system employing optical heterodyning of uncorrelated
lasers for signal generation and RF self-homodyning for phase-insensitive data recovery
offers a simplified and cost-effective solution for high-speed wireless access networks.
Theoretical analyses and experimental demonstrations show the effectiveness of various
unlocked heterodyning schemes. Tolerance of the technique against various laser
characteristics such as RIN, linewidth and frequency drift was examined by modeling the
system in VPI platforms. For clarity, modeling also incorporated a comparison of the
technique with traditional LO based recovery process. Such systems avoid complex
phase/frequency locking, high-speed modulators, LO’s and mixing circuitry in the CS and BS
of mm-wave RoF systems, potentially enabling Gigabit wireless access at competitive prices.
Acknowledgments
Authors thank to National ICT Victoria (NICTA), Australia for funding this research. NICTA
is funded by the Australian Government as represented by the Department of Broadband,
Communications and the Digital Economy and the Australian Research Council through the
ICT Centre of Excellence program.