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Design and Analysis of a WLAN CMOS Power Amplifier Using
Multiple Gated Transistor Technique
Liu Hang, Boon Chirn Chye, Do Manh Anh and Yeo Kiat Seng
Division of Circuits and Systems, School of Electrical and Electronic Engineering,
Nanyang Technological University, Nanyang Avenue, Singapore 639798
Abstract—This paper focused on 5.2GHz highly integrated Power Amplifier for IEEE
802.11a WLAN application. Multiple gated transistor (MGTR) technique was employed to
improve linearity. A new approach for choosing the bias voltage of auxiliary transistor by
analyzing the shift of gate bias is used in the design. The simulated results of the proposed
two-stage differential power amplifier indicate 25.28 dBm P1-dB, 32.87% PAE and 26.18
dBm saturated output power with a 5.2 dB P1-dB improvement compared to conventional
single transistor amplifier.
Keywords: CMOS power amplifer, class-AB, WLAN, high linearity, MGTR.
I.
INTRODUCTION
Linearity plays an important role in Radio Frequency (RF) systems because nonlinearity
causes many problems, such as harmonic generation, gain compression, desensitization, blocking,
cross modulation and intermodulation, etc. [1]. Among various distortions, even-order distortion
caused by even-order nonlinearity can easily be reduced by adopting a differential signal
processing architecture [2]. However, it is difficult to reduce odd-order distortion. The third-order
intermodulation distortion is the most dominant nonlinearity component among odd-order
distortions. The performance measure for this nonlinearity is usually expressed by the 1-dB
compression point (P1-dB) and input/output third order interception point (IIP3/OIP3).
The orthogonal frequency division multiplexing (OFDM) based wireless local area
network (WLAN) standard uses non-constant envelope modulation. For OFDM, the linearity of
the power amplifier (PA) is a key parameter as it is closely related to power consumption and
distortion. Thus class-AB PAs are widely used in wireless transceiver designs due to their high
linearity and relatively high efficiency. This paper utilized the multiple gated transistor (MGTR)
technique in the design of this two-stage differential class-AB PA to improve the linearity,
achieving P1-dB of 25.28 dBm with 32.87% PAE, saturated output power (PSAT) of 26.18 dBm and
OIP3 of 28.86 dBm at simulation level. Comparisons with existing works are presented in section
IV. The relationship between auxiliary transistor gate bias voltage and linearity performance is
carefully studied.
This paper is organized as follows: in section II-A, the definitions of P1-dB and IP3 are
reviewed, and section II-B gives an introduction of the MGTR technique. In section II-C, large
signal effect and gate bias shift is discussed. Section III first gives the description of the proposed
PA, followed by a discussion on selection of bias voltage of auxiliary transistor with
analysis
in section III-A and a brief discussion on other design considerations in section III-B. Simulation
results and comparisons with other works are presented in section IV. Section V summarizes the
paper.
II. ILLUSTRATION OF MGTR TECHNIQUE
A. Review of P1dB and IP3
The nonlinear IDS-VGS relationship of a FET can be modeled by means of Taylor series
expansion around a particular VG bias point as follows [3]:
′
′′
⋯
(1)
where
, ′ and ′′ are first order, second order and third order trans-conductance gain that
can be represented as follows:
′
; ; !
and with
being the load impedance,
voltage of fundamental frequency
,
′′
!
; …
(2)
being the amplitude of the input voltage, the output
is given by
′′
,
(3)
It can be easily observed that the fundamental frequency gain variation is caused by the
third-order term ′′ . When ′′
0 , the gain will compress. The point where the gain is
0, the linearity is
compressed by 1 dB is called the 1-dB compression point (P1-dB). When ′′
not affected by the third order term. When ′′
0, the gain will expand, leading to the gain
waveform bending upward, causing nonlinear gain.
The output third order interception point (OIP3) and third order intermodulation (IM3),
according to [4], can be written as:
OIP
IM
′′
′′
(4)
(5)
B. Introduction to MGTR Technique
Figure 1.
Schematic showing transistors with MGTR technique
In Figure 1, an auxiliary transistor (AT) is placed in parallel to the main transistor (MT) with
VG bias. MGTR technique utilizes this AT together with MT to improve the overall linearity [5]. By
properly selecting the width (WAT) and the offset voltage (VOS) of AT, the overall ′′ can be
compensated for a small voltage range [6]. When the transistors are biased at the zero ′′ region,
the linearity is improved, as shown in Figure 2. WMT represents the width of MT, and both
transistors have 0.18µm channel length.
2.5
W AT= 250um and VOS=300mV
W MT= 1000um
2.0
compensated
1.5
2
gm'' (S/V )
1.0
region of zero gm''
0.5
0.0
-0.5
-1.0
-1.5
0.0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
1.8
2.0
VG (V)
Figure 2.
compensation
The MGTR technique has already been used in low noise amplifier (LNA) and mixer
designs [3, 6-8], where there is no obvious large signal effect and the power delivered to load is
relatively low. This technique has also been used in a few RF PAs [9-10]. In order for this technique
to be suitable delivering large amount of power, the large signal effect has to be considered. With an
exemption to Class-A PA, the conduction angle is less than 360°. The gate bias voltage is not
constant and is easily shifted out of the zero ′′ region. The overall ′′ becomes negative with the
increasing input signal and the gain is compressed. This means that MGTR technique works well
only for a small voltage range, and is generally not suitable for devices with large input signal, like
PA.
C. Illustration of Gate Bias Voltage Shift
The ideal I-V characteristic of a MOSFET consists of two regions, cut-off region with zero
drain current and linear region with constant transconductance G. Figure 3 shows an ideal
MOSFET with a sinusoidal input signal of amplitude V1 and biased at VG. The threshold voltage is
VTH and the peak drain current is IP.
Figure 3.
Operation of transistor [11]
Where
(6)
()
()
According to [12], the drain DC current
can be expressed as:
(9)
Thus, the new bias point
′
Δ
′
and the bias shift
can be written as:
(10)
(11)
In Figure 3,
is shown for illustration purpose. In actual case, this is not
necessarily true. In other words, the equations are still valid when
and
0.
V1 = 0.9 V
V1 = 0.6 V
V1 = 0.3 V
0.5
0.4
VG (V)
0.3
0.2
0.1
0.0
0.0
0.2
0.4
0.6
0.8
1.0
1.2
VG (V)
Figure 4.
Δ vs.
for different signal level
Figure 4 shows a plot of Δ vs.
with
0.4V. It can be seen that larger
leads
to larger Δ , and with same , higher
leads to smaller Δ . When
, the
transistor is always off and Δ is invalid. When
, the transistor is always on and
there is no bias shift. The amplifier enters Class-A mode of operation.
III. POWER AMPLIFIER DESIGN
The proposed Class-AB PA was designed using Globalfoundries’ 0.18µm IC technology
with 1.8V supply voltage. The proposed schematic of the PA is shown in Figure 5. The input and
output matching were achieved by C1, C2, L1. L2. C7, C8 and L7 were for output matching. L3-6 were
RF chokes to feed DC current and bias the transistor. All transistors use resistive gate bias. All
transistors have 0.18µm channel length and the widths are: gain stage transistors M1,2=600µm, MTs
MM1,M2 with WMT=1000µm and ATs MA1,A2 with WAT=800µm. Bias voltages are selected as VBias =
550mV, VG = 800mV and VOS = 380mV. DC consumption is 214mA. The output stage is carefully
designed with a study on the effect of various VOS on linearity. All simulations are performed at
5.2GHz of frequency.
VDD
L3
L5
L6
L4
R3
C1
C5
C7
MA1
MM1
M1
RFIN+
R4
L1
C3
R1
C8
MA2
C6
MM2
RFINL2
L7
R5
C4
R6
RFOUT+
C2
M2
R2
RFOUT-
Figure 5.
The schematic of proposed PA
A. Auxiliary Transistor Offset Voltage Selection
The choice of VOS of AT is as follows: with WMT fixed at 1000µm and VG fixed at 0.8V
(Class-AB mode), VOS is varied and the overall gain is analyzed. WAT can be altered to reduce or
′′
increase the value of
to achieve flat gain.
2.5
Secondary Peaks
2.0
with VOS= 800mV
1000um
800um
600um
400um
200um
1.5
2
gm'' (S/V )
1.0
0.5
with W AT = 800um
0.0
VOS=200mV
VOS=380mV
VOS=600mV
VOS=800mV
-0.5
Main Peak
-1.0
-1.5
0.0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
1.8
2.0
VG (V)
Figure 6.
The relationship between WAT, VOS and
Figure 6 is a plot of the overall ′′ with various WAT and VOS. It can be seen that the main
peak is almost unchanged as WMT is fixed. The position of the secondary peak is determined by VOS,
and the height of the secondary peak is determined by WAT. When
is small, the amplifier is
always on and the bias voltage is not changed. As
increases, the bias voltage is shifted as
illustrated in Figure 4. As a result,
′′
is changed.
VOS = 200 mV
VOS = 380 mV
VOS = 600 mV
VOS = 800 mV
8
Peak gain for
VOS = 380 mV
Gain (dB)
6
4
2
0
-2
-30
-20
-10
0
10
20
30
Output Power (dBm)
Figure 7.
Gain vs. output power for different VOS
Figure 7 shows the gain vs. output power from simulation. The VTH is approximately 0.45V.
It can be seen that for a low VOS = 200mV, the gain and the P1-dB are both improved. However, as
VOS is small, the bias of AT VG VOS is high and the significant increase in DC consumption is
undesirable. For larger VOS where
, the AT is off for small
and there is no
extra DC current. Comparing the remaining three curves, the VOS = 380mV curve is chosen as the
offset voltage used in this design, as it provides higher P1-dB with acceptable gain ripple. For VOS =
600mV, the gain expansion is significant. For VOS = 800mV, the gain compress first, and then
expand and compress again. The gain flatness is poor.
V1=1.3V
V1=1.0V
V1=0.7V
V1=0.4V
V1=0.2V
V1=0V
gm'' value
2.5
2.0
1.5
Secondary peak
shifts to the left
as V1 increases
2
gm'' (S/V )
1.0
0.5
0.0
-0.5
-1.0
-1.5
-2.0
0.7
0.8
0.9
VG (V)
Figure 8.
Shift of VG, effective VOS and
′′
variation at VOS =380mV
Figure 8 illustrates how the gain plot for VOS = 380mV in Figure 7 is generated. It shows the
bias shift of both MT and AT, and the change in overall ′′ . Bias shift in MT is observed as an
increase of the operating point VG, from 0.8V at
0 to about 0.9V at
1.3V. The bias shift
in AT is observed as an increase in VG VOS. From Figure 4 it can be seen that as AT is biased
lower than MT, the bias shift is more significant, which means Δ
and the
effective VOS is decreasing. As seen in Figure 8, the secondary peak is shifting towards left as
is shown with respect to
gm''
1.5
Peak gm'' = 1.208
, and is
1.2
IM3
1.0
0.8
0.5
0.4
Peak IM3 = 0.01625
2
gm'' (S/V )
′′
0.0
0.0
-0.5
-0.4
-1.0
-0.8
IM3 (A)
increases, indicating a decreasing effective VOS. The
redrawn with respect to
in Figure 9.
-1.2
-1.5
0.0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
V1 (V)
′′
Figure 9.
and IM3 vs.
Figure 9 shows a plot of ′′ and IM3 vs. . It can be seen that with increasing , the
varies from positive to negative, causing the overall gain to expand and then compress.
overall
It can be seen that the maximum
occurs at V1 = 200mV which can also be observed in Figure 8.
The maximum gain variation occurs at V1 = 400mV corresponding to Pout = 18.09 dBm, which is
shown in Figure 7 and Figure 12.
If the same analysis is performed on VOS = 600mV, the peak IM3 will be larger than in
Figure 9, and the gain expansion is more significant as seen in Figure 7. If this analysis is
performed on VOS = 800mV, a negative-positive-negative
can be observed, and the gain
compresses, expands and compresses again as shown in Figure 7. By analyzing the dynamics of
and the gain plot, a new design approach is established for AT’s bias voltage selection.
Optimization can be achieved by analyzing the effect of VOS and WAT on
under various .
The conclusion drawn from this analysis is that while the initial value of
with no input is
important for LNA and mixer designs, the value when large signal is applied is far more useful in
PA design. Large signal effect or gate bias shift must be taken into account for PA design when
MGTR technique is applied.
B. Other Design Considerations
The output matching network is designed for maximum power transfer. Load-pull
measurement is performed to find the optimum load impedance and the output matching network
transforms the optimum load impedance to a 100 Ω differential port.
The first gain stage is designed to provide some the voltage gain. Inter-stage matching is
achieved by ac coupling, and input matching is achieved by conjugate matching the input
impedance to 100 Ω differential source port.
IV. SIMULATION RESULTS
The PA is designed using Globalfoundries’ 0.18 µm IC technology. The simulation is
achieved using Cadence and based on the device models provided by the foundry. Capacitors are
metal-insulator-metal (MIM) type and inductors are symmetric, center-tapped with 6 layer metal.
This PA operates at 5.2GHz under 1.8V voltage supply. Figure 10 shows the results of small signal
S-parameter analysis. Good return loss can be achieved for WLAN first band (5.15-5.25 GHz).
S11
S21
S22
20
15
S-Parameter (dB)
10
5
0
-5
-10
-15
-20
-25
2.0
4.0
6.0
8.0
Frequency (GHz)
Figure 10. S-parameter of proposed PA
Figure 11 shows the output power and PAE vs. the input power. An output P1-dB of 25.28
dBm is observed at an input power of 10.5 dBm, with 32.87% PAE.
Output Power
PAE
30
40
Output P1dB=25.2796 dBm
20
10
PAE=32.87%
20
0
PAE (%)
Output Power (dBm)
30
10
-10
-20
0
-30
-40
-30
-20
-10
0
10
20
Input Power (dBm)
Figure 11. Output power and PAE vs. Input power
The gain and DC current vs. output power is shown in Figure 12. In this figure, the gain is
compared with an amplifier utilizing the MT alone with identical size and bias. It can be seen that
both the P1-dB and PSAT are improved, with 18% more DC current at 20 dBm output power. When
the device is delivering relatively small amount of power, it consumes no additional DC current.
This explains why the OIP3 performance is not improved: as the method for determining OIP3 is
based on extrapolation of first and third order terms of small input signal, where the auxiliary
transistor is off, OIP3 remains the same as single transistor amplifier.
MGTR gain
MT alone gain
MGTR current
MT alone current
16.14dB
Peak gain
0.9
16
15.8dB
0.8
15
Gain (dB)
Pout=20.07dBm, Gain=15.14dB
14
1.0
0.7
Pout=25.29dBm, Gain=14.8dB
0.6
13
18% more DC current at 20 dBm
output power with MGTR
12
0.5
0.4
11
0.3
10
0.2
-30
-20
-10
0
10
20
Output Power (dBm)
Figure 12.
Gain vs. output power
30
DC current (A)
17
The performance of the proposed PA is summarized in Table I and comparisons with recent
published works are shown in Table II. All designs are based on 0.18µm IC technology. It can be
seen that for this work, the P1-dB is closest to PSAT, which means the circuit is more linear and thus
the PAE at P1-dB is much higher.
TABLE I. SIMULATED RESULTS
Parameter
Value
Center Frequency (GHz)
5.2
Supply Voltage (V)
1.8
PSAT (dBm)
26.18
P1-dB (dBm)
25.28
PAE @ P1-dB (%)
32.87
OIP3 (dBm)
28.86
Power Gain (dB)
15.8
DC consumption (mW)
385
DC consumption at P1-dB (mW)
994
TABLE II. PERFORMANCE COMPARISONS WITH RECENT PUBLISHED PAS
Frequency
PSAT
P1-dB
OIP3
PAE
(GHz)
(dBm)
(dBm)
(dBm)
(%)
5.25
20.9
18.8
28.6
20.1
2-stage cascoded differential
2.4
[13]
5.25
19.5
16
27
32*
2-stage push-pull differential
3.3
[14]
5
24.1
21.8
N/A
13.1
3-stage differential
1.8
[15]
5
26.5
20.8
N/A
26.7*
2-stage differential
3.3
[16]
2-stage differential
1.8
This work
5.2
26.18
**
25.28
**
28.86
**
32.87
**
Design Features
Supply
Voltage (V)
Refs.
* indicates maximum PAE instead of PAE at P1-dB
** indicates simulation results instead of measured results
CONCLUSIONS
The design and analysis of a low-voltage fully-integrated CMOS Power Amplifier has
been presented. The simulation results show that it is suitable for 5.2GHz WLAN application. The
proposed design offers a power gain of 15.8 dB and P1-dB of 25.28 dBm with 32.87% PAE under
1.8V supply. It is capable of delivering high linear output power with high efficiency. By studying
V.
, a new design approach is established for selecting the
the gate bias shift and change in
auxiliary transistor bias voltage, and thus enhancing the linearity performance.
ACKNOWLEDGEMENT
We wish to acknowledge the funding support for this project from Nanyang Technological
University under the Undergraduate Research Experience on CAmpus (URECA) programme.
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Hang Liu was born in Baoding, Hebei, China, in 1988. He is a final year
undergraduate student of Nanyang Technological University and will be receiving
his bachelor’s degree in 2011.
His bachelor’s degree major is electrical and electronic engineering. His research
focuses on low power and high linearity RF front-end design.
Chirn Chye Boon received the B.E. (Hons.) (Elect.) in 2000 and Ph.D. (Elect.
Eng.) in 2004 from Nanyang Technological University (NTU), Singapore.
In 2005, he joined NTU as a Research Fellow and became an Assistant Professor in
the same year. Before that, he was with Advanced RFIC, where he worked as a
Senior Engineer. He specializes in the areas of radio frequency (RF) circuits and
systems design for Biomedical and Communications applications. He is the Programme Director
for RF and MM-wave research in the SG$50 millions research centre of excellence, VIRTUS
(NTU) since Mar. 2010. He has published more than 40 technical papers and one book in the
areas of RF and MM-wave.
Manh Anh Do received the B.Sc. degree in physics from the University of Saigon,
Ho Chi Minh, Vietnam, in 1969 and the B.E.(Hons.) degree and the Ph.D. degree in
electrical engineering from the University of Canterbury, Canterbury, New Zealand,
in 1973 and 1977, respectively. Between 1977 and 1989, he held various positions,
including as a Design Engineer, Production Manager, and Research Scientist in
New Zealand and a Senior Lecturer with the National University of Singapore, Singapore. He
joined the School of Electrical and Electronic Engineering, Nanyang Technological University
(NTU), Singapore, as a Senior Lecturer in 1989 and obtained the Associate Professorship in 1996
and Professorship in 2001. Between 1995 and 2005, he was Head of Division of Circuits and
Systems, NTU. He was the Director of Center for Integrated Circuits and Systems, NTU from
2007 to 2010. He has been a consultant for many projects in the electronics industry and was a key
consultant for the implementation of the 200-million-dollar Electronic Road Pricing project in
Singapore from 1990 to 2001. He has authored and coauthored 105 papers in leading journals and
140 papers in international conference proceedings. His current research interests include mobile
communications, RFIC design, mixed-signal circuits, and intelligent transport systems.
Prof. Do is a Chartered Engineer. He is a Fellow of the Institution of Engineering Technology
(IET). He was a Council Member of IET, U.K., from 2001 to 2004 and an Associate Editor for the
IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES in 2005 and 2006.
Kiat Seng Yeo received the B.E.(Hons.) degree in electronics and the Ph.D. degree
in electrical engineering from Nanyang Technological University (NTU),
Singapore, in 1993 and 1996, respectively. He is currently the Head of the Division
of Circuits and Systems and a Professor of electrical and electronic engineering
with NTU. He is a recognized expert in CMOS technology and low-power CMOS
IC design. He gives consulting to multinational corporations. He is the holder of 15 patents. He has
published four books (International editions) and more than 250 papers in his areas of expertise.
His research interests include device modeling, low-power circuit design, and RFIC design.
Dr. Yeo serves in the organizing and program committee of several international conferences
as the General Chair, a Co-chair, the Technical Chair, etc.