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IEEE TRANSACTIONS ON NUCLEAR SCIENCE, VOL. 51, NO. 5, OCTOBER 2004
1979
Design and Performance of 0.18-m CMOS Charge
Preamplifiers for APD-Based PET Scanners
J.-F. Pratte, S. Robert, G. De Geronimo, P. O’Connor, S. Stoll, C. M. Pepin, R. Fontaine, and R. Lecomte
Abstract—The CMOS 0.18- m technology was investigated for
two analog front-end projects: the low-power budget rat-head
mounted miniature rat conscious animal PET (RatCAP) scanner,
and the high-performance, low-noise, high-rate PET/CT application. The first VLSI prototypes consisted of 1- and 5-mW charge
sensitive preamplifiers (CSP) based on a modified cascode telescopic architecture. Characterization of the rise time, linearity,
dynamic range, equivalent noise charge (ENC), timing resolution
and energy resolution are reported and discussed. When connected to an APD-LSO detector, time resolutions of 2.49 and 1.56
ns full-width half-maximum (FWHM) were achieved by the 1- and
5-mW CSPs, respectively. Both CSPs make it possible to achieve
performance characteristics that are adequate for PET imaging.
Experimental results indicate that the CMOS 0.18- m technology
is suitable for both the low-power and the high-performance PET
front-end applications.
Index Terms—Avalanche photodiode (APD), complementary metal–oxide–semiconductor (CMOS), charge preamplifier,
computed tomography, front-end electronic, positron emission
tomography, Rat conscious animal positron emission tomography
(RatCAP).
I. INTRODUCTION
R
ECENT studies on deep sub-micron CMOS technologies
have triggered interest for applications where highly
integrated analog front-end electronics is required [1]. In
particular, the low power and high integration of the CMOS
0.18- m technology make it an attractive choice for high-resolution APD-based imaging detection systems like small animal
Positron Emission Tomography (PET) scanners.
The scaling of the power supply down to 1.8 V reduces the
available dynamic range, but the gain in signal-to-noise ratio at
a given dissipated power makes this choice attractive compared
to older technologies [2].
In this paper, the experimental results obtained with two
Charge Sensitive Preamplifiers (CSP) for PET imaging implemented in CMOS 0.18 m are reported. Two specific
applications are considered: a low-power, 1-mW CSP for
Manuscript received November 23, 2003; revised June 3, 2004. This work
was supported in part by grants from the Canadian Microelectronics Corporation
(CMC), Le Fonds québécois de la recherche sur la nature et les technologies
(FQRNT) and the Natural Science and Engineering Research Council of Canada
(NSERC). Studentship awards to S. Robert and J.-F. Pratte from the Institut
des Matériaux et Systèmes Intelligents (IMSI) of Sherbrooke and scholarships
from La Fondation Jacques et Suzanne Lagassé were greatly appreciated. This
work is also supported under a grant from the DOE Office of Biological and
Environmental Research and DOE Contract DE-AC02-98CH10886.
J.-F. Pratte, G. De Geronimo, P. O’Connor, and S. Stoll are with Brookhaven
National Laboratory, Upton, NY 11973-5000 USA (e-mail: [email protected]).
S. Robert, C. M. Pepin, R. Fontaine, and R. Lecomte are with the Université
de Sherbrooke, Sherbrooke, QC J1K 2R2, Canada.
Digital Object Identifier 10.1109/TNS.2004.836120
TABLE I
LSO/APD DETECTOR MODULE SPECIFICATIONS
the low-power budget RatCAP [3], [4], and a high-performance, 5-mW CSP for a high-resolution PET/CT scanner with
high-density multi-crystals (phoswich) detector modules [5].
II. CHARGE SENSITIVE PREAMPLIFIERS
A. Design
Initial specifications for the CSPs were based on the detectors planned for the RatCAP scanner (see Table I), which is a
Hamamatsu S8550 4 8 APD-array coupled to 2 mm 2 mm
5 mm LSO crystals [6], [7]. In this application, the APD is
DC coupled to the CSP input requiring a feedback MOSFET
that provides a pathway for the APD leakage current. The feedback MOSFET is designed to operate above threshold for better
white noise performance and in saturation for better compensation performance [8].
The targeted preamplifier gain of 3.3 mV/fC, selected to provide a maximum output swing of 100 mV, corresponds approximately to an input signal of 190 000 photoelectrons. The APDbased detector modules for the RatCAP are expected to be used
at a gain of 50, producing approximately 115 000 photoelectrons
for 511 keV photons in LSO. A higher signal is expected from
the PET/CT detectors at 511 keV, but an excellent signal-tonoise ratio must also be achieved for the detection of low-energy X-rays.
Fig. 1 shows the modified telescopic cascode topology of the
CSP. Transistor M2 is used as the main current source to produce the desired transconductance in the input transistor M1.
The CSPs were implemented in TSMC CMOS 0.18- m technology, having six metal layers and one poly layer, through the
Canadian Microelectronics Corporation.
B. Noise Optimization
The N-channel MOSFET input device of the CSP has been
optimized with respect to the technology parameters and the detector characteristics at its operating point (capacitance, leakage
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IEEE TRANSACTIONS ON NUCLEAR SCIENCE, VOL. 51, NO. 5, OCTOBER 2004
Fig. 1. CSP architecture. The transistor M2 is used as the main current source to achieve the desired transconductance.
current and gain), using the EKV transistor model [9], to minimize the equivalent noise charge (ENC) [10]
TABLE II
SHAPING FORM FACTORS
(1)
(2)
(3)
(4)
with
(5)
where
,
and
are the series, parallel and
flicker ENC contribution, the Boltzmann’s constant,
the
the form factors of
temperature in kelvin,
is the total capacitance at the
the shaping function [10],
input, the shaping function peaking time, the coefficient of
thermal noise for the input MOSFET,
the source transconthe parasitic resistance in
ductance of the input MOSFET,
the NMOS flicker
series with each transistor electrode,
the gate oxide capacitance per unit area,
noise coefficient,
and the width and length of the input NMOS,
the
the APD gain, a coefficient which
APD leakage current,
the transcondepends on the region of operation, and
ductance of the reset transistor. The size of the input device
was optimized for the derivative of a third order semi-gaussian
shaping function [11] with a peaking time of 70 ns and power
consumption of 1 and 5 mW. Table II presents the shaping form
factor for a CR-RC, CR -RC and the derivative of a 3rd order
semi-Gaussian shaping function [11].
All other transistor dimensions of the CSP are optimized for
minimum white series and
noise contribution on the overall
ENC, mainly set by the input device.
III. EXPERIMENTAL RESULTS
A test printed circuit board (PCB) developed at Brookhaven
National Laboratory (BNL) was used for characterization. It
includes an 1.8-V power and ground plane, a low-noise 1.8-V
voltage regulator and an on-board Analog Devices AD8009
buffer used for both 50 ohms adaptation with the shaping
5 at the CSP output. All
amplifiers and voltage gain of
required high voltage hardware and connectors for the BNL
and Sherbrooke APD detector modules were implemented on
the PCB. Charges were injected with a test pulse applied to an
on-board characterized 1-pF capacitor for each CSP. The chip
was directly wire bonded to the PCB.
The BNL detector module is based on the Hamamatsu S8550
4 8 APD-array coupled to a matched array of 2 mm 2 mm
5 mm LSO crystals. The Sherbrooke detector module is based
on a former PerkinElmer BGO/APD module [12] retrofitted
with 3 mm 5 mm 25 mm LSO scintillators. It has a leakage
current of 45 nA and a capacitance of 22 pF when biased at
350 V.
A. Electronic Characterization
The rise time, linearity and the dynamic range of the CSPs
were measured using a custom nuclear instrumentation module
(NIM) pulser and a Tektronix TDS 784A digital oscilloscope for
input capacitance of 0 to 22 pF. Figs. 2 and 3 present the measured linearity for the 1- and 5-mW CSPs. A gain of 2.7 mV/fC
was deduced for both prototypes. The input capacitance had no
influence on the dynamic range and the linearity.
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PRATTE et al.: DESIGN AND PERFORMANCE OF 0.18- m CMOS CHARGE PREAMPLIFIERS FOR APD-BASED PET SCANNERS
Fig. 2. Gain linearity for 1-mW CSP over operating range.
Fig. 3. Gain linearity for 5-mW CSP over operating range.
The rise time for the 5- and 1-mW CSP was 7 and 9 ns, respectively, without any capacitor added at their input (
pF). With a 10-pF capacitor at the input, a rise time of 13 and
18 ns was measured respectively.
B. ENC Measurements
Gain and noise measurements for the 1-mW CSP were performed for shaping times from 10 ns to 10 s with a custom NIM
bipolar CR -RC shaper. The custom NIM pulser was used to
inject the charges through the 1-pF capacitor. The output amplitude and peaking time of the chain was measured using a
Tektronix TDS 784A digital oscilloscope and the noise using a
Rhode & Schwartz TRUE-RMS meter. For the 5-mW CSP measurements, a combination of the Ortec 579 Timing Filter Amplifier and the Ortec 673 Spectroscopy Amplifier with unipolar
shaping were used, for the same shaping time range. A BNC
BL-2 Pulse Generator was used to inject the charges through
the 1-pF capacitor. The output amplitude and peaking time of
the chain was measured using a Tektronix TDS 784A digital
oscilloscope and the noise using a Fluke 8920A TRUE-RMS.
Figs. 4 and 5 present the measured ENC as a function of
the input capacitance for the two devices. The discontinuities
in the ENC curves in Fig. 5 are due to the use of two different
shaping amplifiers, as mentioned above. The evaluated ENC for
1981
Fig. 4. ENC measurements for the 1-mW CSP.
Fig. 5. ENC measurements for the 5-mW CSP. The discontinuities are due to
the use of two different shapers (Ortec 579 timing filter amplifier and the Ortec
673 spectroscopy amplifier) in order to cover a peaking time from 10 ns to 10 s.
the 1-mW CSP connected to the APD/LSO based detector is
reported in Fig. 6. A minimum ENC of 1117 electrons-rms at
100 ns was measured. The noise parameters of the 1-mW CSP
were fitted on that curve, using (1)–(5) and the theory exposed
is the
in [10] and [13]. They are reported in Table III where
is the input referred noise-voltage
flicker noise coefficient,
spectral density, is the series and parallel noise corner time
is the equivalent stray capacitance seen at the
constant,
CSP input node (including the 1-pF charge injection capacitor
and 0.3-pF feedback capacitor). Fig. 7 is the ENC curve for the
10-pF capacitor, used to fit the noise parameters of the 5-mW
device, also presented in Table III.
C. Timing Measurements
The coincidence timing resolution of both CSPs was measured using standard nuclear spectroscopy techniques [14].
The electronic timing resolution was measured using a
Phillips Scientific 704 leading edge discriminator (LED) connected directly at the output of the CSP without filter, providing
the stop signal to a time digitizer Phillips Scientific 7186.
A LeCroy 9210 pulser, providing a signal amplitude corresponding to 511 keV in an APD/LSO detector, was connected
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IEEE TRANSACTIONS ON NUCLEAR SCIENCE, VOL. 51, NO. 5, OCTOBER 2004
Fig. 6. ENC data fit with APD for the 1-mW CSP. A minimum ENC of 1117
electrons-rms at 100 ns was measured.
Fig. 8. Electronic timing resolution for the 1-mW CSP. The Hamamatsu
APD/LSO detector was connected at the input and biased at a gain of 50.
TABLE III
EXTRACTED NOISE PARAMETERS
Fig. 9. Electronic timing resolution for the 5-mW CSP. The Hamamatsu
APD/LSO detector was connected at the input and biased at a gain of 50.
Fig. 7. ENC data fit with 10 pF for the 5-mW CSP.
at the test input of the CSP. The trigger of the LeCroy pulser,
through another channel of the LED, was used as the start
signal in the time digitizer. The Hamamatsu APD/LSO detector
was connected at the input and biased at a gain of 50 for the
evaluation of both devices. The 1-mW CSP has an electronic
timing resolution of 0.56-ns FWHM, as reported in Fig. 8.
The 5-mW CSP achieved a slightly better electronic timing
resolution of 0.4-ns FWHM, as presented in Fig. 9.
For the detector time resolution, the CSPs were connected
to a CR-RC Ortec 474 Timing Filter Amplifier and an Ortec
934 constant fraction discriminator (CFD). The 1-mW CSP
measurement was performed with the Hamamatsu APD/LSO
detector in coincidence with a fast PMT/BaF detector, using
a Na source. For the measurement with the 5-mW CSP,
a PerkinElmer APD/LSO detector in coincidence with a
PMT/Plastic (Nuclear Enterprise NE-102A) detector and an
F source were used. Coincidence timing resolutions of
2.49-ns FWHM and 1.56-ns FWHM were obtained for the 1and 5-mW devices respectively, as shown in Figs. 10 and 11.
D. Energy Resolution
The energy resolution of the 1-mW CSP was evaluated with
the Hamamatsu detector connected to the input. The output of
the preamplifier was fed to a CR-RC Ortec 579 Timing Filter
Amplifier, with 50-ns integration and derivation time constants.
The best energy resolution was measured with the APD biased
at 380 V, which led to 17.4% FWHM, as presented in Fig. 12.
For the 5-mW device connected to the PerkinElmer detector,
using a Ortec shaper 452 with 250-ns integration and derivation
time constants, an optimum energy resolution of 11.7% which
was fairly insensitive to the APD bias was measured, as reported
in Fig. 13.
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PRATTE et al.: DESIGN AND PERFORMANCE OF 0.18- m CMOS CHARGE PREAMPLIFIERS FOR APD-BASED PET SCANNERS
Fig. 10.
Timing resolution spectrum for the 1-mW CSP.
1983
Fig. 13. Energy spectrum for the 5-mW CSP showing the electronic noise
contribution measured with a pulser.
expected. This can be explained by added parasitic capacitance
and process variation.
The 1- and 5-mW devices are linear over the designed dynamic range, suitable for the Hamamatsu detector. However, in
order to use the 5-mW device with the PerkinElmer APD/LSO
detector, which has a greater light output ( 10,000 electronhole pairs per MeV created in the APD before gain) and is used
at a gain of 100, a larger dynamic range is required for the 5-mW
CSP. Hence the gain of the 5-mW CSP shall have to be lowered
in order to allow a greater maximum input charge.
The measured rise times fit accordingly with the simulations.
The 5-mW device shows a faster rise time than the 1-mW, due
to its higher transconductance.
Fig. 11. Timing resolution spectrum for the 5-mW CSP. The time calibration
was obtained by adding an 8-ns delay to the stop channel.
Fig. 12. Energy spectrum for the 1-mW CSP connected to the Hamamatsu
APD/LSO detector biased at 380 V.
IV. DISCUSSION
A. Electronics
The two devices show excellent linearity. The 2.7 mV/fC
measured gain of the two prototypes is lower than the desired 3.3 mV/fC from the initial design specification, which
means that the actual feedback capacitors are bigger than
B. Equivalent Noise Charge
As expected, the 5-mW CSP has a lower ENC as a function
of the input capacitance compared to the 1-mW device. We notice that the measured minimum ENC is higher than expected
and shifted to a higher peaking time. For instance, the minimum ENC of the 1-mW CSP measured with the detector biased at a gain of 50 is 1117 electrons-RMS at 100 ns, instead
of 726 electrons-RMS at 70 ns as designed. This can be explained by an underestimation of the flicker noise coefficient
during the design, which resulted in a higher contribution from
the
noise, as seen by the flatness of the curves. The main
noise is the input transistor M1, operating besource of
tween weak and moderate inversion for both CSPs. Tsividis
noise
et al. [15] suggest that the provided manufacturer
models are aimed at digital design and usually do not appropriately characterize MOSFET devices operating in weak and
moderate inversion regions. Test transistors were included on
chip, and preliminary characterization of those confirms our assumption. In order to get the best noise performance, a review
of the CSPs design will be necessary, using the appropriate
noise conflicker noise coefficient. Considering the high
tribution of the N-channel MOSFET input device, the use of a
P-channel MOSFET looks appealing in theory. However, the requirement for a current return path between the input MOSFET
and the detector high voltage power supply imposes significant design constraints [16]. Recent studies of deep sub-micron
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TABLE IV
FRONT-END FOR PET IMAGING
CMOS technology indicate that the flicker noise contribution
drops as the transistor gate length is slightly increased above the
minimum dimension [17]. This aspect is currently under study
for the CMOS 0.18- m technology and, depending on the conclusion, the adoption of a longer gate length could be considered.
C. Timing Resolution
The electronic timing resolution of the 5-mW CSP is slightly
better than the one of the 1-mW device. This is consistent with
the theory, as the 5-mW device benefits from a lower noise contribution and a faster rise time due to its higher transconductance, leading to a smaller time jitter in the LED.
The coincidence timing resolution of the prototypes is in
accordance with the previous results, where the 5-mW device presents better performance. The difference between the
achieved coincidence timing resolutions is somehow bigger
than the difference between the electronic timing resolutions
because the PerkinElmer detector has a larger output signal and
better energy resolution than the Hamamatsu detector array due
to their different characteristics (ex: scintillator size, coupling,
APD gain operating point). In both cases, the achieved coincidence timing resolutions using a CFD allows the use of a small
coincidence timing window, which is particularly important
to limit the number of random coincidences in PET imaging
system.
D. Energy Resolution
In scintillator/APD-based detectors, the contribution of the
CSP to the deterioration of the energy resolution is negligible, as
it is mainly the stochastic nature of the scintillation process and
the APD excess noise which determine the energy resolution
[18]. The results obtained with the Hamamatsu-based detector
module on the 1-mW CSP and with the PerkinElmer LSO/APD
detector modules on the 5-mW CSP concur with previous measurements performed in our laboratory and elsewhere, and confirms the negligible contribution of the CSP to the energy resolution.
E. Charge Sensitive Preamplifiers for PET
Table IV compares the key performance parameters of
CMOS charge preamplifiers previously developed for PET
imaging with the 1- and 5-mW CSP investigated in this paper.
As expected, the 5-mW CSP achieves better input referred
noise-voltage and timing resolution compared to the 1-mW
CSP, due to the higher transconductance of the input transistor
M1. Still, the 1-mW device reaches nearly similar performance
at a fraction of the power required by previous CSP designs.
The 1-mW CSP fulfills the stringent power requirements of
the RatCAP where a total power budget of only 3.9 mW per
channel is allowed [3], [4].
V. CONCLUSIONS
Two charge sensitive preamplifiers were designed in a
0.18- m CMOS technology and optimized for LSO/APD-based
detectors for PET imaging. The performance of the low-power
and high-performance devices, which is comparable to that
obtained with previously developed charge sensitive preamplifiers, clearly indicates their suitability for PET front-end
systems. Finally, it has been demonstrated that the CMOS
0.18- m technology is suitable for the realization of low power,
low noise analog front-end electronics.
ACKNOWLEDGMENT
The authors would like to thank V. Radeka, J. Triolo, S.
Rescia, R. Machnowski, C. Woody, A. Kandasamy, and the
RatCAP group from BNL and D. Rouleau, R. Bernier, and L.
Hubert from Université de Sherbrooke for support, advice, and
their precious time, and J. Sondeen who provided excellent
Magic tech-file support. Finally, the authors would like to
acknowledge the assistance of the Canadian Microelectronics
Corporation for providing tools and technical support for the
chip fabrication. This work is part of a joint collaboration
between Brookhaven National Laboratory and Université de
Sherbrooke in Canada.
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PRATTE et al.: DESIGN AND PERFORMANCE OF 0.18- m CMOS CHARGE PREAMPLIFIERS FOR APD-BASED PET SCANNERS
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