Download Burst Mode Switch Mode Power Supply (SMPS) for 320W (peak) Class D

Burst Mode Switch Mode Power Supply (SMPS) for 320W (peak) Class D
Guitar Amplifer Supply.
Intro.
* * * * * * * * * * * * * * * * * * * * * * * * * * * ** * * * * * * * * * * * * *
This describes a very simple & robust type of Switch Mode Power Supply (SMPS)
which is so simple that companies whose primary technology is *not* SMPS, can
benefit from SMPS technology, without having to resort to huge payments to
Consultancies or Contract Engineers to design the power supply, and maintain &
modify the SMPS when their product is in production.
This SMPS also has very good no_load to full_load transient response, without
having typical instability issues.
* * * * * * * * * * * * * * * * * * * * * * * * * * * ** * * * * * * * * * * * * *
A basic simulation in the free LTspice has been sent with this report, to demonstrate
the basic operation.
The SMPS of this report is a Burst Mode, Two Transistor Forward Converter with
peak load of 320W.
…..Its load is a Class D Audio Amplifier for a Guitar.
…..Its input is 230VAC mains. (this could be made universal though)
…..It has no PFC section due to a clause in the regulations (discussed later)
…..No optocoupler in the feedback path.
…..No need for frequency compensated feedback loop
…..Very unlikely to become a victim of instability
…..Very good transient response.
‘Normal’ SMPS’s at this power level (320W) are called “Frequency Compensated
SMPS’s”.
A Burst Mode SMPS is *not* a “frequency compensated SMPS”. (at least not
formally speaking)
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CONTENTS:
Page
3
Contents
Basic Schematic: Burst Mode SMPS
4
Problems with SMPS development & maintenance for
companies who don’t have “SMPS Engineers”
5,6,7
Advantages of Burst Mode SMPS for Class D
Amplifier supply.
8, 9, 10, 11, 12
Disadvantages of Burst Mode SMPS for Class D
amplifier supply.
13
Opto-coupler feedback in ‘frequency compensated’
SMPS’s.
14, 15
PFC for Offline SMPS’s for Class D Guitar Amplifier
supply.
16
High Frequency Radiated Emissions
17
Operation of a Burst Mode SMPS.
18
18
18
18
Improving Burst Mode operation in Burst Mode SMPS’s
Burst Mode SMPS features.
Thermal Testing
Burst Mode SMPS operation in light load.
19
Burst Mode SMPS for Class D amplifier supplied Bass
guitar
20, 21
Perception of Burst Mode SMPS’s
22
Ripple on the primary DC Bus.
23
Improving EMC for a Burst Mode SMPS
23
24, 25
Examples of high power Burst Mode SMPS’s
Two Transistor Forward converter (an SMPS).
26
High-side FET Gate drive for the Two Transistor
Forward Converter
27
Basic Schematic: Burst Mode SMPS
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Basic Schematic: Burst Mode SMPS
This schematic can be better seen by opening the supplied LTspice simulation.
(LTspice is a free simulator fromlinear.com)
This schematic is also shown in “Landscape” on the last page of this report.
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Problems with SMPS development & maintenance for companies who don’t
have “SMPS Engineers”
As you know, components within an SMPS often go obselete and need replacing. If
such a component change affects eg the feedback loop (which is pretty likely), then
this could mean an expensive payment to a Design Consultancy so that they can
replace the component and assure feedback loop stability of the new, ‘updated’
SMPS.
Obselescence is one reason why you may need to change a component in an SMPS,
but other reasons for component changes exist, eg, the SMPS may need to be made
smaller, or a certain component may double in price, so may need replacing with an
alternative.
Also, supposing you have a specification change such that the power level of the
SMPS needs changing, ..its less than ideal if you need to go to a Consultancy and
pay them a huge sum of money to design the new, ‘updated’ SMPS.
What is needed is a robust SMPS that can be designed, maintained & modified by
the company who wishes to use the SMPS in their product.
The simple SMPS described here does not require “frequency compensation” for the
sake of feedback loop analysis. This is a relief for companies who don’t employ
permanent “SMPS Engineers”, since SMPS stability is often a great fear for these
companies. Also, with this Burst Mode SMPS, a Class D amplifier with a different
amount of capacitance added at its input can be used without the fear of deleterious
output voltage oscillations.
The SMPS described here has advantageous properties for high power Guitar
Amplifiers in Class D. This is because Class D amplifiers draw power in
“Pulses”….they often go from no_load to full_load and back repeatedly. Also, the
average power drawn by a Class D guitar amplifier, is extremely low. A Class D
amplifier does not draw its peak power continuously. If it did, then this Burst Mode
power supply would not be recommended, because the Burst Mode operation means
that the RMS current levels are higher than in a typical ‘frequency compensated’
SMPS.
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Advantages of Burst Mode SMPS for Class D Amplifier supply.
*…Simple:
Anyone with a modicum of electronics knowledge, should be able (with the help of
this document and the simulation) to design, modify and maintain a Burst Mode
SMPS design.
*…Quick to develop:
*…Easy & cheap to Maintain & Modify:
*…Does not need “frequency compensation” in the feedback loop:
This can be a good thing concerning SMPS development for companies who don’t
wish to pay large sums to SMPS consultancies or Contract SMPS Engineers. This is
because the feedback loop can be one of those things that appears ok in the lab during
prototype transient response testing, however, if not enough gain and phase margin
exists, then the SMPS may go unstable when with the customer.
A total and thorough mathematical analysis of the feedback loop is fairly extensive.
Also, every time a component of the loop changes, then it needs re-working.
Some of the components have dreadfully poor tolerances, such as the optocoupler, as
well as ESR values of various capacitors.
This is something that you can well do without if your prime concern is the amplifier
(load) and you haven’t got bags of time and money for working with the power
supply.
Frequency analysers do exist to test the feedback loop of ‘frequency compensated’
SMPS’s for phase and gain margin. However, they only test the particular prototype
power supply which you have on the bench in front of you…what about the power
supply that operates after a year or two with the customer and has different batchs of
components inside it and an optocoupler with different characteristics due to the loose
tolerance on otpocoupler parameters. You can mitigate this problem by setting a high
gan and phase margin, but it usually comes at the expense of transient response.
*…Very good absolutely_no_load to absolutely_full_load transient response.
(Explained later in this document.)
*…Virtually no output voltage overshoot. (due to fast comparator feedback via
digital isolator)
*…Not affected by adding capacitance to its output (eg a new amplifier with a bigger
input capacitance could be used )….adding capacitance to a Burst Mode SMPS output
always improves the output voltage ripple…..In a ‘frequency compensated’ SMPS,
adding more capacitance to the output, can cause big problems as per pages 126-133
of the book, “Power Supply Design, Volume 1:Control “ by Dr Raymond Ridley.
*…Uses simple, cheap feedback without needing an optocoupler. (see opto section).
Most offline, ‘frequency compensated’ SMPS’s use an optocoupler to feed the error
voltage across the isolation boundary back to the primary side controller.
5
Alternatively a high frequency transformer can be used, though this is more
expensive.
There is no error voltage as such with a Burst Mode SMPS, so there is only need to
feed the comparator signal back across the isolation boundary, and this can be done
with a cheap digital isolator, such as SI8710 by silabs.com.
Optocouplers as described in previous reports sent to Jesmond , present certain
challenges to the SMPS designer.
*….Transient response testing alone, will suffice to test the “feedback loop” of this
Burst Mode SMPS, -you will not need to do Bode Plots with a Frequency analyser
such as AP300 by Ridleyengineering.com.
If you do a no_load to full_load transient test on this Burst Mode SMPS, then the
delta in the Vout that results will be the worst case delta in Vout that you ever get.
(ensure to do this at the lowest Vin)
*…No slope compensation needed: This is also a bigger advantage than at first sight,
because when slope compensation is added, the primary source current sense resistor
usually needs downsizing in order to make “room” for the added ramp….such
downsizing of the sense resistor denigrates the short circuit protection.
Downsizing of the source sense resistor like this is not necessary in a Burst Mode
SMPS.
*…Can more easily meet Standby Power regulations.
In ‘frequency compensated’ SMPS’s, an optocoupler is usually used for feedback.
When in no_load, the current in the optocoupler diode is maximum.
Since the CTR variation of an optocoupler is very wide over age, temperature and
batch, one always needs to allow “room” for the maximum likely needed opto-diode
current to be able to flow in the opto-diode, with the worst case optocoupler CTR.
This tends to mean that you have a significant amount of current flowing in the optodiode when the product is on “standby”. –So standby dissipation can be unwantedly
high. This problem can be worsened with the high output voltages prevalent in some
Class D amplifier power supplys.
Even if one makes the opto-diode current maximum as low as possible, having low
current in the opto-diode tends to mean its more challenging to achieve a fast transient
response.
The Burst Mode SMPS needs no optocoupler, and so this problem doesn’t exist so
much. All that would be needed would be a mini regulator for supplying the
secondary side feedback comparator and digital isolator.
*….Transient response:
The Burst Mode SMPS can relatively easily have a fast transient response, without
having some of the disadvantages of ‘frequency compensated’ SMPS’s that are
made with a fast transient response.
If a frequency compensated smps has a fast transient response, then one must be
very careful that it is not on the verge of instability….as it could go unstable with a
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new batch of certain electronics components, or tolerances in loosely toleranced
components such as optocouplers.
*…No “Low power burst mode” problems:
When a frequency compensated SMPS goes into no_load, it must reduce its duty
cycle to virtually nothing, or else the output voltage will eventually get badly
overvltaged. This is because “frequency compensated” SMPS’s tend to always switch
the FET ON in every oscillator cycle…..they then remain on for at least the duration
of the “dead time” in the controller. This dead time prevents premature switchoff of
the switching FET during heavy load conditions, where noise at the beginning of a
switching pulse can unfortunately prematurely curtail the FET on time.
Thus to mitigate this in light_load, or no_load, frequency compensated SMPS’s tend
to go into “low power burst mode” or “skip cycle mode”.
In this condition, it can be a problem if there is a sudden onset of full_load, since the
controller often takes a relatively long time to exit the “low power burst mode”…..this
will mean that one sees a larger than usual voltage undershoot at the output voltage
rail if a sudden no_load to full_load transient occurs.
Of course, this wouldnot be a problem for a burst modeSMPS. If a burst mode smps
goes into no_load, then it just doesn’t switch the FET on, because the oscillator pulses
get disabled by the feedback comparator.
Another point about burst mode smps’s is that they have no error amplifier. This has
certain advantages relating to no_load to full_load transients. When a frequency
compensated smps goes into no_load, its error amplifier (due to the action of the
integrator) tends to ‘rail’ its output to the lower supply rail. What this means is that if
a sudden maximum_load transient occurs, then the error amplifier is at first ‘railed’,
and takes a relatively long time to raise its output enough so as to serve the transient
by increasing the FET swictch duty cycle…….thus the voltag eundershoot at the
output voltage rail will be relatively severe.
This problem does not exist in burst mode SMPS’s. A burst mode SMPS is always
virtually immediately ready to service a load transient of any size up to peak load,
even from a no_load starting point.
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Disadvantages of Burst Mode SMPS for Class D amplifier supply.
Before embarking on a Burst Mode SMPS, you would first need to be sure that you
can mitigate the following points….
EMC
In theory, the Burst Mode control will tend to mean that the draw of power will be
somewhat less smooth than in a frequency compensated SMPS.
However, it is the ‘smoothness’ of the frequency compensated SMPS which tends to
make its transient response worse than a Burst Mode type of SMPS.
-Having said that the Burst Mode SMPS is “less smooth” in its draw of power from
its input, it is a fact that in a no_load to full_load transient, the output inductor (this is
a 2TFC converter) is ‘de-fluxed’ at first (no current in it), and since the duty cycle is
limited, the current in the output inductor takes a few cycles to get up to maximum, so
you do actually get a slight degree of “soft start” in this respect. If you view the
simulation, and look at the no_load to full_load transient response, you can actually
see the output inductor current building up to maximum over a few switching cycles.
–so the burst mode is not entirely ‘un-smooth’.
Another point is that it’s a fact that the draw of power in ANY SMPS will be
somewhat un-smooth for Class D amplifier supply, simply because the class D
amplifier is not a smooth load….the Class D amplifer, especially for guitar, is literally
‘banging’ up and down from full_load to no_load continuously, with long gaps in
between the ‘pulsey’ loading, and very low average power.
‘Adjusted Burst Mode’:
In order to mitigate the lack of ‘smoothness’ in a burst mode SMPS, you could make
an adjusment to the burst of pulses by producing the gate drive pulses with a
microcontroller…..then whenever the feedback comparator “called” for power, you
could arrange for the first three (say) pulses to be shorter than maximum….ie
supposing that the maximum duty cycle is a FET gate pulse of 4us…….then you
could make the first gate pulse of a burst be 1us, the second be 2us, the third be 3us,
and then the fourth and remaining pulses be 4us. (you can adjust this “soft-start”
regime as you wish…noting that the undershoot in Vout will be worse, the ‘softer’
you start.)
Of course, if you supply the FET gate pulses with a microcontroller, then you are
free to easily supply a ‘soft-start’ when the power supply is switched on….and then
stop this when the power supply has got up near regulation level.
Another point about Burst Mode SMPS’s, is that they operate by bursts (groups) of
FET gate driver pulses………even if the switching frequency is an easily filterable
100KHz, the “burst frequency” can be very low….say in the audible range. If its
required to filter this out of the input, then it can be more difficult because its lower
frequency…..however, as discussed, the power drawn from the Class D Amplifier is
going to be “bursty” anyway, and this bursting will be in the audible range, so the
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problem of low frequency noise pulses being drawn from the input is going to be a
problem even with a “ frequency compensated” SMPS.
Also, I believe class D amplifier SMPS’s are mainly tested on 1/8th power, so the
problem gets much smaller…..and especially so if you have used the “adjusted burst
mode” method described elsewhere in this report.
Another point is that the Burst Mode SMPS suggested here does not contain a PFC
stage, so it has a large electrolytic capacitor after the mains diode bridge…so at least
they are going to do a good job of filtering out the high frequency switching noise
from the mains.
Regarding EMC and PFC in SMPS, since guitar amplifiers are such low average
power, and relatively low in sales volume, it may be possible to get a “clause” in the
EMC and PFC regulations.
I actually know of one extremely high volume domestic product which uses a mains
transformer, since they get an exemption from standby power and efficiency
regulations.
Larger ferrites:
The ferrites will tend to be larger than an equivalent power , frequency compensated
SMPS.
However, for guitar amplifiers, when good transient response is needed, frequency
compensated SMPS’s are sometimes made with peak power levels well above that
required, and in such cases the burst mode ferrites mightn’t be bigger than the
frequency compensated SMPS ferrites.
Larger RMS & Peak currents
The currents aren’t smooth and continuous in burst mode , so burst mode RMS
currents within the SMPS are likely to be higher than for a frequency compensated
SMPS….however, since the load is a class D amplifier, this isn’t such the big factor
that it would be if the load were constant.
…after all, even a frequency compensated SMPS is going to be operating in a
‘pulsey’ kind of a way when supplying such a ‘pulsey’ load as a Class D Audio
amplifier….because it’s a massively ‘pulsey’ load.
..this is why I have suggested the burst mode smps.
Audible noise:
As you know, buzzing noises in the transformers and inductors of an SMPS for a
audio amplifier are unwelcome.
Since Burst Mode is somewhat less ‘smooth’ in its power draw, and since the burst
frequency can be in the audible range, then audible buzz may possibly be heard.
It can be mitigated somewhat by varnishing or potting the inductive components.
Also,the buzzing would obviously be louder at higher power loading, and since the
guitar noise itself would be loud, then obviously the sound of the buzzing would be
drowned out.
Buzzing is sometimes criticised heavily in SMPS’s…however, the SMPS that
supplies the following LED warning light buzzes significantly, and yet this
“VigiLED” warning light has given superbly reliable operation at a great price to
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customers for over a decade. (it is the warning light seen at the back of dustbin
lorries)……
http://www.visionalert.co.uk/products/productdetails.aspx?id=10853&catid=183&tabi
d=3&menuId=
(Part number ED0001R)
As you can imagine, a dustbin lorry’s noise drowns out the buzzing of the VigiLED.
(and in vigiled’s case, the extra noise, is, if anything, an advantage, as it provides an
audible warning as well as a flashing warning)
Anyway, if audible noise was a problem, there is the opportunity to ‘smoothen’ the
bursts of pulses with the “adjusted burst mode” method described elsewhere in this
report.
In truth, if buzzing occurred, I believe it would be drowned out by the guitar noise.
Ripple in the output voltage.
If a Burst Mode SMPS is left on constant load which is near max load, then the
output voltage will feature a ripple. This ripple is due to the SMPS bursting on and
off. The ripple in the output voltage is to an extent , necessary, to allow the control of
the Burst Mode SMPS.
Note that the amount of ripple in Vout for a Burst Mode SMPS, is likely to be pretty
much the same in light_load as it is when on max load.
This is because the Burst Mode SMPS “reads” the ripple via the divider which feeds
into the hysteretic feedback comparator, and the ripple kind of controls the SMPS.
Sometimes these Burst Mode SMPS’s are actually known as “hysteretic” SMPS’s.
What the user must do, is select the output capacitor, hysteretic resistor, peak FET
current, maximum duty cycle and a few other factors, in order to get this ripple in the
allowable range. For a SMPS with an output of 80V and 4A maximum load, this
ripple can be easily made below 0.5%.
You can easily use the simulation supplied to get a feel of the Burst Mode operation.
Even though the output voltage ripple of a Burst Mode SMPS may be perceived as a
disadvantage…….when you think about it, it’s not as relevant as it would be with
constant power loads….you see, with a Class D amplifier, the Class D amplifier is
constantly ‘banging’ up and down from no_load to full_load……therefore, with a
“frequency compensated” SMPS, it is likely to constantly be ‘suffering’ from the
effects, and will be undershooting to voltages that are possibly well lower than the
ripple of a Burst Mode SMPS.
Also, its worth noting that for an almost instantaneous no_load to full_load transient,
no matter what type of SMPS is used, the initial sudden fall in Vout will be due to the
ESR of the output capacitor, and will not depend on whether a Burst Mode or a
“frequency compensated” SMPS is used.
However, the Burst Mode SMPS, will , generally be able to respond quicker than the
“frequency compensated” SMPS and thus give less undershoot in Vout, and indeed,
present virtually no overshoot whatsoever.
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Immediately following a no_load to full_load transient, a “frequency compensated”
SMPS will likely have its error amplifier’s output being railed to the low rail, and it
will likely take a relatively long time to rise up and regulate the Vout back to
regulation value, thus meaning that more significant undershoot can occur in Vout.
If, as in many of todays “frequency compensated” PWM control IC’s, the IC is in
“Low power burst mode”, (due to light loading) then it may take significantly longer
for the PWM controller to get fully “compus mentus”.
As discussed, the Burst Mode SMPS won’t suffer these problems.
In effect, a Burst Mode SMPS relishes ‘bursting’ loads such as class D
amplifiers…..this is because a Burst Mode SMPS is controlled by a comparator,
which can switch fairly immediately and thus command power to be sent through the
SMPS to its output….also, since the comparator can immediately command the FET
drive to cease, one suffers virtually no overshoot with a Burst Mode SMPS.
The output voltage ripple of a Burst Mode SMPS, will actually be no worse after a
no_load to full_load transient , than when in constant full load, or even constant
light_load.
Noise:
The Burst Mode SMPS has a comparator in the feedback loop, so obviously all the
noise precautions regarding (fast) comparators need taking.
In the Burst Mode SMPS of this document, there is hysteresis on the feedback
comparator. Also, there is a latch on the primary side, such that if the comparator
switchs erroneously at all, then the switching FET will not switch on and off
repeatedly within a switching cycle. The latch ensures that the switching FET only
comes on once per switching period.
Such a latch occurs also in most “frequency compensated” SMPS’s, for the same
purpose.
Of course, “frequency compensated” SMPS’s also contain comparators in the
feedback loop, ie, the “PWM comparator” (this receives the output of the error
amplifier), however, the output of the error amplifier is deemed to be less noisy than
the divider voltage of the feedback comparator of a Burst Mode SMPS.
However, in a “frequency compensated” SMPS, the very noisy primary current sense
resistor voltage is fed into an input of the PWM comparator, so really there is the
potential for big noise problems with comparators even in “frequency compensated”
SMPS’s. In ‘frequency compensated’ SMPS’s, the primary current sense resistor
voltage is often filtered to get rid of the high frequency noise. Alternatively, the PWM
controller may feature a dead-time.
Of course, in a Burst Mode SMPS, the secondary side feedback comparator input may
have a small RC filter connected to it, to reduce noise problems.
As dicussed, with this Burst Mode SMPS, there is mitigation at hand by way of
hysteresis, very small amounts of RC noise filtering and the latch as discussed.
As always, good PCB layout can mitigate noise issues. (Tight current loops etc, being
aware of which traces carry high rise-time switching currents, etc)
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Considering Noise, its worth noting that Current Mode “frequency compensated”
SMPS’s often operate noisily when on no_load or light_load. This is because the
current sense ramp signal is very low indeed when a current mode “frequency
compensated” SMPS is on light_load, and often the current sense signal is not high
enough to get above the noise floor, so the SMPS can operate noisily….so its not just
Burst Mode SMPS’s that suffer from noise by any means.
Indeed, some Current Mode “frequency compensated” SMPS’s have audible noise in
the wound ferrite components due to this problem at light load.
(Voltage Mode “frequency compensated” SMPS’s mitigate the noise issue in light
load, since they use a “synthetic” ramp, from circuitry inside the PWM controller
chip, and this ramp is still high peak-to-peak even in light_load.
However, Voltage Mode “frequency compensated” SMPS’s provide other challenges
such as double poles due to LC resonances in the SMPS. If not dealt with correctly,
then these can give rise to instability.)
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Opto-coupler feedback in ‘frequency compensated’ SMPS’s.
Note, the Burst mode SMPS suggested here contains *no* optocoupler…….the
feedback signal is the output of a comparator with hysteresis, and is sent to the
primary by a cheap digital isolator.
As you know the opto-coupler is often used in ‘frequency compensated’ , isolated
SMPS’s. It usually passes the error signal to the primary side. This job can also be
done by a high frequency transformer, but that is more expensive than an opto.
The opto-coupler’s base_collector capacitance, or rather the pole resulting from it,
and the opto’s CTR (Current Transfer Ratio), are factors which feature in the
feedback loop analysis.
Unfortunately, these parameters are extremely variable, and have poor tolerance
between batchs. Also, CTR varies very widely with component age and temperature.
It is not possible even to get a spread of possible values for the base_collector
capacitance variation between batchs of optocouplers.
This is far less than ideal, -how are you going to get an SMPS with sufficient gain
and phase margin when you don’t even know what some of the parameters are for its
calculation?
You can mitigate the low opto pole frequency problem by casc0ding the
opto…however, some controllers don’t offer the voltage headroom to easily facilitate
casc0ding. (please excuse, my computer is unwantedly turning ‘cascode’ into
‘cascade’)
Also, if the opto is casc0ded, you still need ‘frequency compensation’.
Another point about optocouplers in the feedback loop of a ‘frequency compensated’
SMPS is that really one needs to also consider the dynamic impedance of the diode
inside the opto coupler in the feedback loop calculation…this is especially relevant in
Low Standby Power SMPS’s where the opto-diode current may be very low.
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PFC for Offline SMPS’s for Class D Guitar Amplifier supply.
PFC here refers to an active Power Factor Correction stage.
The situation of PFC in this case has been discussed thoroughly in previous reports
sent to Jesmond.
In a survey of many guitar amplifier companies including all the biggest ones, not
one single guitar amplifier company said that they used PFC in any of their
amplifiers….even those labelled as up to 600W.
I am aware of a “350W” guitar amplifier which is billed as 230VAC input and
definitely contains no PFC, and is sold to the UK market at the very least. I have seen
inside this guitar amplifier, and it contains no PFC, and has 100uF of Electrolytic
capacitance appearing directly after the mains rectifier bridge. The enclosure of this
guitar amp is labelled “350W” but I am told it is 150W “at the load”.
If you wish to add a PFC stage , then feel free. The Burst Mode SMPS can work off
the output of a PFC stage, and indeed, the primary current in the Burst Mode SMPS
would be lower on average.
Having a PFC stage for a Class D amplifier though, is not as useful as PFC in a
320W Constant load power SMPS.
Even if a 320W Class D guitar amplifier were powered via a PFC stage, the mains
current would not likely resemble a sine wave, due to the ‘pulse’ nature of the Class D
Guitar Amplifier loading.
Most of the time, the PFC stage would be on no_load, because the Class D Amplifier
would not be drawing power most of the time, because the guitarist is not
continuously plucking strings.
I t would be very surprising to see the British Government encouraging regulations
which put generally well-intentioned British companies out of business when no real
harm was being done. For one thing, they’ll loose the Tax Revenue on Sales.
It could well be worth applying to your local MP about getting a clause in the
regulations regarding PFC.
Also, regarding EMC, again it could be worth applying for a clause if the SMPS was
just missing out on passing the regulations.
There are just too few guitar amplifiers around for it to be a real issue. Also, those
guitar amplifiers that are out there with customers, consume on average throughout
their lifetime, a miniscule amount of power.
I am already aware of at least one high volume domestic product company (not audio
related) who use a mains transformer to power a 10W (20W peak) product.
As you can imagine, they cannot pass Standby Regulations, and owing to an
enormous storage capacitor, they cannot pass regulations concerning mains harmonic
emissions. However, they are unpenalised for this.
I am aware of a different company who label their (non audio) product at a different
power level than it really is, so as to benefit from easier regulatory requirements…its
quite legal.
One only has to type “LED light bulb tear down danger” into youtube to see a host of
LED lightbulbs sold in the West, which contain far less than the necessary regulatory
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mains voltage clearance. Again, these products don’t get recalled, and again no
penalty…
http://www.youtube.com/watch?v=YLlpoZaBx-g
http://www.youtube.com/watch?v=keaE7QTKTYE
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High Frequency Radiated Emissions
In any SMPS, there are regulatory limits imposed for this.
However, most of the potential emissions problems relating to Burst Mode SMPS’s
are at lower frequencies, ie the overall frequencies of the burst “packets” of pulses.
These lower frequencies are certainly not candidates for radiated emissions. So
therefore one can rest assured that the Federal Radiocommunications Agancy is not
likely to come knocking on the door.
As with all SMPS’s, high frequency radiated emissions problems are countered with
tight PCB layout (small area power switching current loops), and adequate capacitive
decoupling and filtering. (there is also things like common mode input filters, screens
in the switch mode transformer, and keeping high dv/dt traces away from eg metal
such as enclosure or heatinks)
In this Burst Mode power supply’s case, it has no PFC, so has a big electrolytic
capacitor after the mains rectifier bridge, and just before the power switch loop, so
therefore this is going to do a good job of filtering out the high frequency switching
noise form the mains input. –specially when combined with some ceramic capacitance
local to the power switch loop.
One would need to look into lower frequency disturbances, though with the Class D
amplifier being tested on 1/8th power, the problem obviously reduces.
I would be surprised if a SMPS for a Class D amplifier was tested on continuous
maximum (peak) load, since it would not be operated like that.
Indeed, circuitry could be added, such that if , for whatever reason, the SMPS was put
on continuous maximum load, then it could shut down, or just go into standby, or
timed re-try etc.
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Operation of a Burst Mode SMPS.
The basic operation is as seen in the basic simulation which was sent with this report.
The simulation runs in LTspice which is the free download from linear.com
The operation is so simple, that running the simulation and looking at the voltages and
currents will quickly lead you to understand it.
There is no frequency compensated error amplifier in the Burst Mode SMPS.
There are the following basic parts to the Burst Mode SMPS controller………
*…Potential divider: (divides down the output voltage to be regulated)
*…Feedback Regulation comparator:
*…Voltage reference: (put into one port of the regulation comparaotr…determines
output voltage level)
*…Hysteresis resistor: (gives hysteresis to comparator switching, and is variable , up
to user)
*…Digital isolator: (transfers the comparator output signal across the isolation
barrier)
*…Fixed Duty Cycle Oscillator: (on primary side, this is at high frequency, say
100KHz ,and is for driving the FET)
*…Latch: (this ensures that if the feedback comparator switches twice within a
period, then the FET will not be brought back on in that same period, -the Latch is
reset at the start of every oscillator period).
*…Primary Current Limit Comparator: (this makes sure that the primary current
doesn’t go above the limit, and so the transformer doesn’t saturate, -as soon as the
current limit comparator trips, the FET is switched immediately OFF.)
*…Logic: (this commands the signals from the oscillator, the feedback comparator,
and the current limit comparator such that the FET is driven properly.)
All of the “disadvantages” of Burst Mode SMPS’s can to an extent be mitigated by
the finer details of operation which you can control.
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Improving Burst Mode operation in Burst Mode SMPS’s
If a cheap microcontroller is employed, then many of the disadvantages of Burst
Mode can be mitigated to an extent.
For example, for a 320W offline SMPS with no PFC, the primary DC Bus voltage
ripples up and down at 100Hz when loaded. When on no load, the DC bus voltage is
highest…thus the micro could read the bus voltage and adjust the maximum duty
cycle to suit…ie reduce the maximum duty cycle when the DC bus voltage is
high…..the micro would not have to do this terribly quickly, as the DC bus voltage is
relatively slow moving.
This could be done with a couple of duty cycles to suit various DC Bus voltage
ranges.
Burst Mode SMPS features.
Burst Mode SMPS’s are very common indeed, but almost exclusively at the low
power levels.
Of course it is to be remembered (and is highly relevant) that a Class D guitar
amplifier, even if its peak power is very high, it has a very low average power draw.
From a worldwide power consumption perspective, the energy consumed by Class D
guitar amplifiers is of course, pretty much nothing.
Burst Mode type SMPS’s are sometimes known as “ON/OFF controlled SMPS’s”, or
“hysteretic SMPS’s”.
Thermal Testing
Like any SMPS that does 320W, even if only for short intervals, thermal testing is
mandatory.
Burst Mode SMPS operation in light load.
As you know, if the burst “packets” of switching pulses were sent from a
microcontroller, then the micro could (optionally) commence the burst packet with a
low duty cycle, and then gradually increase up to maximum duty cycle. This would be
advantageous at lighter loads, because the burst packets would get shorter in duration
at light load, and so the Burst Mode SMPS would operate with generally lower peak
currents. This would make the burst mode operation less unsmooth.
However, anything other than starting a burst ‘packet’ with maximum duty cycle
pulses would mean an increase of Vout undershoot after transient loading.
One would have to verify what “soft-start” to the packet is acceptable from an
undershoot perspective, and a peak current perspective.
Ultimately, as you know, we are talking about what is the effect on the sound of the
Class D guitar amplifier.
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Burst Mode SMPS for Class D amplifier supplied Bass guitar
As you know, the first harmonic of bass guitar notes are 30Hz to 390Hz.
It is accepted that Bass Guitar amplifiers need to be of higher power.
Since Bass notes are lower in frequency, they spend relatively longer time intervals at
their peaks/troughs.
The half period at 30Hz is about 16.7ms…….so every 16.7ms, there’s obviously a
good few milliseconds where the Class D amplifier is going to be at near maximum
power.
However, it’s still less than half peak power overall, and when you consider that notes
aren’t constantly being plucked , it’s even less power, so really, the disadvantages of
burst mode are not as big as they are said to be in other power supply applications
which involve constant high power loads.
I am not sure how the situation changes with distorted guitar music, so in order to
optimise the attached simulation of a Burst Mode SMPS, I would need to see a few
scope shots of current draw into a class D amplifier supplying a bass guitar.
The great servicing of no_load to full_load transients by Burst Mode SMPS’s is great
news for Class D amplifiers supplying guitars. This is because it’s easy to avoid
considerable undershoot in the amplifier supply rail when a no_load to full_load
transient occurs.
In the sound, one presumes that this means a brighter, more “punchy” sound from the
Class D amplifier.(?)
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Perception of Burst Mode SMPS’s
Using an offline Burst Mode SMPS at 320W is likely to provoke “knee-jerk”
responses from many people in the power supply world.
Burst Mode power supplies are pretty much considered to be a “dummy’s” power
supply.
-And frowned upon for high power applications.
However, it is the particularities of Class D supplied guitar amplifiers, and dare I say
it, the tight margins and lack of money generally in the guitar amplifier market that
brings it into the realm of advantageousness.
Also, there’s little advantage for a guitar amplifier power supply to be small in
physical size. After all, the power supply is going to be sat on top of a huge speaker
cabinet, so there’s little point in making it miniscule. This is especially the case with
Bass guitar speakers.
Of course, there are ways of adjusting the standard Burst Mode operation that can
mitigate to an extent , many of the disadvantages of Burst Mode control.
For example, no PFC stage is used, and so the primary DC Bus voltage is rippling up
and down at 100Hz (when loaded significantly), …..what one could do is use a micro
to sense the primary DC Bus voltage and then adjust the maximum duty cycle of the
FET gate pulses…(ie , suitably reduce the duty cycle the higher the primary DC Bus
voltage) alternatively, one could adjust the reference voltage on the primary current
sense comparator so that the primary peak current was lower at the higher DC bus
voltages.
Also, if the high frequency FET gate pulses were sent by a microcontroller, then the
micro could initiate a burst by starting from a low duty cycle, and gradually and
steadily working up to maximum duty cycle…this would provide much smoother
operation when at light load. (ie 1/8th Max load when the amplifier is going through
regulation testing.)
Certain Electronics consultancies (only a minority) will not consider a Burst Mode
SMPS at 320W due to perhaps academic piety.
Also, a “320W Burst Mode SMPS” is not going to tend to look good on anybody’s
CV.
Also, its not generally something that may impress future customers when seeking to
boast about one’s previous projects.
Also, dare I say it, high power SMPS projects don’t come round that regularly in the
West these days, and so when they do, it’s an opportunity for the designer to “showoff” and produce an exotic resonant SMPS or such like, something that will add to the
CV and enhance the CV , etc etc.
Also, a Burst Mode SMPS is simple enough that a customer company could discover
how it works themselves, and be able to design other versions for themselves…also,
due to the simplicity of a Burst Mode SMPS, the customer company may very well
be able to successfully modify it without needing to go back to the consultancy, and
some consultancies will not like this, as they may well wish to earn repeat fees from
power supply customers for modification work on existing power supplies.
20
I am aware of one consultancy that has four PCB layout software packages in
use….the idea is, as the story goes, that they find out what is the PCB layout
package(s) used by a particular customer, and then use a different one, so that the
customer is less easily able to modify the circuitry, and more likely to go back to the
consultancy and pay them for circuit modifications.
I had a situation with one consultancy, who actually changed from Altium to Cadstar
when they found out that we used Altium, even though they’d actually finished it all
in Altium.
21
Ripple on the primary DC Bus.
As you know, when no PFC stage is used on an Offline SMPS of relatively high
power, there is a large electrolytic capacitor just downstream of the mains rectifier
bridge.
As you know, under no_load or light_load, the voltage on the primary DC bus of the
SMPS will not actually be rippling that much , and will be fairly constant at near the
mains peak voltage,
This is good news for an Class D supplying SMPS because it means that a no_load to
full_load transient will occur with the input at near mains peak voltage….this is good
because the output voltage undershoot will be at its lowest for this transient (ie as
opposed to when the primary DC Bus voltage is at its minimum (ie the minimum
that would be seen if the SMPS was on constant load, which of course it would not be
as this is a Class D amplifier load)
22
Improving EMC for a Burst Mode SMPS
Some would say that the sudden onset of the burst of FET gate drive pulses is
detrimental to EMC passing. However, as discussed, this can be mitigated by all the
usual methods involving tight current loop layout and capacitor decoupling. However,
also, a Burst Mode SMPS can be improved in this situation , as previously discussed,
by making the burst of pulses commence in a smooth manner….eg the duty cycle of
the gate drive pulses can be steadily increased to maximum, and also, to give even
further improvement, the actual frequency of the switching pulses could be started
with a high frequency, and transitioned to a lower frequency.
Also, as discussed, if the switching pulses are sent from a microcontroller which is
sensing the primary DC Bus voltage, then the maximum duty cycle can be made
smaller whilst the primary DC bus voltage is at a higher level , and thus the peak
currents would be smaller. This would easily be possible as the dv/dt on the primary
side DC Bus electrolytic capacitor is relatively slow moving.
There wouldn’t need to be terribly many different duty cycles, -just a few for various
DC bus capacitor voltage ranges.
Another point is that if this SMPS was meant for universal (90-265VAC) operation,
then this adjustment of the maximum duty cycle could be employed to make the Burst
Mode SMPS operation as smooth as possible across the whole supply range.
Examples of high power Burst Mode SMPS’s.
As you know, a few years ago (possibly 2012) I sent Jesmond details of an applicaton
note from powerint.com which demonstrated a full design and test of an Offline
200W (peak power) Switch Mode Power Supply for supplying Half Bridge Class D
Amplifiers.
It actually comprised two 100W Flyback SMPS’s (one for each half of the “split”
output voltage)
Powerint.com say that they deleted this from their website due to the controller being
discontinued or updated.
This was a 200W Burst Mode SMPS.
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Two Transistor Forward converter.
This was chosen due to it being a simple topology.
The advantages of the Two Transistor Forward Converter are stated nicely on page 7
of the book “Power supply design Volume 1: control”, by Dr Ray Ridley.
Dr Ridley states that the Two Transistor Forward Converter is “the most rugged of all
the isolated converters”. Dr Ridley also states that it is good to 1KW of power. That
presumably means continuous power, and of course, continuous power is not a feature
of a Class D guitar amplifier load.
Dr Ridley does not state reasons for stating that the Two Transistor Forward
Converter is “more rugged than the other isolated converters”, however in his
summary of other isolated converters, Dr Ridley lists the following disadvantages, so
one presumes that it is the absence of the following disadvantages that makes the
Two Transistor Forward Converter the ”most rugged of all isolated converters”.
********* ********* ******** ****** **** *******
Half Bridge: Ridley states on page 8 of his book that the Half Bridge SMPS does not
work well with Current Mode Control. Also, that “The current balancing mechanism
of current mode works as counter to the input voltage balancing mechanism
commonly used for the two input voltages to the Half Bridge.”
Dr Ridley states that the Two Transistor Forward Converter is preferable to the Half
Bridge SMPS. Ridley also states that the Two Transistor Forward Converter is more
preferable at higher power levels than the Half Bridge SMPS.
Ridley also sites gate drive problems leading to cross-conduction as another ill of the
Half Bridge. Ridley also slaps the Half Bridge with the problem of ‘transformer flux
balancing’
Full bridge: Ridley lists gate drive noise problems leading to cross conduction, also
transformer flux balancing.
********* ********* ******** ****** **** *******
The Two Transistor Forward Converter has a limit on the duty cycle of 0.45,
however, that does not stop it being used in the application of this document.
Of course,if greater power is required, then two “Two Transistor Forward Converters”
could be used, in interleaved fashion if you wish.
In an Offline converter, if the diode dissipation is too much then synchronous
rectifiers can be used if necessary.
The Flyback SMPS is not a great choice for a Class D amplifier supply with 320W
peak, because even though the average power is very low, the Flyabck suffers the ills
previously described to Jesmond in a previous report.
The Flyback SMPS has high FET drain voltage and of course the RCD clamp
dissipation.
24
Since the Average power draw of the Class D amplifier is very low, the single switch
Flyback could not be ruled out, though some of the modifications & precautions
previously sent to Jesmond would be helpful.
One guitar Amp manufacturer does use a single switch Flyback for a guitar amplifier
labelled at 350W.
Its important to try andkeep the leakage inductance down uin the flyabck
tranasformer, and alsointhe two transsitor forward converter. The leakage in the two
tranasistor forward tramnsformer will mean a slight increase in duty ycle is needed ,
compared to what is calacultated.
The advantageof the Two Transistor Forward Converter is that the magnetising
current is recycled back to the primary DC Bus capacitor, and obviously the leakage
inductor current discharges through the diodes and back into the primary DC Bus
capacitor also.
One has to be mindful of overvoltages on the secondary diodes.
25
High-side FET Gate drive for the Two Transistor Forward Converter
Digital isolator based:
For this Burst Mode SMPS, I would recommend a high-side FET gate drive
comprising a cheap high side power supply from say a Linkzero based flyback (eg
LNK574 by powerint.com).
This would power a gate drive IC, in the high-side.
Then I would couple the low side FET gate drive signal to the high-side using a cheap
digital isolator (eg SI8710 by silabs.com). The output of the digital isolator would
feed into the gate drive IC , which would drive the FET.
The low-side FET would simply be driven by a gate drive IC at the low side.
Pulse transformer based high-side FET gate drive:
-However, a pulse transformer could be used for high-side FET drive, though
attention would need to be made to leakage inductance in it, and the tolerance on this
leakage between parts.
Also, with the type of FET gate drive with pulse transformers and series capacitors, be
aware that if the gate pulses suddenly cease, (which they often do in a Burst Mode
SMPS) then the charge on the series cap is sometimes known to drive the FET on
unwantedly….this is not so bad with a Two Transistor Forward Converter because
both high and low side FETs need to be on simultaneously to get current to flow in
the primary.
So anyway, if a pulse transformer is used instead of the digital isolator type FET
drive, then be aware of the sizing of the series capacitors, and the LC oscillations that
can result in the pulse transformer type FET drives.
Also, if the type of pulse transformer drive without series capacitors is used, be aware
of the reduction in gate drive voltage, as duty cycle increases.
26
Basic Schematic: Burst Mode SMPS
This schematic can be better seen by opening the supplied LTspice simulation. (LTspice is a free simulator from linear.com)
27