Download Compact, Low-Loss, Wideband, and High-Power Handling Phase Shifters With Piezoelectric

IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 60, NO. 6, JUNE 2012
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Compact, Low-Loss, Wideband, and High-Power
Handling Phase Shifters With Piezoelectric
Transducer-Controlled Metallic Perturber
Jing Wu, Student Member, IEEE, Jing Lou, Ming Li, Student Member, IEEE, Guomin Yang, Member, IEEE,
Xi Yang, Student Member, IEEE, Jason Adams, Student Member, IEEE, and Nian X. Sun, Member, IEEE
Abstract—Compact phase shifters with large phase shift, low
loss, and high-power-handling capability are desired for a variety
of applications. In this paper, a new type of compact meander line
phase shifter with metallic perturber controlled by a piezoelectric
transducer (PET) has been designed, fabricated, and tested. The
new phase-shifter design led to a large phase shift of 360 with a
3 dB,
control voltage of 50 V at 3.5 GHz, a low insertion loss of
a wide range of operation frequency of 1 6 GHz, a high-powerhandling capability beyond 30 dBm on a compact meander line
of 18 mm
18 mm, compared with similar phase shifters with
a dielectric perturber which exhibited very limited phase shift at
the -band. In addition, this is a low-cost phase-shifter design that
-band and beyond. Combined with a low
is extendable to the
loss, large phase shift, compact size, and a high-power-handling capability, the new meander line phase shifter with PET-controlled
metallic perturber showed great potential to be used in different
RF/microwave systems.
Index Terms—Meander line, perturber, phase shifter, piezoelectric transducers (PETs).
P
I. INTRODUCTION
HASE shifters are essential microwave components that
provide controllable phase shifts of microwave/RF signals. They are widely used for beam steering and beam forming
for phased arrays, phase equalizers, and timing recovery circuits
[1]. With thousands of phase shifters that are usually required
for a phased-array antenna system, it is crucial to have phase
shifters with small sizes, light weights, and low costs. It is also
important for phases shifter to have low loss, minimized power
consumption, and large power-handling capability.
Different techniques and approaches have been adopted for
achieving phase shift in RF/microwave components, such as
Manuscript received October 24, 2011; revised February 11, 2012; accepted
February 16, 2012. Date of publication April 03, 2012; date of current version
May 25, 2012. This work was supported in part by the Air Force Research Laboratory under Grant UES FA8650-090-D-5037, the National Science Foundation under Grant ECCS—0746810, the Office of Naval Research under Grant
N0001411M0187 and Grant N00014-10-M-0117, and the MIT Lincoln Laboratory.
J. Wu, J. Lou, M. Li, X. Yang, J. Adams, and N. X. Sun are with the
Department of Electrical and Computer Engineering, Northeastern University,
Boston, MA 02115 USA (e-mail: [email protected]; [email protected];
[email protected]; [email protected]; [email protected];
[email protected]).
G. Yang is with the Key Laboratory of Wave Scattering and Remote Sensing
Information, Department of Communication Science and Engineering, Fudan
University, Shanghai 200433, China (e-mail: [email protected]).
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TMTT.2012.2189240
magnetic field-tuned ferrite-based phase shifters [2], ferroelectric varactor-based phase shifters [3], p-i-n diodes [4], field-effect transistor (FET) switches [5], and RF micro-electro-mechanical systems (MEMS) switched line phase shifters [6]. Nevertheless, the state-of-the-art phase shifters listed above have
their own limitations. Ferrite phase shifters have large powerhandling capability, but typically have limited bandwidth, large
size, high power consumption, and slow tuning. FET switches,
p-i-n diodes, and ferroelectric varactor-based phase shifters typically have high insertion loss at the -band and exhibit limited
frequency range. RF MEMS phase shifters show good performance on bandwidth, insertion loss, size, and power consumption [7], [8]; however, they show limited power handling of
1 W 30 dBm . These limitations prevent their applications
in mission-critical phased arrays, such as high-power radars and
electronic warfare.
Chang et al. reported a new type of phase shifter with dielectric perturber controlled by piezoelectric transducers (PET) on
a planar microstrip transmission line such has those reported in
[9]–[12]. With the introduction of the dielectric perturber that
is closely placed above a microstrip transmission line, the characteristic impedance of the line is only slightly altered, while
its effective dielectric constant can be changed significantly,
which leads to phase shift. However, such phase shifters still
have problems, such as limited phase shift, large size, and high
insertion loss when the dielectric perturber is closely placed on
the microstrip for achieving a large phase shifter. For example,
a phase shifter with a size of approximately 30 mm can only
produce a controlled phase shift of less than 80 in Sthe -band
[9], which is far from the typical requirement for a 360 phase
shift.
Most recently, we have reported a similar phase-shifter design with PET-controlled magneto-dielectric perturber, which
compared
leads to significantly enhanced phase change
with a PET-controlled dielectric perturber approach due to the
increased miniaturization factor, which is related to the high permeability of the magneto-dielectric disturber. At the same time,
the increased permeability of the magneto-dielectric disturber
leads to better wave impedance match to the free space and,
therefore, much lower reflection due to the loading of the perturber and lower insertion loss [2]. This leads to high phase shift
per decibel loss of 500 dB insertion loss. However, this approach has its own limited bandwidth of less than 3 GHz due to
the increased loss tangent of the self-biased magneto-dielectric
perturber, and it still could not meet the need for ultra-wideband
(UWB) phased arrays, such as used for electronic warfare.
0018-9480/$31.00 © 2012 IEEE
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IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 60, NO. 6, JUNE 2012
Fig. 2. Equivalent circuit of meander line with piezoelectric bending actuator.
itor. Therefore, the variable phase constant
perturbation can be calculated as
caused by the
(2)
Fig. 1. Schematic and photograph of the meander line phase shifter with PETcontrolled metallic perturber.
In this paper, we present a novel distributed phase-shifter design that is compact, and wideband, low-loss and has high power
handling. This phase-shifter design consists of a meander microstrip line, a PET actuator, and a Cu film perturber, which
has been designed, fabricated, and tested. This compact phase
shifter with a meander line area of 018 mm 18 mm has been
demonstrated at -band with a large phase shift of
at
4 GHz with a maximum insertion loss of
3 dB, and a high
power-handling capability of 30 dBm was demonstrated. In
addition, a UWB low-loss and compact phase shifter that operates between 1 and 6 GHz was successfully demonstrated. Such
a phase shifter has great potential for applications in phased arrays and radars systems.
II. DEVICE CONSTRUCTION AND THEORETICAL ANALYSIS
The variable capacitance
can be tuned electrically by
applying variable voltage on the piezoelectric bending actuator.
Hence, the phase shift can be estimated as
(3)
and
denote the capacitance variance.
where
2) Loss Analysis: For microstrip meander lines, most losses
are contributed by dielectric and conductor losses, given that
the radiation loss is small. The dielectric loss
in dB/cm [13]
caused by the finite conductivity of the dielectric layers is given
by
(4)
,
dewhere the substrate loss tangent
denotes the efnotes the dielectric constant of the substrate,
fective dielectric constant for the microstrip transmission line,
and
denotes the wavelength in the substrate.
[14]–[16] can be obtained from
The conductor loss
Similar to the previous PET phase shifter using a dielectric
perturber, the structure of the designed phase shifter is shown in
Fig. 1. The PET used in the design is a commercially available
piezoelectric bending actuator (PI PICMA PL140.10), which
features a multilayer structure that reduces the voltage needed
for large deflection. The dimension of the PET is approximately
45 mm in length and 11 mm in width and can be deflected up and
down for a total range of 2 mm with a control voltage ranging
from 0 to 60 V.
1) Equivalent Circuit Model for Meander Line With Copper
Perturber: A microstrip meander line structure is widely used
in phase shifter designs due to their broadband, low insertion
loss, and ease of manufacturing. The characteristics impedance
and phase velocity of a typical microstrip transmission line
can be expressed as
denotes the surface impedance,
denotes the width
where
of the strip line, denotes the conductivity, denotes the characteristic impedance, and denotes the angular operating frequency.
With a piezoelectric bending actuator, a variable capacitance
leads to variable characteristics impedance. The return loss
due to perturber
in dB/cm will increase due to the
impedance mismatch to a standard 50- port, which can be
described as
(1)
The final form of the loss calculation is a function of loss metal
thickness, strip width and conductivity, frequency, and distance
to the perturber. The insertion loss in decibels for a perturbed
length of the phase shifter is given by
where and indicates the equivalent capacitance and inductance.
As a distributed transmission line, meander lines with piezoelectric bending actuator can be also modeled as an L-Ccircuit,
as shown in Fig. 2. The variable distance from the copper perturber to the meander line leads to an equivalent variable capac-
dB
cm
(5)
(6)
(7)
3) Device Construction: The meander line was designed to
possess a characteristic impedance of 50 , which has a conductor width of 0.356 mm. As each of the segments of the me-
WU et al.: HIGH-POWER HANDLING PHASE SHIFTERS WITH PET-CONTROLLED METALLIC PERTURBER
Fig. 3. Design dimensions for the meander line phase shifter. The gray area
shows the size and position of the metallic perturber.
ander line is 10.8 mm and each of the corners is 0.71 mm, the
total length of the meander line is about 4.5 in within an area of
12.8 mm 12.8 mm, as shown in Fig. 3. Also shown in Fig. 3
is the dimension and position of the metallic perturber, which is
a 12.8 mm 12.8 mm copper square that covers the majority of
the meander line. Without the metallic perturber, the meander
line structure is essentially a transmission line with a working
frequency range of 0 4 GHz. The maximum insertion loss of
the meander line is less than 1 dB at 4 GHz.
For broadband true-time-delay phase shifters (e.g.,
1
6 GHz), there is an important design tradeoff between
the highest and lowest operating frequencies, that is, the size of
the phase shifter should be smaller than half wavelength at the
highest frequency, e.g., 25 mm at 6 GHz, and a sufficiently large
phase shift should be achieved at the lowest frequency, say 90 .
Clearly, we need to make the phase shifter small enough to fit
size requirement while simultaneously achieving a moderate
phase shift at lower frequencies. A substrate with relatively
high K was used for the meander line design. Rogers TMM
10i has a nominal dielectric constant of 9.8, and a thickness of
0.38 mm was chosen to accomplish both longer length of the
meander line and higher power-handling requirement.
III. SIMULATION RESULTS
Simulations of the device were carried out by High Frequency
Structure Simulator (HFSS) before the meander-line -band
transmission line was fabricated. To match the traveling distance of the PET of 2 mm, the maximum and minimum distances between the metallic perturber and the substrate were set
to be 1.80 and 0.13 mm, respectively.
Fig. 4 shows the transmission coefficient
of the meander-line phase shifter with different distances between the
metallic perturber and the substrate. Clearly, when the metallic
perturber is far away from the substrate (1.8 mm), the insertion
loss of the phase shifter stays at a relatively low level of 1 dB
throughout the entire -band. However, when the metallic
perturber approaches the substrate, the insertion loss starts to
increase due to the impedance mismatch introduced by the
metallic perturber. Nevertheless, the maximum insertion loss
of the phase shifter is less than 2 dB at a 0.13-mm spacing
between the metallic perturber and the meander line.
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Fig. 4. Simulated
of the meander line with different distances between the
metallic perturber and the substrate.
Fig. 5. Simulated
of the meander line with different distances between the
metallic perturber and the substrate.
Fig. 5 shows the reflection coefficient
of the phase
shifter with different metallic perturber distances. As one may
expect, when the distance between the perturber and the substrate is 1.8 mm, the return loss is greater than 20 dB, while,
with the perturber getting closer to the substrate, the return loss
eventually reaches a minimal level of about 8 dB for a 0.13-mm
distance.
The
and
spectra show clear ripples associated with
the meander line structure, as shown in Figs. 4 and 5. The amplitude of the ripples increases with the approaching of the metallic
perturber to the substrate, and their positions as well as their
separations also vary. This is attributed to the change of the
capacitance per unit length
of the transmission line due to
the metallic perturber. This increased leads to changes of the
characteristic impedance of the meander line transmission line
expressed by
, where is the inductance per length
of the meander transmission line and therefore decreased return
loss and increased insertion loss as shown in Figs. 4 and 5. At
the same time, the increased also decreases the phase velocity
of the meander line,
. As a result of such
changes of the phase velocity of the microstrip line, the relative
phase shift changes dramatically as a function of the distance
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Fig. 6. Simulated relative phase shift of the phase shifter with different distances between the metallic perturber and the substrate.
Fig. 7. Measured
PET.
of the meander line with different voltage applied on the
Fig. 8. Measured
the PET.
of the meander line with different voltages applied on
between the metallic perturber and the substrate, as shown in
Fig. 6.
From Fig. 6, it is very clear that the phase shift of the meander line can be readily tuned by varying the distance between
metallic perturber and the substrate, although it is not a linear
function of the distance. For example, the phase shift is only 28
when the disturber–meander line gap is 1.12 mm at 4 GHz and
is 54 when the distance is 0.80 mm. However, the phase shift
reaches a value of 266 and 352 at a gap of 0.20 and 0.13 mm,
respectively.
IV. MEASUREMENT RESULTS
The meander line was fabricated by printed circuit board
(PCB) fabrication technique, and the phase shifter was assembled as schematically shown in Fig. 1. Measurement of the
meander-line phase shifter was done on an Agilent PNA series
vector network analyzer (VNA). With a control voltage applied
on the PET changing from zero to 50 V, the distance between
the metallic perturber and the substrate can be tuned. It should
be mentioned that, due to the difficulty of accurately measuring
the distance between the perturber and the meander line, the
applied voltage should only be used for reference purpose
to compare with the actual distance. However, after careful
calibration, these two values should be able to be preciously
linked to each other.
Fig. 7 shows the transmission coefficient of the meander-line
phase shifter with different voltage applied on the PET. When
the voltage is 0 V, which corresponds to the largest distance between the metallic perturber and the meander line, the insertion
loss shows very flat response with a maximum loss of 1 dB,
which matches well with simulated data shown in Fig. 4. With
the increase of the voltage applied on the PET, the distance
between the metallic perturber and the substrate was reduced,
which led to degraded insertion loss.
Since the performance of the phase shifter is very sensitive to
the distance between the perturber and the meander line, waviness of the perturber surface may introduce additional loss in the
device. As we can see from Fig. 7, compared with simulated results, the insertion loss of the device is slightly larger at higher
voltage. Nevertheless, the overall insertion loss is still less than
2 dB over the entire -band.
Similar to the simulated results, the measured reflection coefficient has the same trend, as shown in Fig. 8. For a control
voltage of 0 V, the return loss stays at very low level of 25 dB.
However, for higher voltages, a maximum return loss of 7 dB
is observed for 50 V of control voltage, which is in close match
with the simulated data.
The maximum travelling distance of the PET is 2 mm for
60-V applied dc voltage. Starting from a 1.8-mm gap with 0 V,
the PET bent down and the gap between the perturber and the
meander can be approximated as
V
mm (PL140
Data sheet, Piezo University), where V is the applied voltage
on the PET. For 50-V dc voltage, the gap is 0.13 mm, where
the measured relative phase shift has a maximum phase shift of
362 at 4 GHz as shown in Fig. 9. HFSS simulation showed a
352 phase shift, indicating a decent match between measurement and simulation results. Also, compared with the published
phase shifter based on a dielectric perturber, this accounts for
enhancement of one order of magnitude [5]. Furthermore, it can
be found that the relative phase shift is very sensitive to the
voltage change at higher control voltages as well. The phase
shift from 40 to 50 V contributes to almost 70% of the total
WU et al.: HIGH-POWER HANDLING PHASE SHIFTERS WITH PET-CONTROLLED METALLIC PERTURBER
Fig. 9. Measured and simulated relative phase shift of the meander-line phase
shifter with different voltage applied on the PET. The symbols indicate simulated results from HFSS.
phase-shift range. This agrees well with the simulated results
that the phase shift is particularly sensitive to the distance between the perturber and the substrate when the distance is small.
This phenomenon leads to the conclusion that it is possible to
use a much smaller tunable distance between the metallic disturber and the meander line, which means that large phase shift
can be achieved with a shorter PET and/or at a smaller voltage
span in order to gain majority of the phase shift capability. As
an alternative, one can start with a smaller distance between the
perturber and the substrate as an initial reference state, and a
much lower control voltage of 20 V can lead to a phase shift
of 300 . This will dramatically reduce the need for high control voltage and is needed to reduce the power consumption
of the device. Compared with other phase-shifter designs, this
phase-shifter design showed significantly enhanced phase shift
and lower loss [9].
Unlike most semiconductor-based planar phase shifters
that can only handle very limited microwave input power of
30 dBm [1]–[4], our phase-shifter design with a PET-controlled metallic disturber on the meander line has the potential
to handle a much larger range of input power since the phase
shifter has just copper and dielectric substrates. As a result,
power handling of such phase shifters will mainly be limited
by Joule heating at large RF/microwave power level. We
measured the insertion loss of our phase shifter at 3 GHz under
different microwave input powers at 3 GHz, with both 0 and
50 V applied to the PET, as shown in Fig. 10. Clearly, the
insertion losses of both cases stay nearly straight at different
microwave input power levels, with only a negligible increase
in the insertion loss at a control voltage of 50 V and at 30 dBm.
The maximum power level was only tested to up to 30 dBm
due to the limited power output level in our laboratories, while
simple extrapolation of the two curves in Fig. 10 indicate that
the phase shifter shows much higher power-handling capability
than 30 dBm. The high microwave power-handling capability
of the meander-line phase shifter is critical for high-power
phased array radars.
V. EXTENDED DESIGN FOR 1–6 GHZ
Some applications, such as satellite communication and radar
system, require controllable phase shifts in a wider band, such
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Fig. 10. Measured insertion loss of the meander-line phase shifter with different input power at 3 GHz.
Fig. 11. Design dimensions for the extended meander-line phase shifter.
as 1 to 6 GHz, which covers L the -band, -band, and part of
the -band. Hence, it is also important for phase shifters to have
a wide working bandwidth and the properties of low profile, low
loss, minimized power consumption, and large power-handling
capability. Fig. 11 shows an extended meander line phase shifter
working from 1 GHz to 6 GHz. The meander line was designed
to have the conductor width of 14 mils. With each of the segments of the meander line being 5.58 mm and each of the corners
being 0.508 mm, the total length of the meander line is about 223
mm within an area of 18 18 mm. The same metallic perturber
has been use to tune the capacitance through different heights.
It should be mentioned that the performance of the phase
shifter is very sensitive to the distance between the perturber
and the meander line. In addition, the bending actuator brings an
inclined copper surface, which leads to additional insertion loss
and nonlinearity of phase shifts. These are more critical at closer
distance. Therefore, in the extended meander-line approach, the
perturber was placed at the closest distance and completely parallel to the meander line, when the voltage is 0 V. Then, it would
be bent up when higher voltages were applied. With the metallic
perturber far away from the substrate, the phase shift due to the
metallic surface will be neglected. Thus, we set the 25-V applied
voltage as the reference point for relative phase shift measurement.
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Fig. 12. Measured relative phase shift of the extended meander-line phase
shifter with different voltages applied on the PET.
Fig. 13. Measured
applied on the PET.
of the extended meander line with different voltage
Fig. 14. Measured
applied on the PET.
of the extended meander line with different voltage
TABLE I
MEASURED RELATIVE PHASE SHIFT OF THE EXTENDED MEANDER-LINE PHASE
SHIFTER AT 6 GHZ WITH DIFFERENT VOLTAGES APPLIED ON THE PET.
Fig. 12 shows the phase shifts of the meander-line phase
shifter with different voltages applied on the PET. It is very clear
that the phase shift of the meander line can be readily tuned by
varying the distance between the metallic perturber and the substrate through variable voltage applied. The measured relative
phase shift showed a maximum phase shift of 367 at the center
frequency 3.5 GHz with a control voltage of 0 V on the PET, 88
at 1 GHz, and 807 at 6 GHz. With the increase of the voltage
applied on the PET, the distance between the metallic perturber
and the substrate was increased. Then, the reduced capacitance
leads to smaller phase shifts. For example, if we set 6 GHz as the
working frequency, we get the following phase shifts as shown
in Table I.
Fig. 13 show the transmission coefficient
of the meander-line phase shifter with different voltages applied on the
metallic perturber. Clearly, with the higher voltage (25 V),
where the metallic perturber was far away from the substrate,
the insertion loss of the phase shifter stays at a relatively
low level of
dB throughout the entire band of 1-6 GHz.
However, when the applied voltage was reduced, the metallic
perturber approaches the substrate. The insertion loss starts to
degrade to 3.8 dB at 6 GHz, which is the maximum insertion
loss throughout the entire band. However, it should be mentioned that 360 phase shift is sufficient for most applications.
In our design, the phase shift exceeded the 360 phase shift
requirement in the frequency band of 3.5–6 GHz, with the majority of bad insertion loss cases. A customized voltage set can
be used to achieve the required phase shift while maintaining
relatively low insertion loss. For example, at 6 GHz, the tuning
range of 8–25 V can achieve 360 phase shift, with a maximum
insertion loss 2.85 dB; at 5 GHz, the tuning range of 3.5–25 V
can achieve 360 phase shift, with a maximum insertion loss
of 3.53 dB.
Fig. 14 shows the return loss
of the meander line phase
shifter with different voltage applied on the metallic perturber.
A high
6.5 dB was observed when the voltage was 0 V,
and the perturber was very close to the meander line. Once the
voltage was increase and the metallic perturber was sufficiently
far and had less impact on the meander line,
went beyond
10 dB.
Compared with the original design (working at 2–4 GHz),
the extended meander-line shifter has a small insertion loss
increase. Loss was then analyzed by applying (4) and (5). The
estimated of the meander line at 6 GHz is 0.1035 dB/cm
for conductivity loss and 0.0262 dB/cm for dielectric loss.
The total effective length of the meander line is 22.2976 cm.
Therefore, the total loss can be estimated as 2.3 dB for conductivity loss, 0.58 dB for dielectric loss, 0.8 dB for metallic
perturber according to the measurement results in Table I, and
the rest 0.12 dB for impedance mismatching of original perfect
conductor meander line. Apparently, the majority of the loss
comes from finite conductivity of copper transmission line,
WU et al.: HIGH-POWER HANDLING PHASE SHIFTERS WITH PET-CONTROLLED METALLIC PERTURBER
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TABLE II
PERFORMANCE COMPARISON OF PHASE SHIFTERS WITH DIFFERENT DEVICE TECHNIQUES
which is also the bottleneck of meander-line phase shifter.
However, it achieved much wider bandwidth (1–6 GHz),
which is very important for some application desired of wide
operation frequency band.
Table II shows the performance comparison of the fabricated
phase shifter in this work with the other reported phase shifters.
The measured degree/decibel low insertion loss of 212 is found
to be better than those of the previously reported phase shifters.
Also, the device size is the smallest among PET phase shifters,
although larger than others.
VI. CONCLUSION
A novel type of phase shifter was proposed and demonstrated utilizing a PET-controlled metallic transducer on a
meander transmission line. Compared with phase shifters with
PET-controlled dielectric or magnetodielectric perturbers,
the phase shifter with a PET-controlled metallic perturber
exhibited significantly enhanced phase shift
and
bandwidth, reduced size, and insertion loss. A compact -band
meander-line phase shifter with metallic perturber controlled
by a PET has been designed, fabricated, and tested. The total
dimension of the meander line is only 18
18 mm . Compared with a dielectric perturber that only exhibits very limited
phase shift at the -band, our design reached a phase shift
of 360 with a low controlling voltage of 25 V at 3.5 GHz,
along with a wide operating bandwidth from 1 to 6 GHz. In
addition, there is no fundamental limit of the frequency range
for such a phase shifter, as the frequency limit is mainly from
the design of the meander line. While the meander line can be
easily designed for frequencies of -band was demonstrated
in this work, similar phase shifter designs can be made for the
-band, -band, -band, and beyond from our simulations,
and even an extremely wideband phase shifter can be achieved
with a straight transmission line and a PET-controlled metallic
disturber. High power handling of 30 dBm has been experimentally demonstrated in a compact -band phase shifter,
with an expected power-handling limit of 50 dBm. With the
combined low insertion loss, large phase change, compact size,
high microwave power-handling capability, and the extended
abilities to other frequency bands, the new meander-line phase
shifter with a PET-controlled metallic perturber shows great
potential for different phased array systems.
ACKNOWLEDGMENT
The authors would like to thank Dr. S. Berkowitz for the highpower measurements.
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Jing Wu (S’08) received the B.Sc. degree in electrical engineering from the University of Science and
Technology of China, Hefei, China, in 2006, and the
M.Sc. degree from Northeastern University, Boston,
MA, in 2009, where he is currently working toward
the Ph.D. degree.
Since 2009, he has been a Graduate Research
Assistant with the Sun Group at Northeastern University, Boston, MA. His research interests include
theory and applications of novel magnetic, ferroelectric and multiferroic materials for RF integrated
circuits (ICs), monolithic mircrwave ICs, and power electronics, different
RF/microwave devices, integrated tunable multiferroic electromagnetic subsystems, waves interactions in complex media, and photonic bandgap structures.
Jing Lou received the B.S. degree in physics from
Nanjing University, Nanjing, China, in 2003, and the
M.S. degree in physics and Ph.D. degree in electrical
engineering from Northeastern University, Boston,
MA, in 2005 and 2010, respectively.
He is a Postdoctoral Research Associate with the
Electrical and Computer Engineering Department,
Northeastern University, Boston, MA. His main
research interests include synthesis, microstructure,
and properties of magnetic and magnetoelectric
materials for applications in RF and microwave
devices. Novel devices based on magnetoelectric concept are also his focus.
Ming Li (S’10) received the B.Sc. degree from
Wuhan University, Wuhan, China, in 2006, and the
M.Sc degree in physics from the Institute of Physics,
Chinese Academy of Sciences, Beijing,China, in
2009. He is currently working toward the Ph.D.
degree in electrical and computer engineering at
Northeastern University, Boston, MA.
His research interests include ultra-wideband
antennas, antenna miniaturization, tunable and
nonreciprocal microwave/RF devices, and magnetic
and ferroelectric materials and their application in
microwave devices design.
Guomin Yang (S’07–M’10) was born in Zhejiang
Province, China, in 1979. He received the B.S.
degree (with honors) in communication engineering
from Xi’an University of Technology, Xi’an, China,
in 2002, the M.S. degree in electronic engineering
from Shanghai Jiao Tong University, Shanghai,
China, in 2006, and the Ph.D. degree in electrical
and computer engineering from Northeastern University, Boston, MA, in 2010.
In 2010, he joined the faculty of School of Information and Technology at Fudan University, where
he is currently an Assistant Professor. His research interests include antenna
miniaturization, magneto-dielectric materials, metamaterials, frequency selective surfaces, UWB filters, UWB antennas, computational electromagnetics, and
inverse scattering problems in electromagnetics. He has authored 16 journal
publications and 14 conference papers.
Dr. Yang was the recipient of National Graduate Student Scholarship in
2006. He is a member of the Editorial Board of the IEEE TRANSACTIONS ON
MICROWAVE THEORY AND TECHNIQUES.
Xi Yang (S’11) received the B.Sc. degree in electrical
engineering from the Shanghai Jiao Tong University,
Shanghai, China, and the M.Sc degree from Jilin University, Changchun, China, in 2008 and 2010, respectively. He is currently working toward the Ph.D. degree at Northeastern University, Boston, MA.
Since 2010, he has been a Graduate Research Assistant with the Sun Group, Northeastern University,
Boston, MA. His research interests include novel
magnetic, ferroelectric, and multiferroic materials
for microwave applications, different RF/microwave
devices, and integrated tunable multiferroic electromagnetic subsystems.
Jason Adams (S’08) received the B.Sc. and M.Sc.
degrees in electrical engineering from Boston University, Boston, MA, in 2006 and 2008, respectively.
He is currently working toward the Ph.D. degree at
Northeastern University, Boston.
Since 2008, he has been an Electrical Engineer
with Raytheon. Additionally, he has been conducting
research in the Sun Group, Northeastern University,
Boston, MA, since 2010. His research interests
include RF/microwave devices and systems, as well
as the theory and application of novel magnetic,
ferroelectric and multiferroic materials for RFIC, MMIC and power electronics.
Nian X. Sun (S’98–M’02) received the Ph.D. degree
from Stanford University, Stanford, CA, in 2002.
He is currently an Associate Professor with the
Electrical and Computer Engineering Department,
Northeastern University, Boston, MA. Prior to
joining Northeastern University, he was a Research
Scientist with IBM and Hitachi Global Storage
Technologies between 2001 and 2004. He has
authored and coauthored over 80 publications and
holds nearly 20 patents and patent disclosures. His
research interests include novel magnetic, ferroelectric ,and multiferroic materials, devices, and subsystems.
Dr. Sun was the recipient of the National Science Foundation CAREER
Award, the Office of Naval Research Young Investigator Award, the U.S. Air
Force Summer Faculty Fellowship, and the first prize IDEMA Fellowship.
One of his papers published in 2009 was selected as the “ten most outstanding
in Advanced Functional Materials”.
papers in the past decade