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A MEMS based electrometer with a low-noise switched reset amplifier for charge measurement G. Jaramillo, D. A. Horsley C. Buffa, G. Langfelder University of California, Davis Berkeley Sensor and Actuator Center Davis, CA, USA Department of Electronics and Information Technology Politecnico di Milano Milano, Italy [email protected] Abstract— Electrostatic charge measurements are at the base of chemical, physical and biological experiments. In this work the authors present an electrometer based on the vibrating capacitance of a microelectromechanical (MEMS) micromachined resonator. We present improvements on the low-noise readout amplifier by reducing input-referred voltage noise and parasitic capacitances. An amplifier has been designed to have a noise corner frequency well below the device’s operating frequency fn. The electrometer geometry allows for charge output signal measurements at 2fn minimizing feedthrough of driving signals. The sensor consists of a set of comb-finger capacitors placed on each side of a moving mass for push-pull driving. Operating at resonance, charge collected on the moving electrode is modulated and the induced voltage is read with a low-leakage very high-input impedance feedback amplifier. Due to the specific readout technique, a switched-reset is used to prevent charge saturation. Reduction of parasitic capacitance and increase in resolution is achieved through the careful selection and placement of discrete electronic components alongside the silicon MEMS chip. I. INTRODUCTION Electrometry is a technique for measuring small electrical currents. Electrometer instruments are commonly used in mass spectrometry, surface charge analysis, and the development and delivery of pharmaceuticals [1]. Electrometer techniques include single-electron transistors, nano-mechanical resonators at cryogenic temperatures [2], graphene resonators [3], and devices based on the vibrating reed technique. The time-varying capacitance electrometer or vibrating reed can be implemented using MEMS surface micromachining of parallel plate sensors and actuators. MEMS based electrometry can play an important role in the field of aerosol technology [4]. The device can be employed for the counting of nanometer-sized aerosol particulate. MEMS technology permits the reduction of the cost, size, weight and power of current commercial electrometers used in the aerosol scientific field. The MEMS electrometer is based on parallel plates to sense charge and comb drive capacitive actuators to generate a time varying capacitance, thereby modulating a slowly-varying input charge signal to a frequency > 1 kHz where the measurement noise is greatly reduced. The need for highly accurate, high precision, low cost electrostatic sensors continues to expand into many fields. This demand drives the development and improvement of electrometer circuits. Electrometer circuits suffer from a variety of error sources, including noise, parasitic, noise mixing and leakage currents [5]. In this paper we tackle two sources of error: parasitic capacitances and noise. II. DEVICE AND PRINCIPLE OF OPERATON The miniaturized electrometer consists of a mechanical sensing element based on MEMS technology. The electrometer works as a variable capacitor coupled to the excitation and readout electronics. When an external amount of charge is deposited onto a conductive electrode electrically connected to the MEMS capacitor, a voltage signal builds up on the capacitor. There is a universal relation between the measured voltage and the detected input charge, . The device, shown in Fig. 1, is manufactured using 15µmthick epitaxial polysilicon by STMicroelectronics ThELMA (Thick Epitaxial Layer for Micro-actuators and Accelerometers) process. It consists of a main shuttle anchored by four springs. At opposite ends of the shuttle, comb-finger actuators placed in the push-pull driving configuration force the shuttle into resonance. The middle of the structure contains a set of parallel-plate capacitors, CCV, that sense the input charge. Figure 1. SEM picture with a close-up of drive comb-finger and readout parallel-plate capacitors. MEMS electrometer device oscillates at fn and generates a detection signal at 2fn which is proportional to the deposited charge, QC, on parallel-plate electrodes. In order to minimize the power consumption, the device is voltage driven at its mechanical resonance frequency fn, designed to be 2.3kHz, since electrostatic forces are amplified by the quality factor Q. The lower driving voltage makes the electrometer compatible with most CMOS integrated circuits and discrete components. A high Q will allow a final system implementation with a self-oscillating closed loop circuit. This first demonstration prototype run in open-loop requires a preliminary mechanical characterization of the device to identify fn. The operating principle of the vibrating reed electrometer was previously proposed [6]. When an external unknown amount of charge QC is deposited on the parallel-plate electrodes, it is converted to an alternating voltage signal as a consequence of the modulation of the resonating capacitor: . (1) where Cpar contains all parasitic components loading the input node. In order to avoid a capacitive feedthrough at fn which might shadow the weak charge signal, each moving parallelplate armature is faced by two fixed and electrically shorted electrodes with the consequent generation of a charge signal at the second harmonic, 2fn, where the feedthrough signal is negligible. The sensitivity, here intended as the variation of modulated voltage per input charge variation, is defined as [6]: √ ⋅ ⋅ , (2) where x is the displacement of the moving mass and g is the capacitance air gap at rest. Once the geometry and rest capacitance of the device are fixed, a highly sensitive system can be obtained by minimizing all parasitic capacitances and maximizing the displacement of the moving mass as long as the resonator remains in a stable linear operating region. A. Electromechanical characterization Preliminary electro-mechanical characterization of the devices used for charge measurements was carried out in order to obtain an applied force vs. displacement curve, to estimate the mechanical offset, resonance frequency and pull-in voltage [4,7]. The actual gap, g = 3.6 µm, was carefully extracted from these electro-mechanical measures and its difference with respect to the air gap drawn on the mask (3.0 µm) is in line with common overetch values for this manufacturing process. The importance of this mechanical characterization is justified by the strong dependence of device sensitivity on the gap. Also the non-linear behavior of the device is estimated by applying increasing voltages on the actuator and measuring the corresponding displacement. Fig. 2 shows the displacement amplitude at resonance with respect to the amplitude of the driving signal. For displacements larger than x = 430 nm, the device enters a non-linear working region. III. ELECTRONICS DESIGN Customized driving and analog readout electronics have been designed based on the mechanical parameters of the device (i.e. resonance frequency, required actuation voltage) using a discrete component solution on a printed circuit board (PCB) and carefully choosing components in order to minimize additional unwanted parasitic capacitances and noise with respect to previous implementations [4]. A. Push-pull driving stage Push-pull drive is widely used in micromachined actuators as the actuation force can be linearly controlled by the amplitude of applied voltage [8]. At the current state of setup development, as the device is not included in an oscillating loop, an external excitation stimulus must be provided with a function generator. In order to split this signal into a phase and an anti-phase component (required for push-pull driving), a differential driving scheme has been designed using two difference amplifiers with unity gain, as shown in Fig. 3. They are based on operational amplifier (OPA445) that is capable to stand high voltage with a low temperature drift coefficient (typical offset drift about 25 µV/°C) and fast slew rate (15 V/µs). Each driving signal consists of a DC component, VOS, which can be set using an onboard potentiometer and an ac component (180° phase shifted in one channel) applied with an external function generator. Considering an input voltage ,! ! ⋅ sin2& ⋅ '( ⋅ provided by an external function generator, the two output signals fed to MEMS driving electrodes are: Figure 2. The normalized displacement of the suspended mass, x0 = x/g, at resonance is plotted with respect to drive voltage amplitude. Non-linear response for displacement larger than x = 430 nm. )*+, ,- . ! ⋅ sin2& ⋅ '( ⋅ , (3) )*+,/ ,- . ! ⋅ sin2& ⋅ '( ⋅ . & , (4) With these signals the electrostatic force applied to the comb-finger actuators in push-pull configuration becomes: 0)*( 12 34 56 ⋅ ,- ⋅ ! ⋅ sin2& ⋅ '( ⋅ , (5) where ε0 is the air permittivity, H the comb electrode height, N the overall number of driving cells and gcf the gap of comb finger capacitors. As H and N are known from design and gcf can be precisely estimated with electromechanical characterization, the total driving applied force is finely controlled by offset and ac voltage amplitudes. package, is done to minimize leakage current. This particular kind of transistor has lower loss and higher isolation capabilities compared to other traditional discrete component transistors. Figure 4. The first stage of analog readout electronics is based on a feedback amplifier and the device output votlage is readout with the positive high impedance input. G = 14.6. Figure 3. Schematic of push-pull dirving circuit. The two stages are used to generate driving signals with 180° phase shift. Signals offset can be on-board selected using two trimming resistors. The frequency response of driving amplifiers, with a precise gain due to the choice of 0.1% tolerance resistances (R3 = 10 kΩ), is flat up to a few tens of kHz, enough to guarantee symmetric driving at the device resonance frequency with a negligible phase shift. B. Low-noise switched reset readout amplifier Readout analog frontend specifications are set by the need to measure low amounts of charge flowing on the vibrating capacitor and converted into a small ac signal. First, a study of the architecture used was done with the final choice of a highinput impedance amplifier based on a non-inverting closedloop scheme. The main advantage of this scheme lies on the high input impedance guaranteed by the operational amplifier input, see frontend schematic in Fig. 4. The operational amplifier was carefully selected to minimize input capacitances, leakage current and noise. AD8067 by Analog Devices has an input voltage noise density 78( 9 10 < ⁄√=> with a noise corner frequency around 1 kHz, a leakage current on the order of pA and an input common mode capacitance CCM = 1.5 pF thanks to the tiny package SOT23-5. The MEMS device, schematically represented as a variable capacitor in Fig. 4, is connected to the positive input of the amplifier with a set gain, G = 14.6, chosen high enough to achieve a phase margin close to 85° for a stable operation and low enough to allow a 15 s charge measurement period before reaching saturation. Indeed all leakage current components are integrated on capacitances (device + parasitic) generating drift of the input voltage. In order to reset this ramp when the amplifier has reached saturation, a reset transistor is connected in parallel to the device. The choice of NXP BF1107, a depletion type field-effect transistor in a SOT23 The choice of the values of gain resistors, R1 = 49.9 Ω and R2 = 680 Ω, is made in order the make their noise negligible with respect to the input voltage noise of the amplifier which is unavoidable. An experimental confirmation of the design is given in Fig. 5: the measured noise is close to theoretical operational amplifier noise after the 1/f noise corner frequency of 100 Hz. Figure 5. Input referred measured noise of readout amplifier. Flicker noise component is well below the device operating frequency, nominally equal to 2.3 kHz. The voltage output of the first stage is then further amplified and filtered using Stanford Research System SR810, selecting the second harmonic (2fn) component. The lock-in amplifier output is acquired using NI PCI 6251 acquisition board for further signal processing. The MEMS device is glued on a small PCB close to the first-stage amplifier’s positive input pin to minimize parasitic capacitance. The same PCB also hosts a reset mechanism and a tiny connector to interface an off-board electrode for charge collection. This PCB is then socketed into a motherboard with the rest of electronics and ±10 V power supply with filtering. A metal box shields the whole system, shown in Fig. 7. Sensitivity calibration is performed using a step voltage applied to a known capacitor Ccal = 0.2 pF and measuring the voltage output variation at 2fn, output signal shown in Fig. 6a. A low leakage depletion FET resets input charge every measurement. The electrometer in the new configuration achieved a sensitivity of 53.25 µV/fC or 5.33 × 1010 V/C, see Fig. 6b. The sensitivity corresponds to almost a 1000x improvement from our previous reported sensitivity. was swept over a range of frequencies surrounding the natural frequency fn. The magnitude and phase of the charge measurement output (Fig. 9) show the characteristic peak and 180 phase dependence of the MEMS resonator. (a) (b) Figure 9. Frequency response showing magnitude and phase measured at the second harmonic of the drive frequency. The amplitude peak and 180° phase transition at 2fn = 4750 Hz are clearly observed. IV. Figure 6. Sensitivity calibration. (a) Amplifier output signal with applied voltage steps. (b) 2fn voltage outout vs. input charge. CONCLUSION AND FUTURE WORK We implemented a switched reset electrometer based on discrete circuitry and a MEMS-based vibrating capacitance electrometer. Parasitic capacitances were reduced by carefully designing board layout and circuit implementation. These modifications achieved a ~1000x improvement in device responsivity. Charge measurements can be conducted using the new electrometer in a scanning mobility particle sizer (SMPS) configuration [4]. These experiments will consist of an aerosol particulate source and a particle selection system which accurately charges and feeds a known particle diameter to the electrode-MEMS electrometer device. REFERENCES [1] [2] [3] Figure 7. Image of the board showing the main components for the MEMS electrometer. The board is placed inside a grounded metal box. [4] [5] [6] Figure 8. Illustration of the charge reset and sampling/integration of the charge using the MEMS electrometer system. To demonstrate the sensitivity of the charge measurement to the MEMS displacement amplitude, the excitation frequency [7] [8] S. Hoe, P. M. Young and D. Traini, “A Review of Electrostatic Measurement Techniques for Aerosol Drug Delivery to the Lung: Implications in Aerosol Particle Deposition,” Journal of Adhesion Science and Technology, vol. 25, no. 4–5, pp. 385–405, Jan. 2011. A. Cleland and M. Roukes, “A nanometre-scale mechanical electrometer,” Nature, vol. 392, no. March, pp. 160–162, 1998. J. S. 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