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Designing Operational Amplifier Oscillator Circuits
For Sensor Applications
Author:
Jim Lepkowski
Microchip Technology Inc.
INTRODUCTION
Operational amplifier (op amp) oscillators can be used
to accurately measure resistive and capacitive sensors. Oscillator design can be simplified by using the
procedure discussed in this application note. The derivation of the design equations provides a method to
select the passive components and determine the influence of each component on the frequency of oscillation. The procedure will be demonstrated by analyzing
two state-variable RC op-amp oscillator circuits.
The ratio oscillator provides an output frequency that is
proportional to the square root of the ratio of two capacitors (i.e., freq. ∝ (C 4 / C3)1/2). The ratio oscillator can
be used to cancel the effect of a fluid-level sensor with
a varying dielectric constant, such as an oil level sensor. An oil level sensor consists of two capacitances
that are formed by tubes where the fluid serves as the
dielectric media. The measurement capacitor (CMEAS)
is partially covered by fluid and detects the level of the
oil in the tank. In contrast, the compensation sensor
(CCOMP) is completely buried in the fluid. When the
ratio CMEAS / C COMP is calculated, the dielectric constant of the oil is canceled. Air pressure and acceleration sensors can also use the ratio sensor to minimize
the error that occurs from the variance of the dielectric
constant over temperature.
SENSOR APPLICATIONS
State-variable oscillators are often used in sensor conditioning applications because they have a reliable
start-up and a low sensitivity to stray capacitance. The
absolute and ratio state-variable oscillators can be
used to accurately detect both resistive and capacitive
sensors. However, this application note will only analyze capacitive applications. The state-variable's three
op-amp topology provides for a more dependable oscillation start-up than a single op amp oscillator. The virtual ground voltage at the inverting terminal of the
amplifiers provides for immunity from stray capacitance, which is important in sensor applications,
because the sensor capacitance is often only 10 to
100 pF. In addition, the state-variable oscillator does
not require matched capacitors or capacitors that have
a terminal connected to ground.
The absolute oscillator provides an output frequency
that is proportional to the square root of the product of
two capacitors (i.e., freq. ∝ (C 1 x C 2)1/2). Absolute
quartz pressure sensors and humidity sensors are
examples of capacitive sensors that can use the absolute oscillator. Also, this circuit can be used with resistive sensors, such as RTDs, to provide a temperatureto-frequency conversion.
 2003 Microchip Technology Inc.
TRANSDUCER SYSTEM
A block diagram of a typical sensor system is shown in
Figure 1. The oscillation frequency can be found by
counting the number of clock pulses (i.e., MHz) in a
time window that is formed by the square wave output
(i.e., kHz) of a comparator circuit. The counter and
comparator circuits can be implemented with a
PICmicro® microcontroller.
The PICmicro microcontroller can be used to provide
curve-fit temperature correction for precision sensing
applications. Temperature correction can be accomplished by implementing a curve-fitting routine with
data obtained by calibrating the sensor over the operating range. The temperature correction data can be
stored in the E2 memory of the PICmicro microcontroller. A silicon IC sensor can provide the temperature of
the sensor.
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C
C
1
2
RC Op-Amp
Oscillator
Comparator
Temperature
Sensor
Algorithm:
Count the number of clock pulses in a time window set by oscillator pulses.
PICmicro®
Microcontroller
Clock
Signal
Oscillator
Signal
FIGURE 1:
Typical RC Operational Amplifier Oscillator Sensor System.
OSCILLATOR THEORY
OSCILLATOR DESIGN PROCEDURE
An oscillator is a positive feedback control system that
generates an output without requiring an input signal. A
sustained oscillation is initiated by factors such as
noise pick–up or power supply transients. Figure 2
shows a block diagram of an oscillator, along with the
definition of the mathematical terms that describe an
oscillator.
Listed below is a procedure to design RC operational
amplifier oscillators. Refer to the design equation section of this application note for additional information on
deriving oscillator design equations.
The design equations of an oscillator are determined
by analyzing the denominator of the transfer equation
T(s) of the circuit. The poles of the denominator of T(s),
or equivalently, the zeroes of the characteristic equation (∆s), determine the time domain behavior and stability of the system. An oscillator is on the border line
between a stable and an unstable system and is
formed when a pair of poles are on the imaginary axis.
Step 1: Find LG and ∆s
The oscillation frequency is determined by finding the
poles of the denominator of the transfer equation T(s),
or equivalently, the zeroes of the numerator N(s) of the
characteristic equation (∆s). Mason’s Reduction Theorem, shown in Appendix A, provides a method of obtaining ∆s. Then ∆s is found by breaking the feedback loop
and obtaining the gain equation at each op-amp in order
to calculate the loop-gain.
The magnitude and phase equations of an oscillator
also must be analyzed. If the magnitude of the loopgain is greater than one and the phase is zero, the
amplitude of oscillation will increase exponentially until
a factor in the system, such as the supply voltage,
restricts the growth. In contrast, if the magnitude of the
loop-gain is less than one, the amplitude of oscillation
will exponentially decrease to zero.
VIN
+
A ≡ Amplifier Gain
+
β ≡ Feedback Factor
FIGURE 2:
VOUT
V OUT
A
A
A
A
T ( s ) = ---------------- = ---------------- = ----------------- = ------ = -----------V
1 – Aβ
1 – LG
∆s
N(s)
IN
----------D(s)
where: A x β = LG ≡ loop gain
∆s ≡ characteristic equation
If VIN = 0, then T(s) = ∞ when ∆s = 0
Oscillator Block Diagram.
 2003 Microchip Technology Inc.
DS00866A-page 2
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Step 2: Solve N(s) = 0
The second step in the procedure determines the
zeroes of N(s). Routh’s stability criterion, shown in
Appendix B, provides a method that determines the
zeroes of the characteristic equation without the
necessity of factoring the equation.
First, the Routh test consists of forming a coefficient
array from N(s). Next, the procedure substitutes s = jωο
for s, with the summation of the row set to zero. If the
row equation produces a non-trivial solution for ωο, the
procedure is complete and the frequency of oscillation
is equal to ωο. If the row equation does not yield an
equation that can be solved for ωο, the procedure continues with the next row in the Routh array. Usually, it is
necessary only to complete the first two or three rows
of the Routh array to produce an equation that can be
solved for ωο.
Step 3: Sub-Circuit Design Equations
The third step in the design procedure analyzes the
sub-circuits formed at each amplifier. The sub-circuit
equations are formed by obtaining the gain equation
and pole/zero locations for each amplifier.
Step 4: VerifyLG ≥ 1
The final step in the procedure verifies that the loopgain is equal to, or greater than, one after the R and C
component values have been chosen. This step is also
required to verify that the amplifiers do not saturate,
which will result in an error in the oscillation frequency.
AMPLIFIER SELECTION CRITERIA
The appropriate op amp to use in a sensor oscillator is
determined by the required accuracy and acceptable
distortion of the oscillation frequency. The design equations assume that the amplifiers are ideal. However, op
amps have a finite gain bandwidth product (GBW), a
limited slew rate (SR) and full power bandwidth (fP).
The non-ideal characteristics of the amplifier will lower
the oscillation frequency at high frequencies and may
also result in a design with poor start-up characteristics. Note that the total harmonic distortion specification
of the amplifiers is critical for oscillators that are used
as sine wave references. However, the shape of the
waveform is not critical in most sensor applications
because only the frequency of the output is measured.
with the MCP6024 with enough design margin that the
non-ideal characteristics of the amplifier can be
neglected.
ABSOLUTE STATE-VARIABLE
OSCILLATOR
Circuit Description
The schematic of the absolute circuit is shown in
Figure 3. The state-variable oscillator consists of two
integrators and an inverter circuit. Each integrator provides a phase shift of 90°, while the inverter adds an
additional 180° phase shift. The total phase shift of
360° of the feedback loop produced by the three amplifiers results in the oscillation. The first integrator stage
consists of amplifier A1, resistor R1 and sensor capacitance C1. The second integrator consists of amplifier
A2, resistor R2 and sensor capacitance C2. The inverter
stage consists of amplifier A3, resistors R 3 and R4 and
capacitor C4. The addition of capacitor C4 helps ensure
oscillation start-up by providing an additional phase
shift.
The absolute oscillator does not require a limit circuit if
rail-to-rail input/output (RRIO) amplifiers are used and
the gain of the inverter stage (A 3) is equal to one
(i.e., R3 = R4). The sinewave output of the signal will
swing within approximately 50 mV of the VDD and VDD
power rails as shown in Figure 5.
A complementary output voltage comparator (A4) is
used to convert the oscillator’s sinewave output to a
square wave digital signal. The comparator functions
as a zero-crossing detector and the switching point is
equal to the virtual ground voltage (i.e., VDD/2). Resistor R9 is used to provide additional hysteresis (V HYS) to
the comparator. Listed below is the hysteresis
equation.
EQUATION:
V
R
8 × (V
= -------------------–V
)
O ( max )
O ( min )
HYS
R8 + R9
R8
V HYS ≅ -------------------- × V DD
R8 + R9
Several general design rules can be used to select an
op amp for an oscillator circuit. First, the GBW should
be a factor of 10 to 100 higher than the maximum oscillation frequency.
Next, the full-power bandwidth,
defined as fP = SR / (2πVP), where VP is the voltage
swing (VO(max) - VO(min)) of the output signal, should be
at least 2 times greater than the maximum oscillation
frequency. For example, the MCP6024 quad amplifier
has a GBW = 10 MHz (typ.), SR = 7 V/µs (typ.) and a fP
of 400 kHz, with VDD = 5V. An oscillator with a
maximum frequency of 100 kHz can be implemented
 2003 Microchip Technology Inc.
DS00866A-page 3
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VDD
R6
VDD/2
R7
C4
C1
R1
C2
A1
V1
VDD /2
FIGURE 3:
R2
R4
R3
A2
VDD/2
V2
R9
A3
V3
VDD/2
R8
A4
V0
VDD /2
Absolute Oscillator Schematic.
–1 A 2 = ---------------sR C
2 2
V1
A
3
=
 R 4 
1
 ------  --------------------------
 R 3 sR 4 C 4 + 1
V3
V2
–1 A 1 = ---------------sR 1 C 1
FIGURE 4:
Absolute Oscillator Signal Flow Diagram.
 2003 Microchip Technology Inc.
DS00866A-page 4
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ABSOLUTE STATE-VARIABLE
The procedure continues by analyzing row s 2 to determine when the equation is equal to zero.
Design Equations
Let s = jωο
2
STEP 1: FIND LG AND ∆S
0 = R4 – R 1R 2 R3 C1 C2 ωo
A
A
A
T ( s ) = ----------------- = ------ = -----------1 – LG
∆s
N(s)
----------D(s)
A2
2
ωo = [ R 4 ⁄ ( R 1R 2R 3 C1 C2 ) ]
A3
V1
V2
2
ωo = [ R4 ⁄ ( R1 R2 R3 C1 C2 ) ]
The loop-gain is found by breaking the loop in the
signal flow diagram of Figure 4, as shown below.
A1
2
a 1 s + a 3 = ( R 1 R 2 R 3 C 1 C 2 ) ( jω o ) + R 4 = 0
V3
1/2
P = 2π ⁄ ω o = 2π ⁄ [ R 4 ⁄ ( R 1 R 2 R 3 C 1 C 2 ) ]
1/2
Note that C4 does not appear in the oscillation equation. C4 and R 4 form a low-pass filter. The gain of amplifier A3 will not be a function of C 4 if the oscillation
frequency is less than the cut-off frequency of the filter.
If:
A 1 = – 1 ⁄ ( sR 1 C 1 )
A 2 = – 1 ⁄ ( sR 2 C 2 )
A3 = –Z 4 ⁄ Z3
= – ( R 4 || C 4 ) ⁄ R 3
= – [ ( R 4 ⁄ R 3 ) ( 1 ⁄ ( sR 4 C 4 + 1 ) ) ]
LG = A × A × A
1
2
3
= [ – 1 ⁄ ( sR C ) ] [ –1 ⁄ ( sR C ) ] [ ( –R ⁄ R ) ( 1 ⁄ ( sR C + 1 ) ) ]
1 1
2 2
4 3
4 4
3
2

= – R ⁄ s R R R R C C C C +s R R R C C 
4 
1 2 3 4 1 2 3 4
1 2 3 1 2
1.
2.
3.
R1 = R2 = R
C1 = C2 = C
R3 = R4
Then:
STEP 3: SUB-CIRCUIT DESIGN EQUATIONS
Integrator A1
∆s = N ( s ) ⁄ D ( s ) = 1 – LG
3
2
= 1 – [ –R 4 ⁄ ( s R 1 R 2 R 3 R 4 C 1 C 2 C 4 + s R 1 R 2 R 3 C 1 C 2 ) ]
3
Integrator A2
2
[ s R 1 R2 R3 R4 C1 C2 C 4 + s R1 R2 R3 C1 C2 + R4 ]
= ----------------------------------------------------------------------------------------------------------------3
2
[ s R1 R2 R3 R4 C 1C 2C 4 + s R1 R2 R3 C 1C 2 ]
3
2
N (s ) = s R1 R2 R3 R4 C 1C 2C 4 + s R1 R2 R3 C 1C 2 + R 4
STEP 2: SOLVE N(s) = 0
N(s )
a0
a1
a2
a3
=
=
=
=
=
Gain A 1 = – 1 ⁄ ( 2πfR 1 C 1 )
Pole f p1 = 1 ⁄ ( 2πR 1 C 1 )
Gain A 2 = – 1 ⁄ ( 2πfR 2 C 2 )
Pole f p2 = 1 ⁄ ( 2πR 2 C 2 )
Integrator A3
Gain = – [ ( R 4 ⁄ R 3 ) ( 1 ⁄ ( sR 4 C 4 + 1 ) ) ]
Gain ≅ – R 4 ⁄ R 3
STEP 4: VERIFY LG ≥ 1
The zeros of the characteristic equation are determined
by using the Routh stability test.
3
P = 2πRC
2
a 0 s + a 1 s + a 2 s + a3
R1 R2 R3 R4 C1 C2 C 4
R1 R2 R3 C1 C2
0
R4
Routh Stability Test Coefficient Array
row s3 a0 + a2 = 0
row s2 a1 + a3 = 0
Row s3 produces a trivial solution (ωο = 0):
3
a 0 ( jω o ) + 0 = 0
 2003 Microchip Technology Inc.
Assume:
1.
2.
3.
R1 = R2 = R
C1 = C2 = C
R3 = R4
A1 =
LG =
V1 =
V2 =
V3 =
A 2 = A3 = 1
A1 × A2 × A3 = 1
A1 × V3
A2 × V1
A3 ×V2
Note that a voltage limit circuit should be added if rail-torail input/output operational amplifiers are not used, or if
the gain of the inverter is not equal to one (i.e., R3 ≠ R4,).
A limit circuit is required to prevent the frequency error
that will result from the saturation delay time of the
amplifiers.
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ABSOLUTE STATE-VARIABLE
Test Results
The components used in the evaluation design are
listed below. Note that the capacitive sensor (i.e., C1
and C2) was simulated with discrete capacitors.
The measured and calculated oscillation frequency is
shown in Table 1. Figure 5 shows the oscillation waveform when C1 = C2 = 220 pF. The error in the measured
oscillation is attributed to the accuracy of the test
equipment.
R1 = R2 = R 6 = R 7 = 32.7 kΩ
R3 = R4 = 10 kΩ
R8 = 1 kΩ
R9 = 1 MΩ
C1 = C2 = see Table 1
C4 = 18 pF
VDD = 5.0V
A1, A2, A3 ≡ MCP6024 (quad RRIO,
GBW = 10 MHZ)
A4 ≡ MCP6541Push-Pull Output
Comparator
TABLE 1:
ABSOLUTE OSCILLATOR TEST RESULTS
Capacitor
Values
Calculated Oscillation
Period (Frequency)
Measured Oscillation
Period (Frequency)
C1 = C 2 = 47 pF
9.66 µs (103.6 kHz)
10.0 µs (100.0 kHz)
C1 = C 2 = 56 pF
11.5 µs (86.9 kHz)
12.0 µs (83.3 kHz)
C1 = C 2 = 82 pF
16.9 µs (59.4 kHz)
16.6 µs (60.2 kHz)
C1 = C2 = 100 pF
C1 = C2 = 150 pF
20.6 µs (48.7 kHz)
21.0 µs (47.6 kHz)
30.8 µs (32.6 kHz)
30.0 µs (33.3 kHz)
C1 = C2 = 220 pF
45.2 µs (22.1 kHz)
46.0 µs (21.7 kHz)
Output of A3 Amplifier (V 3)
Comparator Output (VO)
FIGURE 5:
Absolute Oscillator Test Results (C1 = C2 = 220 pF).
 2003 Microchip Technology Inc.
DS00866A-page 6
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RATIO STATE-VARIABLE
OSCILLATOR
A Bode plot of the differentiator stage is provided in
Figure 8. The values of resistors R 3, R4 and R5 are
selected to set the break frequencies of the differentiator stage so that the gain of the stage is equal to -C3/
C4 at the oscillation frequency. Resistor R5 is also
used to provide a DC current path around capacitor C3
in order to initiate oscillation at power-up.
Circuit Description
The schematic of the ratio circuit is shown in Figure 6.
This circuit consists of two integrators and a differentiator circuit. The integrators formed by amplifiers A1 and
A2 are identical to the integrators used in the absolute
circuit. The differentiator stage is formed by amplifier
A3, resistors R3, R4 and R5, and the sensor capacitors
C3 and C4 to provide a 180° phase shift. The comparator (A4) used to convert the sinewave output to a
square wave digital signal is identical to the absolute
oscillator circuit.
VDD /2
VDD
Q1
C1
R1
C2
A1
V1
VDD/2
C4
R2
VDD/2
R6
C3
VDD /2
R4
R3
A2
R5
V2
VDD/2
R7
A3
V3
R9
R8
A4
VDD/2
FIGURE 6:
Ratio Oscillator Schematic.
V1
–1
A 2 = ----------------sR C
2 2
A
-
3
=
R ( sR C + 1 )
4
5 3
--------------------------------------------------------------------------------------( sR 3 C 5 C 3 + R 3 + R 5 ) ( sR 4 C 4 + 1 )
V3
V2
–1 A 1 = ---------------sR 1 C 1
FIGURE 7:
Ratio Oscillator Signal Flow Diagram.
 2003 Microchip Technology Inc.
DS00866A-page 7
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Gain (dB)
Assumptions
 C 3
20log 10  -------
 C 4
1 F z1 = -------------------2 π R5 C3
1.
2.
3.
Oscillation
Range
0
 R4 
20log  -------------------10  R + R 
3
5
FIGURE 8:
C3 > C4
R3 < R4 < R5
R
R3 < < R5 and R4 ≈ -----510
Frequency (Hz)
1
F p1 = --------------------2πR C
4 4
F
p2
1 = -------------------2π R 3C 3
Bode Plot of Differentiator Amplifier.
Voltage Limit Circuit
The ratio oscillator uses a limit circuit to accommodate
the varying gain requirement of the circuit. It may be
necessary to add a voltage limit or clamp circuit to the
oscillator to prevent the amplifiers from saturating and
avoid slew rate limitations. The voltage limit circuit
formed by PNP transistor Q1 is used to create the maximum voltage limit. The clamping voltage of the limit
circuit is provided below:
EQUATION:
V Max_Limit = V Q1_base + V Q1_base-to-emitter
V Max_Limit ≅ V Q1_base + 0.7V
 2003 Microchip Technology Inc.
In single-supply applications, it is not necessary to use
both maximum and minimum limit circuits. Only one of
the limit circuits is required due to the symmetry of the
sinewave that is centered around the virtual ground
voltage at the non-inverting terminal of the amplifiers
(VDD/2). In the reference design of Figure 6, VDD is
equal to 5V and VQ1-base is equal to 2.5V. Thus, the
oscillation waveform at V2 will swing from 1.8V to 3.2V
or 2.5V ±0.7V.
Note that the transistor adds a small capacitance (CQ1)
to the integrator capacitor of the circuit (C2). If C2 is relatively small, the effective capacitance of the limit circuit (CLimit) can be reduced by connecting a diode in
series between the emitter junction and the output of
the amplifier (i.e. 1/CLimit = 1/CQ1 + 1/CDiode).
DS00866A-page 8
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RATIO STATE-VARIABLE
Design Equations
STEP 1: FIND LG AND ∆S
A
A
A
T ( s ) = ----------------- = ------ = -----------1 – LG
∆s
N(s)
----------D(s)
The loop-gain is found by breaking the loop in the signal flow diagram of Figure 7, as shown below.
A1
A2
V1
A3
V2
V3
A 1 = – 1 ⁄ ( sR 1 C 1 )
A 2 = – 1 ⁄ ( sR 2 C 2 )
A 3 = Z4 ⁄ Z 3
= – [ ( R 4 || C 4 ) ⁄ ( R 3 + ( R 5 || C 3 ) ) ]
= – [ R 4 ( sR 5 C 3 + 1 ) ⁄ ( sR 3 R 5 C 3 + R 3 + R 5 ) ( sR 4 C 4 + 1 ) ]
LG = A1 x A 2 x A 3:
[ – R 4 ( sR 5 C 3 + 1 ) ]
LG = [ – 1 ⁄ ( sR 1 C 1 ) ] [ – 1 ⁄ ( sR 2 C 2 ) ] ---------------------------------------------------------------------------------( sR 3 R 5 C 3 + R 3 + R 5 ) ( sR 4 C 4 + 1 )
4
3
2
= – ( sR 4 R5 C3 + R 4 ) ⁄ s R1 R 2 R 3 R 4 C1 C2 C 3 C 4 + s ( ( R 1 R 2 C1 C 2 ) ( R 3 R 5 C 3 + R 3 R 4 C 4 + R4 R5 C 4 ) ) + s ( R1 R2 C 1 C 2 ) ( R3 + R5 )
∆s = N(s) / D(s) = 1 - LG:
 s 4 R R R R R C C C C + s 3 ( R R C C ) ( R R C + R R C + R R C ) + s 2 ( R R C C ) ( R + R ) + sR R C + R
4 5 3
4

1 2 3 4 5 1 2 3 4
1 2 1 2
3 5 3
3 4 4
4 5 4 
1 2 1 2
3
5
∆s = ------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------ s 4 R R R R R C C C C  + s 3 ( ( R R C C ) ( R R C + R R C + R R C )) + s 2 ( R R C C ) ( R + R )

1 2 3 4 5 1 2 3 4
1 2 1 2
3 5 3
3 4 4
4 5 4
1 2 1 2
3
5
STEP 2: SOLVE N(s) = 0
The zeroes of the characteristic equation
determined by using the Routh stability test:
Ns
a
0
a1
a2
a
3
a4
=
=
=
=
=
=
are
4
3
2
a 0 s + a1 s + a2 s + a 3 s + a4
R R R R R C C C
1 2 3 4 5 1 2 4
( R1 R 2C 1 C2 )( R 3 R5 C3 + R3 R4 C 4 + R 4 R5 C4 )
( R1 R 2C 1 C2 )( R 3 + R 5)
R R C
4 5 3
R4
Row s 4 produces an equation that cannot be solved
with simple algebra. Therefore, the next row is
analyzed:
4
2
a 0 ( jω o ) + a 2 ( jω o ) + a 4 = 0
Routh Stability Test Coefficient Array
row s4 a0 + a2 + a 4 = 0
row s3 a1 + a3 = 0
 2003 Microchip Technology Inc.
DS00866A-page 9
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The procedure continues by analyzing row s3 to determine when the row equation is equal to zero.
3
2
a s + a s = s a s + a  = 0
1
3
 1
3
STEP 3: SUB-CIRCUIT DESIGN EQUATIONS
Integrator A1
Gain A 1 = – 1 ⁄ ( 2 π fR 1 C 1 )
Pole f = 1 ⁄ ( 2 π R 1 C 1 )
Let s = jωο
2
jω o  – a 1 ω o + a 3 = 0


a
2
3
ω o = -----a1
R R C
4 5 3
= ---------------------------------------------------------------------------------------------------------------------( R1 R2 C1 C 2 ) ( R 3 R5 C3 + R 3 R 4 C4 + R 4 R 5 C 4 )
( R 4R 5C3 )
ω o = --------------------------------------------------------------------------------------------------( R 1 R2 C1 C2 ) ( R3 R 5 C3 + R 3 R4 C4 + R4 R5 C4 )
Integrator A2
Gain A 2 = – 1 ⁄ ( 2 π fR 2 C 2 )
Pole f = 1 ⁄ ( 2 π R 2 C 2 )
Differentiator A3
DC Gain = – R 4 ⁄ ( R 3 + R 5 )
C3
Gain at Oscillation = – -----C4
Pole f p1 = 1 ⁄ ( 2 π R 4 C 4 )
Pole f p2 = 1 ⁄ ( 2 π R 3 C 3 )
Zero fz = 1 ⁄ ( 2 π R 5 C 3 )
1/2
If:
R1 = R2 = R
C1 = C2 = C
Then:
1.
2.
STEP 4: VERIFY LG ≥ 1
2π
P = -----ωo
Assume:
R 3C4
C4
R3
R 5C3
C3
R
P = 2πRC ----------- + ----- + ----If:
1.
2.
R5 >> R3
R4 >> R3
4
1/2
1.
2.
R5 >> R3 and R4 >> R3, then A3 = - C 3 / C4
V2 = VMax_Limit (i.e. place limit circuit at A 2)
Next, calculate the voltages at the output of each
amplifier starting at V 2.
V 2 = V Max_Limit
Then:
C
P ≅ 2πRC -----4
C3
1/2
C
V 3 = A 3 × V 2 = -----3- × V Max_Limit
C4
1 -×V
V 1 = A 1 × V 3 = --------------------3
2 π fR 1 C 1
Oscillation will be sustained if:
A 2 × V 1 ≥ V Max_Limit
1
 --------------------- × V 1 ≥ V Max_Limit
 2 π fR C 
2 2
 2003 Microchip Technology Inc.
DS00866A-page 10
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RATIO STATE-VARIABLE
The measured and calculated oscillation frequency is
shown in Table 2. Figure 9 shows the oscillation waveform when C3 = 100 pF and C4 = 47 pF. Note that the
voltage limit circuit adds distortion to the waveform of
amplifier A2. In most sensor applications, waveform
distortion is inconsequential because the measurement is proportional to frequency and not the amplitude
of the oscillation.
Test Results
R1 = R2 = R 6 = R 7 = 32.7 kΩ
C1 = C2 = 220 pF
C3 = C4 = see Table 2
R3 = 5 kΩ
R4 = 3.3 MΩ
R5 = 10 MΩ
R8 = 1 kΩ
R9 = 1 MΩ
VDD = 5.0V
A1, A2, A3 ≡ MCP6024 (quad RRIO,
GBW = 10 MHZ)
A4 ≡ MCP6541 Push-Pull Output
Comparator
Q 1 ≡ 2N3906
TABLE 2:
RATIO OSCILLATOR TEST RESULTS
Capacitor
Values
Calculated Oscillation
Period (Frequency)
Measured Oscillation
Period (Frequency)
C3 = 47 pF
C 4 = 47 pF
45.2 µs (22.1 kHz)
47.0 µs (21.3 kHz)
C3 = 47 pF
C4 = 100 pF
65.9 µs (15.2 kHz)
68.0 µs (14.7 kHz)
C3 = 47 pF
C4 = 220 pF
97.8 µs (10.2 kHz)
98.4 µs (10.2 kHz)
C3 = 56 pF
C 4 = 47 pF
41.4 µs (24.2 kHz)
43.0 µs (23.3 kHz)
C3 = 56 pF
C4 = 220 pF
89.6 µs (11.2 kHz)
92.0 µs (10.9 kHz)
C3 = 100 pF
C4 = 47 pF
31.0 µs (32.3 kHz)
33.0 µs (30.3 kHz)
C3 = 100 pF
C4 = 220 pF
67.0 µs (14.9 kHz)
70.0 µs (14.3 kHz)
Output of A 2 Amplifier (V2)
Output of A 3 Amplifier (V3)
FIGURE 9:
Ratio Oscillator Test Results (C3 = 100 pF, C4 = 47 pF).
 2003 Microchip Technology Inc.
DS00866A-page 11
AN866
APPENDIX A: MASON’S REDUCTION
THEOREM
APPENDIX B: ROUTH STABILITY
TEST
The oscillation frequency is determined by finding the
poles of the denominator of the transfer equation T(s)
or equivalently the zeroes of the numerator N(s) of the
characteristic equation ∆(s). Mason’s theory is especially useful for analyzing oscillators that have multiple
feedback loops.
The Routh Stability Test [5] can be used to test the
characteristic equation to determine whether any of the
roots lie on the imaginary axis. Routh’s test consists of
forming a coefficient array from N(s). Next, the procedure substitutes s = jωο for s, and the summation of the
row is set to zero. If the row equation produces a nontrivial solution for ωο, the procedure is complete and the
frequency of oscillation is equal to ωο. If the row equation does not yield an equation that can be solved for
ωo, the procedure continues with the next row in the
Routh array. This technique arranges the numerator of
the characteristic equation (i.e., denominator of the
transfer equation) into the array listed below.
Mason’s theorem [5] states that the transfer function
from input X to output Y is:
EQUATION:
Σ i P i ∆s i
Σ i P i ∆s i
Y
T ( s ) = --- = ------------------- = ------------------X
∆s
N (s )
----------D (s )
EQUATION:
N(s) = aosn + a1sn-1 + a2sn-2 + a 3sn-3 +...
+ an-1s + an
Where:
Pi = the direct transmittance or path form input X to
output Y
∆si = the system determinant. (∆si = 1 if P i touches
all of the loops)
Note for simplicity, only the first three rows of the Routh
coefficient array are shown below.
∆s = 1 - ΣLj + ΣLkLl - ΣLmLnLo +....
ΣLj = the sum of all loops (i.e. loop gains)
ΣLkLl = the sum of products of pairs of
non-
touching loops
sn
a0
a2
sn–1
a1
a3
a4.... an
a5.... an–1
sn–2
b1
b2
b3.... b.n–2
where the coefficients b1, b2, b3, etc., are defined as:
ΣLmLnLo = the sum of products of gains of non–
b1 = (a1a2 - a0a3) / a1
touching loops taken three at a time.
b2 = (a1a4 - a0a5) / a1
b3 = (a1a6 - a0a7) / a1
The Routh stability criterion states:
1.
2.
3.
4.
 2003 Microchip Technology Inc.
A necessary and sufficient condition for stability
is that the first column of the array does not contain sign changes.
The number of sign changes in the entries of the
first column of the array is equal to the number
of roots in the right half s–plane.
If the first element in a row is zero, it is replaced
by ε, and the sign changes when ε → 0 are
counted after completing the array.
The poles are located in the right half plane or
on the imaginary axis if all the elements in a row
are zero.
DS00866A-page 12
AN866
CONCLUSION
BIBLIOGRAPHY
Operational amplifier oscillators can be used to produce a frequency that is proportional to resistive and
capacitive sensors. Design equations defining the
oscillation frequency are readily available for several
common oscillators, such as Wein bridge and phase
shift oscillators. However, detailed design equations
that show the relationship of the resistors and capacitors are generally not available. Thus, there is a need
for a design procedure that derives the equations in
order to select the resistor and capacitor components
that maximize the accuracy of the oscillation frequency.
The design procedure was demonstrated by analyzing
two state-variable oscillators for capacitive sensing
applications.
1.
2.
3.
4.
5.
6.
 2003 Microchip Technology Inc.
Celma, C., Martinez, P., and Carlosens, A.,
“Approach to the Synthesis of Canonic RC–
Active Oscillators Using CCII”, IEE Proc. Circuits, Devices and Systems, Vol. 141, No. 6,
December 1994, pp. 493–497.
Lepkowski, J. and Young, C, “AND8054 Designing RC Oscillator Circuits with Low Voltage Operational Amplifiers and Comparators
for Precision Sensor Applications”, ON
Semiconductor, Phoenix, Arizona, 2002.
Martinez, P., Aldea C. and Celma, S., “Approach
to the Realization of State-Variable Based Oscillators”, IEEE International Conference on Electronics Circuits and Systems, Vol. 3, 1998,
p.139–142.
Sidorowicz, R., “An Abundance of Sinusoidal
RC–Oscillators”, Proc. IEE, Vol. 119, No. 3,
March 1972, pp. 283–293.
Truxal, J., “Introductory System Engineering”,
McGraw–Hill, N.Y., 1972.
Van Valkenburg, M., “Analog Filter Design”,
Saunders College Publishing, Fort Worth, 1992.
DS00866A-page 13
AN866
NOTES:
 2003 Microchip Technology Inc.
DS00866A-page 14
Note the following details of the code protection feature on Microchip devices:
•
Microchip products meet the specification contained in their particular Microchip Data Sheet.
•
Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the
intended manner and under normal conditions.
•
There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our
knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip's Data
Sheets. Most likely, the person doing so is engaged in theft of intellectual property.
•
Microchip is willing to work with the customer who is concerned about the integrity of their code.
•
Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not
mean that we are guaranteeing the product as “unbreakable.”
Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our
products. Attempts to break microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts
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Information contained in this publication regarding device
applications and the like is intended through suggestion only
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No representation or warranty is given and no liability is
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DS00866A-page 15
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