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Transcript
ESSCIRC
A LowVoltage Fully Balanced OTA with Common Mode Feedforward and Inherent Common
Mode Feedback Detector
Ahmed Nader Mohieldin Edgar SnchezSinencio Jos SilvaMartnez
Department of Electrical Engineering, Texas AampM University, College Station, Texas ,
USA Abstract
A pseudo differential fully balanced CMOS OTA architecture with inherent common mode
detection is proposed. Through judicious arrangement, the commonmode feedback circuit
can be economically implemented. The OTA achieves a good CMRR and is suitable for low
voltage operation. As an example of the applications of th the proposed OTA, a MHz order
linear phase OTAC filter is presented. The measured group delay ripple is , the filter dynamic
range is dB. It consumes .mW per complex poles. proposed. It is shown that a separate
CMFB circuit can be avoided with appropriate arrangement of cascaded pseudo differential
OTAs, for example in a filter. This approach takes advantage of the OTAs used for the
differential mode operation to render low common mode gain without extra circuitry. The
proposed OTA has inherently the common mode detector, hence the CMFB is economically
realized.
. Proposed OTA architecture
In general operational transconductance amplifiers can be divided into two main groups Fully
Differential FD and Pseudo Differential PD. FD is based on a differential pair with tail current
source and PD is based on two independent inverters without tail current source. Avoiding
the voltage drop across the tail current source, in a PD structure, allows achieving wider
linear input range and makes the architecture attractive for low voltage applications.
Removing the tail current source, however, results in a poor rejection to common mode
noise. Using a common mode feedforward CMFF technique, ACM at low frequency is in the
order of unity . A separate transconductance is often used for common mode detection . This
approach adds more load to the driving stage due to the connection of two input transistors,
one for the differential transconductance and the other one for the common mode
transconductance, thus doubling the input capacitance. The CMFF improves the rejection to
common mode signals but it is incapable to fix the DC common mode output voltage. Figure
shows the circuit implementation of the proposed OTA. It can be described as a conventional
three current mirror single ended OTA, without the tail current, plus the additional branches
dashed lines. Thus the OTA becomes fully balanced and fully symmetric with enhanced
features to be described.
. Introduction
In general, a fully differential structure has an improved dynamic range over its single ended
counterpart. This is due to better common mode noise rejection, better distortion
performance, and increased output voltage swing. One important disadvantage of fully
differential structures is the need of an extra common mode feedback CMFB circuit. A CMFB
circuit is needed for two reasons To fix the common mode voltage Vcm at different high
impedance nodes. to suppress the common mode signal components, on the whole band of
differential operation that tends to saturate different stages. A CMFB circuit is classically
performed by means of a CMFB loop. The CMFB loop has to be designed carefully to avoid
potential stability problems. This often increases the complexity of the design, the power
consumption, and the silicon area used. The frequency response of the differential path is
also degraded due to the added parasitic components involved in conventional CMFB
schemes. In some cases and due to the system configuration, the CMFB requirements can
be relaxed. For instance, if the common mode gain is sufficiently small, as would be the case
of a conventional differential pair based OTA with a tail current, the CMFB bandwidth might
be reduced. Its primary function will be to set the common mode voltage and it will not
consume much power. Nevertheless, for good power supply rejection, a wide bandwidth
CMFB is needed. In other cases, when the output is loaded by low impedance, the DC
output voltage is well defined and assuming that the common mode gain is small enough,
there may be no need for the CMFB circuitry as would be the case of filters built using lossy
integrators . In this work, a practical pseudo differential OTA is
. Inherent common mode detection
It can be shown, using the quadratic IV characteristics of a transistor neglecting all short
channel effects, that current I I in Figure provides the information about the common mode
level of the inputs as follows v K W I I . P V dd V icm VTP d L where Vi Vicmvd/, and Vi
Vicmvd/.
the effective carrier mobility eff is no longer constant and is a function of both the
longitudinal and transversal electric fields. CMFB can be implemented by judiciously
connecting at least two of the proposed OTAs.V . eff/E/EC. we have I VSS Figure . CMFB
and CMFF arrangements Although the CMFF provides rejection to the common mode
components at the output. For short channel devices. sat is the saturation carrier drift
velocity. with minimum length L. due to the cross product of differential and common mode
signals. Note that the value of in must be modified to include the effect of the transversal
electric field. the drain current I of a transistor in saturation region is given by Vout I I I I Vi I M
M ViI VoutI I I I I I M M M M VZ M M VSS Figure . Thus the excess phase is given by gm s
tan nd M M VX M M Note that CZ as well as gm are functions of W . OTA with CMFF and
CMFB MBM . yields . Due to transistor mismatches. is the lowfield mobility. Linearity issues
Note from that the linearization of the output current depends on the cancellation of the
quadratic components of the individual currents I and I. VDD I I I I . even neglecting all
mismatches. L can be chosen to be minimum to reduce the excess phase although it will
sacrifice the DC gain. Cox is the oxide capacitance per unit channel area. Using taylor series
expansion. Thus. and fedback by M . with VdVPeakcost. and /LEC. This can be achieved by
adding the four transistors M M and M M. Proposed OTA architecure . then VSGVTPVovvd/
VSDVDCkvd/ where VovVddVicmVTP is the overdrive voltage. is the output impedance
constant. E is the longitudinal electric field. Considering the degradation of mobility due to
these effects and the channel length modulation effect. and EC is the critical electric field. In
the proposed OTA architecture. as shown in Figure . The signal common mode components
are suppressed by the action of the CMFF. L is the device electrical channel length.V and .
this cancellation is not perfect and some second order distortion will result. thus involves an
iterative design procedure. Frequency response and excess phase The differential mode
transconductance of the proposed OTA. yielding for MBM I I K W I B P Vdd Vicm VTP vd L K
W I od I I B P Vdd Vicm VTP vd Gm v d L VDD iod gm g m g m vd g m sCZ s / nd where CZ
is the total parasitic capacitance at node VZ. to be . can be written as K W V DC V ov v d / vd
I P L V ov v d / where /Vov.Note that after extracting the common mode information. The DC
level of the output is sensed at node VX of the next OTA. VDD VDD VX from next stage VX
from next stage Vref M M M M VX M M M VDD Vout I Vi M M Vi I Vout I VY I M M M M M M
VZ M M M VY K P W V SG VTP VSD L V SG VTP where KPeffCox. for MM. and k/VDC. the
common mode current I I/ is subtracted at the output nodes of the OTA. we still need to fix
the DC common mode output voltage. The main nonlinearity contribution for the proposed
OTA structure is due to short channel effects. If a differential signal is applied. respectively.
Substituting and in . the common mode signal has to be suppressed as much as possible .
ECsat/. Even order harmonics can also appear. This arrangement has the advantage that
the differential mode signals and the common mode signals share basically the same loop. id
is then compared with the proper current id fixing the DC output level to the required value.
and the transconductance of the CMFB loop is the same as the differential mode
transconductance. and k is the gain from the input to node Vz see Figure . The value of and
have been determined for the used technology by a best fit to the simulated device
characteristics.
g m g sC X ACM s g m sCZ g m sC X GCMFB s g sC L A direct tradeoff between linearity
and frequency response excess phase is observed from . CMFB CMFB g m sC Z g m sC X g
o sC L g o sC L Note that the flow of the common mode information from one stage to the
other is done in a natural way. Increasing Vov also improves the linearity at the expense of
power consumption. Using and . . Consider only the integrated thermal noise power. yields
HD .m CMOS process. .Vov Vov Vov . Note that ACM. ACMFB should be made as high as
possible at DC. where CX and CZ is the total parasitic capacitance at node VX and node VZ
respectively. the last stage of the first biquad. is much less than unity. This can be
accomplished by increasing the length of the channel /L which at the same time increases
the parasitic capacitances. and consequently the SNR. and the last stage of the second
biquad do not need to have a CMFB since their outputs are common with other OTAs which
will automatically fix the output DC level of the output. the common mode gain. where fMHz
and fdB. lt. at low frequency. Measurement results The filter has been fabricated in AMI . The
transconductance gm needs also to be maximized to reduce the noise contribution of
subsequent transistors. The CMFB is idealy transparent to differential signals and acts at low
frequencies as a resistor of value /gCMFB gCMFBgm is the small signal CMFB
transconductance for common mode signals. the output common mode level needs to be
fixed only once for any number of OTAs sharing the same output. which is also used for
differential mode operation.BW . Filter architecture For a given HD. some of the fully fledged
OTAs with CMFB and CMFF may be replaced by ones with CMFF only as that shown in
Figure . The smaller is. In this design B is chosen to be unity to maintain the ability for high
frequency operation. the maximum input voltage. where V nKT. Increasing B will reduce the
noise. Similarly. For instance. CL is the load capacitance.HD . CMFFCMFB g m A C A g m A
CMFF Vin g m A g m A VBPCommon Mode Information VLP where BW is the equivalent
noise bandwidth.VPeak Vov Vov Vov .BW .BW/gm. ACM. Increasing gm will also reduce the
noise.xf.KT . the payment is in the power consumption in this case. It . m g m g m g m B g m
B C A CMFF Fig. gmB. Thus. Noise performance The encircled transistors in Figure
contribute to common mode noise only due to the symmetric configuration and thus their
effect can be neglected.HD Vov Vov Vov V Peak Neglecting channel length modulation
effect for k. increase the effective transconductance and consequently gCMFB. is computed
as g m. The requirement is a group delay error less than for frequencies up to . m g m VBP
CMFF g mB VLP CMFFCMFB CMFF CMFFCMFB g mB g m B CB g mB C B VLP VBP
Figure . VBP VLP Vin CMFFCMFB Fig. CMFB loop design considerations The open loop
gain of the CMFB. for MM. . The CMFB loop . The accuracy of the current mirror is also less
for large values of B. can increased by increasing VOV.g m SNR log log g g m g m V n rms .
and its bandwidth should be as high as the differential mode bandwidth. The chip micrograph
is shown in Figure . The block diagram of the filter with the required CMFB arrangement is
shown in Figure .xf.Vin rms Vov Vov Vov is compensated for stability purposes by the load
capacitance CL. Thus for MM. and VDCgtVov and substitute the value of in . Filter
architecture A fourth order linear phase BesselThompson OTAC filter has been
implemented. the input referred noise becomes V n rms g g m g m KT . gmA. the following
expression for the SNR can be obtained Vin rms . g is the output conductance. the wider the
linear range for a given HD. is given by g m g m g m G s A s . but it directly deteriorates the
excess phase as CZ increases. this is a result of the action of both CMFB and CMFF circuits.
. The common mode level is sensed only once per output.
at MHz. The proposed OTA features very attractive linearity properties and inherent
common mode detection. Chip micrograph Ripple on lt group delay f lt . no. vol. The group
delay ripple is for frequencies up to the dB frequency. November . pp. dB. IEEE J.J. The
same principle can be applied to any OTA with CMFF to incorporate the detection of the
common mode information of the previous stage for CMFB stabilization. Measured results
assured the good performance of the proposed transconductance building block and its
suitability for high frequency and low voltage operation. . no.V . Nov. th A MHz order linear
phase OTAC filter has been implemented in . IEEE J. Spectrum for a mVpp MHz input signal
area of . The filter consumes mA from a . .fdB Max Input mVpp THD dB Output noise .MHz
GmC LPF. IEEE J. SolidState Circuits. .. and W. Equirriple This Work th Order Bessel f dB
MHz MHz lt f lt fdB mVpp dB N/A gt dB MHz lt f lt fdB mVpp dB Vrms dB Figure . pp. IEEE
Custom Integrated Circuits. and is dB. July . . Castello. March . The measured CMRR. pp. . .
The measured group delay ripple is about for frequencies up to MHz. V power supply.m
CMOS process and occupies an A. A V MHz BiCMOS pseudo differential continuoustime
filter. respectively. can be tuned by changing Vicm. . pp. IEEE International Symposium on
Circuits and Systems. equiripple linear phase filter. Steyaert. Group delay .. V. Shoval. W.
Avoiding common mode feedback in continuoustime GmC filters by use of lossy integrators.
Snelgrove and A. . Equirriple th Order .. M. Figure shows the measured phase response of
the filter. Rezzi. A balanced . Tanimoto.mm only... Sansen. N. Dehaene. A V MHz
continuoustime seventh order . no. References . Phase response Figure . Baschirotto. no.
Arai. SolidState Circuits.m CMOS Figure . and T. A MHz standard CMOS pulse equalizer for
hard disk read channels. respectively. The transconductance. Figure shows the filter output
spectrum for a differential input signal of amplitude mVpp and frequency MHz. Filter
Performance Parameters Filter type and order th Order . V mA . vol.m CMOS Include filter
boost V mA . SolidState Circuits. PSRR . Table contains a summary of different filter
performance parameters compared with previously reported works in and . dB. and R. and
dB. Figure . A Vpp linear input range fully balanced CMOS transconductor and its application
to a . IEEE J. Contreras. Conclusions A pseudo differential fully symmetric fully balanced
OTA architecture has been presented. T. H.. pp. M. vol.S. Schaumann. pp. . and R. . This is
done at minimum cost of area and power consumption. This corresponds to dB of dynamic
range DR for . Table .occupies a small area of x m . . The total inband integrated output
noise is about Vrms. Balan. SolidState Circuits. and consequently the dB frequency of the
filter. Itakura. vol. THD at MHz. A. and PSRR at MHz is dB. This is achieved by changing the
CMFB reference voltage Vref see Figure . F. and dB. Ueno. W. It has been demonstrated
that the structure made it easy to incorporate the CMFB arrangement.m BiCMOS . The
group delay is shown in Figure . . m CMOS transconductanceC filter tunable over the VHF
range. T. Rao. mVrms DR dB Power supply V Current drain mA Technology . vol.