Download Gerhard Mercator Universität Gesamthochschule Duisburg

Gerhard Mercator
Universität
Gesamthochschule
Duisburg
Department of
Optoelectronics
Annual Report 1996/97
Gerhard-Mercator-Universität
Gesamthochschule Duisburg
Fachbereich Elektrotechnik
Fachgebiet Optoelektronik
ZHO
Lotharstr. 55
D - 47057 Duisburg
Germany
Head:
Prof. Dr. rer. nat. D. Jäger
Tel:
+49-203-379-2340
Fax:
+49-203-379-2409
Email:
URL:
[email protected]
http://www.oe.uni-duisburg.de
Editor: R. Buß
4
The Center for Solid-State Electronics and Optoelectronis (ZHO)
After only nine month of construction the topping-out-ceremony of the new Center for Solid-State
Electronics and Optoelectronics (Zentrum für Halbleitertechnik und Optoelektronik, ZHO) has been
celebrated on June, 4th, 1997.
The center is planned to be the new “home” of the Department of Optoelectronics and the SolidState Electronics Department in the beginning of October 1998. It consitst of two parts: The cleanroom building with an area of approx. 470 m2 and the building for the offices and laboratories with an
area of approx. 1200 m2.
Pictures of the Center for Solid-State Electronics and Optoelectronics taken in Dec. 1997
5
Table of Contents
1 Foreword
7
2 Members of the Department
9
3 Research
3.1 Optical Networks
3.1.1
3.1.2
3.1.3
3.1.4
3.1.5
High-speed, high-power travelling-wave photodetectors
Fabrication and characterization of a travelling-wave photodetector
Simulation of the microwave generation of a travelling-wave photodetector
Determination of RF-equivalent circuit elements of travelling-wave photodetectors using network analysis
Polarization insensitive waveguide modulators on InP
3.2 Optical Interconnects and Processors
3.2.1
3.2.2
3.2.3
3.2.4
3.2.5
3.2.6
Neurotechnology: Retina Implant
Analysis of the optical energy and signal transfer module for an artificial vision
prosthesis
Development of an optical signal and energy transmission system
Infrared data link for rotating display-systems
Nonlinear hybrid GaAs/AlGaAs multilayer-heterostructures for high-speed
information processing
8 x 8 LED arrays integrated with 64 channel Si-driver circuits
3.3 Millimeterwave Electronics
3.3.1
3.3.2
3.3.3
Picosecond pulse generation on monolithic nonlinear transmission lines using
high-speed InP-HFET diodes
Millimeter wave power generation nonlinear transmission lines
Nonlinear RTD circuits for high-speed A/D conversion
3.4 Optical Sensor Systems
3.4.1
3.4.2
3.4.3
MQW-Electroabsorption-Modulator for application in a fiberoptic fieldsensor
Photovoltaic cells for fiber optic EMC - Sensor power supply
Time- and frequency-domain electro-optic field mapping of nonlinear transmission lines
11
11
11
14
17
20
25
29
29
32
35
38
41
48
51
51
55
59
62
62
64
66
6
TABLE OF CONTENTS
3.4.4
3.4.5
Characterization of monolithic microwave integrated circuits by heterodyne
electro-optic sampling
Development of an experimental setup for field probe measurements on
nonlinear transmission lines
3.5 Technologies for Optoelectronic Components and Systems
3.5.1
3.5.2
3.5.3
3.5.4
Development of a measurement system for the optical characterization of
full-colour-LED-displays
Opinion poll on the evaluation of the legibility of LED-based displays
Evaluation of possible improvements to enhance the UV-power efficiency of
a xenon flashlamp system
Construction of a flip chip device for bonding integrated circuits
4 Teaching activities
71
74
77
77
80
83
89
93
4.1 Lectures, excercises, and practical studies
93
4.2 Seminars and colloquia
95
4.3 Doctoral, Diploma, and Graduate theses
100
5 Publications and presentations
103
6 Guide to the Department of Optoelectronics
107
7
1 Foreword
The “Fachgebiet Optoelektronik” at the Gerhard-Mercator-Universität Duisburg has a tradition of research and teaching excellence dating
back to its establishment in 1989/90. This Report provides a summary of major research involvements and teaching activities reviewing
also publications and presentations by the members of the institute .
The last two years were characterized by a
further-broadening of our scientific networks and
additional projects funded by various external
institutions. A first key research area is microwave photonics with special emphasis on travelling-wave photodetectors, electro-absorption
modulators and nonlinear optoelectronic devices, where system aspects played a continuously increasing role. Special emphasis has further
been laid upon the two-dimensional electro-optical characterization of monolithic microwave
integrated circuits and high-speed devices. As
a third topic, nonlinear optics and optoelectronics in III-V-heterostructures and pulse compression in nonlinear MMICs were studied in detail.
Additionally to our activities within the Sonderforschungsbereich 254, mayor funding has been
provided by the “Retina Implant” project (EPIRET) and the collaborative programme on the
development of an EMC-field sensor, where our
Fachgebiet acts as the coordinator. We are already proud of looking back to a relevant exhibition during the Laser ‘97 Fair in Munich. Another remarkable event was the “International
Topical Meeting on Microwave Photonics
(MWP)” held at the moated castle “Schloß Hugenpoet” in September 1997 which was organized by our Fachgebiet, D. Jäger being simultaneously the Chair of the International Steering
Committee on MWP (details on next page).
As a result of their remarkable research work,
Dr.-Ing. G. David received a fellowship of the
Alexander von Humboldt-Stiftung, funding a twoyears stay at the University of Michigan. Moreover, Dr.-Ing. A. Stöhr was awarded a grant to
carry out research work at the Communications
Research Laboratories, Ministry of Posts & Telecommunications in Tokyo. Further, D. Jäger received the title “Professor Onorific” from the
University of Brasov/Romania and became the
Chair of the German IEEE/LEOS Chapter.
Besides the usual and obligatory courses, the
Fachgebiet Optoelektronik offered in 1997 a new
lecture “Einführung in die Multimediatechnik Technologien, Systeme, Anwendungen”. Moreover, the Institute was involved in the “Duisburg
Summerschool for Women”, the “Tag der Forschung” and the organisation of research activities on photonic bandgap materials in the framework of the “Forum Materialforschung” in our
university. Finally, we note with great pleasure
that T. Alder and D. Kalinowski have been
awarded the University Price for excellent diploma theses.
I wish to thank all friends inside and outside
the university for their continuous encouragement and assistance. Also, I would like to express my sincere thanks to all members of the
Institute for their efforts and contributions to our
success in optoelectronics.
Duisburg, September 1998
8
T
he 1997 International Topical Meeting
on Microwave Photonics (MWP’97)
has been held from September 3 through 5
in the historical buildings of the majestic 17th
century moated Castle “Schloß Hugenpoet”,
situated in the beautiful Ruhrtal countryside
in the south of Essen near Duisburg.
This 7 th Topical Meeting in the series on this
subject followed those in Cernay-la-Ville, France
(1994), Keystone, U.S.A. (1995), and Kyoto,
Japan (1996). It was the first one held in Germany and has been organized by the GerhardMercator-Universität-Duisburg. On September
3, the Meeting started with a Workshop entitled
“Photonic technologies for phased array antennas” where 6 invited papers addressed recent
results in this continously growing field of research. In the Plenary Session on September 4,
3 invited speakers, Dr. R. Heidemann, Alcatel
SEL AG, Stuttgart, Germany, Dr. M.J. Wale,
GEC-Marconi Materials Technology Ltd,
Northampton, U.K., and Dr. D. Novak, The University of Melbourne, Australia, presented lectures on the topic “Microwave Photonics:
Present and Future”. The regular conference
program consisted of 7 sessions on topics such
as Optical generation of microwave signals,
Optoelectronic modulators, mixers, and receivers, Microwave photonic systems, Fibre radio
networks, Modelling in microwave
photonics, and Microwave photonics
for measurements. Each session has
been opened by invited senior technologists from France, Hong Kong,
Japan, U.K., U.S.A., and Germany
having provided additional impetus to
this multi-disciplinary research area
of microwave photonics.
The whole program included furtheron a video session via internet with KDD in
Tokyo, Japan, a poster session and was completed by a postdeadline session. More than 100
papers have been submitted from 13 countries
showing the increasing interest of scientists and
engineers in this area. After careful evaluation
the Technical Program Committee has recommended 69 papers - 54 oral and 15 poster - for
presentation. Additionally, 4 papers have been
selected during the conference for the postdeadline session. 140 scientists and engineers have
registered for the Topical Meeting. In addition
to the technical program, a Partner’s Program,
a Welcome/Barbecue Party and a Gala Dinner
have been organized.
The Meeting has been sponsored by the
Deutsche Forschungsgemeinschaft (Bonn, Germany), Hewlett-Packard GmbH (Ratingen, Germany), Institut für Mobil- und Satellitenfunktechnik GmbH (Kamp-Lintfort, Germany), Lucent
Technologies (Allentown, P.A., U.S.A.), and the
Gerhard-Mercator-Universität Duisburg. Moreover it has been cooperatively sponsored by the
IEEE MTT-S and LEOS including the German
Chapters.
The organizers of MWP’97 look back to a very
fruitful conference and look forward to the upcoming meetings MWP’98 in Princeton, N.J.,
U.S.A., and MWP’99 in Melbourne, Australia.
9
2 Members of the Department
Department of Optoelectronics
ZHO, Lotharstr. 55
47057 Duisburg, Germany
fon: +49 203 379-2340
fax: +49 203 379-2409
Head of the Department
Jäger, Dieter
Prof. Dr. rer. nat.
Secretary
Gappa, Ulrike
Tempel, Karin
Optoelectronics
SFB 254
Scientists
Alles, Martin
Alder, Thomas
Braasch, Thorsten
Buß, Rüdiger
David, Gerhard
Groß, Matthias
Heinzelmann, Robert
Hülsewede, Ralf
Jäger, Irina
Kalinowski, Dirk
Knigge, Steffen
Kremer, Ralf
Redlich, Stefan
Schmidt, Manuel
Stöhr, Andreas
Wingen, Georg
Zumkley, Stefan
Dipl.-Ing.
Dipl. Ing.
Dipl.-Phys.
Dipl.-Ing.
Dr.-Ing.
Dipl.-Phys.
Dipl.-Ing.
Dipl.-Phys.
Ph. D.
Dipl.-Ing.
Dr.-Ing.
Dr.-Ing.
Dipl.-Ing.
Dipl.-Phys.
Dr.-Ing.
Dipl.-Phys.
Dr. rer. nat.
Guest Scientists
Dragoman, Mircea
Johnson, Roger
Lee, Chi
Mezentsev, Vladimir
Wendrix, Veronique
Prof. Dr.
Dipl.-Ing.
Prof. Dr.
Ph. D.
Dipl.-Ing.
Technicians
Mang, Sabine
Schedwill, Veronique
Slomka, Heinz
Students
Appenrodt, Nils
Balci, Senay
Baumeister, Thomas
Berger, Oliver
Boscher, Guido
Brings, Ludger
Bussek, Peter
Christoffers, Niels
Einweck, Michaela
Engel, Thomas
Ervens, Jutta
Hedtke, Ralph
Heinzdorf, Michael
Jabs, Mirco
Kampermann, Claus
Kreuder, Andreas
Lüdeke, André
Lotz, Oliver
Manh-Duc, Ngo
Meininger, Mark
Moeck, Jens-Peter
Neuhaus, Birgit
Ponellis, Bernd
Reintjes, Stefanie
Rogall, Michael
Spiegeler, Britta
Wenning, Michael
Weimann, Uwe
Ing. grad.
10
3 RESEARCH
3.1 Optical Networks
3 Research
3.1 Optical Networks
3.1.1 High-speed, high-power travelling-wave photodetectors
M. ALLES
R
ecently, new communication systems combining the advantages of
wireless transmission and fiber optics have
been proposed. Applications are transmission of traffic information, multimedia, or Internet access. These systems operate
usually at an optical wavelength of 1.55µm
at frequencies up to 60GHz. Since the photodetector should generate as much electrical power as possible, the travelling-wave
photodetector under investigation has to fulfill these requirements.
Introduction
Novel wireless millimeterwave communication systems have been proposed by several
groups, see for example [1-4]. These systems
use fiber optics to transmit the signals over large
distances. Fiber optic cables have very low attenuation of about 0.8dB per km, which is also
independent from the signal frequency, whereas loss of coaxial cables is about 100dB per km.
An optical source generates a heterodyne
signal at 1.55µm, where the difference frequency of the two optical carriers corresponds to the
electrical millimeterwave frequency. The data
signal is modulated on one optical carrier. The
optical heterodyne signal is distributed to an
antenna station using usual fibre optic components. At this antenna station, the photodetector converts the optical signal to a millimeter-
11
wave, which is amplified and transmitted to a
remote station. These communication systems
should be used for traffic information, digital video, multimedia, or Internet access.
This system approach leads to some important requirements for the photodetector. The
photodetector has to work at an optical wavelength of standard fiber optics, i.e. at 1.3/1.55µm.
In a further point, the photodetector should be
capable to operate in the high frequency regime
to generate electrical signals in the millimeterwave regime. Additionally, the photodetector has
to generate as much electrical output power as
possible to reduce the requirements of the
electrical amplifier used in the antenna station.
Usually, high-speed photodetectors are fabricated as lumped devices which are RC-time
limited. This means that high bandwidths can
only be reached if the device size is scaled down
to the micrometer regime. Dimensions of photodetectors operating at 100GHz are about 10µm2
[5]. Due to the small device size the photodetectors can only operate at low optical input powers to avoid saturation effects in the small volume. This limitation can be overcome if the
travelling-wave concept is considered for the
development of high-speed photodetectors [6].
High-speed travelling-wave photodetectors
The travelling-wave photodetector is fabricated as an optical waveguide which is coupled to
an electrical waveguide due to an optical absorption layer. The capacitance of the electrical
waveguide is compensated by the inductance
of the transmission line resulting in an electrical
bandwidth which is not RC-time limited. This
concept avoids scaling down the device dimensions. In contrast, the travelling-wave photodetector can be fabricated as a distributed device
in order to reduce optical saturation effects.
12
3 RESEARCH
The structure of the fabricated travelling-wave
photodetector is depicted in Fig. 1. The device
consists of an active region and a taper at the
end of the structure, used for hybrid integration
with electrical millimeterwave amplifiers. The
photodetector is MBE-grown on a semi-insulating InP-wafer for operation in the 1.3/1.55µm
regime. An InGaAlAs layer is used as an optical
waveguide. An InGaAs absorbing layer, leakage coupled to the waveguide, generates electron-hole pairs. Finally, an InAlAs-layer as a cladding layer and an InGaAs/InGaAlAs superlattice
as a Schottky-barrier enhancement layer are
grown.
Electrical waveguiding is achieved using a
coplanar transmission line. The outer conductors form ohmic contacts to the n+ doped region
of the optical waveguide, whereas the center
conductor is fabricated as a Schottky contact.
The depletion layer underneath the center con-
ductor separates the photo-generated electronhole pairs in the absorption region.
Since the travelling-wave photodetector uses
the interaction of optical and electrical waves,
the optical and the electrical phase velocity
should be equal (phase-matching condition).
This can be achieved due to the fact, that the
Schottky-contact generates slow-wave effects
on the electrical transmission line [7].
The efficiency of the travelling-wave photodetector can be calculated numerically using a
distributed equivalent circuit model for generation and propagation of electrical waves on the
coplanar transmission lines (see “Calculation of
the electrical millimeterwave generation of a travelling-wave photodetector“ in this annual report).
A distributed current source describes the impressed photocurrent per unit length due to electron-hole generation in the depletion layer. The
numerical calculation of the output power leads
Fig. 1: Sketch of the travelling-wave photodetector. Popt and Pel are the input optical and output electrical powers,
3.1 Optical Networks
to an electrical millimeterwave power of 18.4dBm at 40GHz for a travelling-wave photodetector with an active length of 1mm. The optical wavelength is 1.3µm and the input power is
0dBm per optical carrier.
The electrical output power of the travellingwave photodetector has been measured using
an optical heterodyne setup. Two tunable 1.3µm
Nd:YAG-lasers generate an optical heterodyne
signal with beating frequencies in the millimeterwave regime. The generated electrical signal is measured using a coplanar on-wafer probe
and a spectrum analyzer.
The fabricated travelling-wave photodetector
leads to an electrical output power of -19.7dBm
at a reverse bias voltage of 12V and a frequency of 40GHz using the heterodyne measurement
setup, which is in good agreement with the theoretically determined value.
Since the capacitance of this device is 1.2pF,
the resulting RC -time constant in a 50W system would lead to a 3dB-frequency of about 2.5
GHz, which is far below the measured frequency. This shows the validity of the travelling-wave
concept to overcome RC-time limitation.
13
ing the epilayer and for fabrication of the travelling-wave photodetector.
References
[1] R. Heidemann, R. Hofstetter, H. Schmuck,
“60GHz fibre-optic distribution technology for traffic information and multimedia“, IEEE, MTT-S
and LEOS Topical Meeting on Optical Microwave
Interactions, Proc. pp. 133-136, Abbaye des
Vaux de Cernay, 1994
[2] J. Park, K.Y. Lau, “Millimetre-wave (39GHz) fibre-wireless transmission of broadband multichannel compressed digital video“, Electron.
Lett., pp. 474-476, Vol. 32, 1996
[3] D. Wake, C.R. Lima, P.A. Davies, “Transmission
of 60-GHz signals over 100km of optical fibre
using a dual-mode semiconductor laser
sources“, IEEE Photon. Techn. Lett., pp. 578580, Vol. 8, 1996
[4] E. Boch, “High bandwidth mm-wave indoor local area networks“, Microwave Journal, pp. 152158, 1996
[5] K. Kato, A. Kozen, Y. Muramoto, Y. Itaya,
T. Nagatsuma, M. Yaita, “110-GHz, 50%-efficiency mushroom-mesa waveguide p-i-n photodiode for a 1.55-µm wavelength“, IEEE Photon.
Conclusion
Future wireless millimeterwave communication systems using optical heterodyne techniques are described. The requirements for highspeed photodetectors are discussed. Since high
bandwidth and large electrical output power are
needed simultaneously, travelling-wave photodetectors are under investigation. Up to now,
an electrical output power of -19.7dBm at a frequency of 40 GHz could be measured.
Acknowledgment
The author would like to thank U. Auer (Fachgebiet Halbleitertechnik/-technologie) for grow-
Techn. Lett., pp. 719-721, vol. 6, 1994
[6] D. Jäger, “Optical Information technology“, ed.
S.D. Smith and R.F. Neale, Springer-Verlag, pp.
328-333, 1193
[7] D. Jäger, “Slow-wave propagation along variable
Schottky contact microstrip line“, IEEE Trans. Microwave Theory and Techn., pp. 566-573, vol.
24, 1976
14
3 RESEARCH
3.1.2 Fabrication and characterization of a travelling-wave photodetector
Fabrication of travelling-wave
photodetectors
For the fabrication of a travelling-wave photodetector several etch steps, a polyimide step
V. WENDRIX AND M. ALLES
and metalization steps are needed [2]. The MBEgrown wafers, which contain the necessary layecently, high-speed travelling-wave
ers are depicted in Fig. 1 are fabricated in the
photodetectors are under investigaDepartment of Optoelectronics. The travellingtion as optoelectronic power converters used
wave photodetector contains in general three
in future communication systems. In this
different layers grown on a semi-insulating InP
work the fabrication of high-speed travellingwafer.
wave photodetectors is described. The fabAn InGaAlAs layer acts as optical waveguide,
ricated devices are characterized using
an InGaAs quantum well layer provides optical
standard measurement techniques. The elecabsorption, and, finally, an InAlAs layer is used
trical millimeterwave generation is deteras a cladding layer. The metalization of the travmined using an optical heterodyne setup with
elling-wave photodetector is fabricated as an
two 1.3µm Nd:YAG lasers.
electrical coplanar waveguide. The center conductor forms a Schottky contact to the InAlAs Introduction
layer. The outer metalization is evaporated on
For future communication systems combinthe InGaAlAs-layer. This metalization is alloyed
ing fiber optic links with wireless transmission
in order to form ohmic contacts to the n-doped
techniques, high-speed photodetectors are
semiconductor. The taper is fabricated at the
needed. These photodetectors should be used
output end of the travelling-wave photodetecfor hybrid integration with millimeterwave amtor. In order to reduce millimeterwave attenuaplifiers. A taper at the output end of the phototion and for a characteristic impedance of 50W,
detector facilitates flipchip- or wire-bonding with
the metalization of the taper has to be fabricatadditional devices. The fabrication of travellinged directly on the semi-insulating InP-wafer.
wave photodetectors with tapers is described in
Therefore, an insulation of the edge of the methis work [1].
sas is necessary to prevent a short circuit between the center conductor and the outer metalization.
The processing of the travellingwave photodetector starts with two
InAlAs
etch steps. The etching defines the
InGaAs SQW/MQW
lateral dimension of the optical
InGaAlAs : Si
waveguide and of the absorbing layInP
er, cf. Fig. 2.
R
Fig. 1: Travelling-wave photodetector layer structure.
3.2 Optical Networks
InP
15
(a)
(b)
Fig. 2: Etching of the two mesas, (a) cross-section, (b)
top-view.
InP
(a)
(b)
Fig. 3: Polyimide step, (a) cross-section, (b) top-view.
InP
(a)
Fig. 4: Evaporation of the metalization for the ohmic
contacts, (a) cross-section, (b) top-view.
(b)
408), Fig. 3. The polyimide is processed
as a negative light sensitive resist and developed with a polyimide developer. A hard
bake process at 350°C makes the polyimide resistant for the following process
steps.
In the next step, the metalization for the
ohmic contacts is evaporated on top of the
n-doped InGaAlAs-layer, Fig. 4. The metalization consists of Ge (30nm), Ni (5nm),
and Au (300nm). The metallic layers are
alloyed at about 550°C. This leads to ohmic contacts with low impedances between
the metalization and the semiconductor.
The fabrication of the travelling-wave
photodetector finishes with the metalization of the center conductor and the taper,
Fig. 5. Since the center conductor should
form a Schottky contact to the semiconductor, it is necessary to evaporate Pt/Ti/
Pt/Au or Cr/Au.
Measurements
The characterization of the travellingwave photodetector is first done using current-voltage and capacity voltage measurements.
The
current-voltage
measurement gives information about the
InP
(a)
(b)
functionality of the Schottky-contact diode
Fig. 5: Processing of center conductor and taper, (a)
formed between the center conductor and
cross-section, (b) top-view.
the outer metalization. As can be seen
from Fig. 6, the fabricated devices show
The fabrication of the mesa structure is done
the typical current-voltage characteristic of a diusing wet chemical etching with a liquid etchant
ode.
consisting of H3PO 4:H 2O2 :H2 O (1:1:40). This
In forward direction the current raises expoetch system has almost no effect on InP.
nentially with increasing voltage. With reversed
The insulation of the mesa edge is done usbias, the diode shows a high dark current which
ing a non-conducting polyimide step (Probimid
increases with the voltage applied to the device.
The build-in voltage of this diode is about 0.6V.
16
3 RESEARCH
with the area of the depletion layer A, the
dielectric constant er e0, the density of donor dopants ND, the build-in voltage UB, the
applied bias voltage U, Boltzmann’s constant k, the temperature T, and the electron charge q.
If the capacity is known, it is possible to
calculate the density of donor dopants:
100
10
1
0.1
0.01
ND = −
0.001
-2
-1
0
U (V)
1
2
Fig. 6: Current-voltage measurement of a travelling-
The capacitance-voltage measurements allows the determination of the doping level of the
fabricated structures. The capacitance C of a
depletion layer is given by:
C=A
Laser 1 lens
Laser 2 lens
ε r ε0 N D

kT 
2 − U B − U −


q 
2
(
dU
qε r ε 0 A 2 d 1 C 2
)
The capacity-voltage measurement
leads to a doping level of about 1018cm-3
in the InGaAlAs-layer and less than
1017cm-3 in the InGaAs and the InAlAs-layer.
Optical heterodyne setup
For the measurement of the optoelectronic
conversion efficiency, a heterodyne setup with
two Nd:YAG-lasers operating at 1.3µm is used.
The wavelength of the lasers is adjustable by
detuning the temperature of the laser head. Up
to now, the two lasers illuminate the travellingwave photodetector via free space and a microscope lens. To facilitate
the optical coupling to the
Spektrumphotodetector the use of
analyzer
optical fibers has been
investigated. This measurement setup is shown
in Fig. 7.
DUT
Each laser is coupled
on-wafer
Bias-Tee
to a monomode fiber.
probe
Both fibers are coupled
to a third monomode fiMultiber using a GRIN-lens.
meter
Finally, this fiber is direct-
Fig. 7: Heterodyne measurement setup using monomode fibers.
3.1 Optical Networks
ly coupled to the device under test (DUT).
The electrical measurement of the optoelectronically generated millimeterwave is achieved
using a coplanar on-wafer probe, a bias-tee to
separate the high-frequency and the dc-signals,
a multimeter for photocurrent, and an spectrum
analyzer to measure the amplitude of the millimeterwave in frequency domain.
Conclusion
In this report, the fabrication of travelling-wave
photodetectors is described. Photodetectors
have been processed successfully. Characterization of these devices is done using currentvoltage and capacity-voltage measurements.
Finally, the heterodyne measurement setup has
been investigated in order to facilitate optical
coupling to the photodetectors using monomode
fibers.
References
[1] V. Wendrix, “Fabricatie en karakterisatie van een
17
3.1.3 Simulation of the microwave
generation of a travelling-wave photodetector
A. LUEDEKE AND M. ALLES
I
n this report the photoelectronic microwave generation of a coplanar InGaAlAs/InP Schottky-contact travelling-wave
photodetector (TWPD) is analyzed by numerical solution of the wave-equations. Computational simulations of the electrical behavior
of the travelling-wave photodetector are carried out using an equivalent circuit model.
Introduction
The electrical behavior of the travelling-wave
photodetector can be described using the equivalent circuit model of Fig. 1 [1]. The travellingwave photodetector is fabricated as an electrical millimeter waveguide. Therefore all elements
are per unit length.
Traveling-Wave Photodetector“, Diploma thesis,
Department Toegepaste Natuurkunde, Vrije
Universiteit Brussel in cooperation with
V
R’m
dc
L’
Fachgebiet Optoelektronik, Gerhard-MercatorUniversität Duisburg, 1996
[2] R. Haupt “Experimentelle Untersuchungen zur
Integration von Schottky-Kontakt-Varaktordioden
R’hl
für den Einsatz in periodischen Leitungs-
G’rlz
C’rlz
strukturen“, Graduate thesis, Fachgebiet
I’p
C’L
Optoelektronik, Gerhard.Mercator-Universität
Duisburg, 1995
C’b
G’b
Fig. 1: Equivalent circuit model of the travelling-wave photodetector. All elements are per
unit length.
18
3 RESEARCH
The impedances R’m and R’hl describe the
longitudinal ohmic losses in the metalization and
the semiconductor, respectively. The inductance
of the electrical waveguide is taken into account
with the inductance L’. The conductance G’rlz
and the capacity C’rlz consider the Schottky-contact depletion layer. The two elements C’b and
G’b describe the behavior of the bulk material.
An additional capacity C’L is introduced for the
electric field in air above the photodetector. The
impressed current source I’Ph describes the optoelectric conversion in the absorbing quantum
well layer.
’
’
’
’
’
’
∂ 2u
’ Y1 ⋅Y 2 + Y1 ⋅Y 3 + Y 2 ⋅Y 3
= u ⋅W ⋅
∂ z2
Y 1’ + Y ’2
I ’p ( z ) ⋅
+
W ’ ⋅ Y ’2
where W’ ist the impedance and Y’1 , Y’2 and Y’ 3
the admittances of the transmission line. I’ p(z)
is given by
I ’p ( z) =
q ⋅ ηopt ⋅ (1 − R ) ⋅ α opt ⋅ Popt
h ⋅ν
⋅e
Numerical solution
The wave-equations of the travelling-wave
photodetector are derived by using a summarized equivalent circuit model, shown in Fig. 2,
which based on the model above. The equation
of the complex voltage amplitude along the
transmission line can be shown as
− (α opt + jβopt ) ⋅ z
(2)
where hopt is the internal quantum efficiency, aopt
is the optical absorption coefficient, hn is the
photon energy, Popt is the incident light power,
R is the reflection of interface device/air and bopt
is the optical phase coefficient.
i + ∂ i dz
∂z
W ’dz
i
(1)
Y 1’ + Y ’2
iB
iA
u1
Y 1’dz
I ’p ( z) dz
u3
u
Y ’3 dz
u
u+∂
∂z
i A + I ’p ( z) dz
Y ’2 dz
u2
Fig. 2: Summarized equivalent circuit model of the travelling-wave photodetector. All elements are per
unit length.
3.1 Optical Networks
19
Using the finite differences method [2], the
numerical solution of equation (1) is given by
u k +1 + u k −1 − h 2 ⋅ I ’p ( z k ) ⋅
uk =
2 + h ⋅W ⋅
2
’
W ’ ⋅ Y 2’
Y1 + Y 2
’
’
Y 1’ ⋅ Y 2’ + Y 1’ ⋅ Y 3’ + Y 2’ ⋅ Y 3’
Y 1’ + Y 2’
(3)
with
h =
b−a
N
(4)
where a is the beginning and b the end of the
transmission line. N is the number of discrete
points, where the voltage is calculated
(0 £ k £ N).
The loads at the two ends of transmission line
are defined as Z1and Z2, the characteristic resistance of the transmission line is Z. According
to microwave theory there exist reflections r1 and
r2 at the two ends z = 0 and z = l of the device.
The boundary condition for this problem is de-
termined by the current voltage relation at the
specific load resistance.
u ( z = 0) = u 0
=
Z 1 + r1 1
⋅
⋅ ⋅ [ u1 − u − 1 ]
W ’ 1 − r 1 2h
(5)
u (z = l ) = u N
= −
Z 1+ r2 1
⋅
⋅ ⋅ [u N +1 − u N −1 ]
W ’ 1 − r 2 2h
(6)
The solution of equation (3) is computationally calculated.
Voltage (mV)
Simulation
For the simulation of photoelectrical microwave generation a frequency of 40GHz is taken. The wavelength of the optical sources is
1.3µm and the light power of the two beams are
both 1mW. In the following calculation the quantum efficiency h opt
of photoelectrical conversion is
assumed to be 1 and the incident
40
light energy is fully and uniformly
35
coupled to the active layer, so the
30
optical reflection R is 0.
Fig. 3 shows the voltage dis25
tribution along z direction. The
20
reflection factor at z = 0mm is 1
15
and at z = 1mm the reflection is
0.
10
The dashed line shows the re5
sult by using a simplified equivalent circuit model, in which C’b, G’b
0
0
0,2
0,4
0,6
0,8
1
and C’L are neglected. An existz (mm)
ing simulation program, which
calculates the solution analyticalFig. 3: Voltage distribution along z direction for a simplified
ly, is based on this model. The
(interrupted line) and the fully equivalent circuit model.
20
3 RESEARCH
Voltage (mV)
Conclusion
This report presents the numerical solution of the photoelectrical microwave generation of a
coplanar travelling-wave photodetector. The wave equations are
solved by the finite differences
method. The results of the simulations have shown, that the efficiency is reduced by 10% relative
to using a simplified equivalent
circuit model and that a further
optimization is possible.
60
50
40
30
20
10
0
0
0,1
0,2
0,3
0,4
0,5
z (mm)
References
[1]
Fig. 4: Voltage disribution along z direction for Z 2 = 50W
D. Jäger, R. Kremer, ”Trav-
elling-wave optoelectronic devices
(dashed line) and Z2 = Z (solid line) for a device length of
for microwave applications”, Proc.
IEEE, MTT-S and LEOS Topical
Meeting on Optical Microwave Interactions, pp.
comparison with the numerical simulation, which
is based on the complete equivalent circuit model, shows, that the efficiency is reduced by 10%.
One possibility to optimize the device relative to the output-voltage is to reduce the length
of the transmission line. In the following simulation the length of the device is set to 0.5mm.
Fig. 4 shows the results by using Z 2 = Z (solid
line) and Z2 = 50W (dashed line). In case of adaptation the efficiency is raised by 44% relative
to a device length of 1mm. In case of mismatching there is a high voltage at the end of the device, but the efficiency has not raised because
of higher load at the device´s end.
Another possibility of optimization is to reduce
the value of R’m. Simulations have proved that
the efficiency is raised by 20%, if R’m is set at
0W (device length is 1mm).
11-14, 1994, France
[2] D. Marsal, ”Finite Differenzen und Elemente:
numerische Lösung von Variationsproblemen
und partiellen Differentialgleichungen”, SpringerVerlag, Berlin/Heidelberg, 1989
3.1.4 Determination of RF-equivalent
circuit elements of travelling-wave
photodetectors using network analysis
O. BERGER AND M. ALLES
I
n this report, the determination of the
elements of the equivalent circuit model
of coplanar waveguides is described. Since
the longitudinal and the transverse complex
3.1 Optical Networks
impedance of the equivalent circuit model
can be calculated from the characteristic impedance and the propagation coefficient
measured with a network analyzer, it is possible to compute the equivalent circuit
straight forward. This method has been used
to determine the equivalent circuit model of
travelling-wave photodetectors. Results and
comparison to theory and other measurements are shown.
21
cuit model directly from network analyser measurements.
Equivalent Circuit Model
The coplanar waveguide structure of the travelling-wave photodetector can be described using the distributed equivalent circuit model
shown in Fig.1 [1]. Note that all elements are
per unit length. The impedances R’M and R’
describe the longitudinal ohmic losses in the
metalization and the semiconductor, respectiveIntroduction
ly. The inductance of the electrical waveguide
Recently, 60GHz-travelling-wave photodetecis considered by the inductance L’. The depletors are under development in the Fachgebiet
tion layer of the Schottky-contact is taken into
Optoelektronik. For characterization and further
account with the conductance G’ and the caoptimization of the device, network analyzer RFpacitance C’ while G’B and C’B characterize the
measurements are analyzed in a new way in
bulk material of the semiconductor. The addiorder to determine the high-frequency equivational capacitance C’L is associated with the
lent circuit elements of the device. The impleelectrical field in air above the device. Finally,
mented method determines the equivalent cirthe impressed current source I’Ph is introduced
to describe the optoelectronic conversion within the absorbing layer.
To determine the equivalent
circuit elements it is necessary to
make some simplifications. Usually, network analyser measurements take place without optical
illumination of the device, therefore I’Ph can be neglected. The
two capacitances C’ L and C’ B
have much less influence on the
behavior of the device in comparison to the Schottky-capacitance
C’ and can be neglected.
One can conclude, that in the
longitudinal and in the transverse
part of the equivalent circuit three
elements are to be considered:
two real impedances and one
Fig. 1: Equivalent circuit model of the travelling-wave photodeimaginary one. The longitudinal
tector.
22
3 RESEARCH
and the transverse part of the
equivalent circuit can be described separately with
W ’=
R ’( R’M + jω L’)
R ’+ R’M + jω L’
and
.
G’ (G’+ jωC’)
Y ’= B
G’B + G’+ jωC ’
Using separate Smith charts
for the longitudinal and the transversal part of the equivalent circuit to display the frequency dependence of W’ and Y’,
semicircles with two intersections
with the real axis for ω = 0 and
theory
low-frequency
measurements
high-frequency
measurements
Ω 
R ’M 
 mm 
6.58
6.21
13.8
Ω 
R ’
 mm 
707
-
887
S 
G’
 mm 
6.7×10 -14
750×10 -12
1.3×10-3
874
1.1
2.54
nH 
L’
 mm 
0.466
-
0.62
pF 
C’
 mm 
1.39
1.17
0.91
S 
G ’B 
 mm 
ω → ∞ can be determined. The
intersections with the real axis are
Tab. 1: Values for the elements of the equivalent circuit model
of a travelling-wave photodetector.
Z ’(ω = 0) =
R ’R ’M
,
R ’+ R ’M
Z ’(ω → ∞ ) = R ’
for the longitudinal part and
Y ’(ω = 0) =
G’G ’B
,
G ’+ G’B
Y ’(ω → ∞) = G’B
for the transverse part.
Determination of the equivalent circuit
elements
The network analyser measures S-parameters from a device under test (DUT). The characteristic impedance Z and the propagation coefficient g are determined from this data.
The new method calculates the impedance
of the longitudinal part and the admittance of
the transverse part using the relations W’ = g Z
and Y’ = g / Z. The frequency dependence of
both parts is displayed in separate Smith charts.
An statistic-based algorithm fits a semicircle to
the measured data. The intersections with the
real axis are used to determine the real elements
of the equivalent circuit model. With these results, it is possible to calculate the imaginary
elements L’ and C’ for each frequency. This
method has been implemented to an easy-tohandle windows-program with graphic features.
The calculation results are displayed instantly
on-screen.
Results
The equivalent circuit elements calculated
using this method have been compared with
results of low-frequency measurements and theoretical determined values [2], Tab. 1. As can
3.1 Optical Networks
be seen from this table, the measurements are in good agreement with theoretically determined values. Only the
two admittances show different values.
In case of G’ the network analyzer measurement leads to a higher value. An
explanation is, that the accuracy of the
network analyzer makes it impossible
to measure these low admittances.
The theoretical value of the bulk admittance G’R is much larger than the
measured values. A reason is, that the
metal-semiconductor resistance is neglected in the derivation of G’B.
Taking measurements at various
bias voltages one can see that only the
elements C’ and G’ regarding the depletion layer show major bias dependence while the other ones keep constant over the bias voltage.
Fig.2 shows the characteristic impedance, Fig. 3 the phase and attenuation coefficient calculated from the
measured data. The real part of the
characteristic impedance rises at frequencies above 35GHz indicating that
probably the major part of the electrical quasi-TEM wave on the structure
changes from TEM to TM.
The smithcharts with the measured
data and the semicircle calculated as
an approximation for the longitudinal
part and the transversal part of the
equivalent circuit are shown in Fig. 4.
As is visible, the frequency dependence
of the measured data leads to semicircles, indicating that the device under
test can be described with the simplified equivalent circuit model described
above.
23
40
30
re{Z}
20
10
0
im{Z}
−10
−20
−30
0
10
20
frequency (GHz)
30
40
Fig. 2: Characteristic impedance of a travelling-wave
photodetector.
4
4
3
3
2
2
1
1
0
0
10
20
30
frequency (GHz)
Fig. 3: Phase coefficient and attenuation coefficient of a
travelling-wave photodetector.
0
40
24
3 RESEARCH
(b)
(a)
frequency
frequency
Fig. 4: Smithcharts for the longitudinal part (a) and for the transversal part (b).
The frequency dependence of the inductance
L’ and the capacitance C’, shown in Fig. 5, can
be determined directly with the program. Due to
a suggested change of the quasi-TEM-Mode to
a TM-mode, the inductance rises and the capacitance decreases with higher frequencies.
Conclusions
A method to determine the RF-equivalent elements directly from network analyser measurements has been developed. This method approximates the frequency dependence of the
1.0
measurement data in a Smith chart graphically
without any further knowledge about the values
to be determined. The measured results fit well
with both theoretically determined and low-frequency measured values. It is also possible to
examine the frequency dependence of the inductance and the capacitance.
Acknowledgement
The author would like to thank U. Auer (Fachgebiet Halbleitertechnik/-technologie) for growing the epilayer and for fabrication of the travelling-wave photodetector.
1.0
References
[1] M. Alles, T. Braasch, D. Jäger,
0.8
0.8
“High-speed coplanar Schottky travelling-wave photodetectors“, Int. Conf.
0.6
0.6
on Integrated Photonics Research,
Proc. pp. 380-383, Boston, USA, 1996
0.4
0.4
[2] O. Berger, “Bestimmung der HFErsatzschaltbildelemente von Photo-
0.2
0.2
detektoren
mit
Hilfe
der
Netzwerkanalyse“, Graduate thesis,
0.0
0
10
20
30
frequency (GHz)
Fig. 5: Inductance and capacitance versus frequency.
0.0
40
Fachgebiet Optoelektronik, GerhardMercator-Universität Duisburg, 1997
3.1 Optical Networks
3.1.5 Polarization insensitive
waveguide modulators on InP
T. ALDER AND R. HEINZELMANN
E
lectroabsorption modulators using
strained multiple quantum well (MQW)
structure have been designed, fabricated and
characterized. Utilizing the Quantum Confined Stark Effect (QCSE) due to high electric field underneath a Schottky-electrode,
the absorption coefficient of the optical
waveguide can be changed. The use of
strained quantum wells enables an operation
of the device, with almost no sensitivity to
different polarisation. With this device an on/
off-ratio of 18.5dB has been achieved.
25
Introduction
In general, the absorption change in a MQW
structure is strongly polarization dependent [1].
From the viewpoint of system applications, a
polarization insensitive or at least polarization
independent modulator is desirable. Appropriate structures can be designed using quantum
wells with tensile strain [2-3]. In this paper electroabsorption waveguide modulators using a
strained InGaAs/InAlAs MQW structure in the
electrooptical active region will be presented.
Device structure and principle of operation
A schematic diagram of the modulator structure is shown in Fig. 1. The modulators investigated utilize a nin-structure containing Si-doped
InAlAs top and bottom cladding layers with thickness of 570nm and 1120nm, respectively.
Fig. 1: Schematic diagram and cross section of the modulator.
26
3 RESEARCH
The doping concentration of the bottom cladding layer is ND = 1×1017cm-3, whereas that of
the top cladding layer is ND = 1×10 16cm-3. The
non intentionally doped guide consists of 19 ×
6nm thick InGaAs MQW’s separated by
19 × 7.7nm thick InAlAs barriers. The structure
was grown using the MBE machine of the Department of Optoelectronics. To examine the
dependence of the device behaviour on contact
geometry, a number of devices were fabricated
with chrome-gold-Schottky-electrodes of different width, ranging from 8µm to 16µm. The Schottky-electrodes were manufactured by thermal
evaporation of chrome and gold in ultra high
vacuum. The waveguide structure was formed
using wet chemical etching after evaporation of
the Schottky contacts. The second contact,
shown in Fig. 1 is carried out as Ohmic contact
and was manufactured by thermal evaporation
of germanium, nickel and gold in ultra high vacuum.
As a reverse bias is applied to the Schottkyelectrode, there will be a high electric field underneath the Schottky-contact within the depletion region. This increases the absorption
coefficient of the guide due to the quantum confined Stark effect (QCSE). In this way, the optical output power can be controlled electrically.
Experimental results
Fig. 2 shows the nearfield-pattern and the
lateral profile of the optical waveguide mode at
different reverse biases. From this figure, it can
be seen, that the output becomes weaker as
the applied reverse bias is increased. This behavior is due to the increased absorption coefficient in the optically guiding region.
As the previous result shows, it is possible to
control the optical output power by a reverse
bias. In the following, systematic results on transmission changes will be presented. In Fig. 3 the
transmission is plotted as a function of different
reverse biases. From this figure, it can be seen,
that within the range from -4.3V to -7.4V the
transmission changes almost linear with the
applied bias. The ratio between maximum and
minimum transmission is 18.5dB. Furthermore
it can be seen, that the change in transmission
from 0V to about -4V is very low. This indicates
that quantum wells were grown smaller than they
were designed.
Fig. 4 shows the transmission as a function
of reverse bias for TE- and TM-polarization. It is
evident from the figure, that for TE- and TM-po-
100
position
0V
position
-2V
position
-4V
position
-6V
position
-8V
position
transmission [%]
80
60
= 1,2mm
40
w = 14µm
-10V
20
0
-10
Fig. 2: Nearfield-pattern and lateral profile of
the optical waveguide mode at different
reverse biases.
-8
-6
-4
voltage [V]
-2
Fig. 3: Transmission as a function of different
reverse biases.
3.1 Optical Networks
27
maximum transmission change [dB]
transmission [a.u.]
larization the change in transmission is almost equal. While for TE1
polarization an on/off-ratio of
18,5 dB
18.5dB could be measured a
0,8
slightly smaller value of 17.2dB
17,2 dB
appeared for TM-polarized light.
0,6
Considering only the „linear
range“ (the voltage range be0,4
tween -4.3V and -7.4V bias voltage) of the transmission characTE - polarisation
TM - polarisation
teristic, for TE- as well as
0,2
TM-polarization an on/off-ratio of
8.2dB is measured.
0
-10
-8
-6
-4
-2
To examine the dependence
voltage [V]
on device dimensions modulators
with different Schottky-electrode
Fig. 4: Transmission as a function of different reverse biases for
widths from 8µm to 16µm were
TE- and TM-polarisation.
investigated. The results are
shown in Fig. 5. As can be seen,
there is no recognizable influence from contact
ther for TE-polarization nor for TM-polarization.
width on the maximum transmission change, neiThis is of major importance, as with a change in
the contact width the propagation
properties of the electrical
waveguide can be fit to those of
20
the optical waveguide, for highspeed operation the travelling16
wave concept [4] can be applied.
12
TE - polarisation
TM - polarisation
8
4
0
7
9
11
13
contact width [µm]
15
Fig. 5: Maximum transmission change as a function of different
reverse biases for TE- and TM-polarization.
17
Conclusions
Electroabsorption waveguide
modulators based on strained InGaAs/InAlAs-MQW have been
designed, fabricated and characterized. A maximum on/off-ratio of
18.7dB has been achieved. It
could be shown, that the polarisation influence on the transmission behaviour was small, due to
the influence of the strain in the
quantum well region. Additionally
28
3 RESEARCH
no influence of the contact dimensions on the
transmission was observed.
References
[1] T. Aizawa, K. G. Ravikumar, R. Yamauchi,
„Polarisation Independent Refractive Index
Change In InGaAs/InGaAsP Tensile Strained
Quantum Well “, Electronics Letters, Vol. 29, No.
1, pp. 21 - 22, January 1993
[2] H. W. Wan, T. C. Chong, S. J. Chua, „Considerations For Polarisation Insensitive Optical Switching and Modulation Using Strained InGaAs/
InAlAs Quantum Well Structure”, IEEE
Photonics. Techn. Lett., Vol. 3, No. 8, pp. 730 732, August 1991
[3] J. Shimizu, T. Hiroshima, A. Ajisawa, M.
Sugimoto, Y. Ohta, „Measurement of the
polarisation dependence of field induced refractive index change in GaAs/AlAs multiple quantum well structures”, Appl. Phys. Lett., Vol. 53,
No. 2, pp. 86 - 88, 1988
[4] D. Jäger, R. Kremer, and A. Stöhr, „Travellingwave optoelectronic devices for microwave applications“, IEEE MTT-S 1995 International Microwave Symposium, Vol. 1, pp. 163-166, 1995
(invited paper)
3.2 Optical Interconnects and Processors
29
3.2 Optical Interconnects and
Processors
retina layer, and a wireless signal and energy transfer from RE to RS.
The task of the Department of Optoelectronics in this project is the development of a
device for optoelectronic signal and energy
transfer into the eye. To achieve this, a prototype consisting of a laserdiode as transmitter and a receiver consisting of a
monolithically integrated photovoltaic cell
array and a photodiode, together with driving and receiving electronics, was manufactured.
3.2.1 Neurotechnology: Retina Implant
M. GROSS AND R. BUSS
T
he Department of Optoelectronics is
a member of a consortium of 14 German expert groups, working on the project
EPI-RET: Retina Implant. This interdisciplinary project, funded by the Federal Ministry
for Education, Science, Research and Technology (BMBF) in Germany, is developing a
retina implant.
This device is a neural prosthesis, designed
for patients blinded by a disease where the
outer retinal layer degenerates (retinitis pigmentosa or macula degeneration). It consists
of three parts: a so-called retina encoder (RE)
outside the eye, simulating the function of
the retina, the retina stimulator (RS), a microchip placed on the retina with electrodes
stimulating the ganglion cells in the outer
Fig. 1: Signal and energy transmission into the eye
Introduction
The signal and energy transmission line described here is part of a technical system functioning as a vision aid for blind people who have
lost their vision due to retinal degenerations,
especially retinitis pigmentosa [1]. An often appearing kind of blindness is the partially degeneration of the retina, e.g. the disease retinitis
pigmentosa, which is leading to blindness
through following steps: The typical begin is the
loss of the rod photoreceptors, causing night
blindness. Next the cone photoreceptors are
30
dying off, beginning at the outer perimeter of vision. This leads to a tunnel vision and finally to
total blindness, when the cones in the fovea are
lost.
However, while the photoreceptors are dying
off, the nerve cells in the retina and subsequent
parts of the central visual system are remaining
mostly intact [1]. This leads to the possibility of
developing a visual prosthesis with the ability to
replace main parts of the retina that gives sight
back to the visually impaired [2].
System description
The whole system sketched in Fig. 1 consists of three main parts:
> a retina encoder (RE), consisting of a CMOS
camera and an artificial neural network (encoder) for image data processing,
> a retina stimulator (RS), a flexible chip, epiretinal affixed, with µ-electrodes on the back,
> a wireless signal and energy transmission line
from the retina encoder to the retina stimulator.
The system works as follows: First the high
dynamic range CMOS camera generates a picture. This dataset is then reduced by an artificial neural network and transformed into digitally coded pulse trains (nerve signals), to which
the ganglion cells can react. This corresponds
to the data reduction from 120 million photoreceptors to 1 million ganglion cells by the human
retina. The dataset is then optically transmitted
at a rate of 1 Mbit/s to a microcontact foil on the
retina (receiver), where eye movements of up
to +/- 15°, measured from looking straight ahead
have to be compensated. Together with this information transfer an optical bias is transmitted,
supplying the driving circuit with 5 mW electrical power. The retina stimulator is a soft microcontact foil which is implanted adjacent to the
3 RESEARCH
ganglion cell layer on the outer retinal limit. The
µ-electrodes are stimulating the ganglion cells
of the retina, thus transmitting the signals via
the optic nerve to the visual cortex in the brain.
Results
The signal and energy transmission has been
realized in a first prototype using a laser diode
as transmitter and a photovoltaic cell array together with a photodiode as receivers for energy and signals, respectively. In the first step we
designed the parts for the optical transmission,
considering the boundary conditions given by
the human eye and the technical demands of
the whole device. For surgical reasons optical
fibres cannot be used to connect transmitter and
receiver directly. Therefore, a light source (e.g.
a pigtailed laser diode) has to be fixed in front of
the eye, transmitting the signal and energy onto
the retina, using free space optics. A micro-lens
system was developed mapping the light homogeneously on the retina in a spot with a diameter of about 5 mm. This assures that the receiver is illuminated for eye movements of up to +/
- 15°. The material used for the receiver is GaAs,
mainly to achieve high conversion efficiencies
with the photovoltaic cell array [2]. This has several advantages:
1. The fibre has an gaussian beam profile, while
laser diodes have strong astigmatism that has
to be corrected with a microoptic in front of
the eye.
2. The heat that the laser diode produces is led
away very easily.
3. The high frequency modulation of the laser
diode for the signal transmission is made far
away of the eye avoiding problems with electromagnetic compliance.
Latest results are shown in Fig. 2 and Fig.3:
Fig. 2 schematically depicts the system design.
3.2 Optical Interconnects and Processors
31
Fig. 2: System design
In Fig. 3(a) the I-V characteristics of a single photovoltaic cell and of an array with 5 cells connected in series are plotted. At a wavelength of
l = 800 nm this array delivers up to 5 mW electrical power with a conversion efficiency of about
23%, which turns out to be a great improvement
as compared with the results published in [3]. It
should be noted, however, that this efficiency
was obtained without any antireflection coating.
Experiments have shown that the efficiency can
be increased up to almost 31% by encapsulating the cell array with a biocompatible antireflection coating consisting of SiO2 /Si3N4 multilayers.
Moreover,
Fig. 3(b)
shows
measurements of the signal transmission: The
signals at the output of the encoder, curve (1),
are transmitted optically into the eye at a rate of
1 Mbit/s. The output of the receiver is plotted in
curve (2) of Fig. 3(b) together with the recovered
clock, curve (3), in Fig. 3(b).
Conclusion
In this report the progress in the work for an
optoelectronic signal- and energy transmission
line for use in a visual prosthesis is presented.
A concept is developed and a prototype is described. This prototype currently has the capability of delivering 5 mW electrical power together
with digitally coded signals at a rate of 1 Mbit/s
simultaneously, thus meeting the current sys-
Fig. 3: (a) I-V characteristic of photovoltaic cells (PVCs), (b) digitally coded signals before (1) and
after (2) transmission, and (3) recovered clock signal.
32
3 RESEARCH
tem requirements. However, the optical link described here is capable of transmission rates up
to 1 Gbit/s, suitable for optically powered highspeed data links.
Acknowledgement
The authors would like to thank the Federal
Ministry for Education, Science, Research and
Technology for financial support and all members of the EPI-RET team for fruitful discussions.
References
[1] R. Eckmiller “Retina implants with adaptive retina
encoders”, Proc. of the 1996 RESNA Research
Symp., Salt Lake City, pp. 21-24, 1996
[2] M. Groß, T. Alder, R. Buß, R. Heinzelmann, M.
Meininger, and D. Jäger, “Micro Photovoltaic Cell
Array for Energy Transmission into the Human
Introduction
The Retina Implant project was founded in
1995 as part of a young and interdisciplinary
area of research: Neurotechnology.
The goal of this ten year project is the development of both an artificial eye implant (retina
stimulator), stimulating the ganglion cells of the
human retina from patients, who lost their eyesight due retinitis pigmentosa or macula degeneration and an encoder transforming the signals
coming from a video system like human retina
does.
One of the coming tasks is to develop a system transporting the electrical power for this
implant and the signals from the output of the
encoder into the eye. This system analysis
shows the technical preferences for this transport by IR-rays.
Eye”, Proc. of the 14th European Photovoltaic Solar Energy Conference, Barcelona, Spain, vol.
1, pp. 1165-67, 1997
[3] J. Rizzo, J. Wyatt, “Silicon retinal implant to aid
patients suffering from certain forms of blindness”, Proc. of the 1996 RESNA Research
Symp., Salt Lake City, pp. 1-3, 1996
3.2.2 Analysis of the optical energy
and signal transfer module for an
artificial vision prosthesis
T. BAUMEISTER, M. G ROSS, AND R. BUß
W
ithin the scope of the Retina Implant Project supported by the German government the possibility of a wireless
transfer of signal and energy into the eye of
human patients was analyzed.
Criteria catalogue
The base of each scientific analysis is a criteria catalogue being a decisive help for the
evaluation of possible alternatives. The following criteria were found:
1. The fundamental criteria are the dimensions
of the implant. Due to surgical reasons the
maximum length is limited to 1.5 mm.
2. The Efficiency of the power transmission is a
criterion of great importance for any implanted system, because of the absence of possibility to cool any part of system inside the eye.
3. The reliability is fundamental too, because it
is nearly impossible to repair any failure and
the exchange of the whole system is more
dangerous for the patient as the first time implantation.
4. The biocompatibility is one more very important criterion for the long time function of any
3.2 Optical Interconnects and Processors
5.
6.
7.
8.
implant. This may be given by a biocompatible coating of any material, but there is the
risk of damage during the implantation and
fixation of the innerocular part. Further, we
should be aware of the influence of overgrowing the receiving part of the signal and energy transport system.
The receiver can´t be placed at all places on
the retina, because the fixation of it may damage axons of stimulated ganglion cells.
The technical availability and the need of
development of parts of the system are important due to the cost and the time to market of the system.
The possibility of extension, especially the
number of stimulating electrodes, is an important criterion for the future.
The acceptance of the whole system by the
patient and his milieu is quiet important too.
System analysis
First the need of bandwidth and power for the
stimulation of a given amount of stimulating electrodes were analyzed. The first version of the
retina stimulator will consist of an
array of twenty stimulating electrodes.
The bandwidth needed for the
stimulation with twenty electrodes
is nearly 100kbit/s, for 400 electrodes we found a bandwidth of
approximately 25Mbit/s.
This calculation of the bandwidth includes the scheme shown
in Fig. 1, the number of bits needed to address the electrodes, the
number of bits needed for encoding the stimulating pulseform, the
rate of neuro impulses, and fac-
33
tor needed for encoding the signal by wireless
transmisson.
The power needed using a single amplitude
modulated IR-laser for simultaneous transport
of energy and signals is 190mW of optical power in a worst case analysis with twenty electrodes
stimulating in one time frame. The worst case is
defined here by a constant electrical stimulation power of 750µW at the electrodes.
Next, the possibilities of signal encoding were
analyzed. The whole signal of amplitude modulated laser beam includes an AC signal, the encoded signal for stimulation and addressing the
electrodes, and a DC offset for the energy transfer.
The CMI-code (see Fig.2) also known as 2AMI-1-code or modified FSK-code is the best
compromise between the technical expense of
signal and clock recovering and the need of a
DC free signal in this application. This coding is
mentioned with a factor of two in analysis of
bandwidth [1]. Considering this we found the
best way to transfer the energy and signals is to
Fig. 1: Stimulation scheme
34
3 RESEARCH
Fig. 2: Calculation of the power needed
use a single amplitude modulated IR-laser diode.
Fig. 3: Example of a DC-free encoded signal
added with an offset for transmision of power
Analysis of the optics
To avoid a complicated, heavy and cost intensive eyetracking system focussing the laserbeam exactly on the position of the energy and
signal receiving photodiode, we decided to illuminate an area on the retina great enough to
compensate eyemovements about 10° in each
direction. The position of this area is vertical
ahead the macula. At this position the beam is
least influenced by the eye lid and the greatest
eye movement possible. The best way to lower
the reflection of the beam at the iris and thereby
to expand the possible angle of movement is to
focus the beam in the hole of the iris. For a nearly
uniform illumination of area, needed for an uniform supply of power of the stimulation electronic, we have developed a ring-shaped focus of
the beam. Thus, we solved the problem how to
3.2 Optical Interconnects and Processors
35
day’s industrial optics including one standard
micro lens.
Fig. 4: Example of CMI encoding
get a uniform illumination on the screen of a good
imaging optical system i.e. human eye. To solve
this problem we simulated the whole optical path
with the ray tracing program ZEMAX SE for human eye and additional for the rabbit eye, see
Fig. 5. For the simulation of the human eye we
used a slightly modified model of the eye from
Helmholtz. We completed this model with the
axes of movement [2]. For the eye of the rabbit
we used a similar model [3].
The optics consists of a two lens beam expander and one complex collecting lens to get
the above mentioned ring shaped focus (see
Fig.5). The optics is not modular, i.e. it is not
possible to use parts of the optics designed for
the human eye in the optical system for the rabbit eye.
The optical source can be the end of an optical fiber or a laser diode with a micro lens that
corrects the astigmatism of the laser. The lenses used in our model are all in the range of to-
Rabbit eye
Conclusion
Here is shown, that, principally, it is possible
to transfer enough power and a signal with sufficient bandwidth into the eye by using an IR
light beam. Further it is shown, that it is preferable to use a single amplitude modulated laser
diode according to the above mentioned catalogue of criteria. This paper shows that a transport of power and signal from the signal processing unit outside the eye to the implanted
microelectronics by optical means can be realized without the need of an eye tracking system.
3.2.3 Development of an optical signal and energy transmission system
R. HEDTKE AND M. G ROSS
A
n optical transmission system to power and provide a retinal implant with
energy and digitally coded information is developed.
Human eye
Fig. 5: Optical analysis of the rabbit eye and the human eye
Introduction
The “EPI-RET: Retina
Implant“ project is part of the
Neurotechnology Program
of the Federal Ministry for
Education, Science, Research and Technology (BMBF). The implant
system is evolved in co-operation with several interdisciplinary project partners as
a vision aid for people who
3 RESEARCH
23
POPT
OPTICAL OUTPUT-POWER (POPT)
36
TIME
TIME
I
I
th
DRIVING CURRENT (I )
Fig. 1: Principle of the modulation of the
laser diode
are suffering from retinal degenerative defects
like retinitis pigmentosa and macula degeneration. With the help of this optical transmission
system the retinal implant is provided with the
needed information and energy.
Transmitter
The driving current, as shown
in Fig.2, is regulated by a current
control unit to secure a constant
Receiver
The modulated laser light hits the photodiode
and the photovoltaic cell array (see Fig.3). The
photodiode generates the detector signal which
is transformed to TTL-level by a signal processing unit. A clock recovery unit is used to generate the clock signal. The other part of the laser
CURRENT ADJUSTMENT
ACTUAL CURRENT VALUE
CURRENT
CONTROL
RESISTOR
AND
SIGNAL
CONVERSION
PHOTO
DIODE
AND
SIGNAL
CONVERSION
DR IVING C URRENT
DRIVING
UNIT
SIGNAL
MODULATOR
+
MOD ULATION C URR ENT
Principle
For the transmission to the retinal implant the information is
modulated onto the laser light,
which transmits the energy. The
kind of modulation that is used is
shown in Fig. 1. A continuos energy transmission into the implant
is guaranteed by modulating the
driving current around a mean
operating point (OP). The driving
current and the optical outputpower for the laser diode is
sketched for an stimulation with
a rectangular-signal.
optical output power of the laser diode. This control unit compares the preset current value with
the actual and provides the control signal for the
driving unit. To generate the actual current value a photodiode integrated into the laser diode
[1] (alternatively: an external shunt-resistor) is
used. With the help of a signal conversion the
measured signal is processed for the following
current control unit. The modulation of the driving current occurs in the previously described
way.
ALTERNATIVE GENERATION OF
ACTUAL CURRENT VALUE
LASER
DIODE
Fig. 2: Block diagram of the transmitter
O PTICAL OUTPUT-POW ER
3.2 Optical Interconnects and Processors
37
CLOCK
RECOVERVY
MODULATED LASER LIGHT
PHOTO
DIODE
DETECTOR
SIGNAL
CLOCK
SIGNAL
PROCESSING
SIGNAL
PHOTOVOLTAIC
CELL
ARRAY
ENERGY
Fig. 3: Block diagram of the receiver
light is converted into electrical energy by the
photovoltaic cell array [2].
Measured results
Fig. 4 shows the driving current of the laserdiode for a stimulation with a 1 MHz rectangular
wave. The used mean operating point is 400
mA [3] or rather 190 mW optical power. To code
the signal the Manchester code [4] is used. The
main advantage of this coding are the well-balanced relation of logical high an low states (important for a constant energy transmission) and
the easy recovery of the clock signal. The transmitted and the received Manchester coded signal and the recovered clock are shown in Fig. 5.
800
I/mA
600
400
TRANSMITTED SIGNAL
RECEIVED SIGNAL
CLOCK
0
5
10
15
20
t/µs
Fig. 5: In- and output signals of the transmission system
Conclusion
At the Fachgebiet Optoelektronik (GerhardMercator-Universität Duisburg) an optical signal
and energy transmission system was developed
to provide a retinal implant with digitally coded
signals. To realize this aim a transmission and
a receiving unit were designed.
References
200
[1] SDL, Laser Diode Operator’s Manual & Techni0
0
1
2
3
4
t/µs
5
cal Notes, SDL Inc., San Jose/USA, 1994
[2] M. Meininger, “Entwicklung photovoltaischer
Fig. 4: Modulated driving current for the
Zellen zur Energieversorgung einer künstlichen
laserdiode
Sehprothese (Retina Implantat)“ Diploma thesis,
38
3 RESEARCH
Fachgebiet Optoelektronik, Universität Duisburg,
1997
[3] T. Baumeister, “Systemanalyse des optischen
Energie- und Signalübertragungsmoduls für eine
(dongle) for notebook-PCs without an builtin infrared interface and a special data transmission software protocol is described in the
second part of the report.
künstliche Sehprothese (Retina Implantat)“ Diploma thesis, Fachgebiet Optoelektronik,
Universität Duisburg, 1996
[4] R. Mäusl, Digitale Modulationsverfahren,
Hüthing, Heidelberg, 1991
3.2.4 Infrared data link for rotating
display-systems
U. WEIMANN AND R. B Uß
T
he ‘magicball’ is a recently designed
rotating LED display system for texts
and graphics. In this report two methods of
wireless programming of the ‘magicball’ by
using either an infrared remote control or a
notebook-PC are presented. In the first instance, a polymethylmethacrylate (PMMA)body is designed and a new software is
developed allowing the programming via an
IR-remote control unit. An infrared interface
Introduction
The ‘magicball’ display-system is used as an
“eye-catcher” in stores or at exhibitions amongst
others. Its design and operation principles are
shown in Fig. 1. The 16 LEDs are fixed at the
end of the rotating arm. The microcontroller creates moving titles on the surface of the ‘magicball’ by switching the LEDs on and off individually. The microcontroller and the LEDs on the
rotating carrier and arm are powered by a generator situated in the socket of the ‘magicball’.
The data transmission was formerly achieved
by a serial PC cable and rubbing contacts. By
designing a wireless infrared data link between
the programming unit and the display, programming of the ‘magicball’ is simplified and allows
customizing. In addition, the life-span of the
product is increased while running costs decrease.
IR Data Link: Remote Control => Magicball
The data transmission via infrared remote control unit is unidirectional from the programming
unit to the display. The infrared
signal of the remote control is biphase coded , similar to the standard RC 5 code, and consists of
a pre-signal and the main signal
[1]. A photodetector - with an integrated preamplifier, a demodulator and a filter - is used for detecting the infrared signal of the
remote control unit [2]. Since the
receiver module is placed on the
Fig. 1: Side and top view of the ‘magicball’-display-system
3.2 Optical Interconnects and Processors
remote IR
-bea
m
control
39
15 mm
epoxy glue
IR-Detektor
10 mm
Fig. 2: Exemplary infrared signal beam
15 mm
PMMA-body
IR Data Link: Notebook <=> Magicball
For the data transmission to the “magicball”
and vice versa , the notebook-PC needs an infrared interface (built-in or additional) which operates according to the SIR (Serial Infrared) standard of the IrDA (Infrared Data Association) [3].
The SIR standard enables a data transmission
range of up to 3 m, at angles of up to 15° and at
a data rate of 115.2 kbit/s. For notebook-PCs
without a built-in infrared interface a dongle fitting to the serial port of a PC has been designed.
The encoder/decoder chip and the transceiver
module with an additional IR-LED for wider transmission ranges support the infrared data trans-
rotating parts of the ‘magicball’,
an optical system (e.g. a special
mirror) is necessary to ensure an
uninterrupted connection between transmitter and receiver.
This is why a dynamically balanced body made of polymethylmethacrylate (PMMA) has been
developed.
Fig. 2 shows the path of an exemplary infrared ray from the reFig. 3: Block diagramm of the developed infrared dongle
mote control unit through the
PMMA-body onto the detector. The loss of inmission based on the SIR standard. The contensity caused by the reflection and transmisverters allow a power supply of the dongle with
sion involving the PMMA-body is less than 10%.
the serial RS232-port. The principle block diaTests show that the configuration consisting of
gram of the dongle is shown in Fig.3.
the remote control unit, PMMA-body and phoIn contrast to the remote control unit configutodetector enables a transmission range of up
ration, the notebook - ‘magicball’ infrared data
to 10 m. In addition to the hardware components,
link is bidirectional. Because of the bidirectionspecific software has been developed to proality and the lower radiation intensity using the
cess the incoming infrared bi-phase coded sigSIR standard, the PMMA-body cannot be used
nal. After sampling the signal the microcontrolhere. Therefore, a simple transmitter-receiver
ler reassembles the keycode of the remote
system has been chosen, omitting the optical
control unit and displays the new character on
system used before. This configuration is illusthe surface of the ‘magicball’.
40
3 RESEARCH
Conclusion
In this report, two
transmitter
methods of the
reception area
wireless programem
iss
ming of a ‘magSe
ion
nd
ew
ang
receiver
ink
le
icball’ via infrared
el
data transmission
are discussed. In
Fig. 4: Transmitter - rotating receiver - configuration
the first instance,
programming is
trated in Fig.4 . Because of the symmetry of the
done with an IR remote control using a PMMA
infrared data link, transmitter and receiver as
body with integrated photodetector as an optishown on the illustration are interchangable. The
cal medium on top of the rotating arm. In the
bi-phased character of the data link remains in
second instance, a customized program has
place.
been developed to enable a bidirectional IR data
Due to the rotation of the detector and belink between a notebook-PC and the “magicball”
cause of the angle of detection/emission, we
without an additional body. Many tests have
obtain a periodically recurring optical contact
shown the possibilty of infrared data transmisbetween transmitter and receiver. Because of a
sion on such a rotating receiver system.
rotation of about 3600 rpm and a detection anIn terms of practicability, the remote control
gle of 15° the optical contact time (time window)
allows a wider transmission range and a lower
is nearly 1 ms, while the non-contact time is 15
level of noise interference. On the other hand
ms. The software we have developed takes adthe comfortable editor for the notebook-PC is a
vantage of the resulting time window, thus enbig advantage for the second solution, especially
abling the data exchange between the notebook
if large ammounts of text has to be transmitted.
and ‘magicball’. The software program first partitions the information into blocks of data with
Acknowledgement
equal size depending on the time window. After
We would like to thank the LUMINO Licht Elebeing received, the single data blocks are reasktronik GmbH for the possibilty of working on
sembled. The data security of the transmission
the display system ‘magicball’ and the support
is safeguarded by parity bits on the one hand
during this work.
and by feedback of “magicball” on the other hand
( acknowledge / non-acknowledge). For examReferences
ple, a bad data block effects a non-acknowledge
[1] Adaptive Micro Systems „Infrared Communication Theory of Operations“ Abstract, 13.06.88
response and the block will be transmitted again.
[2] TEMIC Semiconductors „TFMx IR Detector
The software for this type of data transmission
Photomodules“ Design Guide, June 1996
and the infrared dongle have been successfully
[3] St. Williams, I. Millar The IrDA Platform HP
tested for functionality.
receiver path
n
t io
tec e
de angl
Labaratories, Bristol, 1996
3.2 Optical Interconnects and Processors
3.2.5 Nonlinear hybrid GaAs/AlGaAs
multilayer-heterostructures for highspeed information processing
C. KAMPERMANN, A. KREUDER, AND S. REDLICH
I
n this report we present theoretical and
experimental results on the nonlinear
optical, electrical, electrooptical and optoelectronic properties of hybrid GaAs/AlGaAs
multilayer heterostructures. These structures exhibit fast nonlinear properties and
high sensitivity which can be used for highspeed information processing in microwavephotonics.
Introduction
In recent years optical nonlinearity and bistability in multilayer heterostructures (MLHS) have
received increasing attention because of their
potential use in all-optical high-speed information processing systems. Applications are foreseen in the areas of photodetectors and modulators with internal amplification as well as fast
optical switching and memory devices. In 1991
He et.al. [3] achieved all-optical bistability in a
30 period GaAs/AlAs structure at an optical intensity of 10kW/cm² [1]. Switching intensities in
Fig. 1: Sketch of the device
41
the range of kW/cm², however, are orders of
magnitude too high for optical information processing. In the presence of an applied electric
field perpendicular to the layers of a MLHS, the
Franz-Keldysh effect together with the accumulation of photocarriers in the GaAs layers and
the voltage dependence of the current through
the device are used to decrease the switching
intensities by five orders of magnitude [2]. These
kind of hybrid MLHS exhibit the lowest switching intensities in comparison to other device
concepts. Appropriate designed MLHS also exhibit s-shaped negativ differential conductivity
(SNDC), based on bistability between tunneling
and thermionic emission across the heterobarriers. Calculations and experiments have shown
that selfsustained voltage oscillations up to 100
GHz occur, if the MLHS is driven in an external
resonator. In this report we present theoretical
and experimental results concerning these novel
kind of devices.
Device Structure
Fig. 1 shows the cross section of the device
containing a periodical GaAs/ Al0.45Ga 0.55As
MLHS. The MLHS consists of 20 bilayers with
nominal thicknesses of 58 nm (GaAs) and 69
nm (AlGaAs). The layers are grown by usual
42
3 RESEARCH
MBE on s.i. GaAs substrates. A 1000nm n+GaAs contact layer is introduced for sufficient
high values of the bulk conductance. A number
of different process techniques such as wet etching and evaporation are used to form the device. Conventional photolithography is applied
to pattern structures on the wafer. The transmission line is made of a TiPtAu multilayer metalization, which is applied to the wafer by evaporation. To prevent a short circuit between the
center conductor of the transmission line and
the contact layer at the bottom of the MLHS the
edges of the mesas are coated with polyimide.
A voltage can be applied to the coplanar
transmission line to get a high electric field concentrated in the MLHS. With coplanar transmission lines used as electrical contacts, operation
up to millimeterwave frequencies is possible.
The optical input and output ports are defined
by a via hole in the center conductor of the transmission line.
Theory
To simulate the device it is necessary to analyse the optical wave propagation as well as the
transport and the accumulation of charge carriers in the MLHS. Additionally, the electrooptical
and the optoelectronic interactions between the
optical and the electrical subsystems, namely
the Franz-Keldysh effect and the generation of
photocarriers in the GaAs layers have to be considered.
Optical properties
For an optical wave propagating in a periodically layered medium like a MLHS it has been
shown that, there exists an optical resonance
effect when the optical wavelength is close to
the optical stopband. This means that the intensity distribution in the MLHS depends on the
optical wavelength and strongly increases when
approaching the resonance. This effect is essential for the behaviour of the device and we
had to take this into consideration for our simulations. In the linear case one can use the transfer-matrix method (TMM) to calculate the intensity distribution in the layered medium. In the
nonlinear case of the MLHS the standard TMM
cannot be applied because of the intensity dependent refractive index of the GaAs layers. To
overcome this problem we used a generalised
Fig. 2: (a) Linear reflectivity spectrum of a InGaAlAs/InAlAs MLHS and average optical intensity in the
InGaAlAs layers of the structure.( b) Reflectivity over incident optical intensity of the same MLHS for
two different wavelength.
3.2 Optical Interconnects and Processors
form of the TMM [3] where the GaAs layers are
divided into a number of sublayers. Assuming
that the optical intensity in each sublayer is constant one can determine the intensity dependent
refractive indices by using the boundary conditions of the wave amplitudes in two adjacent layers. Then, specifying the field at the end of the
structure one can apply the standard TMM to
calculate the nonlinear characteristics of the
MLHS. Figure 2(a) shows the linear reflectivity
spectrum of a MLHS consisting of 40 pairs of
InGaAlAs/InAlAs bilayers, lattice matched to InP.
As can be seen at a glance, by using the InGaAlAs/InP system one can shift the operation
wavelength of the device towards 1.5µm, an
important wavelength for applications in optical
communication systems. Like demonstrated for
the AlGaAs/GaAs system by He et.al. all-optical bistability based on the nonlinear refractive
coefficient n2 without an applied electric field can
also be observed. The nonlinear reflectivity-versus-intensity characteristics of the structure,
calculated with the above method, are shown in
Fig.2(b). The curves are calculated for two different wavelengths at the long-wavelength side
of the stopband. Both are Z-shaped and exhibit
a bistable hysteresis loop. As mentioned above
we have experimentally shown that by applying
an electric field perpendicular to the layers of
the MLHS switching intensities below 10mW/cm²
can be achieved. Therefore, besides the optical
properties, the electronic, optoelectronic and
electrooptical properties of the MLHS are from
special interest.
Electrical properties
Our model of the transport and the accumulation of charge carriers in the MLHS is based
on an analytical model of a heterostructure hot
electron diode (HHED) by Wacker et.al.[4]. The
43
HHED shows S-shaped negative differential resistance (NDR) or differential gain and consists
of two undoped adjacent heterolayers (GaAs/
AlGaAs) with ohmic contacts. In this structure
two conduction mechanisms exist: At low fields
the current is limited by tunneling through the
AlGaAs layer (low conductance state on
theFig.3(a)). At higher fields the charge carriers
are heated up to sufficiently high energies so
that thermionic emission over the barrier becomes dominant (high conductance state on the
Fig. 3(b)). The extremely fast transition between
these conduction modes leads to NDR or differential gain. We extended the model of Wacker
et.al. to calculate the electronic properties of
MLHS. The physical processes of the charge
transport in the MLHS are sketched in Fig. 4(a).
As an additional effect, the cooling of the charge
carriers, which means the capture by the GaAs
wells is included. The numerically obtained current-density-voltage characteristics (see Fig.
4(b)) are in good agreement with the results of
Monte-Carlo simulations published by Reklaitis
[5]. Fig. 4(b) further elucidates a pronounced Sshaped NDR for (GaAs 100nm / AlGaAs 70nm)
and for thicknesses used in our device structure merely a preliminary stage of NDR.
W
a)
W
GaAs
AlGaAs
b)
GaAs
AlGaAs
Fig. 3: Schematic conduction band structure
of a GaAs/AlGaAs heterostructure with a
perpendicular electric field. The two possible
conduction states are shown.
44
3 RESEARCH
current density in kA/c
of the refractive indi1600
ces in the GaAs layEmission
ers. It has been shown
1200
TOTAL
CURRENT
that the same change
800
Heating
of the refractive index
400
can be achieved at
0
W
much lower optical in2,0
3,0
4,0
5,0
6,0
a)
b)
voltage in V
tensities as for the intrinsic optical nonlinFig. 4: (a) Schematic view of the conduction band structure of a MLHS with
earity. Including these
a perpendicular electric field and the physical processes of the charge
mechanisms the feedtransport. (b) Current density-vs.-voltage characteristic of a heterostructure
back, which deterfor two different layer thicknesses.
mines the optical bistability of the hybrid MLHS can be described
Interaction
as follows: An incident optical intensity leads to
Fig.5 shows schematically the system of a
an optical intensity in the MLHS, where a part of
hybrid MLHS [5]. The optical and the electrical
the light will be absorped. The generated carrisubsystems are coupled by two interaction
ers give rise to a photocurrent. This in turn leads
mechanisms, the generation of photocarriers
to a change of the voltage drop across the GaAs
(optoelectronic) and the Franz-Keldysh effect
layers and a change of the refractive indices of
(electrooptical). The photocarriers are generatthis material. This variation finally changes the
ed by optical absorption in the GaAs layers. Due
reflectivity of the hybrid MLHS and in turn the
to the nonlinear electrical properties of the MLHS
absorped optical intensity. Thus, a feedback loop
a small photocurrent leads to a strong variation
exists. The interaction mechanisms and the
of the internal voltage distribution and, by the
feedback loop are also implemented in our modFranz-Keldysh effect (FKE), to a large change
el of the hybrid MLHS. Thus, the experimentelly
observed device characteristics including the opi0
tical, electrooptical, optoelectronic and optically
induced electrical bistability of a hybrid MLHS
could be verified (Fig. 6).
Cooling
GaAs 100nm / 0.45
Al Ga0.55As 70nm
GaAs 58nm / Al
0.45Ga0.55As 69nm
C
Z
V0
Fig. 5: Cross section of the MLHS with I the
current flow and P the optical wave propagating through the device.
Experimental results
Comprehensive measurements of the optoelectronic properties of hybrid MLHS have
shown a high optical sensitivity of these devices. At a reverse bias of V=30V we have measured photocurrents of around I=1mA and dark
currents of merely 20nA (see Fig. 7).
A bias voltage also changes the reflectivity,
as shown on theFig. 8(a) , where the electroop-
3.2 Optical Interconnects and Processors
45
current in A
tical modulation
near the bandgap wavelength
is due to the
Franz-Keldysh
effect. A modulation contrast of
about 6dB could
be reached at a
voltage change
Fig. 6: (a) Measured current-voltage characteristic of a hybrid MLHS under
of 20V. Time and
illumination. (b) Calculated I-V curve of the same device and operation pafrequency dorameters.
main measurements were carried out to investigate the dynamic
cut-off frequencies of up to 420 MHz, measured
properties of the MLHS. We have determined
from MLHS devices based on another contact
geometry. The frequency response of these devices is RC limited, therefore higher cut-off fre-2
10
quencies should be reached by using the transdark
-3
10
Popt = 1mW
mission line design. The coupling of the
-4
10
-5
electrooptical modulation and the optoelectron10
-6
ic properties of the device leads to optical bista10
-7
10
bility, which has been measured at optical in-8
10
tensities below 10mW/cm² (Fig. 9).
-30
-20
-10
0
10
20
30
voltage in V
Conclusion
In this report new theoretical and experimental results concerning hybrid MLHS are present-
Fig. 7: Measured current-voltage character-
6
-10 V
-20 V
4
2
0
-2
860
880
wavelength in nm
900
rel. modulation in dB
modulation contrast in
istics in the dark and illuminated case.
4
A = (600µm)²
0
-4
-8
10 5
10 6
10 7
10 8
frequency in Hz
Fig. 8: (a) Modulation contrast over wavelength at different reverse biases. (b) Relative modulation
characteristic as a function of frequency.
46
3 RESEARCH
3.2 Optical Interconnects and Processors
47
48
3 RESEARCH
Fig. 2: Photograph of the array with one LED
illuminated .
above 1 MHz are to be expected. In comparison, the results differ only slightly from those
obtained with commercially available superluminescence diodes.
64 Channel Silicon Driver Circuit
The TTL-compatible silicon chip was designed to power the above mentioned array of
8 x 8 LEDs, with each LED driven independently by an output current adjustable in the range
of 0 - 10 mA. Based on the requirement of a
maximum input current for the IC of 1 mA, a
current amplification with a factor of 10 must be
achieved. The realized circuit consists of the
following main components: (i) two 3 to 8 demultiplexers for binary coded x- and y-selection
of each LED driver, (ii) a current mirror to tune
the gain of 10 and to shorten the input I in if Izero
is set to zero, and (iii) 64 independently selectable LED drivers containing xy-selection units
and a capacitor providing a constant output current Iout during regeneration cycles. In Fig. 3 the
layout of the chip, consisting of 64 identical
amplifier cells, bondpads, and demultiplexing
circuits, is sketched. Each cell requires an area
of 200 * 200 µm², leading to a total square surface of 1.6 * 1.6 mm² and consequently to a pixel
density of 127 DPI. Together with control circuitry and pads for external wire bonding, a total chip dimension of only 2.3 * 1.6 mm² is
achieved.
The electrical characterization of the silicon
circuit by measuring the pulse response with a
sampling oscilloscope was leading to rise and
fall times less than 250 ns. This results in a cutoff frequency of fc ³ 1.45 MHz. Due to the fact
that the I-V characteristic referred to the input of
the circuit shows strong non-linear behaviour,
an interface circuit (voltage driven current
source) was applied, leading to a
decrease of the cut-off frequency. Together with a D/A converter computer board the system
shown in Fig. 4 was built, providing a good linearity between input voltage and output current.
Fig. 3: Layout of silicon integrated circuit with detail of LED
drivers.
Hybrid integration In Fig. 5 the final device consisting of the silicon driver IC
bonded to the LED array is depicted. A composition of almost
3.2 Optical Interconnects and Processors
49
photonic IC, are established using
wire bonding technique.
Fig. 4: Computer controlled silicon driver circuit.
eutectic solder (60 wt % Sn, 40 wt % Pb) is evaporated onto the metal contacts of the LED array. After the reflow process, at
200° C for 10 seconds, both the
array and silicon driver IC are adjusted and bonded together.
Since the PbSn layer thickness is
much smaller than electroplated
PbSn due to the evaporation process, this bonding technique is a
mixture of thermocompression
and soldering. Following the flipchip process the ground contact
for the LED array, together with
connections for packaging of the
Applications
With this system presented
here several applications can be
realized. One example is a special kind of vision aid for blind
persons with a blurred cornea. In
Fig. 6 one possible realization of
this vision aid is sketched. Under
various circumstances (accidents
where the cornea is damaged, e.g
in explosions or by erosion due to acid) a number of people loose their sight although their ret-
Fig. 6: Vision aid for people with blurred cornea.
ina is fully intact. A photodetector array converts
images into digital information wirelessly transmitted to a miniature display like our proposed
model, implanted into the lens. This display
projects a very simple image onto the retina, offering a primitive vision.
Fig. 5: Cross-sectional view of the silicon
driver circuit bonded to the LED array.
Conclusion
Experimental investigations of both the silicon
circuit and the LED array show cut-off frequen-
50
3 RESEARCH
cies beyond 1 MHz, leading to the conclusion that
this hybrid integrated circuit is a promising subsystem not only for parallel optical information
processing systems but also for a novel application of photonic integrated circuits in the field of
neurotechnology.
Acknowledgements
This work was financially supported by the
Federal Ministry for Education, Science, Research, and Technology (BMBF) in the frame of
the “EPI-RET: Retina Implant” project under
contract number 01 IN 501 G. The authors would
like to thank G. Sixt (TEMIC Telefunken, Heilbronn) for providing the GaAsP/GaP wafer and
R. Klinke (Fraunhofer Institut-IMS, Duisburg) for
helping designing the silicon chip. Thanks goes
also to K. Heimann (Uni-Augenklinik, Köln) for
giving an insight into several ophtalmological
problems.
References
[1] H.F. Bare et al., IEEE Photon. Technol. Lett., 5,
2, pp.172, ‘93
[2] G.W. Turner et al., IEEE Photon. Technol. Lett.,
3, 8, pp.761, ‘91
[3] A.J. Moseley et al., Electron. Lett., 27, 17,
pp.1566, ‘91
[4] K. Werner, IEEE Spectrum, pp.30-39, Jul. ‘94
[5] W.R. Imler, et al., IEEE Trans. Compon., Hybr.
Manufact. Technol., 15, 6, pp.977, ‘92
[6] Y. Nitta et al., IEEE Photon. Technol. Lett., 4, 3,
pp.247, ‘92
[7] H. Yonezu et al., Electron. Lett., 25, 10, pp.670,
‘89
[8] M.A. Brooke et al., Optics & Photonics News,
pp.26, Jun. ‘93
[9] M. Wale et al., IEEE Circuits & Devices, pp.25,
Nov. ‘92
3.3 Millimeterwave Electronics
51
3.3 Millimeterwave Electronics
L/2 R/2 Ik
3.3.1 Picosecond pulse generation
on monolithic nonlinear transmission lines using high-speed InPHFET diodes
R. HÜLSEWEDE
C(V)
Vk
G
Fig. 2: Equivalent circuit for one element of
the NLTL
E
lectrical pulses with transients less
than 5 ps are generated and compressed on monolithic InP-HFET diode nonlinear transmission lines. The transients are
measured by time domain electro-optic sampling technique and the waveforms show
good agreement with numerical results. Additionally, frequency domain measurements
and numerical simulations reveal that the
nonlinearities work with frequencies higher
than 400GHz for 20µm x 20µm InP-HFET diodes. Instead of costly ion implantation technology a chemical recess is used to isolate
the active structures.
Introduction
Monolithic nonlinear transmission lines
(NLTL) are circuits with an alternating arrangement of coplanar waveguides and Schottky diodes as shown in Fig. 1.
The capacitance-voltage characteristic of the
Schottky diodes in combination with the low pass
filter characteristic of the periodic structure leads
to the generation of shock waves and the formation of pulses with (sub-) picosecond transients [1,2]. Therefore, these circuits are important for novel measurement and characterization
methods for new high-speed devices.
For numerical simulations the equivalent circuit as shown in Fig.2 is used leading to a difference equation for current and voltage at each
element of the NLTL.
Applying a transition to a differential equation one obtains the following wave equation
which considers separately the influences of the
nonlinearity of the diodes, the periodic structure
and the losses of the transmission line (for details see [3,4]):
∂V
 C(V )  ∂V

+
= 1 −
∂x
C0  ∂ t

+
D
D
D
D
Fig. 1: Monolithic NLTL with periodic array of
Schottky diodes
R/2 L/2
C0 ∂ 2V
L ⋅ C0 ∂ 3V
R
V
−
+
G ∂t2
L
12 ∂ t 3
(1)
Here the nonlinearity of the diodes is considered by the normalized capacitance-voltage
dependence C(V)/C 0 , where C0 is the capacitance at the operating point of InP-HFET diode
In order to improve the nonlinearity of the
Schottky diodes in NLTLs d-doped diodes based
52
3 RESEARCH
C(V)/Co
10
Schottky contact
Ohm contact
InGaAs-channel
InAlAs
InGaAlAs
InP
δ -Si
Fig. 3: Schematic profile of an InP-HFET
diode (for details see [5])
on InP-HFET layer structures are used ( see
Fig.3 and [5]). High electron concentration in the
d-doped layer (4.9 1012cm-2), maximum mobility (10900 Vs/cm-2), and the 2 dimensional electron gas (2-DEG) in the InGaAs channel are
special features of this layer structure at T=300K.
The strong nonlinearity of the HFET-layer structure is shown in Fig. 4 where the normalized capacitance-voltage characteristic of an InP-HFET
diode is sketched. Over the 0.5V bias range
around the working-point a 2200% change of
the capacitance is achieved. This is a 20x greater nonlinearity than d-doped GaAs Schottky diodes used in NLTLs described in [6]. One reason for this strong nonlinear behaviour is the
depletion of the 2-DEG underneath the negative biased Schottky contact. Using InP-HFET
diodes in NLTLs the nonlinear interaction of the
propagating waves is increased and thus the
line-losses are decreased due to shortening of
the line length. Another advantage is the application of a C4H6O4,H2O2,NH3 recess [7] for electrical isolation of the InP-HFET diodes in NLTLs.
Thus, no costly ion implantation process is needed and no preparation of an accelerator is required.
1
0.1
0.01
-3
-2
-1
0
Bias voltage(V)
Fig. 4: Normalized capacitance-voltage
characteristic of InP-HFET diodes
InP-HFET NLTL
In a first step a 10 diode periodic InP-HFET
NLTL was fabricated in order to verify experimental and numerical results. For that purpose
frequency domain measurements along the center conductor are shown in Fig. 5 (see also [8]
and P. Bussek et al., Time- and frequency domain electro-optic field mapping of nonlinear
transmission lines, in this annual report).The
electro-optic signal of the excited input wave
(15GHz, 25dBm) decreases from input to output of the NLTL, whereas the generated harmonic signals at 30GHz, 45GHz and 60GHz increase. Using the nonlinearity of InP-HFET
diodes (Fig.4) and a FFT the nonlinear wave
propagation on this NLTL is simulated. The result is shown by the grey lines in Fig. 5. With
respect to the -128dB noise level and the +/-5dB
accuracy of the sampling signal both results are
in good agreement. This agreement and the
numerical value for C0/G = 3,5 10-13s indicates
that the nonlinearity works with frequencies higher than 400GHz for the 20µm ´ 20µm InP-HFET
diodes.
Thereupon different NLTLs are simulated based
on these perceptions in order to generate one
3.3 Millimeterwave Electronics
53
Signal voltage (V)
Signal (dBm)
(a)-(d). The frequency of the input
signal is 6.5GHz with an amplitude
of 3.5V (measured at 50W). Clear-80
15GHz
ly the steeping of a shock wave
Input
signal:
15GHz,27dBm
-90
(k=11) and the generation of a sinexperiment
simulation
-100
30GHz
gle pulse (k=31) with FWHM of
-110
10ps and a fall time of about 5ps
45GHz
is observed. Thus, picosecond
-120
60GHz
pulse generation on NLTL using
-130
high speed InP-HFET diodes is
-140
shown for the first time. Addition2500
2000
0
500
1500
1000
x (µm)
ally, the numerical results (Fig.6)
at the corresponding points of
Fig. 5: Generation of harmonic signals on periodic InP-HFET
measurements are plotted in Fig. 7
NLTL. The structure of NLTL is sketched at top of this figure.
(grey lines). The agreement of
(Data of simulation: L = 120 pH , C0 = 1.6 pF,
waveforms is satisfying demon− 13
R / L = 2 ⋅1010 s −1 , C 0 / G = 3.5 ⋅ 10 s )
strating that the fundamental
mechanisms of nonlinear wave
single pulse per period of the sinusoidal input sigpropagation on NLTLs have been considered in
nal. The simulation in Fig.6a demonstrates the
equation (1).
generation and compression of single pulses out
of the exiting input signal (6.5GHz, 2.5V) on a
Conclusion
graded NLTL with increasing values of L and C0
In this work the generation and compression
in direction of the propagating microwave. The
of picosecond pulses on InP-HFET NLTL is demtransient with minimum 10-90% fall time of 4ps
onstrated. Advances have been achieved by
and an amplitude of 1.8V is shown in Fig. 6b.
applying high speed InP-HFET diodes exhibitAfter fabrication of this graded InPHFET NLTL using self aligned opSignal voltage
1
tical contact lithography process(b)
0
es [7] the electro-optic sampling
set-up was modified making time
1V
-1
x
domain measurements (see P.
-2
Bussek et al., Time- and frequen0
80
120
40
cy domain electro-optic field map10ps
Time
(ps)
t
(b)
(a)
ping of nonlinear transmission
lines, in this annual report). In
Fig. 6: Simulation of pulse generation on a graded InP-HFET
Fig. 7 a top view of the graded
NLTL; (a) development of a sinusoidal signal along the NLTL,
InP-HFET NLTL is figured includ(b) transient with minimum fall time (Data of simulation:
ing the four points of measurement
L = 960 pH , C = 1.76 pF, R / L = 2 ⋅1010 s −1 ,
0
10 − 1
C 0 / G = 3.5 ⋅ 10 − 13 s , grading: 0.89αx , α = 8 .4 ⋅ 10 s )
54
3 RESEARCH
(a)
(b)
(c)
signal voltage (a.u.)
input
(a)
-80
-80
k = 11
-40
0
time (ps)
40
(c)
-80
k = 15
-40
0
time (ps)
signal voltage (a.u.)
(b)
(d)
signal voltage (a.u.)
signal voltage (a.u.)
output
40
k=7
0
-40
time (ps)
(d)
-80
40
k = 31
-40
0
time (ps)
experiment
40
simulation
Fig. 7: Pulse compression on graded InP-HFET NLTL (top view of the processed NLTL in the upper
left side of this figure, (a)-(d): points of electro-optic measurements)
ing strong nonlinearities. The numerical simulation has been improved by considering separately the influence of nonlinearity, periodic structure and losses to nonlinear wave propagation
on NLTLs.
optoelectronics“, Proc. IEEE, Vol. 82, No. 7, 1994,
pp. 1037-1059
[2] D. Jäger, “Characteristics of travelling waves
along nonlinear transmission lines for monolithic
integrated circuits: A review”, Int. J. Electron.,
Vol. 58, 1985, pp. 649-669
Acknowledgement
The author would like to thank Dipl.-Phys.
U. Auer (Fachgebiet Halbleitertechnik / -technologie) for processing of the transmission lines and
Dr. D. v.d.Weide (at that time: Max-Planck-Institut für Festkörperforschung, Stuttgart) for lending a suitable mask for optical contact lithography processes.
[3] D. Jäger, “Pulse generation and compression on
nonlinear transmission lines“, workshop on Picosecond and Femtosecond Electromagnetic
Pulses: Analysis and Applications, MTT-S Symp.
Dig., 1993, pp. 37-57
[4] R. Hülsewede et al, “CAD of pulse compression
on nonlinear transmission lines“, Proc. MIOP’ 95,
Sindelfingen, 1995, pp. 511-515
[5] U. Auer et al, „InP based HFETs with high qual-
References
ity short period InAlAs/InGaAs Superlattice
[1] M.J.W. Rodwell et al, “Active and nonlinear wave
Channel Layers“, J. o. Crystal growth, vol. 146,
propagation devices in ultrafast electronics and
1995
3.3 Millimeterwave Electronics
[6]
D.W. van der Weide, “Delta-doped Schottky diode nonlinear trans-mission lines for 480-fs, 3.5V transients”, Appl. Phys. Lett. Vol. 65, No. 7,
1994, pp.881-883
[7] C. Heedt et al, „On the Optimisation and Reliability of Ohmic- and Schottky Contacts to InAlAs/
InGaAs HFET“, Proc. 4th InP & Related Materials Conference, Newport, USA, 1992
[8] Report on the Special Collaborative Programm
SFB 254, 1993-1995, Gerhard-MercatorUniversität - GH - Duisburg, 1995
3.3.2 Millimeter wave power generation on nonlinear transmission lines
R. HÜLSEWEDE , V. K. MEZENTSEV, AND
I. V. R YJENKOVA
I
n this paper nonlinear transmission
lines are described which are used to
generate millimeterwave signals with high
efficiencies. In particular, arrays of monolithic varactor diodes loading a coplanar
waveguide are studied which can be applied
for travelling wave harmonic generation
where special phase matching and filter
structures give rise to high conversion efficiencies. A second transmission line consisting of any array of resonant tunneling diodes
is used as a distributed active device which
can generate millimeterwave power at frequencies as determined by a resonance condition of the resonator structure under study.
In this paper theoretical and numerical results are presented based upon experimental data.
55
Introduction:
The generation of millimeter waves by harmonic frequency generation and active wave
propagation along nonlinear transmission lines
(NLTLs) has recently become a subject of major research activities [1-4]. However, the power efficiencies achieved so far are small because
millimeterwave power is converted into undesired frequency components when the dispersion
and filter characteristics of the NLTL are not
designed in a suitable way. In this paper, firstly
we describe the bi-modal NLTL which uses concepts of nonlinear optics aiming towards achieving phase matching condition between the frequency components under study [5,6]. In
particular, we study a bi-modal NLTL where, as
an example, the phase velocity of the second or
third harmonic equals that of the fundamental
wave and where other components are suppressed by a suitable filter structure leading to
a distinct cut-off frequency [7-9]. Secondly, we
discuss the characteristics of a travelling-wave
tunneling-diode transmission line resonator capable of generating high power millimeter wave
signals [7-9].
In Fig. 1, the basic structure of an NLTL is
sketched consisting of a suitable array of nonlinear devices D in a passive coplanar waveguide [2,3]. As nonlinear elements, we have
studied Schottky diodes, quantum barrier var-
D
D
D
D
Fig. 1: Sketch of a nonlinear transmission
line
56
3 RESEARCH
actor structures (QBV), as well as resonant tunneling diodes (RTDs).
The second characteristic feature of the circuit in Fig. 1 is the dispersion which is mainly
determined by the arrangement of the diodes,
which can be periodic, bi-periodic, graded etc.
The dispersion itself controls the phase velocities of different spectral components and hence
the strength of interaction and the superposition in the time domain. Reflections at input and
output ports further determine the resonance
behavior of the whole structure.
Harmonic frequency generation:
We have studied nonlinear
wave propagation along NLTLs of
Fig. 2. In a first example the parameters used are those of an
experimental device on InP-HFET
substrate as discussed in [10].
Input frequency and power are
100GHz and 11dBm as delivered
into a small-signal characteristic
impedance of 50W. As a numerical tool we have used a continuum approximation on the basis of
a corresponding nonlinear evolution equation as described in
[10,11] and compared the results
with a CAD model based on a discrete representation of the NLTL
, cf.[10].
The results of our numerical
calculations are plotted in Fig. 3
showing the spatial distributions
of the amplitudes of the fundamental and second harmonic
wave. As can be seen, the amplitudes of the two waves at the in-
put are comparable, which leads to power efficiencies > 70% for second harmonic generation (SHG), here at 200GHz.
In a second numerical experiment we have
studied third harmonic generation (THG) on a
bi-modal NLTL of Fig. 2(b) assuming special
quantum barrier varactor diodes [12] with a symmetric capacitance voltage relationship. Fig. 4
presents the numerical results. Note that in this
case of THG phase matching is achieved between frequencies f1 = 72GHz and f3 = 216GHz.
As can be seen from Fig. 4, the 72GHz input
signal is converted into millimeter wave power
(a)
InGaAs
250 nm
n+
InGaAs
250 nm
n
InGaAs
25 nm
InAlAs
25 nm
undoped
InGaAs
25 nm
undoped
InGaAs
250 nm
n
500 nm
350 µm
n+
InGaAs
InP
undoped
C
Co
C
Co
(b)
InAlAs
50 nm
InGaAs
50 nm
InGaAs
300 nm
InGaAs
800 nm
InP
350 µm
nn+
nn+
C
Co
C
Co
(c)
InGaAs
500 nm
InGaAs
40 nm
n+
undoped
InAlAs
7.2 nm
undoped
InGaAs
4.3 nm
undoped
InAlAs
7.2 nm
undoped
InGaAs
40 nm
undoped
n+
InGaAs 500 nm
InP 350 µm
RTD
RTD
RTD
RTD
Fig. 2: Monolithic NLTL on InP substrate for millimeter wave
generation. (a) Bi-modal NLTL with Schottky varactor diodes for
SHG, (b) bi-modal NLTL with QBV for THG, and (c) RTD-NLTL
for a distributed oscillator.
3.3 Millimeterwave Electronics
57
1.2
0.6
1
0.5
f1
0.8
0.4
0.6
0.3
f = 2f1
0.4 2
0.2
0.2
0.1
0
0
20
40
60
80 100 120
number of elements, k
140
160
f1
f
3
0
0
20
40
80 100 120
60
number of elements, k
140
Fig. 3: Spatial distribution of amplitudes at
Fig. 4: Spatial distribution of amplitudes at
frequencies f1 and f2 for the NLTL of Fig.2(a).
frequencies f1 and f3=3f 1 for the NLTL of
160
Fig.2(b)
at 216GHz with an efficiency of about 25%. Again,
the third harmonic is available at the input of the
NLTL because the propagation characteristic is
that of a backward wave [5,6].
cillator and the frequency f(n, N) of the self-generated oscillations where n = 1, 2, …, N defines
the mode. Fig. 6 shows the results where the
dots represent the results of the simulations.
Clearly, a decreasing N leads to an increasing
f(n,N) because the wavelength, as given by
2 x N, decreases. In Fig. 6 a comparison with
analytical results is additionally carried out where
f(n, N) is given by
amplitude
voltage, V
Tunneling diode NLTL:
Very recently, another type of NLTL has become known where resonant tunneling diodes
are used as nonlinear elements [3]. However,
such RTD-NLTLs can also be used for nonlinear active wave propagation effects leading to a travelling wave
oscillator, when a transmission
1.6
line with limited length, provided
by short circuits at input and out1.2
put ports, for example, is used
0.8
[7,8]. In a numerical experiment
0 100 200 300 400
we have studied the generation
frequency, GHz
0.4
of millimeter waves in a RTDNLTL resonator. The results are
0
plotted in Fig.5 revealing self-gen-0.4
erated oscillation at 170 GHz after about 700 ps which is the
0
100
200
300
400
500
600
700
800
switch-on time.
time, ps
In a further numerical example
we have studied the relationship
Fig. 5: Generation of a 170 GHz signal on a tunneling diode
between the „length“ N of the osNLTL. The spectrum is shown in the inset.
58
3 RESEARCH
References
[1]
600
propagation in electronics”, John Wiley &
n=1
numeric
theory
500
400
A. Scott, “Active and nonlinear wave
Sons, New York, 1970
[2]
D. Jäger, “Characteristics of travelling
waves along nonlinear transmission lines for
300
monolithic integrated circuits: A review”, Int.
J. Electron. 58, 649-669 (1985) (invited pa-
200
per)
100
[3]
0
0
5
10
15
N
20
25
30
M.J.W. Rodwell, S.T. Allen, R.Y.Y. Yu,
M.G. Case, U. Bhattacharya, M.Reddy, E.
Carman, M. Kamegawa, Y. Konishi, J. Pusl,
R. Pullela, “Active and nonlinear wave propa-
Fig. 7: Oscillation frequency vs. number N of elements
gation devices in ultrafast electronics and op-
of the RTD-NLTL, theory according to eq. (1)
toelectronics”, IEEE Proc., vol. 82, no. 7, pp.
1037-1059, 1994 (invited paper)
[4]
f ( n, N ) =
1
π
1
LC
E. Carman, M. Case, M. Kamegawa, R. Yu, K.
Giboney, and M.J.W. Rodwell, “V-band and W-
sin( π2 Nn ) , n = 1,2...,N
band broadband, monolithic distributed frequency
multipliers,” in: 1992 IEEE MTT-S Digest, pp.
(1)
as calculated from the dispersion relation for a
cascaded LC - chain neglecting losses.
819-822, 1992
[5]
B. Wedding and D. Jäger, “Phase-matched second harmonic generation and parametric mixing
on nonlinear transmission lines”, Electron. Lett.
17, 76-77 (1981)
Conclusion
In this paper, specific NLTLs are presented
which are capable to generate millimeter waves
with high conversion efficiencies. The NLTLs are
compact, easily fabricated using standard InP
technology, suitable for monolithic integration,
and can provide high output powers. We therefore conclude that the travelling wave concept
under study can provide a solution to the problem of realizing efficient millimeter wave signal
sources.
[6] D. Jäger, “Nonlinear slow-wave propagation on
periodic Schottky coplanar lines”, IEEE Microwave and Millimeter-Wave Monolithic Circuits
Symposium, St. Louis 1985, Symp. Dig., 15-17
(1985)
[7] I. V. Ryjenkova, V. K. Mezentsev, S.L.Musher,
S. K. Turitsyn, R. Hülsewede, and D. Jäger, “Millimeter Wave Generation on Nonlineat Transmission Lines”, Proc.1996 International Workshop
on Millimeter Waves, April 11-12, Orvieto, Italy,
1996
[8] V. K. Mezentsev, S.L.Musher, I. V. Ryjenkova, S.
K. Turitsyn, R. Hülsewede, D. Jäger, “Travelling
wave generation of millimeter waves in bi-modal
3.3 Millimeterwave Electronics
NLTLs”, Proc. 26th European Microwave Conference 1996, Prague
[9] I. V. Ryjenkova, V. K. Mezentsev, S.L.Musher, S.
K. Turitsyn, R. Hülsewede, and D. Jäger, “Millimeter Wave Generation on Nonlineat Transmission Lines”, Ann. des Telecomm., Special Issue
(submitted).
[10] R. Hülsewede, U. Effing, I. Wolff, and D. Jäger,
“CAD of pulse compression on nonlinear transmission lines” , Proc. MIOP ’95, Sindelfingen, pp.
511-515
[11] M. Dragoman, R. Kremer, and D. Jäger, “Pulse
generation and compression on a travelling-wave
MMIC Schottky diode array”, in: Ultra-Wideband,
Short-Pulse Electromagnetics, H.L. Bertoni, L.
Carin, and L.B. Felsen, eds., Plenum Press, New
York, pp. 67-74, 1993
[12] M.A. Frerking, J.R. East, “Novel heterojunction
varactors”, IEEE Proc., vol. 80, no. 11, pp. 18531860, 1992
3.3.3 Nonlinear RTD circuits for
high-speed A/D conversion
I. JÄGER
I
59
range [1]. The underlying characteristic is a nonlinear N-shaped current voltage relationship even
at millimeterwaves. The lack of these very interesting devices, however, is the low power conversion efficiency and the small output power
levels [2]. Up to now the only solution to the latter
problem which has become known is the use of
a series i.e. distributed connection of several
RTDs using MMIC technology [3,4]. Such a RTD
nonlinear transmission line (NLTL) can further
provide the basis of very interesting microwave
signal processing devices as has been predicted
by Crane already in 1962 [5].
In this paper, we discuss first the fundamental
concept of nonlinear active wave propagation
effects along monostable RTD-NLTLs utilized to
generate a set of spikes from anelectrical input.
The idea of such a transmission line, where losses are exactly compensated by distributed amplification, dates back to the so called ´neuristor´
[5] as a line-analog of axons in the nervous system, where the information of an input signal is
converted into a number of output spikes, travelling in a stationary way for arbitrarily long distances. In a second step, we describe an electrical circuit, where the monostable RTD-NLTL is
the main building block to realize a n-bit A/D con-
n this report a novel nonlinear MMIC
structure based upon monostable resonant tunneling diodes (RTDs) is studied. For
the first time, it is shown that an input signal
can be converted into a set of output spikes
to be used for GHz A/D conversion.
LG
Introduction
A huge amount of work has recently been
dedicated to the study of resonant tunneling diodes (RTDs) which can provide gain and can
directly be used as the key components for oscillator circuits approaching the THz frequency
RTD
LG
RTD
Fig. 1: Sketch of a nonlinear array of
monostable resonant tunneling diodes in a
coplanar transmission line
60
3 RESEARCH
verter at GHz rates, similar to the lumped RTD
A/D converter described in [7-9].
RTD-circuit
The array of monostable RTDs is sketched
in Fig.1. One can see a coplanar transmission
line which is periodically loaded, cf.[3,4], with
RTDs shunted by LG circuits -here air bridges- in
order to provide monostable behavior, see [6].
The cross section of the MMIC structure in Fig. 1
has been described in [3,4].
The simulation carried out in this paper is
based upon a suitable equivalent circuit, as
shown in Fig.2. Each section consists of an Tequivalent representation of the transmission
line. The nonlinear element in Fig.2 is determined by the RTD current voltage relationship
approximated by , where an external bias current and V1,V2 > 0 have been assumed.
The basic idea of the circuit in Fig. 2 is roughly
the following. An input current source charges
the capacitance C up to a threshold value given
by J(V) of the RTD. A switching up occurs which,
however, will be inverted due to the LG time
constant. As a result, a spike is produced and
after an RC time constant another switching
occurs. Hence the spiking period is determined
by the amplitude of the input current. The trans-
R
C
J(V)
V
Fig. 2: Equivalent circuit of a
monostable RTD-NLTL
L
G
mission line itself ensures the generation and
propagation of identical pulses - such as solitons - formed after a few diodes.
Results
Fig.3 shows a numerical result for an input
sinusoidal wave of 25 GHz. As can be seen, the
monostable RTD-NLTL produces a set of 6 puls-
1,0
0,5
0,0
-0,5
-1,0
10
20
Time40
30
50
60
70
Fig. 3: Generation of five pulses
per period of sinusoidal input wave (dashed
line)
es per period. When the input frequency or the
input amplitude are changed, the number,
phase, and position of the spikes are altered in
a characteristic way.
Fig.4 shows an example where the width and
amplitude of a rectangular input signal have been
changed. As a result, the generated spiking as
obvious from the regions with different shadings
is a characteristic pattern for the input signal. In
particular, we observe that the number of spikes
per time depends linearly on the applied current
amplitude providing a linear voltage-frequency
conversion.
Such a NLTL can be used to realise a high
speed n-bit A/D converter similar to the lumped
version in [7-9]. Correspondingly, we propose
3.3 Millimeterwave Electronics
61
Resonant-Tunneling Diodes“, Appl.Phys.Lett.,
vol.58, no. 20, pp. 2291-2293, 1991
10.00
[2] R.Sun, O.Boric-Lubecke, D.-S.Pan, and T.Itoh,
9.00
„Considerations
8.00
and
Simulations
of
Subfrequency Excitation of Series Integrated
Resonant Tunneling Diodes Oscillator“,
7.00
IEEE Trans. Microwave Theory Techn., vol. MTT
6.00
- 43, no.10, pp. 2478-2485, 1995
5.00
[3] I.V.Ryjenkova, V. K.Mezentsev, S.L.Musher,
S.K.Turitsyn, R.Hülsewede, and D.Jäger, “Milli-
4.00
meter Wave Generation on Nonlineat Transmis-
3.00
2.00
2.00
sion Lines”, Proc.1996 International Workshop
3.00
4.00
5.00
6.00
7.00
8.00
on Millimeter Waves, April 11-12, Orvieto, Italy,
1996
Fig. 4: Contour plot of generated
number of spikes (see text)
[4] [4] I.V.Ryjenkova, V.K.Mezentsev, S.L.Musher,
S.K.Turitsyn, R.Hülsewede, and D.Jäger, “Millimeter Wave Generation on Nonlineat Transmission Lines”, Ann. Telecomm., Special Issue,
to realise a common n-channel (for n bits) coplanar signal devider to provide input amplitudes by
powers of 2. Hence each channel delivers a spike
train to the output array establishing a Gray code
as in Ref.[7]. In the present case, the LC time
constant will determine the bandwidth which exceeds 100 GHz in the device under test.
vol.52, No 3-4, pp. 134-139, 1997
[5] H.D.Crane, ‘Neuristor - A Novel Device and System Concept’, Proc. IRE, vol.50, pp. 20482060, 1962
[6] J.Nagumo, S.Arimoto, and S.Yoshizawa, „An Active Pulse Transmission Line Simulating Nerve
Axon“, Proc. of the IRE, vol.50, p.2061, 1962
[7] T.-H..Kuo, H.C.Lin, R.C.Potter, D.Shupe, „ A
Conclusion
In conclusion, a novel monostable RTD-NLTL
in MMIC technology is proposed which can generate a characteristic pulse pattern for a given
input signal. The application of such a NLTL for
an ultrafast A/D conversion is discussed in a
second step. The use of RTDs in the presented
circuit is expected to yield a bandwidth in excess of 100 GHz.
Novel A/D Converter Using Resonant Tunneling Diodes“, IEEE Journal of Solid-State Circuits,
vol.26, No.2, pp.145-149, 1991
[8] S.-J.Wei, H.C.Lin, R.C.Potter, D.Shupe, „ Dynamic Hysteresis of the RTD Folding Circuit and
ist Limitation on the A/D Converter“, IEEE Transaction on Circuits and Systems II: Analog and
Digital Signal Processing, vol.39, No.4, pp.247251, 1992
[9] S.-J.Wei, H.C.Lin, R.C.Potter, D.Shupe, „ A Self-
References
Latching A/D Converter Using Resonant Tunnel-
[1] E.R.Brown, J.R.Söderström, and T.C.McGill, „
ing Diodes“, IEEE Journal of Solid-State Circuits,
Oscillations up to 712 GHz in InAs/AlSb
vol.28, No.6, pp.697-700, 1993
62
3 RESEARCH
3.4 Optical Sensor Systems
3.4.1 MQW-Electroabsorption-Modulator for Application in a fiberoptic
fieldsensor
M. SCHMIDT, R. HEINZELMANN , AND A. STÖHR
I
n this report we present electroabsorption waveguide-modulators for an operation wavelength of 1.55µm. These devices
are fabricated for an application in a fiberoptical E-field sensor system [1], [2]. In this
system the task of the modulator is to convert electrical signals with frequencies up to
6 GHz into optical signals.
Introduction
In recent years there has been an increasing
interest in electrooptical modulators. The main
application of these devices is in fiberoptical
communication systems for the external modulation of laserdiodes. Electroptical modulators
have been realised in lithium niobate as well as
in semiconductors using the Franz-Keldish-effect in bulk materials and the quantum confined
starck effect in multiple quantum well structures.
As MQW structures exhibit the strongest electrooptic effect they allow the use of smaller electrodes than the other mentioned modulator principles. The resulting lower capacitance has the
advantage of higher cut off frequencies. Furthermore MQW modulators can be made insensitive to the polarisation of the modulated light by
introducing tensile strain to the quantum wells.
This avoids the need for expensive polarisation
maintaining fibers.
Layer structure:
contact
n InAlAs
nid InAlAs
InGaAs/InAlAs - MQW
nid InAlAs
n+InAlAs
s. i. InP
Fig. 1: Sketch of the electroabsorption waveguide modulator
3.4 Optical Sensor Systems
0
100 %
63
Field (106 V / m)
10
5
waveguide is formed by wet chemical etching
with cytric acid down to the n+ layer. Afterwards
the electrical contact are produced by vacuum
coating. The ohmic contact on the n+ layer is
realised in GeNiAu, for the Schottky contact on
the n- layer we use CrAu.
The design of the device structure was supported by BPM-Simulations and by calculations
of the electrooptical behaviour. For the BPMSimulations we used the comercial BPM-Software BPM-Cad. The main aim of this simulation
was to determine the optical confinement factor, i. e. which part of the guided modes overlaps with the absorbing MWQ layer. We calculated a confinement factor of 14 %. For the
calculation of the absorption coefficient of the
MQW-material in dependence of the electrical
field we used a transfer matrix method. From
the obtained results we calculated the optical
absorption of the modulator in dependence of
the applied electric field as shown
in Fig. 2.
15
80 %
Absorption
Slope: 0,38 / V
60 %
40 %
20 %
Wavelength: 1.55µm
Device lenght: 500 µm
0%
0
2
4
6
Voltage [V]
Fig. 2: Calculated electrooptical behaviour
of a modulator device.
Device structure
In Fig. 1 a sketch of the device is shown. The
modulators are grown by MBE in the ternary
material system InGaAs/InAlAs on InP subtrates.
The device structure is n + in -. The ridge
1580
PL-Measurement at 12 K
Simulation at 12 K
Simulation at 300 K
1560
Excitonic wavelength [nm]
1540
1520
1500
1480
1460
1440
1420
Mod 05
Mod 07
Mod 09
Mod 08
1400
Mod 13
1380
Shg 05
Shg 07
Shg 03
Shg 04
Shg 06
Mod12
Shg 08
1360
1340
6
7
8
9
Thickness of quantum-wells [nm]
Fig. 3: Excitonic wavelength of the MQW material determined
by photoluminescence measurements compared with calculated
values.
10
Experimental results
The epitaxial wafers were characterized by photoluminescence
measurements. The point of interest was the spectral position of
the excitonic peak of the MQW
material, which indicates the position of the absorption edge. A
comparison between the experimental results and the calculations is shown in Fig. 3.
The optical transmission of the
modulator is characterized by
coupling the light of an erbium
doped fiber laser into the
waveguide and detecting the
transmitted light on the other fac-
64
3 RESEARCH
[2] Heinzelmann, A. Stöhr, M. Groß, D. Kalinowski,
0,5
T. Alder, M. Schmidt, and D. Jäger, “Optically
Powered Remote Optical Field Sensor Sys-
Modulation [a.u.]
0,4
tem using an Electroabsorption-Modulator“,
0,3
IEEE MTT-S International Microwave Sympo-
λ = 1550 nm
sium, Conference Proceedings, Baltimore, June
0,2
1998
0,1
0,0
0
-1
-2
-3
-4
Voltage [V]
Fig. 4: Modulation of the MQW modulator
versus applied voltage.
et of the waveguide. The optical tranmission is
measured in dependence of the applied voltage
as shown in Fig. 4.
Conclusions
An electrooptical MQW waveguide modulator has been designed. The epitaxial layers have
been grown by MBE and characterised by photoluminescence measurements. The position of
the measured exciton peaks was in good agreement with the calculated values. Modulator devices have been processed from these wafers
by wet chemical etching and vacuum coating of
the electrical contacts. The devices have been
characterized by optical transmisson measurements.
References
[1] Stöhr, R. Heinzelmann, T. Alder, M. Schmidt, M.
Groß, and D. Jäger, “Integrated Optical E-Field
Sensors
using
TW
EA-Modulators“,
Interational Topical Workshop on Contemporary
Photonic Technologies CPT’98, Technical Digest, Tokyo, January 1998
3.4.2 Photovoltaic cells for fiber optic
EMC - Sensor power supply
D. KALINOWSKI
P
hotovoltaic cells play an important
role in power supply of hybrid sensors. A photovoltaic cell array is under construction to supply an active fiber optic hybrid
sensor head. A prototype with first
experimental results will be shown.
Introduction
Due to more and more restrictive laws regulating the electromagnetic compatibility (EMC)
of electronic equipment the necessity of developing precise and reliable sensors to measure
electromagnetic fields steadily increases. One
request for such sensors is non invasiveness.
Hence, our approach to reach this goal is to
develop a hybrid fiber optic fiels sensor. This
concept takes advantage of the fact, that optical fibers do not interfere with the electromagnetic field that is to measure, but that they are
capable to transmit optical information. By this
distortion of the E-field is minimized. The photovoltaic cell array (PVC) described in this article
is part of this optical E-field sensor which is
shown in Fig. 1.
3.4 Optical Sensor Systems
65
active region is 600
µm, i.e. the diameter of
the core of the multimode fiber used. Thus,
each quarter of this
array, i.e. each PVC, is
illuminated uniformly
leading to a maximum
generation of electrical
power by this configuration [1].
Fig. 1: Sketch of the optically powered integrated optical field sensor
Device
One requirement to be matched by the PVC
is a high efficient conversion of optical into electrical power. Therefore, special effort has to be
laid upon the layout and the composition of the
heterostructures. Since cheap and powerful laser diodes are available in the 800 to 850 nm
wavelength regime and since GaAs has its optimum photovoltaic response at about 800 nm,
the PVCs are designed as AlGaAs/GaAs pindiodes. Fig. 2 shows a photograph of a cell array consisting of 4 cells. The diameter of the
Fig. 3: Cross section of the photovoltaic cells
The GaAs and AlGaAs layers are MBE grown
on semi-insulating GaAs substrate. The layer
structure, illustrated in Fig. 3, consists of an 100
nm n+-AlGaAs contact layer, a 2 µm i-GaAs absorption layer and a 100 nm p+-AlGaAs window
layer. A 10 nm p+-GaAs contact layer offers an
aluminium protection from oxidation.
The metallic contacts act as ohmic contacts.
GeNiAu is used for the n-contact and PtTiPtAu
for the p-contact.
Fig. 2: Perspective view of the photovoltaic
cell array consisting of 4 cells
66
3 RESEARCH
3.4.3 Time- and frequency-domain
electro-optic field mapping of nonlinear transmission lines
P. BUSSEK, TH. BRAASCH, AND G. DAVID
W
Fig. 4: Photovoltaic cell efficiency
Results
The cell array is illuminated by a laser with a
emission wavelength of 800 nm. Measurements
of the max. efficiency show results up to 28%
depending on the optical input power (Fig. 4).
Conclusion
This report presents a photovoltaic cell array
for power supply of our hybrid EMC - Sensor.
The design and a first result are shown.
References
[1] M. B. Spitzer, et. al., “Monolithic series-connected gallium arsenide converter development”,
Proc. 22nd IEEE Photovoltaic Specialists Conference, Las Vegas, USA, 1991
e report on the measurements of
electric field distributions in monolithic microwave integrated circuits (MMICs)
using electro-optic probing techniques. In
1996 the activities have been focused on the
analysis of microwave propagation in nonlinear transmission lines (NLTLs). The measurements have been performed in frequency
domain as well as in time domain as they
have been done in one dimension as well as
in two dimensions. As an example, in this
documentation we present experimental
results of periodic NLTLs demonstrating the
generation of higher harmonics on these
devices and the formation of shock waves.
Introduction
In recent years, the complexity of monolithic
microwave integrated circuits (MMICs) expanded necessitating the development of measurement techniques which keep abreast of the increased demands of an appropriate
characterization of these devices. So far, network-analyzers (NWA) are mostly used for onwafer microwave characterization of MMICs.
This measurement technique is well established
but its application is limited due to the fact that
the on-wafer probes needed for this technique
only allow the access to external ports. Thus,
no circuit-internal measurement or local failure
test of the device nore the observation of wave
propagation effects is possible using NWAs.
In contrast, electro-optic sampling has become a sophisticated technique to study quan-
3.4 Optical Sensor Systems
titatively field distributions and wave propagation
effects insight microwave and millimeter-wave
devices [1-3]. This technique can be performed
in frequency as in time domain enabling the
detection of the amplitude and phase of a
microwave signal as of the temporal evolution
of this signal. The spatial resolution of this method is measured to be down to less than 0.5 µm
[4]. Hence, each MMIC component can be tested and evaluated noninvasively up to millimeter-wave frequencies. By combining the direct
electro-optic probing with 2D scanning of the
laser beam two-dimensional field mappings of
the device under test (DUT) are possible [5].
Thus, wave propagation effects can be studied
which gain a growing interest by circuit designers for two reasons. On one hand, these effects
influence the electrical behaviour of MMICs resulting in a limitation of their bandwidth or the
67
generation of unwanted modes [3]. On the other
hand, novel types of integrated circuits such as
nonlinear transmission lines (NLTLs) can be
designed which make use of these effects, e.g.
to generate short electrical pulses or to excite
higher harmonics. This has been observed particularly in periodic NLTLs and will be shown in
this report.
Experimental setup
The experimental setups used in this project
are sketched in Fig. 1. An actively modelocked
Nd:YAG laser (wavelength = 1064 nm, pulse
repetition rate = 82 MHz) combined with a fibergrating pulse compressor provides short pulses
of 5 ps FWHM (full width at half maximum) corresponding to a bandwidth of the setup in excess of 80 GHz. The device under test (DUT) is
illuminated from the backside, i.e. the direct elec-
Fig. 1: Experimental setup, (a) for the frequency domain measurements, (b) for the time domain
measurements
68
3 RESEARCH
tro-optic sampling is applied since the
linear electro-optic effect in the sub-80
15 GHz
strate itself is used for the modulation
-90
of the polarization. To convert this po-100
30 GHz
larization modulation into an intensity
-110
modulation, polarizers and a quarterwave plate are implemented in the op-120
45 GHz
60 GHz
tical pathway. The reflected intensity is
-130
detected by a small area photodiode.
-140
Due to the combination of a small area
0
500
1500
2500
photodiode and the confocal arrangepropagation distance (µm)
ment of the setup out-of-focus-light is
Fig. 2: Electro-optic signal of the fundamental microwave
suppressed improving the spatial resat 15 GHz and of its higher harmonics along the center
olution of the measurement system
conductor of a periodic NLTL from input to output.
down to less than 0.5 µm [4]. For 2D
scans the probe stage is movable in
For the measurements performed in time
the x- and y-direction.
domain some modifications of the experimental
Fig 1(a) illustrates the experimental setup for
setup are needed. As depicted in Fig. 1(b), the
frequency domain measurements. Here, a specoptical pulses themselves generate the electritrum analyzer working as a tunable bandpass is
cal signal in order to establish a phase locking
used for the detection of the signal amplitude.
between the probe pulses and the electrical miThe intermediate frequency is set to several
crowave signal. A small part of the output beam
MHz, since in this regime the high speed avaof the Nd:YAG laser is separated via a beam
lanche photodiode used exhibits a maximum
splitter and chopped at about 4 kHz. The photosensitivity. The microwave synthesizer, the modcurrent of a fast photodiode detecting this outelocker synthesizer of the laser system and the
put signal then traverses a mechanical delay line
spectrum analyzer are phase stabilized via a
that periodically shifts the phase of the signal
phase locked loop (PLL). For the phase meawhile the observation point is kept constant. The
surements the spectrum analyzer is replaced by
photodiode has to be changed by a slow Gea lock-in amplifier (not shown in Fig. 1(a)). In a
diode to apply lock-in techniques at the chopsecond mode this setup is used to receive an
ping frequency.
optical image of the measurement region by simply detecting its front surface reflectivity. Thus,
Frequency domain measurements
the electro-optical signal can be normalized to
The results presented here all have been
the particular reflectivity of the device, and the
done with periodic nonlinear transmission lines.
absolute value of the voltage between the deFor a more detailed description of the examined
vice‘s top and bottom surface can be determined
samples see R. Hülsewede, Investigations of
[6].
3.4 Optical Sensor Systems
69
plitudes of the
higher harmonics increase indicating that
they
are
generated
along the transmission line.
The obvious
standing wave
patterns are
caused by an
impedance
mismatch at
the end of the
line and phase
mismatching of
the harmonics.
Tw o - d i m e n sional field
mappings of
an NLT L are
shown in figs.
3. Here, the
frequency of
the fundamental is 6 GHz,
Fig. 3: Nonlinear transmission line; (a) metallization structure; results of 2D field
and the metalmappings (b) at the fundamental at 6 GHz, (c) at the second harmonuc at 12 GHz
lization strucand (d) at the third harmonic at 18 GHz.
ture of the depulse compression on nonlinear transmission
vice, the electro-optic signal of the fundamental,
lines, in this annual report. Fig. 2 depicts the
the second harmonic at 12 GHz and the third
spatial distribution of the incident fundamental
harmonic at 18 GHz are presented in Figs. 3(a)
electrical signal at 15 GHz and the amplitudes
- (d), respectively. These Figs. show the decrease
of the second, third, and fourth harmonic with
of the fundamental signal and the increase of the
frequencies up to 60 GHz. As can be seen the
harmonics while propagating along the NLTL as
amplitude of the fundamental signal decreases
Fig. 2 does, but additionally they reveal an unin the direction of propagation whereas the amsymmetrical distribution of the electro-optic signal
70
3 RESEARCH
k=1
k=5
(a)
-80
-40
0
40
time (ps)
k = 10
(b)
80
-80
-40
0
time (ps)
40
80
k = 14
higher harmonics. In the
time domain, this formation of a shock wave is the
counterpart to the
generation of harmonics in
the frequency domain.
The presented results
validate that the electrooptic probing technique is
capable of studying and
demonstrating this effect
as well.
Conclusion
In summary, electrooptic measurement tech(d)
(c)
niques have been used
-80
-40
0
40
80
-80
-40
0
40
80
to internally investigate
time (ps)
time (ps)
wave propagation effects
along periodic nonlinear
Fig. 4: Electro-optic signal of the fundamental microwave at 15 GHz and
transmission lines enof its higher harmonics along the center conductor of a periodic NLTL
abling circuit-designers to
from input to output.
get an insight into the inthat can not be detected with one-dimensional
circuit electrical characteristics of complex milinescans as is the case in Fig. 2. We contribute
crowave devices. The generation of harmonics
this behaviour to the excitation of parasitic propand the formation of shock waves have been
agation modes [3].
demonstrated showing, that this method is suitable to examine internal field distributions in
Time domain measurements
MMICs in both, frequency domain and time doIn time domain measurements there is a fixed
main.
phase relation between each particular measurement point. Thus, the evolution of a periodReferences
ic signal can be observed as is elucidated in
[1] K.J. Weingarten, M.J.W. Rodwell, and D. Bloom,
Figs. 4(a) to 4(d) for the development of a sinu“Picosecond optical sampling of GaAs integrated
soidal electrical signal of 6 GHz propagating
circuits”, IEEE J. Quantum Electron., vol. QEalong a periodic NLTL at the 1st, the 5th, the
24, (1988), pp. 198-220
10th and the 14th diode, respectively. As can
[2] G. David, S. Redlich, W. Mertin, R.M. Bertenburg,
be seen, shock waves are generated with fall
S. Kosslowski, F.J. Tegude, E. Kubalek, and D.
times down to 5 ps due to the interaction of the
Jäger (1993), “Two-dimensional direct electro-
3.4 Optical Sensor Systems
optic field mapping in a monolithic integrated
GaAs amplifier”, Proc. 23rd EuMC 1993, Madrid,
71
dyne electro-optic measurement setup and
present first experimental results.
Spain, 1993, pp. 497-499
[3] G. David, R. Tempel, I. Wolff, and D. Jäger,
“Analysis of microwave propagation effects using 2D electro-optic field mapping techniques”,
Optical and Quantum Electronics, Special Issue
on Optical Probing of Ultrafast Devices and Integrated Circuits, vol. 28, 1996, pp. 919-931
[4] G. David, P. Bussek, U. Auer, F.J. Tegude, and
D. Jäger, “Electro-optic probing of RF signals in
submicrometre MMIC devices”, Electron. Lett.,
1995, Vol. 31, No. 25, pp. 2188-2189
[5] M.G. Li, E.A. Chauchard, C.H. Lee, and H.-L.A.
Hung, “Two-dimensional field mapping of GaAs
microstrip circuit by electrooptic sensing”, Proc.
OSA Int. Top. Meeting `Picosecond Electronics
and Optoelectronics`, March 13-15, 1991, Salt
Lake City, USA, pp. 54-58
[6] G. David, W. Schröder, D. Jäger, and I. Wolff,
“2D electro-optic probing combined with field
theory based multimode wave amplitude extraction: a new approach to on-wafer measurement”,
Symposium Digest 1995 IEEE MTT-S International Symposium, May 15 -19, 1995, Orlando,
USA, pp. 1049-1052
3.4.4 Characterization of monolithic
microwave integrated circuits by
heterodyne electro-optic sampling
TH. BRAASCH
T
he propagation of electric signals in
the millimeter- and microwave regime
along monolithic microwave integrated circuits (MMICs) can be studied by electro-optic measurement techniques. In this paper,
we describe the implementation of a hetero-
Introduction
Using common network analyzer methodes
(NWA) for the characterization of MMICs, the
device under test (DUT) is measured as an integral device and no in-circuit measurements are
possible [1]. The increasing working frequencies
of integrated circuits up to the millimeter- and
microwave region [2-4] necessitate measurement techniques that allow an insight into the
device. In recent years, electro-optic sampling
(EOS) has become a sophisticated technique
to observe field distributions in MMICs with a
spatial resolution down to less than 0.5µm [5,
6]. Quantitative characterizations have been
carried out, and one- as two-dimensional measurements are possible in time-domain as in frequency domain [7, 8]. Thus, circuit-designers get
a knowledge of circuit-internal parameters, which
is of increasing importance since the complexity of the devices expands. Nevertheless, so far
the EOS has been mostly performed with a
pulsed laser. Here, to convert the microwave
down to frequencies, where spectrum analyzer,
lock-in amplifier and the photodiode used are
able to detect the electro-optic signal, the n-th
harmonic of the repetition frequency of the
pulsed laser is used to interact with the electric
signal. The electric bandwidth of these setups
is limited by the pulse width of the laser pulses.
In frequency domain, measurements on nonlinear transmission lines up to 100 GHz are reported [9, 10], and fall times down to 1.5 ps have
been measured in time domain [11]. However,
utilizing the n-th harmonic of the repetition frequency of the pulsed laser for the down conversion of the electric signal leads to a reduction of
the signal to noise ratio of the electro-optic sig-
72
3 RESEARCH
of the beat frequency between
the two lasers.
After passing a l/
4 polarization
control the light of
the first laser is
back-side coupled into the device under test
where the stray
field of the microwave in the substrate interacts
with the laser
light via the Pockels-effect. The
Fig. 1: Sketch of the heterodyne electro-optic measurement setup
reflected light is
again coupled
nal since the phase noise of the setup increases
into the fiber and traverses the circulator and a
with the measurement frequency. To circumvent
second polarization control. Thus, the polarizathis restriction two cw lasers can be used where
tion modulation due to the Pockels-effect is
the second laser acts as a local oscillator. Here,
converted into an intensity modulation directly corphase noise of the setup only depends on the
related to the strength of the electric field at the
stability of the two lasers [12]. In this paper, we
particular point of measurement. The DUT is
present our measurement setup and show first
placed on a translational stage enabling twoexperimental results.
dimensional field mappings of the field distribution. Due to the confocal arrangement of the setup
Experimental setup
it can also be used as an optical microscope. In
We operate with two identical Er-doped fiber
this mode, the front surface reflectivity of the
lasers exhibiting a linewidth < 10 kHz. Both ladevice can be detected and afterwards the elecsers are continuously tunable between 1530 nm
tro-optic signal can be normalized to the particuand 1560 nm leading to beat frequencies up to
lar reflectivity at each particular measuring point.
4 THz. They deliver > 20 mW optical output and
As a consequence, the absolute value of the
show almost no mode-hops once they reached
voltage between the device‘s top and bottom side
thermal equilibrium. The degree of polarization
can be determined [8]. Via a fiber coupler the
is > 99%. Fig. 1 demonstrates the configuration
local oscillator, i.e. the second laser, is
of our setup. A small percentage of both lasers
superposed to the reflected light from the DUT
is coupled into a Fabry-Perot optical spectrum
carrying the information of the microwave signal
analyzer with a finesse >150 for the detection
applied to the DUT. The intermediate frequency
3.4 Optical Sensor Systems
73
Fig. 2: (a) Sketch of a coplanar waveguide structure, (b) reflected intensity measured with the heterodyne setup at 8.5 GHz.
between the second laser and one sideband of
the first laser, i.e. f1 ± fm with f1 the frequency of
the first laser and fm the microwave frequency,
can now be adjusted by the frequency f2 of the
second laser. This intermediate frequency is detected by a fast travelling-wave photodetector
[3]. Hence, any microwave frequency within the
tuning range of the two lasers can be converted
to some MHz or GHz only affected by the inherent
phase noise of the two lasers but independently
of the frequency.
Results
A coplanar waveguide structure (CPW) was
used to demonstrate the feasibility of the setup
as a scanning microscope. Fig. 2 depicts the
surface reflectivity of the CPW. The difference
frequency between the lasers was arbitrarily set
to 8.5 GHz since at this value they worked extremely stable and the detected signal of the
spectrum analyzer was about 50 dB larger than
the noise floor. In the next step, a microwave
has now to be applied to the device and the elec-
tro-optic signal has to be detected as in [6], [8]
or [10] with the pulsed laser system.
Conclusion
In summary, to bypass the phase noise restrictions of a pulsed electro-optic measurement
setup two narrow linewidth tunable cw lasers
have been implemented in the configuration.
Thus, the phase noise only depends on the characteristics of the lasers but is not affected by
the measurement frequency. Owing to the tuning range > 30 nm of the Er-doped fieber lasers
used heterodyne detections of electric signals
up to 4 THz should be possible. As a first result,
the surface reflectivity of a coplanar waveguide
structure detected at 8.5 GHz difference frequency is presented.
References
[1] D.J. Bannister and M. Perkins, Tracebility for onwafer s-parameter measurements, IEE Proc. A,
vol. 139, 5, 1992, pp. 232-233
74
3 RESEARCH
[2] M.J.W. Rodwell, S.T. Allen, R.Y. Yu, M.G. Case,
[9] R. Majidi-Ahy, B.A. Auld, and D.M. Bloom, 100
U. Bhattacharya, M. Reddy, E. Carman, M.
GHz on-wafer s-parameter measurements by
Kamegawa, Y. Konishi, J. Pusl, and R. Pullela,
electro-optic sampling, IEEE MTT-S, 1989, pp.
Active and nonlinear wave propagation devices
299-302
in ultrafast electronics and optoelectronics, Proc.
IEEE, vol. 82, 7, 1994, pp. 1037-1060
[10] Th. Braasch, G. David, R. Hülsewede, U. Auer,
F.-J. Tegude, and D. Jäger, Propagation of mi-
[3] M. Alles, Th. Braasch, R. Heinzelmann, A. Stöhr,
crowaves in MMICs studied by time- and fre-
and D. Jäger, Optoelectronic devices for micro-
quency-domain electro-optic field mapping, Proc.
wave and millimeterwave optical links, Proc.
“Trends in Optics and Photonics Series (TOPS)”
MIKON‘96, Workshop Optoelectronics in Micro-
of OSA 1997 Spring Topical Meeting “Ultrafast
wave Technology, Warsaw, Poland, 1996 (in-
Electronics and Optoelectronics”, 1997, Lake
vited)
Tahoe, USA
[4] I.V. Ryjenkova, V.K. Mezentsev, S.L. Musher,
[11] K.S. Giboney, S.T. Allen, M.J.W. Rodwell, and
S.K. Turitsyn, R. Hülsewede, and D. Jäger, Mil-
J.E. Bowers, Picosecond measurements by free-
limeter wave generation on nonlinear transmis-
running electro-optic sampling, Phot. Tech. Lett.,
sion lines, Ann. des Telecomm., Special Issue,
vol. 6, 11, 1994, pp. 1353-1355
1996 (invited)
[12] S. Loualiche and F. Clerot, Electro-optic micro-
[5] K.J. Weingarten, M.J.W. Rodwell, and D. Bloom,
wave measurements in the frequency domain,
Picosecond optical sampling of GaAs integrated
Appl. Phys. Lett. 61, (18), 1992, pp. 2153-2155
circuits, IEEE J. Quantum Electron., 1988, QE24, pp. 198-220
[6] G. David, P. Bussek, U. Auer, F.J. Tegude, and
D. Jäger, Electro-optic probing of RF signals in
submicrometre MMIC devices, Electron. Lett.,
1995, Vol. 31, No. 25, pp. 2188-2189
3.4.5 Development of an experimental setup for field probe measurements on nonlinear transmission
lines
[7] M.G. Li, E.A. Chauchard, C.H. Lee, and H.-L.A.
Hung, Two-dimensional field mapping of GaAs
microstrip circuit by electrooptic sensing, OSA
Proc. Picosecond Electronics and Optoelectronics, March 13-15, 1991, Salt Lake City, USA, pp.
54-58
[8] G. David, R. Tempel, I. Wolff, and D. Jäger,
Analysis of microwave propagation effects using 2D electro-optic field mapping techniques,
Optical and Quantum Electronics, Special Issue
on Optical Probing of Ultrafast Devices and In-
D. KALINOWSKI
AND
R. HÜLSEWEDE
A
n experimental setup for field probe
measurements has been established.
The electrical fields on nonlinear transmission lines has been measured up to 60GHz.
Theoretical considerations have been done
up to 200GHz. Experimental results have
been compared with results using electrooptic testing.
tegrated Circuits, 1996, 919-931
Introduction
In recent years there has been a great
progress in the development on nonlinear trans-
3.4 Optical Sensor Systems
75
mission lines making them to key structures for
future microwave circuits [1]. To characterize
these structures measurements at external ports
are not sufficient. Different noncontacting probes
give a chance to take a look at the field distribution on the transmission lines. The probes are
the electro-optic probe [2], the magnetic field
probe [3] and the electric field probe [4]. Such
an electric field probe has been established. Its
potential has been demonstrated.
Experimental setup
A sketch of the experimental set-up is shown
in Fig. 1. The nonlinear transmission lines
(NLTLs) are supplied by a microwave synthesizer and a DC-voltage source. They can be
loaded with variable resistance. The probe detects the electric field above the NLTL. Its signal is evaluated by a spectrum analyzer. A computer controls the position of the probe and
stores the data measured by the spectrum analyzer. By that way two-dimensional field mappings can be done.
Theoretical results
The probe can be described by an equivalent circuit which is shown in Fig.2 [5]. The voltage Ul depends on the electric field strength.
y
U
probe
The probe impedance is given by RV and RS which
characterize the thermal and radiation losses and
X A describing the open line.
The relation between the indicated power at
the spectrum analyser and the square of the
measured field intensitity is shown in Fig. 3. It
demonstrates the steady increasing probe sensitivity versus frequency. Thus the probe can
be used over the whole frequency range.
Measurements confirm this equivalent circuit.
Fig. 4 shows the relation between the indicated
signal on the spectrum analyser and the signal
frequency. The experimental results are given
by the dots (·), the theoretical results by the line
(-). A good correspondence is given between
these results. So the equivalent circuit can be
used to determine the electric field strength by
the spectrum analyzer signal.
0,8
x
synthesizer
z
load
bias tee
dc - source
0,6
0,4
0,2
0
prober
jX
A
Fig. 2: Equivalent circuit for the electric field
pA
)
Vm
mixer
Rs
Ul
sensitivity (
spectrumanalyser
Rv
NLTL
Fig. 1: Sketch of the experimental Setup
0
40
80
120
160
frequency (GHz)
Fig. 3: Sensitivity versus frequency.
200
76
3 RESEARCH
rel. signal (dB)
-26
-30
-34
-38
-42
theory
measurement
-46
-50
5
10
15 20 25 30
frequency (GHz)
35
40
Experimental results
One experimental result achieved with this
setup is presented in Fig. 5. A 4 µm NLTL has
been examined. The sketch of this line is shown
in Fig. 5c. Whereas one end has been connected with a synthesizer, the other end has been
unloaded. The synthesizer has supplied the line
with a 7.4 GHz microwave signal. The field distribution at this frequency is shown in Fig. 5a.
The 3rd harmonic generated on the line is shown
in Fig. 5b. The amplitude increases in the direction of propagation indicating the generation
along the transmission line. Furthermore an
asymmetrical transversal distribution is revealed.
Conclusion
Measurements up to 60 GHz have been done
successfully. Field distributions on transmission
lines with electrode widths down to 12 µm has
been shown. Comparisons with theoretical results and electo-optic probing are in good agreement.
References
[1] M.J.W. Rodwell, et al., “Active and nonlinear
wave propagation devices in ultrafast electronics and optoelectronics”, IEEE Proc., Vol. 82, No.
7, pp. 1037-1059, 1994
[2] P.Bussek, G. David, “Quantitative analysis of
two-dimensional electro-optically measured field
distributions in MMIC-structures”, Annual Report,
Gerhard-Mercator-Universität - GH - Duisburg,
Fachgebiet Optoelektronik, 1995
[3] Y. Gao, I. Wolff, “A new miniature magnetic field
probe for measuring three-dimensional fields in
planar high-frequency circuits”, IEEE Transactions on Microwave Theory and Techniques, Vol.
Fig. 5:2-dim. field mapping of an NLTL (a)
signal with 7.4 GHz, (b) generated 3rd
harmonic at 22.2 GHz c) Sketch of the NLTL
44, No. 6, June 1996, pp. 911-918
3.5 Technologies for Optoelectronic Components and Systems
[4] D. Kalinowski “Entwicklung eines Feldsondenmeßplatzes zur zweidimensionalen Analyse
elektromagnetischer Feldverteilungen auf
77
3.5 Technologies for Optoelectronic Components and
Systems
nichtlinearen Leitungen”, Diploma thesis,
Gerhard-Mercator-Universität Duisburg, 1996
[5] R. Geißler, et al., “Taschenbuch der Hochfrequenztechnik Band 2: Komponenten”, Springer-
3.5.1 Development of a measurement
system for the optical characterization of full-colour-LED-displays
Verlag, Berlin-Heidelberg, 1992
M. WENNING, R. BUß,
AND
A. STÖHR
F
or the optical characterization of a fullcolour-LED-display, two measurement
systems have been developed. The first one
is to determine the spatial distribution of a
LED and the second one is to receive the
spectrum of a LED or a LED-pixel. The x, y, z
colour coordinates of the CIE chromaticity
diagram are evaluated from the spectrum. By
using the measurement system, an
optimization of a full-colour LED-display was
performed.
Introduction
Among the various types of flat panel displays
(i.e., CRT, VFD, PDP, LCD, LED and EL), LED
displays are widely used as information boards
and as transportation terminal displays due to
their excellent reliability, service life and visibility.
Particularly as a result of the remarkable
progress made with high-brightness blue and
green LEDs, full-colour displays can now be
established for outdoors. Any colour can be produced using the three primary colours red, green
and blue. In this report, measurement systems
are developed to characterize a full-colour LEDdisplay. Furthermore, the LUMINO XTralux ML
4 C-Pixel and an optimized Pixel has been characterized.
78
3 RESEARCH
Measurement systems
The spatial distribution of a LED is measured
with the setup shown in Fig. 1. The LED is fixed
on a LED-holder and is driven by a constant
current. By rotating the
swivel-arm in 1°- steps,
the data of the spatial
distribution is received.
The Fig. 2 shows
the setup to determine
the spectral distribution of a LED or LEDpixel. After the spectum is measured, the
x, y, z colour coordinates are determined
[1,2].
The CIE (Commission Internationale de
l’Eclairage) diagram is
the standard colourimetric system. The x, y, z axis of this diagram
are based on three colour-matching functions,
each of which is related to the spectrum of red,
green and blue. A sequence of single-wavelength
computer
IEEE
light stop with aperture
chopper
Lock-in-amplifier
In Ref.
detector =
lens + photodiode
monochromator
with
stepper motor
lens
Fig. 2: Measurement setup - spectrum
hole
single-LED or
LED-Pixel
3.5 Technologies for Optoelectronic Components and Systems
colours can be expressed as a curve in the x, y,
z space of the CIE diagram and the projection of
the curve on the x, y plane is a horseshoe-shaped
pattern (Fig. 3). Any colour can be expressed as
a point inside of this horseshoe-shaped curve.
0,8
single-wavelenght colours
0,6
yellowish green
red
0,4
D65
Experimental results
The colour coordinates of all measured LEDs
are shown in Fig. 3 which are determined from
the spectrum of the LEDs. In Fig. 4 (a) every
colour in the triangle region, of which the vertices indicate the three primary colours of the
LUMINO -Xtralux LEDs, can be radiated by adjusting the luminous intensity of each LED. As
blue
0,2
0,0
0,0
0,2
0,4
0,6
0,8
x
Fig. 3: CIE diagram of all LEDs
(b)
(a)
0,8
0,8
single-wavelenght colours
0,2
0,0
0,0
0,6
green
0,4
D65
0,2
0,4
0,6
0,4
D65
0,2
red
blue
single-wavelenght colours
green
y
y
0,6
blue
0,0
0,0
0,8
red
0,2
0,4
0,6
0,8
x
x
Fig. 4: (a) CIE diagram of LUMINO Xtralux-pixel and (b) of the optimized pixel.
(b)
(a)
0,8
0,8
single-wavelenght colours
single-wavelenght colours
0,6
0,6
D65
y
0,4
0,4
20°
40°
0°
80°
40°
20°
0°
80°
90°
0,2
0,2
0,0
0,0
D65
60°
60°
y
y
green
79
0,2
0,4
0,6
0,8
0,0
0,0
0,2
x
Fig. 5: (a) Colour-shift of LUMINO-Xtralux-pixel and (b) of the optimized pixel.
0,4
x
0,6
0,8
80
3 RESEARCH
(a)
(b)
0,8
0,8
relative intensity
1,0
relative Intensity
1,0
green
0,6
blue
0,4
red
0,2
0,0
green
0,6
red
blue
0,4
0,2
0
20
40
60
viewing angle /°
80
0,0
0
20
40
60
viewing angle /°
80
Fig. 6: (a) Spatial distribution of the LUMINO-Xtralux-pixel and (b) of the optimized pixel.
seen in the diagram, it is not possible to obtain
the standard white D65, which is outside the triangle. The colour coordinates of an optimised
Pixel are shown in Fig. 4 (b). The LEDs are wellchosen to obtain a greater triangle region. In
Fig. 5 the colour coordinate variation versus the
viewing angle of these two Pixels is shown. The
XTralux ML 4 C-Pixel (Fig. 5 (a)) has a large colour shift to the red primary colour, which can
explained with a wider spatial distribution of the
red LED (cf. Fig. 6 (a)).
The small shift of the optimized Pixel was
performed by using a red, green and blue LED
having nearly the same spatial distribution. This
was obtained by modification of the lense-form
(encapsulation) of the LED. Furthermore, the
LED-surface is roughend.
Conclusions
Within the scope of this thesis, two measurement setups have been developed. Furthermore, LEDs and the LUMINO XTralux ML 4 CPixel have been characterized. With the
knowledge of colour metrics and the colour coordinates of the LEDs, an optimized Pixel has
been assembled. A greater colour range and a
nearly constant colour coordinate versus viewing angle of the pixel has been achieved.
References
[1] Heinweg Lang, Farbmetrik und Fernsehen,
R.Oldenburg München Wien, ISBN 3-48620661-3, 1977
[2] DIN 5033, Farbmessung
3.5.2 Opinion poll on the evaluation
of the legibility of LED-based displays
R. HEDTKE AND R. BU ß
n this report the legibility of LED-based displays is evaluated on the base of
interviews with passengers of the public
local traffic. To reach this aim a model explaining the causal connection between
several external influences and the legibility
is made up according to DIN 1450. Based on
this model a questionnaire for the interviews
I
3.5 Technologies for Optoelectronic Components and Systems
81
the legibility of those display-systems (cf. Fig. 1) are acquired.
The causal connection
The causal connection based
on DIN 1450 is modified referring
to the requirements of the questionnaire. The result is shown in
Fig. 2 where the arrows describe
the influences. It can be seen that
Fig. 1: LED-based display used in the public local traffic.
there are eight major points having an influence on the legibility:
is developed. The gained data of the interThe type face and size, brightness and colour,
views is evaluated using statistic methods.
the distance between each letter, the distance
of view, the light conditions, and personal influIntroduction
ences.
Today, the permanent availability of information has become of increasing importance,
The used questionnaire
where the transmission of visual information cerBased on the causal connection, the questainly becomes to play a more and more importionnaire shown in Fig. 3 has been developed
tant role. Due to the permanent development in
to proof and determine the level of influence of
the area of LED-technology it has become poseach point. The first three questions are so called
sible to produce so-called super-luminescence
icebreaker-questions to start the conversation
light emitting diodes (SLED), having a very high
with the person to be interviewed. The following
brightness. In the public local traffic LED-based
questions are related to the legibility of the disdisplays find an increasing application. In coplayed text. Legibility, type size, distance beoperation with the company LUMINO/Krefeld, a
tween each letter, type face, brightness, and
producer of such displays, methods to improve
colour are assessed using marks between one
and five. This type of assessment has been chosen because everyone
knows this marks from
school. This causes
TYPE FACE
BRIGHTNESS
COLOUR
that comprehension
problems are avoided.
DISTANCE BETWEEN
TYPE SIZE
LEGIBILITY
LETTERS
Furthermore, tendencies like small, right,
DISTANCE OF VIEW
LIGHT
PERSONAL
INFLUENCES
and large where recorded if possible. Finally, the other points
Fig. 2: Model of the causal connection.
82
3 RESEARCH
Fig. 3: The used questionaire
3.5 Technologies for Optoelectronic Components and Systems
of influence are recorded after the main part of
the interview.
Interview
For the interviews, locations in the following
cities are selected according to the practicability of the interviews and considering the local
conditions: Duisburg, Essen, Düsseldorf; Oberhausen, Leipzig, Stuttgart.
At each of these locations interviews with 50
passengers have been carried out. To reach a
high comparability of the gathered data, the interviews only took place on platforms for the public local traffic.
Conclusion
The Fachgebiet Optoelektronik at the Gerhard-Mercator-Universität Duisburg has investigated the judgement of the legibility of
LED-based displays in co-operation with the
company LUMINO/Krefeld, a producer of such
displays. The aim of this study was the recording of subjective opinions of the users of the
public local traffic.
The legibility of the analysed LED-based displays has been valued by up to 90% of the interviewed passengers with well or very well.
Especially the displays based on yellow LEDs
in the city of Leipzig have been rated very positively. The majority of the passengers have felt
the brightness to be right. As well in Leipzig, the
valuation has been the same, even if the display was exposed to direct sunlight. As a rule,
type size, letter distance, and type were also
valued as right. Up to 80% of the passengers
have felt as being informed very well by those
displays.
In summary, the users of the dynamic passenger-information-system are contented with
the quality and information provided by these
83
systems. It should be pointed out that the LEDtechnology serves the high requirements referring to the demands in the public local traffic.
Even under unfavourable conditions like direct
sunshine the judgement is good or very good.
References
[1] DIN 1450, Beuth, Berlin, Juli 1993
[2] V. Dreier, Datenanalyse für Sozialwissenschaftler, Oldenbourg, München-Wien, 1994
[3] K. Holm (Hrsg.), Die Befragung 1, Franke, München, 1975
[4] E. Noelle, Umfragen in der Massengesellschaft,
Rowohlt, Reinbek bei Hamburg, 1963
3.5.3 Evaluation of possible improvements to enhance the UV-power efficiency of a xenon flashlamp system
B. NEUHAUS AND A. STÖHR
T
his paper will present experimental
results of the characterisation of a
xenon flashlamp system with a flat aluminium rear reflector fabricated by Bläsing Elektronik GmbH. The spatial distribution of the
UV-radiation is determined. Studies about
typical gas components and some materials
of the discharge tube result in possibilities
to optimize the arc lamp. Furthermore the
shape of a single flash is recorded and the
efficiency is measured in dependence of the
pulse repetition rate. To optimize this system several rear reflectors with different geometrics are tested and the reflection index
of five different materials is measured in the
UV and NIR wavelength range. For all these
examinations different measurment setups
are worked out.
84
3 RESEARCH
Introduction
The UV-drying process is of great significance
to the industry working with printing procedures
e.g. the silk-screen printing. New UV-paints are
free from solvents and they harden only by irradiation with UV-light [1]. Therefore powerful
lamps with great emission in the UV-range are
very important. Pulsed UV-lamps offer several
advantages compared with conventional UVburners. With flashlamps the intense heat emission can be reduced; they radiate only for a short
pulse duration but the hardening process is more
effective because the radiation output of
flashlamps is greater than of conventional lamps
[2]. The “UV-flash drying process” will become
an economical and nonpolluting alternative process in the future.
The intention of the following investigations
is to characterize such a flashlamp system and
to optimize it based on the experimental results.
Arc lamp construction and priniciple of a
flashlamp system
Fig.1 shows the construction of a typical arc
lamp. The electrodes are set in a clear quartz
tube. The type of quartz depends on the desired
output spectrum. The electrodes are made of
tungsten to enhance
electron emission. Arc
+
lamps are filled with inert
gas under several atmoanode
spheric pressure or a
mixture of gas and a deffilling gas
inite amount of mercury.
cathode
tube
The internal pressure in
the tube increases during
operation to 15-75 bar,
depending on the lamp
Fig. 1: Contype.
The flashlamp system operates by sending
an electric charge from a pulse generator to the
gas filled lamp. The gas absorbs the energy by
storing it in its atoms and subsequently it releases the energy by emitting photons, which
results in a high intensity flash of light [3]. Light
emssions in all directions can be guided by installing a reflector behind the lamp which collects the light and reflects it onto a surface which
should be treated. Caution: these lamps produce
high intensity UV radiation and ozon. Precautions are necessary during operation mode.
Experimental setups
Fig. 2 and Fig. 3 show the two basic experimental setups used for the measurements. In
order to determine the spatial distribution of
emitted radiation and to determine the UV-efficiency of the flashlamp in the UV-range the setup in Fig.2 is used. The pulse generator supplies the flashlamp with electrical impulses. The
pulse repetion rate (1,56Hz, 3,12Hz, 6,25Hz,
12,5 Hz, 25 Hz) can be adjusted by a rotary
switch. The light pulses, radiated by the
struction of arc
Fig. 2: Experimental setup to measure the
lamps [2].
relative optical output of an arc flashlamp.
3.5 Technologies for Optoelectronic Components and Systems
Fig. 3: Experimental setup to measure the
reflection index.
85
UVL-1500/TP1 fabricated by Bläsing Elektronik GmbH. The pulse shape has a fast rise time
t r(10% −90%) of 31,5ms and slower decay. The
pulsewidth t FWHM is 520ms. These are typical
values for flashlamps [2]. Further experiments
have shown, that the flash energy as well as
the pulswidth stay constant for all frequencies
in the range from 1,56Hz until 25Hz.
b) In order to determine the spatial distribution
of the radiation from the flashlamp type UVL1500/TP1 the SiC-photodiode is led along a
semicircle around the lamp as shown in Fig.
5. One can record the radiant intensity (relative) for a number of angles. The radius is r =
100cm. The values are recorded in steps of
optical power (rel.)
flashlamp, are detected by a special
silicon carbide (SiC; spectral range: 210380nm) photodiode.
Fig. 3 shows the setup to determine
0
t r(10%− 90%) = 31,5µ s
the UV-reflectivity of different reflector
materials. The light beam of a mercury
t FWHM = 520 µs
lamp is deflected by a beam splitter in
definite directions. The reflector reflects
the incoming light beam which is then
500 µs / div
focused on a SiC photodiode. By us− 2.5ms
2.5ms
0.0s
ing the second detector it is possible
time t
to compare the input power with the reflected power in order to determine the
Fig. 4: Relative optical output power of the flashlamp
reflection index.
UVL-1500/TP1.
For all investigations in the NIR
wavelength range a monochromator
one degree and they are related to the 90°and a silicon photodiode (BPW20; 375-1100nm)
direction. Fig. 6 shows the results (normal
are used to receive spectral values; the light
curve) given in a polar coordinate system. It
source is a halogene lamp. To record a single
can be seen that the distribition of the radiapulse an oscilloscope is used.
tion is symmetrical to the axes. Besides, it
can be noticed that over an angle of a= 168°
Experimental results
fifty percent of the radiant power compared
a) Fig. 4 shows the relative optical output powwith the power in direction of 90° is still radiater with respect to time of the flashlamp type
86
3 RESEARCH
η ⋅ ( 10 −9 )
flashl amp
UVL 1500-TP/1
efficiency
r = 100cm
210 − 380
ηUV
− 280
ηU270
V
pulse repetition rate fPu ls(Hz)
Si C-detector
JEC-1
210-380nm
Fig. 5: Principle to record the spatial distribution of the radiant.
ed by the flashlamp. So the angle of radiation
is very wide. Mostly such a wide angle is undesirable because one “looses” most of the
radiant in the borderlands when no object is
placed very closed to the radiant source. For
plenty of applications a distribution like a club
(dashed curve) is desired. It is possible to
achieve such a distribution with special rear
reflectors.
Fig. 7: UV-efficiency in addiction to the pulse
repetition rate.
c) Fig. 7 represents the UV-efficiency in addiction to the pulse repetition rate from the
flashlamp type UVL-1500/TP1. The values
are calculated for a wavelength range from
210-380nm as well as for the range from 270280nm. The detector was about two meters
away from the light source and the measurements were repeated several times for all repetition rates which are adjustable. From these
values a mean value is formed for each frequency. This mean value is used for the calculation of the efficiency for flashlamps. The
illustration shows that the efficiency stays
nearly constant for each frequency. The con-
45
40
35
IR-reflectivity
30
aluminium
UV-mirror
reflector III
reflector II
reflector I
25
20
15
10
5
0
-5
780
800
820
840
860
880
900
920
940
960
980
1000
wavelength (nm)
Fig. 6: Principle to record the spatial distri-
Fig. 8: .Spectral IR-reflection index of differ-
bution of the radiant.
ent materials.
3.5 Technologies for Optoelectronic Components and Systems
87
groups and experimental results the reflector
clusion is, that the pulse repetition rate has no
coatings I and III are most suitable to optimize
essential influence on the efficiency at least at
the xenon flashlamp system.
small frequencies below f PULS = 25Hz.
d) The printing industry is interested in rear reProposals for improvements
flectors with a high reflection index in the UVThe proposals to optimize the flashlamp sysrange for great UV-efficiency and a low retem can be divided in four groups:
flection index in the IR-range because of the
1. gas filling
undesirable heat, which is produced by infra2. tube material
red radiation. Therefore five different materi3. reflector material
als are tested for the reflectivity in the UV
4. reflector form
(210-380nm)- and IR-(780-1000) wavelength
range. These five materials and their UV-reGas filling
flection index are:
Four typical fillings for arc lamps are xenon
- highly polished aluminium, used so far by
(Xe), mercury (Hg), the mixture mercury-xenon
Bläsing Elektronik GmbH; the UV-reflec(Hg(Xe)) and deuterium (D 2 ). To compare the
tion index is ρUV = 90,24%.
- a special UV-glass-mirror that transmits the
effectiveness of these fillings in the UV-area the
IR-radiation and reflects the UV-radiation;
efficiency as a function of the input power of difthe UV-index results in ρUV = 97,17%.
ferent arc lamps with these gases is calculated
- three metallic reflectors I, II and III with difover a range from 210-380nm. In Fig. 9 this comferent coatings; The compositions of these
parison is shown. To achieve these results some
three coatings and their designation are
spectral irradiance curves from the L.O.T.-Oriel
unkown.
company [2] catalog are evaluated. The xenon
During the measurements it was apparent,
gas lamps are plainly worse than the other
that the three reflectors I, II and III are in all problamps. Increasing the lamp power does little inability diffused and diffused/directed reflecting
fluence to the UV-efficiency of the xenon burnmaterials with the consequence
that the absolute UV-reflection
index could not be determined.
D2-deuterium
210 −380
But a second measurement methηUV
Xe-xenon
od could prove, that all five mateHg-mer cury
rials could increase the radiant
Hg(Xe )-mercury/xenon
power better than the aluminium η *10
reflector.
Fig. 8 shows the spectral IRreflectivity of these five materials
over a wavelength range from
780-1000nm. The aluminium relamp power (W)
flector is worse than the other four
210 − 380
materials. In comparison with all
Fig. 9: UV-efficiency ηUV
of different discharge lamps.
−9
88
3 RESEARCH
r eflectance
moplastic resin and it is thermally
stable to > 350°C. The reflectance
is > 95% over this range. Surface
contamination only decreases the
reflectance at the lower ends of
the spectral range. B) metallic reflectors are very sensitive to surface contamination and to overheating. These facts can decrease
the reflectance greatly as well as
the permanent irradiation with UVlight.
wavelength (nm)
Fig. 10: Spectral reflection grade of spectralon [4]
er. The mixture Hg(Xe) provides the best optical
radiation power but to get the optimal power a
warm up time of about 15 minutes is necessary.
Tube material
Quartz is undoubtedly the best material for
the discharge tube and it is normally used by
the industry. Quartz guaranties the mechanical
and thermal durability. The type of quartz depends on the desired UV-output. There exist
several special quartz types: a) UV grade quartz
that transmits the output to below 200nm, and
b) “ozone free” quartz which absorbs short wavelengths to prevent ozone generation. Above
280nm the special types do not offer advantages compared with standard quartz variants.
Reflector material
Literary investigations give some new information about reflector materials to optimize the
radiation power. a) Labsphere Ltd. company
developed a diffuse reflecting material, spectralon, with reflectivity over the range from 2502500nm shown in Fig. 10. Spectralon is a ther-
Reflector form
Fig.11 shows how the spatial
distribution of radiation could be
changed when two aluminium reflectors with different forms are used. These forms are described
in Fig.12 and Fig.13. The influence on the distribution is tested when the distance between the
tube and the reflector varies. The values are
related to the 90°-direction and to the values of a
100
80
120
60
140
40
160
180
20
0
1,0 1,25 1,5
1,5 1,25 1,0
angle (°)
without reflector
R1 1,5 cm
R1 4,0 cm
R1 6,5 cm
R2 1,5 cm
R2 4,0 cm
R2 6,5 cm
Fig. 11: Spatial distribution of the radiation
with the reflectors I/II in dependence on the
distance between tube and reflector.
3.5 Technologies for Optoelectronic Components and Systems
measurement without a reflector. So you are in
position to determine the increase of the radiant
intensity as well. The reflector R2 could increase
the radiant power best, with nearly 40% compared with the
values without a
reflector. The distance between
the tube and the
Fig. 12: Cross-section
reflector is imporof the reflector R1.
tant as well. The
reflector R1 produces the best
results at a distance of d=4,0cm
whereas
the
second reflector
was best at a distance
of
d=1,5cm.
Fig. 13: Cross-section
of the reflector R2.
Conclusion
The analysis of
the flashlamp system of Bläsing Elektronik GmbH
provided the following results: The flashlamp radiates only for the duration of a pulse. The pulse
repetition rate up to 25Hz has no signifcant
influence on the radiation intensity. The spatial
distribution of the radiation is extremly wide. Over
an angle of 168° fifty percent of the radiant power
compared to the power in direction of 90° is still
radiated by the flashlamp.
The investigations to optimize this system
provided the following possibilties: The mixture
mercury-xenon has the best UV-efficiency and
it is suitable to use for fillings in arc lamps. The
change of the reflector form and material could
increase the radiant power as well. First experimental measurments point to the assumption that
e.g. the combination of the reflector material III
89
with the form R2 which could both increase the
intensity best, would provide a better efficiency.
References
[1] Erhardt D. Stiebner, “Bruckmann´s Handbuch
der Drucktechnik”, Bruckmann, München 1992
[2] L.O.T. ORIEL catalog Vol.II, “Light Sources,
Monochraomators & Spectrographs, Detectors
& Detection Systems, Fiber Optics”, Oriel Corporation, USA, 1994
[3] Polygon flashlamps http://www.polygon1.com/
technology.html
[4] Labsphere catalog; “Diffuse reflectance Coatings
And Materials”, Labsphere, North Sutton, 1996
3.5.4 Construction of a flip chip device for bonding integrated circuits
J. ERVENS AND R. BUß
o bond integrated circuits in flip chip technology a heating device is required, enabling
precise adjustment and soldering of the solder bumps. This device was designed,
built and put into operation. Furthermore, an alternative process to electroplating [1] was tested to put solder bumps onto
microelectrodes. Therefore, testing chips
were constructed on which solder bumps
were evaporated. In the completed device test
soldering points were carried out and analyzed.
T
Introduction
In the age of space-saving integration of semiconductor circuits the use of flip chip technology is getting more and more interesting. A special advantage is the possibility to connect silicon
technique to III-V-compound semiconductors,
90
3 RESEARCH
Fig 1: Self-adjustment of solder bumps [2].
which are of high importance to optoelectronic
applications.
Optoelectronics have many components with
vertical radiation. Therefore a direct connection
of the silicon substrate to the light emitting chip
is very useful for the third dimension. Another
point of interest is the self-adjustment of the
chips by the surface tension of the melted solder bumps. As shown in Fig. 1 an alignment error of several micrometer in the adjustment can
be balanced out.
Description of the mode of operation of the
heating device
The complete equipment consists of
- heating unit
- adjustment unit
- temperature control
- temperature measuring instrument
The heating unit is shown at the top of Fig. 2.
It comprises the halogen radiators and the mirror reflectors and serves for fastening the sample holding device. In the device, which consists
of 10 millimeter thick aluminium plates, the chips
get soldered, lying on top of each other. A membrane pump sucks the top chip to a heat-resistant pane of glass as shown in Fig. 3. The bottom chip is put down on an appliance fastened
to the adjustment unit which consists of a rotary
table, that can be moved by hand, and an x-y-zmanipulator, see Fig. 3. The adjustment of the
chips can be observed with an ocular from the
top.
After the adjustment is completed in x- and in y-direction the
bottom chip is brought into contact with the top chip by the z- manipulator. Then the nitrogen valve
is opened and a low pressure is
adjusted by a manometer, to
heat the chips in a nitrogen atmosphere. This
disables oxidation of the surfaces of the solder
bumps. The halogen radiators are started and
controlled by the temperature control unit, which
consists of a phase-angle control, that influences the electric power. By means of a rotary potentiometer the temperature of the radiators is
also influenced. A fast-response thermocouple
juts as a measuring sensor into the heating unit.
This digital measuring instrument shows the predominant temperature inside. After about ten minutes the chips are soldered and after cooling
down thay can be taken out off the heating unit.
Fig. 2: Front view of flip chip device.
3.5 Technologies for Optoelectronic Components and Systems
91
References
diminished pressure
panes of glass
[1] G. Sadowski, D. Zeidler: “Mikrogalvanik für die
Herstellung lötfähiger Bumpsysteme”, me Bd 6,
1992, pp 358-361
[2] M. Wale, M. Goodwin: “Flip-Chip Bonding Opti-
chip 1
chip 2
sealing ring
rotatable and in
x-y-z-direction
shiftable appliance
z
x
y
Fig. 3: Principle of chip arrangement.
Construction of test chips
At first the substrate is covered with a steel
resist pattern mask and fastened in the deposition apparatus. Then a gold layer is evaporated, because the adhesive bond strength of gold
on the substrate is significantly higher than that
of soft solder. The soft solder layer is evaporated in several steps of the operation.
Results of the test series
The temperature in the heating unit is sufficient in order to melt the solder bumps and to
connect the chips. In case of stronger mechanical demand the gold layer dissolves off the substrate, which means the soldering of the solder
bumps was successful. Self-adjustment can not
be recognized, because the layer thickness
achieved in the evaporation process is far too
thin. If a higher layer thickness can be achieved,
evaporation will be applicable, but regarding today’s knowledge electroplating is preferred.
mizes Opto-ICs”, Circuits and Devices, pp 2531, Nov. 1992
92
4 TEACHING ACTIVITIES
4.1 Lectures, excercises, and practical studies
93
4 Teaching activities
4.1 Lectures, Excercises, and
practical studies
Technical Electronics 3: Optoelectronics
D. Jäger and A. Stöhr
The course „Technical Electronics 3: Optoelectronics“ covers the basic theory and technology of modern semiconductor photonic devices as well as applications of these devices in
optoelectronic integrated circuits (OEICs). The
course starts with the fundamental physical phenomenon of light-material interaction in semiconductors, such as fundamental absorption,
spontaneous and stimulated emission. Subsequent lectures deal with the theory and technology of photoconductive devices, photodiodes,
modulators, light emitting diodes (LEDs), and
laser diodes. Special attention is given to modern quantum well waveguide laserdiodes and
their applications in optical communication systems, medicine, and material processing.
Ultra High Frequency Transmission
Techniques: Optical Signal Transmission
D. JÄGER AND R. BUß
The course “Ultra High Frequency Transmission Techniques: Optical Signal Transmission”
starts with the propagation of electromagnetic
waves considering the features of optical waves
at surface boundaries, such as reflection and
refraction. Proceeding with the description of
such fundamental physical effects like scattering, absorption and dispersion, optical wave
propagation in various types of dielectric
waveguides is discussed. Based on this fundamentals the design, properties and technological realization of waveguides based on III/V compound semiconductors are discussed. Another
main part of this course deals with fiber optic
waveguides: Wave propagation in graded index
fibers as well as in stepped index fibers is derived where both advantages and disadvantages of each type are elucidated. Problems such
as signal distortion in fibre optic waveguides are
analyzed and solutions to avoid them are given. Following the topic of wave propagation, the
most important devices for optical and optoelectronic integrated circuits (OEIC) are presented.
The properties and technological realization of
waveguide laser diodes, vertical cavity surface
emitting laser diodes (VCSEL), modulators, and
detectors are discussed. Finally, economical
aspects of optical communication techniques
and future prospects like „fiber to the home“ are
touched
Special Areas of Optoelectronics:
Lasers
D. JÄGER AND A. STÖHR
The first lectures within the course „Lasers“
cover the basic principles and the mathematical description of electromagnetic waves. The
course proceeds with the quantum mechanical
interactions between electromagnetic waves
and atomic materials resulting in the two most
important requirements for light amplification by
stimulated emission of radiation (laser). Special
attention is then given explaining the basic concepts, the functionality, and the characteristic
94
specifications of different laser sources of importance, such as the Helium-Neon laser, the
Ar-ion laser, Excimer lasers, the Ti:Sapphire laser, semiconductor laser diodes etc.. Finally,
examples of laser applications in various industrial areas (medicine, communication, material
processing etc.) are discussed together with future trends.
Optical Signal Processing
D. JÄGER AND R. BUß
The course “Optical Signal Processing” starts
with the basic theory of non-linear optical effects
both in dielectric materials and in semiconductors. The causes for optical bistability are described and principles like optical switching are
applied to the realization of optical memories
and logic elements. Within the next section of
this course, the phenomenon of opto-electronic
bistability is introduced. It is shown that the integration of a light modulator and a photodetector
is leading to so-called self-electro-optic effect
devices (SEED), showing various forms of
switching behaviour which can be controlled
both optically and electrically. Finally, the main
advantages of optical signal processing are
pointed out while discussing applications such
as optical switching networks, image processing systems, optical neural networks, optical
phased array antennas, optical computing, and
optical interconnects.
Multimedia-Techniques
D. JÄGER AND R. BUß
This course elucidates “Multimedia” from
three different points of view: The optoelectron-
4 TEACHING ACTIVITIES
ic area, the informatic area and the area of data
processing. Starting with optoelectronic devices and interfaces for fiber-optic networks (LAN,
WAN, FDDI), multiplexing (TDM, WDM) and
routing techniques in the optical domain are introduced. Problems of high capacity data storage using optical techniques and mobile connections to the internet are discussed. The
second part deals with modern techniques for
data compression, coding and security problems
together with the discussion of pattern recognition using neural networks. Large electronic
databases, techniques for data retrieval, video
indexing methods and electronic data interchange are presented. The last part of this
course elucidates today’s computer hard- and
software such as Pentium MMX technology,
multimedia PCs , WWW, Internet phone, electronic mail and more. Next, various types of network protocols (ATM, FDDI, Ethernet, TCP/
IP, ...) suitable for multimedia applications are
discussed. Finally applications such as
teleteaching, teleworking, edutainment (Education and Entertainment), video on demand, world
wide web and video conferencing are treated.
Information Technology 1 + 2
D. JÄGER AND
CO-WORKERS
Practical studies for students with emphasis on
“Information Technology” (E3 I/IT)
Exp. 1: Optical Transmission
Exp. 2: Optical Signal Processing
Exp. 3: Optoelectronic Sensors
Exp. 4: Optical Neural Signal Processing
4.2 Seminars and Colloquia
95
4.2 Seminars and Colloquia
Seminar on Optoelectronics
D. JÄGER AND
CO-WORKERS
M. Groß, “Dimensionierung und Entwicklung
eines thermooptischen Schalters im polynmeren Materialsystem”, Apr. 1996
M. Alles, “Tagungsberichte: IPRM ‘96,
22.04.-25.04., Schwäbisch Gmünd und IPR ‘96,
29.04.-02.05., Boston”, May 1996
H. Slomka, “Reinstwassererzeugung für die
Optoelektronik”, May 1996
R. Hülsewede, “Nichtlineare Leitungsstrukturen zur Frequenzerzeugung und Frequenzvervielfachung”, May 1996
R. Buß, “ZEMAX: Ein Softwarepaket zur Simulation optischer Systeme”, May. 1996
M. Engel, “2D-Simulation von INT-HEMT für
OEIC“, Jun. 1996
M. Wenning, “Entwicklung einer Meßtechnik
zur Bestimmung der Physikalischen und fotometrischen Eigenschaften von LED-basierten FullColor-Displays”, Jun. 1996
S. Redlich, “Nichtlineare Vielschichtheterostrukturen für die Microwellenphotonik”, Jun.
1996
R. Hedtke, “Demoskopische Untersuchung
und Beurteilung der Leserlichkeit von LED-basierten Anzeigesystemen”, Jun. 1996
B. Neuhaus, “Aufbau einer Meßtechnik zur
Charakterisierung und Optimierung der UV-Ausbeute von Blitzlampen für den Einsatz in der
Druckindustrie”, Jul. 1996
D. Kalinowski, “Entwicklung eines Feldsondenmeßplatzes zur 2-dimensionalen Analyse
elektromagnetischer Feldverteilungen auf nichtlinearen Leitungen”, Jul. 1996
V. Wendrix, “Herstellung und Charakterisierung eines Wanderwellen-Photodetektors”, Jul. 1996
A. Kreuder, “Ankopplung von Sende- und
Emfpangsmodulen an eine faseroptische Übertragungsstrecke”, Oct. 1996
S. Redlich, “Ladungsträgertransport über
Heterobarrieren - Simulationsmethoden”, Oct.
1996
M. Groß, “Stand des Projekts EPI-RET”, Oct.
1996
A. Lüddecke, “Simulation der Millimeterwellengeneration eines Wanderwellenphotodetektors”, Nov. 1996
J. Evens, “Aufbau einer Flip-Chip Apparatur
zur Verbindung integrierter Schaltungen”, Nov.
1996
P. Karioja, “Overview on activities at VTT”,
Nov. 1996
96
V. Mezentsev, “Nonlinear Problems Related
To The Modern Optical Communication”, Nov.
1996
M. Meininger, “Entwicklung photovoltaischer
Zellen zur Energieversorgung einer künstlichen
Sehprothese”, Dec. 1996
I. Ryjenkova, “Millimeterwave propagation in
nonlinear transmission lines“, Dec. 1996
O. Berger, ”Bestimmung des HF-Ersatzschaltbildes von Photodetektoren mit Hilfe
der Netzwerkanalyse”, Jan. 1997
4 TEACHING ACTIVITIES
T. Braasch, “Bericht über die Messe Laser
1997”, Jul. 1997
J. Ervens,” Experimentelle Untersuchungen
zum Ladungsträgertransport über eine GaAs/
AlGaAs-Barriere”, Oct. 1997
R. Heinzelmann, “Bericht über die OFS’97
in Williamsburg”, Nov. 1997
R. Hedtke,”Entwicklung einer optischen Energie- und Signalübertragungsstrecke”, Nov.
1997
T. Baumeister, ”Systemanalyse des
optoelektronischen Energie- und Signalübertragungssystems im Projekt EPI-RET”, Jan. 1997
C. Kampermann, ”Implementierung eines
analytischen Modells zur Simulation der optischen Eigenschaften nichtlinearer Halbleiter-Heterostrukturen”, Nov. 1997
A. Kreuder, “Untersuchung der dynamischen
Eigenschaften von nichtlinearen Vielschichtheterostrukturen”, Jan. 1997
B. Ponellis, “Simulation des optischen Konversionswirkungsgrades von WanderwellenPhotodetektoren”, Dec. 1997
M. Groß, “Stand des Projekts EPI-RET“, Feb.
1997
O. Lotz, “Entwicklung einer Seelaterne in
LED-Technik in den Farben Rot und Grün”, Apr.
1997
R.S. Johnson, “Silicon Motherboards for Fibre-Chip Coupling”, Apr. 1997
M. Schmidt, “Elektronische Eigenschaften
von Bor”, May 1997
I. Ryjenkova, “Nichtlineare Leitungen für das
Millimeterwellengebiet”, Jun. 1997
4.2 Seminars and Colloquia
97
Colloquium on Optoelectronics
D. JÄGER AND LECTURERS WITH EMPHASIS ON OPTOELECTRONICS
Prof. Dr. H.G. Schuster, Universität Kiel,
“Komplexe Adative Systeme”, Jan. 1996
Prof. Dr. M. Dragoman, “Time-frequency
characterization of optical pulses”, Oct. 1996
Dipl.-Phys. G. David, Universität Duisburg,
“Elektrooptische Feldverteilungsmessungen zur
Höchstfequenz-Charakterisierung von monolithisch integrierten Mikrowellenschaltungen”,
Feb. 1996
Dipl.-Ing. R. Buß, Fachgebiet Optoelektronik, Duisburg, “Optoelektronik in der Neurotechnologie”, Oct. 1996
Dr. A.L. Ivanov, Universität Frankfurt,
“Switching Kinetics of a Low-Intensity ElectroOptical Element due to Intrinsic Photoconductivity”, May 1996
Dr. J.-Uwe Meyer, Fraunhofer-Institut St. Ingberg, “Mikrotechnologien zur Kontaktierung von
biologischen Zellen und Geweben”, May 1996
Dipl.-Ing. R. Heinzelmann, Universität Duisburg, “Elektrooptische Wellenleitermodulatoren für optische Übertragungssysteme”, May
1996
Dipl.-Ing. S. van Waasen, Universität Duisburg, “20 Gb/s Wellenleiter-pin/Wanderwellenverstärker OEIC: Jüngste Ergebnisse”, Jun.
1996
Ass. Prof. Dr. A. Driessen, Univ. of Enschede, Netherlands, “Advanced Micro-Systems for
Optical Networks (AMON)”, Jun. 1996
Dr. N. Vodjdani, THOMSON CSF, Orsay Cedex, France, “Integrated Optoelectronics for
Optical Microwave Links and Optical Communications”, Oct. 1996
Dr.-Ing. M. Martin, Hahn-Meitner-Institut,
Berlin, “Entwicklung von GHz Komponenten am
Hahn-Meitner-Institut”, Nov. 1996
Dipl.-Ing. M. Alles,Fachgebiet Optoelektronik, Duisburg, “60 GHz Wanderwellen-Photodetektoren für optische Millimeterwellenverbindungen”, Dec. 1996
Dipl.-Phys. T. Braasch, Fachgebiet Optoelektronik, Duisburg, “Elektrooptisches Testen zur
on-wafer-Charakterisierung von MMICs” , Jan.
1997
Dipl.-Ing. A. Brennemann, Fachgebiet Halbleitertechnik/-technologie, Duisburg, “Neuartige
Photoreceiver auf Basis einer Kombination von
pin-Diode und Permeable Junction Base Transistor (PJBT)”, Jun. 1997
Prof. Dr. W. Sohler, Universität Paderborn,
“Integrierte Optik in LiNbO3: neue Entwicklungen”, Jun. 1997
Prof. Dr.-Ing. R. Schwarte, Universität
Siegen, “Neuartiges optisches 3D-Meßsystem
für die schnelle Formerfassung“, Jul. 1997
98
4 TEACHING ACTIVITIES
4.2 Seminars and Colloquia
99
100
4 TEACHING ACTIVITIES
4.3 Doctoral, Diploma, and
Graduate theses
Doctoral theses
Diploma theses
Gerhard David, “Höchstfrequenz-Charakterisierung von monolithisch integrierten Mikrowellenbauelementen und -schaltungen durch
zweidimensionale elektrooptische Feldverteilungsmessungen”
Ludger Brings, “Implementierung eines
rechnergestützten Syntheseverfahrens zur Realisierung monolithisch integerierter periodischer
Hochfrequenzleitungen”
Steffen Knigge, “Nichtlineare Optische
Eigenschaften von Vielschichtheterostrukturen”
Andreas Stöhr, “Entwicklung und Realisierung elektrooptischer Wellenleiter-Schalter
für photonische Systeme im Wellenlängenbereich um 1 µm”
Ralf Kremer, “Optisch gesteuerte Koplanarleitungen als III-V-Halbleiter-Bauelemente für die
Mikrowellen-Signalverarbeitung”
Stefan Zumkley, “Vertikale elektrooptische
Modulatoren für optische Verbindungstechnik im
Gbit/s-Bereich”
Peter Bussek, “Quantitative Auswertung von
zweidimensionalen elektrooptischen Feldverteilungsmessungen zur Charakterisierung von
monolithisch integrierten Mikrowellenschaltungen”
Thomas Alder, “Herstellung und Charakterisierung von Wellenleitermodulatoren für den
Wellenlängenbereich um 1,3 µm”
Michael Wenning, “Entwicklung einer
Meßtechnik zur Bestimmung der physikalischen
und fotometrischen Eigenschaften von LEDbasierten Full-Color-Displays”
Thomas Engel, “2D-Simulation von InPHEMTs für Verstärker in Empfänger-OEICs”
Dirk Kalinowski, “Entwicklung eines Feldsondenmeßplatzes zur zweidimensionalen Analyse elektromagnetischer Feldverteilungen auf
nichtlinearen Leitungen”
Thomas Baumeister, “Systemanalyse des
optischen Energie- und Signalübertragungsmoduls für eine künstliche Sehprothese”
Andreas Kreuder, “Untersuchung der
dynamischen Eigenschaften nichtlinearer
Vielschichtheterostrukturen”
4.3 Doctoral, Diploma, and Gratuate theses
101
Graduate theses
Mark Meininger, “Entwicklung photovoltaischer Zellen zur Energieversorgung einer künstlichen Sehprothese (Retina Implant)”
Michael Heinsdorf, “Herstellung und Charakterisierung von Wanderwellen-Photodetektoren auf InP-Substrat”
Oliver Lotz, “Entwicklung einer Seelaterne
in LED-Technik in den Farben Rot und Grün”
Ralph Hedtke, “Demoskopische Untersuchungen zur Beurteilung der Leserlichkeit von
LED-basierten Anzeigesystemen”
Jutta Ervens, “Experimentelle Untersuchungen zum Ladungsträgertransport über eine
GaAs/Alx Ga1-xAs-Heterobarriere”
Uwe Weimann, “Entwicklung einer InfrarotDatenübertragungsstrecke auf rotierenden BildText-Systemen”
Claus Kampermann, “Implementierung
eines analytischen Modells zur Simulation der
optoelektronischen Eigenschaften nichtlinearer
Halbleiter-Heterostrukturen”
Birgit Neuhaus, “Aufbau einer Meßtechnik
zur Charaktierisierung und Optimierung der UVAusbeute von Blitzlampen für den Einsatz in der
Druckindustrie”
Jutta Ervens, “Aufbau einer Flip-Chip-Apparatur zur Verbindung integrierter Schaltungen”
André Lüdecke, “Simulation der Millimeterwellengeneration eines Wanderwellen-Photodetektors”
Oliver Berger, ”Bestimmung des HF-Ersatzschaltbildes von Photodetektoren mit Hilfe
der Netzwerkanalyse”
Bernd Ponellis, “Simulation des optoelektronischen Konversionswirkungsgrades von Wanderwellen-Photodetektoren”
102
5 PUBLICATIONS AND PRESENTATIONS
103
5 Publications and presentations
[1]
G. David, R. Tempel, I. Wolff, and D. Jäger,
Analysis of microwave propagation effects
using 2D electro-optic field mapping techniques, Optical and Quantum Electronics,
Special Issue on Optical Probing of
Ultrafast Devices and Integrated Circuits,
1996, pp. 919 - 931
[2]
G. David, P. Bussek, and D. Jäger, High
resolution electro-optic measurements of
2D field distributions inside MMIC devices,
Proceedings of CLEO 96, Anaheim, USA,
1996, pp. 450-451
[3]
G. David, R. Tempel, I. Wolff, and D. Jäger,
In-circuit electro-optic field mapping for
function test and characterization of
MMICs, 1996 IEEE MTT-S Int. Microwave
Symp., June 17-24, San Francisco, USA,
1996, pp. 1533-1536
[4]
[5]
R. Kremer, S. Redlich, L. Brings, and D.
Jäger, A novel type of constant impedance
travelling wave phase shifter for InP- based
MMICs, 1996 IEEE MTT-S Int. Microwave
Symp., June 17-24, San Francisco, USA,
1996
M. Alles, Th. Braasch, and D. Jäger, Highspeed coplanar Schottky travelling-wave
photodetectors, Int. Conf. on Integrated
Photonics Research, Conference Proceedings pp. 380-383, Boston, USA 1996
[6]
G. David and D. Jäger, Analysis of in-circuit electro-optic measurements of MMICs,
XXVth General Assembly of the URSI,
August 28 -September 5, 1996, Lille,
France
[7]
M. Alles, Th. Braasch, R. Heinzelmann, A.
Stöhr, and D. Jäger, Optoelectronic devices for microwave and millimeterwave
optical links, 11th Int. MIKON ’96, Conference Proceedings, Workshop ‘’Optoelectronics in Microwave Technology’’, pp. 68
- 76, Warsaw, 27-30 May 1996 (invited
paper)
[8]
M. Alles, R. Heinzelmann, R. Hülsewede,
R. Kremer, S. Redlich, A. Stöhr, and D.
Jäger, Wave propagation in planar structures for travelling wave semiconductor
devices, Progress In Elektromagnetic Research Symposium PIERS ’96, Conference Proceedings, p. 136, July 1996,
Innsbruck, Austria (invited paper)
[9]
P. Berini, A. Stöhr, K. Wu, D. Jäger, Normal Mode Analysis and Characterization
of an InGaAs/GaAs MQW Field-Induced
Optical Waveguide Including Electrode Effects, IEEE/OSA J. Lightwave Technol.,
Vol. 14, No. 10, pp. 2422 - 2435, October
1996
104
[10] D. Jäger, M. Alles, T. Braasch, R.
Heinzelmann, and A. Stöhr, Integration
Technology for Microwave Photonic Devices, Interaction technology of Microwaves and Light-Waves-Systems and
Devices, XXVth General Assembly of the
URSI, August 28 -September 5, 1996, Lille,
France (invited paper)
[11] D. Jäger, R. Hülsewede, I.V. Ryjenkova,
V.K. Metzentsev, S. L. Musher, Microwave
Propagation on Nonlinear Transmission
Lines”, XXVth General Assembly of the
URSI, August 28 -September 5, 1996, Lille,
France
[12] D. Jäger, V.K. Metzentsev, I.V. Ryjenkova,
S. K. Turitsyn and R. Hülsewede, Microwave Propagation on Nonlinear Transmission Lines, Proceedings of PIERS’96,
Hong Kong
[13] M. Alles, T. Braasch, and D. Jäger, Travelling Wave Photodetector for Optical Generation of Microwave Signals, Indium
Phosphide and Related Materials IPRM
‘96, Proc. Part II, pp. 30 - 31, Schwäbisch
Gmünd, 1996 (post deadline paper)
[14] R. Hülsewede, V.K. Mezentsev, S.L.
Musher, I.V. Ryjenkova, S.K. Turitsyn, and
D. Jäger, Travelling wave generation of
millimeter waves in bi-modal NLTLs, 26th
European Microwave Conference EuMC
’96, Prague
5 PUBLICATIONS AND PRESENTATIONS
[15] R. Heinzelmann, A. Stöhr, Th. Alder, R.
Buß, and D. Jäger, EMC measurements
using electrooptic waveguide modulators,
International Topical Meeting on Microwave Photonics MWP ’96, Conference
Proceedings, Technical Digest, December
3-5, 1996, Kyoto, Japan
[16] R. Hülsewede, V.K. Mezentsev, S.L.
Musher, V. Ryjenkova, S.K. Turitsyn, and
D. Jäger, Millimeter Wave Generation on
Nonlinear Transmission Lines, 1996 Int.
Workshop on Millimeter Waves Digest,
1996, Orvieto, Italy
[17] D. Jäger, Optically Controlled Microwave
Devices, International Topical Meeting on
Microwave Photonics MWP ’96 technical
digest, December 3-4, 1996, Kyoto, Japan
[18] D. Jäger, V.K. Mezentsev, S.L. Musher and
I.V. Ryjenkova, Millimeter wave power generation on nonlinear transmission lines,
Asia Pacific Microwave Conf. APMC ’96,
New Delhi, India
[19] I. Ryjenkova, V.K. Mezentsev, S.L.
Musher, S.K. Turitsyn, R. Hülsewede and
D. Jäger, Nonlinear Transmission Lines for
Millimeter Wave Applications, INMMC ’96,
Duisburg
[20] Th. Braasch, G. David, R. Hülsewede, U.
Auer, F.-J. Tegude and D. Jäger, Propagation of Microwaves in MMICs Studied by
Time- and Frequency-Domain Electro-Optic Field Mapping, Spring Topical Meeting
„Ultrafast Electronics and Optoelectronics“,
March 17-19, 1997, Lake Tahoe, USA
105
[21] Th. Braasch, G. David, R. Hülsewede, U.
Auer, F.-J. Tegude and D. Jäger, Frequency and time domain characterization
of nonlinear transmission lines using
electro-optic probing techniques, MIOP
’97, April 22-24, 1997, Sindelfingen, Germany
[27] I. Ryjenkova, V.K. Mezentsev, S.L.
Musher, S.K. Turitsyn, R. Hülsewede, and
D. Jäger, Millimeter Wave Generation on
Nonlinear Transmission Lines, Publication
in „annales des télécommunications“ (special issue), Vol. 52, No. 3-4, 1997, pp. 134139
[22] M. Alles, U. Auer, F.-J. Tegude and D.
Jäger, Millimeterwave Photodetectors, Microwaves and Optronics, MIOP ’97, April
22-24, 1997, Sindelfingen, Germany
[28] M. Alles, U. Auer, F.-J. Tegude, and D.
Jäger, High-speed Travelling-Wave Photodetectors for Wireless Optical Millimeter
Wave Transmission,MWP ’97, Sep. 3-5,
1997, Duisburg/Essen, Germany
[23] M. Alles, U. Auer, F.-J. Tegude and D.
Jäger, High-Speed Travelling-Wave Photodetectors for optical Millimeterwave
transmission operating at 1.55 µm, Workshop Mobile Millimeter Communications
MMMCom, Dresden, 12-13. Mai 1997
[24] S. Redlich, A. Kreuder, and D. Jäger, Dynamics of nonlinear electro-optical GaAs/
AlAs multilayer-heterostructures, International Conference on Low Dimensional
Structures (LSDS) ’97, May 19-21, 1997,
Lissabon, Portugal
[25] A. Stöhr, Heterostructure Semiconductor
Photonic Devices and Systems,
Euroconference on Advanced Heterostructures, July 1997, Grenoble, France
[26] I. Ryjenkova, M. Alles and D. Jäger, Nonlinear travelling wave photodetector for millimeter wave harmonic frequency generation, Journal of Communications Special
Issue, Microwave Photonics, Vol. 48, Aug.
1997, pp. 14-17
[29] Th. Braasch, G. David, R. Hülsewede, and
D. Jäger, 1D- and 2D-elektro-optic field
mapping to study nonlinear effects in
NLTLs, MWP ’97, Sep. 3-5, 1997,
Duisburg/Essen, Germany
[30] S. Redlich, and D. Jäger, Nichtlineare
Vielschichtheterostrukturen für die Mikrowellen-Photonik, Photonik Symposium,
Oct. 7-9, 1997, Schwäbisch Hall, Germany
[31] S. Redlich, C. Kampermann, and D. Jäger,
Vielschichtheterostrukturen:
Neue
Materialien für die Mikrowellen-Photonik,
Photonik-Symposium, Oct. 8-10, 1997,
Würzburg, Germany
[32] S. Redlich, C. Kampermann, and D. Jäger,
Modeling and simulation of nonlinear hybrid AlGaAs/GaAs Bragg reflectors, 10th
III-V Semiconductor Device Simulation
Workshop, Oct. 16-17, 1997, Torino, Italy
106
[33] A. Stöhr, R. Heinzelmann, T. Alder, W.
Heinrich, T. Becks, D. Kalinowski, M.
Schmidt, M. Groß, and D. Jäger, Optically
Powered Integrated Optical E-Field Sensor, 12th International Conference on Optical Fiber Sensors, Conference Proceedings, Oct. 1997, Williamsburg, Virginia,
USA
[34] M. Groß, T. Alder, R. Buß, R. Heinzelmann,
M. Meininger, and D. Jäger, Micro Photovoltaic Cell Array for Energy Transmission
into the Human Eye, EPVSEC14, 1997,
Barcelona, Spain, Vol. 1, pp. 1165 - 1167
[35] M. Alles, U. Auer, F.-J. Tegude, and D.
Jäger, High-Speed Travelling-Wave Photodetectors for Optical Generation of
Millimeterwaves, APMC ‘97, Dec. 2-5,
1997, Hongkong, China
[36] I. Ryjenkova, D. Jäger, Nonlinear RTD Circuits for High-Speed A/D Conversion,
APMC ‘97, Dec. 2-5, 1997, Hongkong,
China
5 PUBLICATIONS AND PRESENTATIONS
107
6 Guide to the Department of Optoelectronics
Travel by car - The Department of Optoelectronics, now located in the Center for Solid-State Electronics and Optoelectronics (ZHO), can easily reached by car via various highways: A3 from the
north and south, A40 from the east and west. Exit at Duisburg-Kaiserberg or Duisburg-Wedau, see
map for details.
Travel by plane - From Düsseldorf International Airport take the city-train (S-Bahn) S1 to Duisburg main station (Hauptbahnhof, Hbf).
Travel by train - From Duisburg main station (Hauptbahnhof, Hbf) it is a 20 min. walk to the Department of Optoelectronics and the ZHO. You can either go by Taxi or take the bus 933 or 936 to
“Universität” or take the tram 901 to station “Zoo/Uni”.
108
Notes: