Download Practical Considerations for High Frequency Inductive Links

Practical Considerations for High Frequency Inductive
Links
Anthony N. Laskovski , Mehmet R. Yuce and Tharaka N. Dissanayake
The University of Newcastle, University Drive, Callaghan, Australia;
ABSTRACT
Inductive power links are a popular method of wirelessly transferring power to small devices. High efficiency
power transmitters such as the Class-E transmitter are the preferred choice for frequencies up to the low MHz
range, however at frequencies above 100 MHz, the circuit does not produce a high enough efficiency. This
paper investigates the consideration of parasitic elements of the transmission coils in the circuits, showing
an improvement in the circuit’s performance.
Keywords: Inductive Power Transfer, High Frequency, Self-Resonance
1. INTRODUCTION
The latest trends in technology indicate that electronic devices are becoming increasingly sophisticated, less
power hungry, smaller and more flexible, as evidenced by the surging development in mobile communications,
personal devices, as well as medical telemetry and prosthetics. In fact, a number of interesting projects are
taking place involving biomedical gadgets, many of which have been inspired by the Bionic Ear. Retinal
Prosthesis has become a major field of research in the past twenty years and ideas such as wireless body
sensor networks with implantable nodes and tongue-controlled devices.1
The size of a device dictates a number of factors, one of which is power, and research is constantly being
conducted to improve the quality, efficiency and flexibility in supplying power to various electronic devices
within a shrinking physical space. Batteries have been the traditional method of powering smaller devices,
however they may take a considerable amount of room and require frequent replacement. This has led to
investigations into wireless power transfer, which is the focus of this article. Figure 1 shows the basic blocks
that will be discussed, from the power transmitter to the rectifier.
One particular area of interest is the supply of power to implantable electronic devices. Power was traditionally supplied to implantable electronic devices such as Pacemakers by the use of Lithium ion batteries.2
Further author information: (Send correspondence to A.N.L.)
A.N.L: E-mail: [email protected], Telephone: +61 249 217 842
M.R.Y: Email: [email protected], Telephone: +61 249 215 204
T.N.D: E-mail: [email protected], Telephone: +61 249 215 292
Figure 1. Basic blocks of inductive power transfer
Table 1. Comparison of different powering methods for implantable devices
Powering Method
Device
Frequency
Energy
Ref.
Inductive Transfer & Battery
Cochlear Implant
150 kHz
75 mAh
4
Inductive Transfer & Battery
General
4 MHz
6.15mW
5
Direct Inductive Transfer
Retinal Implant
1 MHz
100mW
6
Direct Inductive Transfer
Animal Implant
4 MHz
4.1 mW
7
These batteries need to be surgically removed and replaced periodically due to discharging. A significant
development in implantable power supply was the use of inductive coupling to charge an implantable rechargable battery. The concept may be understood by considering two windings of a weakly coupled transformer,
where the core is a large air gap.3 The primary winding represents the transmission coil, and the secondary
winding the receiving coil. Numerous applications have been developed along this wireless battery charging
idea, varying in size, frequency and coil structure4 .5 The concept of wireless battery charging extended
to solely wireless power transfer, which is a real-time powering system which saves a considerable amount
of implant space, however it creates the requirement of supplying constant wireless energy to the implant.
Table 1 gives an indication of different applications with their associated frequencies of operation and energy
requirements.
Considering that most new devices are small, little room is left for antennas. This increases the frequency
required for any wireless links within a given space, which therefore increases the frequency required for a
power transmitter. However, the transfer of energy becomes less efficient as the frequency of transmission
increases,8 presenting a dilemma that size restrictions increase the possible frequency of operation, yet
transmission efficiency is reduced as the frequency increases.
Device complexity is also growing, and a greater demand is placed on achieving higher data rates, which
require higher transmission frequencies9 .10 This forms yet another factor in selecting the correct operation
frequency for a wireless device. Given the tradeoffs implied by these factors, it has been determined that
in order to achieve more complexity without drastically compromising the power transfer efficiency, power
and data must be transmitted in dual-band operation at low and high frequencies respectively96 .7 From the
perspective of power transfer, Table 1 indicates that wireless power transfer is generally being implemented
from the kHz to MHz range, however as the size of biological devices decreases, the power frequency band
will simply have to increase.
1.1 Power Amplifiers
Upon increasing pressure to transmit power at higher frequencies, efficiency becomes a more significant
consideration, and in the area of high frequency transmission, switched-mode amplifiers are quite popular.
The Class-D amplifier was a popular option for high frequency operation. It consists of a pair of p and
n-channel transistors arranged as a voltage-switching inverter. The square-voltage output of this inverter is
directed through a simple RLC band pass filter, R being the load. This amplifier was popular because of its
high efficiency in practice, ranging from 70 to 80%, however its theoretical efficiency was 100%. The reason
why the theoretical efficiency was unattainable, was because of the parasitic capacitance existent between
transistor terminals, the effect of which increased with frequency.11
The Class-E power transmitter shown in Figure 2 was the next development in this field, and it has been
successful for several reasons. It operates by switching its transistor while there is zero voltage accross C1 .
Its robustness also comes from the fact that dv/dt at this point is also zero, meaning that very little energy
is wastefully discharged1213 .14 The individual elements in the circuit also perform specific functions. C1
absorbs the parasitic capacitance between the transistor’s terminals, C2 blocks DC energy from reaching the
load, L2 represents the primary side of an inductive link and R represents the load seen at the primary side
of the transformer. L1 is simply an inductive choke. Basically, the transmitter was designed in such a way
Figure 3. Small signal model of the Class-E
power transmitter
Figure 2. Class-E power transmitter
that its load network created favourable switching conditions, and the circuit elements absorbed parasitic
capacitance at higher frequencies. These aspects allowed for almost perfect efficiency to be achieved in
practice, and it was introduced as a power transmitter for use in implantable electronics to become the
current norm.15
Figure 3 shows a small signal model of the Class-E power transmitter shown in Figure 2. Expressing the
load seen by the transistor in the Fourier domain will provide interesting information with regards to the
circuit’s natural resonance.
1
1
||
+ sL2 + R
ZL = sL1 ||
sC11
sC2
s2 L2 C2 + sRC2 + 1
(1)
=
2
+ sL1 1
s3 L2 C1 C2 + s2 (RC1 C2 ) + s C1 + LL2 C1 2 + C2 + RC
L1
If the choke inductor L1 is assumed to be large, the resonance of the circuit is expressed as (2)
1
ω= p
L2 C1 ||C2
(2)
1.2 Energy Transfer
The inductive transfer of energy forms a vital element of supplying power to a secondary device, and it is
a natural point of interest to focus on efficient energy transfer, especially with regards to the self-resonance
of inductive transfer coils. A thorough analysis has been conducted, where inductive wireless links for
implantable electronics have been modelled.16 The analysis shows that the self-capacitance and self-resonance
of multi-strand inductors play a significant factor in determining the Q factor of inductors. Hardware results
supported these findings in the low MHz region.
1.3 Rectification
The next phase of energy transfer is to rectify the received power and supply the device with DC energy. In
higher power applications, this is as straightforward as implementing a diode bridge and a voltage regulating
unit, however achieving this at higher frequencies with lower power levels is difficult. The first obstacle is
the diode, which is the basic building block of any rectifier. It has an immediate voltage drop ranging from
0.1 to 0.7V, depending on the type of diode, meaning that it is favourable to use less diodes in most medical
inductive transfer applications5177 . As mentioned in Section 1.1, the highest frequencies for inductive
transfer are currently in the order of MHz, and most designs send more power to compensate for the forward
voltage drop of rectification diodes.3
An interesting development in this field has been the Class-E rectifier. It is designed for high efficiency
operation at high frequencies due to its zero switching properties. The Class-E rectifier is mostly used in
DC/DC converters as shown in Figure 418 .19
Figure 4. Class-E DC/DC Converter18
Figure 5. Active Diode20
Figure 6. Class-E power transmitter with parasitic capacitor C3
Figure 7. Small signal model of a Class-E power
transmitter with parasitic capacitor C3
Another concept aims to remove the voltage drop on a rectifier all together by using comparators to sense
the initiation of the diode’s forward-bias, then sending a one-shot pulse to a switch, allowing the energy to
bypass the diode and supply the load, as shown in Figure 5. The concept has been tested, with an input
voltage signal at 5 MHz and the target implant supply voltage set at 3V with variable loads of 2 kΩ and 10
kΩ. The results from these investigations show up to a 70% increase in received power compared with an
on-chip passive rectifier.20
2. PRACTICAL CONSIDERATIONS
2.1 Power Amplifiers
It was mentioned in Section 1.1 that a trend of decreasing size in implantable devices will increase power
transfer frequency. While the Class-E power transmitter has proven to be very successful at the highest end
of power transfer frequencies, there may be limitations as to how well the transmitter will perform at even
higher frequencies. As the frequency of operation increases for the Class-E power transmitter the size of
capacitors and inductors decreases, and at frequencies above 100 MHz typical capacitor sizes are in the low
pF region. Higher operation frequency also gives rise to more influential parasitic capacitance across the
inductive transfer windings, and the traditional equations used to determine peak switching conditions are
no longer suffice12 .14 If the parasitic capacitance is introduced to the Class-E transmitter, it would appear
as C3 in Figure 6 .
In a similar approach to Section 1.1, the load network was expressed in the Fourier domain so as to
obtain an expression for the circuit’s resonant frequency.
Table 2. Hardware results of Class-E power transmitters at different frequencies with different Inductors
Frequency
Traditional Class-E
Parasitic Consideration
20MHz
60mW
80µW
133MHz
11mW
85mW
403MHz
45µW
112µW
1
1
1
||
+
||(sL2 + R)
sC1
sC2
sC3
s3 L1 L2 (C2 + C3 ) + s2 RL1 (C2 + C3 ) + sL1
=
4
(C1 C2 + C1 C3 + C2 C3 )(s L1 L2 + s3 RL1 ) + s2 (L1 C1 + L1 C2 + L2 C2 + L2 C3 ) + sR(C2 + C3 ) + 1
(3)
ZT = sL1 ||
If the choke inductor L1 is again assumed to be large, the resonant equation of the circuit appears as (4).
1
ω=p
L2 (C1 ||C2 + C3 )
(4)
Comparing this expression with (2) shows that (4) is a more detailed expression for the circuit’s resonance.
If the circuit is simplified to exclude parasitic capacitance, the resonant frequency expression will revert to
(4).
The consideration of the transmitting inductor’s parasitic capacitance lead to the question of how to
use the Class-E transmitter at frequencies above 100 MHz. Given that the parasitic capacitance of the
transmitting inductor plays a more important role in the Class-E power transmitter as frequency increases,
it may imply that the self-resonance and parasitic element of the inductor is also worth considering at these
frequencies.
Three Class-E transmitters were implemented in hardware to investigate this point of interest. One was
designed to operate at 20MHz and the others at 133 MHz and 403 MHz, each according to established
prescriptions.14 The output power was recorded by measuring the voltage accross the resistor R shown in
Figure 2, and the curves are presented in Figures 8, 10 and 12. Photos of three of the inductor coils are
shown in Figures 14, 15 and 16.
... including their parasitic capacitance...
The circuits were then redesigned by replacing the existing inductive coils, including their parasitic
capacitance, which was determined by (5), where εo is the permeability of free space, r is the radius of the
coil, a is the thickness of the wire, N is the number of turns and l is the length of the wire.21
C=
εo 2πrN.2a
l/N
(5)
√
The inductance was also calculated22 and the resonant frequency determined by ω = 1/ LC. The original
inductors were replaced with the inductors, self-resonant at the circuit’s operation frequency. The output
signal produced by the 20 MHz, 133 MHz and 403 MHz circuits are shown in Figures 9, 11 and 13 respectively.
A comparison of these results in Table 2 indicates that the 20 MHz circuit produces more power using the
original inductor as specified by Class-E design principles when compared with the inductor optimised for its
self-resonance at the operation frequency. The 133MHz circuit on the other hand produces a lower output
when using the prescribed Class-E inductor, compared to the output achieved when the parasitic capacitance
is considered to design a self-resonant inductive coil. The 403 MHz circuit produced considerably lower power,
however the power of the circuit designed to consider parasitic capacitance was higher. A possible reason
for this was the use of a lower power transistor SOT-343 for the 403 MHz experiments.
Figure 8. Voltage accross 10 Ω resistor in a Figure 9. Voltage accross 10 Ω resistor in a
20MHz Class-E power transmitter with tradi- 20MHz Class-E power transmitter with self restional specifications
onant inductor
Figure 10. Voltage accross 10 Ω resistor in a Figure 11. Voltage accross 10 Ω resistor in a
133MHz Class-E power transmitter with tradi- 133MHz Class-E power transmitter with self
tional specifications
resonant inductor
Figure 12. Voltage accross 10 Ω resistor in a Figure 13. Voltage accross 10 Ω resistor in a
403MHz Class-E power transmitter with Tra- 403MHz Class-E power transmitter with self
ditional specifications
resonant inductor
Figure 15. A coil used for 133 MHz
inductive transfer
Figure 16. A coil used for 403 MHz
inductive transfer
Figure 14. A coil used for 20 MHz
inductive transfer
3. CONCLUSION
This paper outlined a number of developments in the area of inductive power transfer, from power transmitters to rectification. Considerable focus was placed on the use of power transmitters at higher frequencies
using the Class-E power transmitter. It was determined that it is important to consider parasitic elements
in the Class-E power transmitter when using it as an inductive power transmitter above 100 MHz. These
considerations will improve the output power of the inductive links at higher frequencies in addition to
assisting in the reduction of device size and perhaps the transmission distance.
REFERENCES
[1] Huo, X., Wang, J., and Ghovanloo, M., “Wireless control of powered wheelchairs with tongue motion
using tongue drive assistive technology,” Engineering in Medicine and Biology Society, 2008. EMBS
2008. 30th Annual International Conference of the IEEE , 4199–4202 (Aug. 2008).
[2] Tsukamoto, H., “Hermetically sealed lithium rechargeable batteries for high reliability applications:
medical, aerospace and other specialties,” Battery Conference on Applications and Advances, 2002. The
Seventeenth Annual , 129–134 (2002).
[3] Sauer, C., Stanacevic, M., Cauwenberghs, G., and Thakor, N., “Power harvesting and telemetry in cmos
for implanted devices,” Circuits and Systems I: Regular Papers, IEEE Transactions on [Circuits and
Systems I: Fundamental Theory and Applications, IEEE Transactions on] 52, 2605–2613 (2005).
[4] Lim, H., Yoon, Y., Lee, C., Park, I., Song, B., and Cho, J., “Implementation of a transcutaneous charger
for fully implantable middle ear hearing device,” Engineering in Medicine and Biology Society, 2005.
IEEE-EMBS 2005. 27th Annual International Conference of the , 6813–6816 (2005).
[5] Li, P. and Bashirullah, R., “A wireless power interface for rechargeable battery operated medical implants,” Circuits and Systems II: Express Briefs, IEEE Transactions on [see also Circuits and Systems
II: Analog and Digital Signal Processing, IEEE Transactions on] 54(10), 912–916 (Oct. 2007).
[6] Wang, G., Liu, W., Sivaprakasam, M., Zhou, M., Weiland, J. D., and Humayun, M. S., “A dual band
wireless power and data telemetry for retinal prosthesis,” in [Engineering in Medicine and Biology
Society, 2006. EMBS ’06. 28th Annual International Conference of the IEEE], 4392–4395 (2006).
[7] Zimmerman, M. D., Chaimanonart, N., and Young, D. J., “In vivo rf powering for advanced biological
research,” in [28th Annual International Conference of the IEEE], in Medicine, E. and Society, B., eds.,
2506–2509 (Aug 2006).
[8] Vaillancourt, P., Djemouai, A., Harvey, J., and Sawan, M., “Em radiation behavior upon biological
tissues in a radio-frequency power transfer link for a cortical visual implant,” Engineering in Medicine
and Biology Society, 1997. Proceedings of the 19th Annual International Conference of the IEEE 6,
2499–2502 vol.6 (Oct-2 Nov 1997).
[9] Gosalia, K., Lazzi, G., and Humayun, M., “Investigation of a microwave data telemetry link for a retinal
prosthesis,” Microwave Theory and Techniques, IEEE Transactions on 52, 1925–1933 (Aug. 2004).
[10] Soora, S., Gosalia, K., Humayun, M., and Lazzi, G., “A comparison of two and three dimensional dipole
antennas for an implantable retinal prosthesis,” Antennas and Propagation, IEEE Transactions on 56,
622–629 (March 2008).
[11] El-Hamamsy, S.-A., “Design of high-efficiency rf class-d power amplifier,” Power Electronics, IEEE
Transactions on 9(3), 297–308 (May 1994).
[12] Sokal, N. and Sokal, A., “Class e-a new class of high-efficiency tuned single-ended switching power
amplifiersw,” Solid-State Circuits, IEEE Journal of 10, 168–176 (Jun 1975).
[13] Kazimierczuk, M., “Class e tuned power amplifier with nonsinusoidal output voltage,” Solid-State Circuits, IEEE Journal of 21, 575–581 (1986).
[14] Sokal, N., “Class-e switching-mode high-efficiency tuned rf/microwave power amplifier: improved design
equations,” Microwave Symposium Digest., 2000 IEEE MTT-S International 2, 779–782 vol.2 (2000).
[15] Troyk, P. and Schwan, M., “Closed-loop class e transcutaneous power and data link for microimplants,”
Biomedical Engineering, IEEE Transactions on 39(6), 589–599 (Jun 1992).
[16] Yang, Z., Liu, W., and Basham, E., “Inductor modeling in wireless links for implantable electronics,”
Magnetics, IEEE Transactions on 43, 3851–3860 (Oct. 2007).
[17] Chaimanonart, N. and Young, D., “Remote rf powering system for wireless mems strain sensors,”
Sensors Journal, IEEE 6, 484–489 (April 2006).
[18] Kazimierczuk, M. and Jozwik, J., “Resonant dc/dc converter with class-e inverter and class-e rectifier,”
Industrial Electronics, IEEE Transactions on 36, 468–478 (Nov 1989).
[19] Reatti, A., Kazimierczuk, M., and Redl, R., “Class e full-wave low dv/dt rectifier,,” Circuits and Systems
I: Fundamental Theory and Applications 40, 73–85 (1993).
[20] Lehmann, T. and Moghe, Y., “On-chip active power rectifiers for biomedical applications,” Circuits and
Systems, 2005. ISCAS 2005. IEEE International Symposium on , 732–735 Vol. 1 (May 2005).
[21] Ludwig, R. and Bratchko, P., [RF circuit design: theory and applications], Prentice Hall (2000).
[22] Lee, Y., RFID Coil Design. Microchip Coil Design (1998).