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Vol. 35, No. 4
Journal of Semiconductors
April 2014
High efficiency and low electromagnetic interference boost DC–DC converter
Li Yajun(李亚军)1; Ž , Lai Xinquan(来新泉)1 , Ye Qiang(叶强)2 , and Yuan Bing(袁冰)2
1 Institute
2 Key
of Electronic CAD, Xidian University, Xi’an 710071, China
Laboratory of High-Speed Circuit Design and EMC, Ministry of Education, Xidian University, Xi’an 710071, China
Abstract: A synchronous boost DC–DC converter with an adaptive dead time control (DTC) circuit and antiringing circuit is presented. The DTC circuit is used to provide adjustable dead time and zero inductor current
detection for power transistors and therefore, a high efficiency is achieved by minimizing power losses, such as
the shoot-through current loss, the body diode conduction loss, the charge-sharing loss and the reverse inductor
current loss. Simultaneously, a novel anti-ringing circuit controlled by the switching sequence of power transistors
is developed to suppress the ringing when the converter enters the discontinuous conduction mode (DCM) for low
electromagnetic interference (EMI) and additional power savings. The proposed converter has been fabricated in
a 0.6 m CDMOS technology. Simulation and experimental results show that the power efficiency of the boost
converter is above 81% under different load currents from 10 to 250 mA and a peak efficiency of 90% is achieved
at about 100 mA. Moreover, the ringing is easily suppressed by the anti-ringing circuit and therefore the EMI noise
is attenuated.
Key words: DC–DC; dead time; discontinuous conduction mode; electromagnetic interference; anti-ringing
DOI: 10.1088/1674-4926/35/4/045002
EEACC: 2570
1. Introduction
The boost DC–DC converter, which is able to deliver a
high output voltage with a very low power supplyŒ1 3 , is
widely used for mobile applications owing to its favorable
characteristics, such as high efficiency, low quiescent current,
small size, wide range of input voltage and excellent driving capacityŒ4 7 . However, the DC–DC converter operates in switch
mode, especially when the converter works in discontinuous
conduction mode (DCM), ringing occurs. On one hand, ringing lowers the power efficiency; on the other hand, ringing
causes large switching noise and serious electromagnetic interference (EMI). EMI noise may be radiated in free space and
damage the radio frequency (RF) circuits. Now, some passive
and active EMI filtering methodsŒ8; 9 are presented to attenuate the EMI noise, but those techniques increase the size and
the cost of a printed circuit board (PCB), which is not suitable
for portable applications. In Ref. [10], a time varying resistor network, which is connected between the switching node
and the power supply of the DC–DC converter, is proposed
to solve this problem. Unfortunately, the time varying resistor
network should be carefully designed to achieve its best function, which requires a complicated control circuit and the die
size will be increased. Meanwhile, the light load efficiency is
decreased when this method is used.
In this paper, a novel boost DC–DC converter with an
adaptive dead time control circuit and anti-ringing circuit is
presented. By this technique, the EMI noise is effectively suppressed and high power efficiency is achieved. Section 2 describes the formation of ringing when boost DC–DC converter
operates in DCM. Section 3 analyzes the working principle of
the proposed boost converter thoroughly. The experimental results are presented in Section 4 and the conclusions are given
in Section 5.
2. Ringing formation
Figure 1 shows the schematic of a conventional synchronous boost DC–DC converter: VIN represents the power
supply and VO is the output; L is an energy transferring inductor; CO is an output filtering capacitor; RL is the load resistor;
CX is the parasitic capacitor at node VLX ; and IL stands for the
current through the inductor L. Normally, to achieve low onresistance and high efficiency, the power switches MN and MP
are always very large, so CX cannot be ignored. There are three
main stages when the boost converter works in DCM. Figure 2
illustrates the timing diagram and equivalent circuits.
During stage 1 (t0 < t < t1 /, MN is on and MP is off. VLX
and CX are shorted to ground. IL ramps up at a slope of VIN =L
from zero and the energy stored in inductor L also increases.
Fig. 1. Schematic of a conventional synchronous boost DC–DC converter.
* Project supported by the National Natural Science Foundation of China (No. 61106026) and the Fundamental Research Funds for the
Central Universities of China (No. K50511020028).
† Corresponding author. Email: yajunli@stu.xidian.edu.cn
Received 27 July 2013, revised manuscript received 9 November 2013
© 2014 Chinese Institute of Electronics
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Li Yajun et al.
Fig. 2. (a) Timing diagram. (b) Equivalent circuits.
During stage 2 (t1 < t < t2 /, MN is off and MP is on. VLX
is equal to VO and CX is paralleled with output capacitor CO .
IL ramps down at a slope of (VIN – VO /=L and flows into the
output VO . The energy stored in inductor L depletes gradually.
At the end of this stage, IL decreases to zero.
During stage 3 (t2 < t < t3 /, both MN and MP are off.
Inductor L and parasitic capacitor CX form a resonant tank.
Ringing will appear at the VLX node, causing large EMI in the
system. In this stage, the following equations can be obtained:
IL .t/ D CX
VIN
dVLX .t/
;
dt
VLX .t/ D L
dIL .t/
:
dt
(1)
(2)
3. Proposed Boost DC–DC converter
3.1. Architecture
d2 VLX .t /
1
1
C
VLX .t/ D
VIN :
2
dt
LCX
LCX
(3)
Moreover
VLX .t2 / D VO ;
(4)
1
IL .t2 / D 0:
CX
(5)
From Eqs. (3)–(5), the resonant voltage at node VLX can be
given by
VLX .t/ D VIN C .VO
VIN / cos !0 .t
t2 / ;
t 2 6 t 6 t3 ;
(6)
and the resonant current can be derived as
IL .t/ D
VIN VO
sin !0 .t
!0 L
where !0 is calculated by
t2 / ;
(8)
Today and in the future, small-value off-chip inductorsŒ11; 12 or even on-chip inductorsŒ13 are and will be widely
used for portable applications to achieve small size and low
cost. Thus, the oscillation frequency at the VLX node gets higher
and higher. Large ringing and EMI noise may seriously affect
RF circuits and decrease the power efficiency. In addition, the
resonant current sinks and sources current from VIN repeatedly,
which therefore results in poor stability of the power supply.
According to Eqs. (1) and (2), we can get
0
VLX
.t2 / D
1
!0 D p
:
LCX
t 2 6 t 6 t3 ;
(7)
In order to suppress the ringing and improve the power efficiency, as shown in Fig. 3(a), a novel boost converter with
a dead time control (DTC) circuit and an anti-ringing circuit
is proposed. D is the body diode of the P-channel metal-oxide
semiconductor (PMOS) switch MP. The substrate of the MP
should be connected to the highest voltage of the boost converter VO and the substrate of the N-channel metal-oxide semiconductor (NMOS) switch MN should be tied to the ground to
make sure the normal operation of DC–DC converter. A compensation blockŒ14 18 ensures the stability of the converter
and the pulse width modulator (PWM) module provides the
duty cycle. The soft-start signal VSTART minimizes the inrush
current and the output overshoot during normal startup. Timing diagram of the proposed converter is shown in Fig. 3(b),
which is different from the timing diagram of a conventional
converter shown in Fig. 2(a) in two aspects.
(1) Dead time, during which the power transistors MP and
MN are both off, is added to reduce shoot-through current loss,
body diode conduction loss and charge-sharing loss; the DTC
circuit provides an optimal dead time. Moreover, the DTC cir-
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Li Yajun et al.
Fig. 3. (a) Architecture. (b) Theoretical waveforms.
Fig. 4. (a) Body diode conduction loss. (b) Charge-sharing loss.
cuit can avoid the reverse inductor current from lowering the
power efficiency when the inductor current reaches zero.
(2) During stage 3, ringing is eliminated by the anti-ringing
circuit based on the switching sequence of power transistors.
Therefore, low EMI and high efficiency can be achieved.
3.2. DTC circuit
The power switches MN and MP always have very low onresistances for the purpose of obtaining high efficiency, so MN
and MP should not conduct at the same time to avoid a large
shoot-through current that greatly degrades the efficiency and
causes large glitches in the inductor current and output voltage.
To solve this problem, a dead time between stages 1 and 2 is
needed. During the dead time, both MN and MP are off. Inductor current IL charges the parasitic capacitor CX and VLX goes
up from zero. It should be noted that the optimal dead time TOPT
is related to the load condition and should satisfy the equation
as follows:
TOPT D
VO CX
;
IPEAK
(9)
where IPEAK is the peak inductor current, which is proportional
to the load current and thus TOPT is finally determined by the
load current when VO and CX are given. If the dead time is fixed
as T and T is designed according to the medium load of
the converter, extra power losses caused by the charge-sharing
and the body diode conduction will be introduced, which can
easily be understood as follows.
(1) Body diode conduction loss. As shown in Fig. 4(a),
assuming the load current is very large and hence the fixed dead
time T is much longer than the optimal dead time TOPT , the
body diode D will conduct and VLX will be clamped to VO C
VD (forward voltage drop of diode D). Since the voltage drop
across the body diode is much higher than that across the power
switch MP, conversion efficiency is decreased due to the body
diode conduction loss.
045002-3
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Li Yajun et al.
Fig. 5. Schematic of the proposed DTC circuit.
(2) Charge-sharing loss. As shown in Fig. 4(b), if the load
current is very small and hence T is much shorter than TOPT ,
MP will be turned on before VLX reaches VO . Energy stored in
the output capacitor CO will charge CX until VLX goes up to VO .
That is to say, the energy stored in CO for the load is depleted
because of the charge-sharing loss. As a result, a dynamic dead
time optimization is required.
In this design, a DTC circuit that can provide an adaptive
dead time is presented. Compared with dead time optimization
methodsŒ19; 20 that need complex algorithms to find out the optimal dead time, the proposed DTC circuit can easily adjust the
dead time depending on the load current with a very compact
structure. As shown in Fig. 5, the DTC circuit is composed of
transistors M1–M9, a driver and a constant current source IB .
The sources of M2 and M5 are connected to the node VLX while
the sources of M1, M3 and M4 are connected to VO . The aspect
ratios of M1–M9 are designed as follows:
W
W
D K1
;
(10)
L M2; 3
L M1
W
W
D
;
(11)
L M4
L M5
W
W
D K2
;
(12)
L M6; 9
L M7; 8
where K1 > 1 and K2 > 1.
Figure 6 shows the simulation results of the proposed DTC
circuit. As mentioned before, at the end of stage 1, MN is turned
off immediately and VLX goes up from zero. Once VLX > VO ,
current I2 goes larger than current I3 because source-to-gate
voltage of M2 is higher than that of M3. Since I6 D K2 I2
and I5 D K2 I3 , I6 > I5 . Moreover, I5 > I4 , I6 is thus much
larger than I4 and the voltage at node VA drops to zero rapidly,
as shown in Fig. 6(a). MP will be turned on and then stage 2
starts. Therefore, adaptive dead time control is achieved. On
the contrary, at the end of stage 2, if a reverse inductor current
happens, VLX < VO and the VA node voltage pulls up to high
Fig. 6. Simulation results of the proposed DTC circuit.
Table 1. State transition table of the proposed DTD circuit.
Stage
VN VP Q
VC MN
MP
MS
Stage 1
1
1
1
1
ON
OFF
OFF
Dead time
0
1
1
1
OFF
OFF
OFF
(T )
Stage 2
0
0
0
1
OFF
ON
OFF
Stage 3
0
1
0
0
OFF
OFF
ON
quickly, as shown in Fig. 6(b). MP is turned off and then stage
3 begins, which prevents the reverse inductor current loss from
lowering the power efficiency.
3.3. Anti-ringing circuit
From Fig. 2(a), ringing occurs during stage 3, causing large
EMI. To overcome this drawback, a novel anti-ringing circuit
is developed. As shown in Fig. 7, the circuit consists of two
level shifters, a PMOS switch MS, a buffer and a dead time
detection (DTD) circuit. The substrate of MS is biased by VO .
When both MN and MP are off in stage 3, MS is turned on. The
oscillation loop is broken by shorting the inductor L and hence
ringing is effectively suppressed.
The difficulty of designing the anti-ringing circuit is to
generate a proper drive signal to make sure MS is turned on
only in stage 3 without affecting the operation of the converter
in other stages. It is obvious that both MN and MP are also
off during the dead time. If control signal VC1 is decided only
by the states of MN and MP, MS will conduct in dead time
and the energy stored in inductor L will be wasted by the onresistance of MS instead of transferring to VO , which results in
large power loss. To overcome this problem, a DTD circuit is
used. From Fig. 7, output signal VC of the DTD circuit is related
to the state of Q besides the states of MN and MP. VC is first
converted by the level shifter and then buffered to get VC1 to
control MS. Table 1 shows the state transition table of the proposed DTD circuit. It can be seen from the table that though
045002-4
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Li Yajun et al.
Fig. 7. Schematic of the proposed anti-ringing circuit.
Fig. 8. Simulation results of the proposed anti-ringing circuit.
Fig. 9. Level shifter.
Fig. 10. Simulation results. (a) A boost converter without anti-ringing
circuit. (b) Proposed boost converter with anti-ringing circuit.
both MN and MP have the same states in stage 3 and dead
time, the state of Q is different and therefore MS is switched
on only in stage 3. Furthermore, the proposed DTD circuit is
very compact and nearly has no time delay. Figure 8 shows the
simulation results of the proposed anti-ringing circuit. From the
results, during the operation period T , drive signal VC is low
only in stage 3, which agrees with the theory analysis above.
It should be noted that drive signals VC1 and VP should be
powered by VO , which is the highest voltage of the converter.
However, VN and the DTD circuit are powered by VIN , so a
level shifter is needed to achieve the conversion of different
voltage levels. The schematic and timing diagram are shown
in Fig. 9.
A boost converter with the proposed anti-ringing circuit
is simulated and the simulation is conducted with the following typical application condition: VIN D 3.3 V, VO D 4.6 V,
L D 1.5 H, CO D 4.7 F, clock frequency fS D 1.6 MHz
and temperature TA D 25 ıC. A high switching frequency ensures a small-value off-chip inductor can be used. In order to
verify the advantages of the proposed boost converter, simula-
045002-5
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Li Yajun et al.
Table 2. Component values and output voltage.
Name
Value
Name
Value
VIN
3.3 V
L
1.5 H
VO
4.6 V
CO
4.7 F
CIN
4.7 F
RFB1
422 k
CREF
1 F
RFB2
150 k
Fig. 11. Micrograph of the proposed boost DC–DC converter.
Fig. 12. Test circuit of the proposed boost DC–DC converter.
tion of a boost converter without an anti-ringing circuit is also
performed and the results are shown in Fig. 10(a). From the
results, ringing occurs when the converter operates in DCM.
Figure 10(b) shows the results of the proposed boost DC–DC
converter and that the ringing is effectively suppressed with the
developed anti-ringing circuit.
Fig. 13. Experimental results. VIN D 3.3 V, VO D 4.6 V, IO D 250 mA.
4. Experimental results
The designed boost DC–DC converter with both the proposed DTC circuit and the proposed anti-ringing circuit has
been fabricated using the 0.6 m CDMOS process. Figure 11
shows the micrograph of the proposed converter and the die
size is about 2.2 1.5 mm2 . The total area of the DTC circuit
and anti-ringing circuit is only 0.03 mm2 , which is less than
1% of the die size. The converter is able to provide a maximum
load current of 250 mA at an input voltage ranging from 2.5 to
4.5 V. The output voltage can be externally set from 4 to 8 V
with a resistor divider. Digital soft-start, cycle-by-cycle current
limit and thermal shutdown are also integrated in this circuit in
order to improve the reliability of the boost converter. The test
circuit of the proposed boost DC–DC converter is shown in
Fig. 12. The component values and output voltage are shown
in Table 2.
Figure 13 presents the experimental results of the proposed
boost DC–DC converter under the condition of VIN D 3.3 V,
VO D 4.6 V and IO D 250 mA (maximum load). The converter
works in continuous conduction mode (CCM) and the output
Fig. 14. Experimental results. VIN D 3.3 V, VO D 4.6 V, IO D 30 mA.
voltage ripple is only 35 mV.
Figure 14 presents the experimental results of the proposed
boost DC–DC converter under the condition of VIN D 3.3 V, VO
D 4.6 V and IO D 30 mA. The converter works in discontinu-
045002-6
J. Semicond. 2014, 35(4)
Li Yajun et al.
Fig. 15. Experimental results. (a) Reference voltage. (b) Line regulation. (c) Load regulation. (d) Measured efficiency.
ous conduction mode (DCM) and the ringing appeared at node
VLX when both MN and MP are off has effectively been suppressed. The measured waveforms agree with the simulation
results and theory analysis perfectly. Furthermore, by carefully
designing the DTC circuit, the on time of body diode is only
about 15 ns and the duration of reverse inductor current is only
about 50 ns owing to the inherent propagation delay in the DTC
circuit and power transistors.
Figure 15(a) shows the reference voltage of the proposed
boost converter under the condition of VIN D 3.3 V, VO D
4.6 V, IO D 0 mA and TA D 40 to 125 ıC. The variation
of the reference voltage is only 2.6 mV under a wide range of
temperatures.
Figure 15(b) shows the line regulation of the proposed
boost converter under the condition of VIN D 2.5–4.5 V, VO
D 4.6 V, IO D 5 mA and TA D 25 ıC. The variation of the
output voltage is only 1 mV and the line regulation is less than
0.03% under a wide range of input voltages.
Figure 15(c) shows the load regulation of the proposed
boost converter under the condition of VIN D 3.3 V, VO D
4.6 V, IO D 0–250 mA and TA D 25 ıC. The variation of the
output voltage is only 12 mV and the load regulation is less
than 0.3% under a wide range of load currents.
Figure 15(d) shows the power efficiency of the proposed
boost converter under the condition of VIN D 3.3 V, VO D
4.6 V, IO D 10–250 mA and TA D 25 ıC. By using the internal synchronous rectification, a maximum power efficiency
of 90% is obtained at about 100 mA load current. Also, by
Table 3. Performance comparison summary of the chips.
Parameter
EUP2512 This work
Input voltage range (V)
2.5–4.5
2.5–4.5
Output voltage (V)
4–8
4–8
Maximum output current (mA)
250
250
Peak current limit (A)
1.2
1.5
Inductor L (H)
4.7
1.5
Capacitor CO (F)
4.7
4.7
Turn-on resistance of MP (/
0.6
0.6
Turn-on resistance of MN (/
0.5
0.5
Switching frequency (MHz)
1.4
1.6
Line regulation @ 40 to 85 ıC (%)
0.5
< 0.3
Load regulation @ 40 to 85 ıC (%)
1
< 0.5
Power efficiency @ 10 mA, VIN D
67
81
3.3 V, VO D 4.6 V (%)
Power efficiency (peak) @ VIN D
83
90
3.3 V, VO D 4.6 V (%)
improving the switching timing of power transistors with the
proposed DTC circuit and anti-ringing circuit, the power efficiency of the boost converter is above 81% under different
load currents from 10 to 250 mA. Table 3 shows the performance comparison of the proposed boost DC–DC converter
and the EUP2512 under room temperature. From the comparison results, high efficiency is achieved by using the proposed
technique.
045002-7
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Li Yajun et al.
5. Conclusions
In this paper, a synchronous boost DC–DC converter with
proposed DTC circuit and anti-ringing circuit is analyzed and
successfully achieved. The DTC circuit provides adaptive dead
time control and zero inductor current detection, thus helping in improving the power efficiency by minimizing the
shoot-through current loss, the body diode conduction loss, the
charge-sharing loss and the reverse inductor current loss. Additionally, the proposed anti-ringing circuit with a DTD cell
not only attenuates the EMI noise but also further enhances the
power efficiency by suppressing the ringing when the boost
converter works in DCM. Experimental results show that the
designed boost DC–DC converter achieves high efficiency and
low EMI and that it is suitable for portable devices.
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