Download Class-AB Amplifier

1
Class-D
Audio Power Amplifiers:
Operation
Efficiency
EMC
2
Class-D Amp Block Diagram
OUT+
feedback
Vcc
PGA
Vin
-
-
-
+
+
+
PWM
LOGIC
H-Bridge
Vcc
feedback
OUT-
Most TI class-D amplifiers are fully differential.
Single-ended bridge implementations are possible.
3
H-Bridge Output Waveforms
• Short-term average equals desired output.
ON
Vcc
Q2
Q1
+ +
- -
Q4
Q3
off
ON
off
Vcc
Q1
+
Duty cycle determines the short-term
average, the amplitude of the output
Positive
Output
Polarity
ON
Q2
-
4
off
- +
Q4
Q3
ON
off
Negative
Output
Polarity
AD Modulation Ripple Current
• AD modulation ripple current in the load
reduces efficiency and increases required
speaker power handling capacity.
Ripple current is limited only by
inductance of the loudspeaker,
typically 20 to 60 uH.
At a switching frequency of
250kHz, approximate peak ripple
current with no output signal is
Vdd * 1uS / L.
For Vcc = 12V and L = 30uH,
peak ripple current is
~ 0.4A !!
This represents nearly a quarter
of a watt drawn from the power
supply, fed to the loudspeaker!
5
BD (Filter-Free®) Modulation
• BD modulation switches the outputs nearly
in parallel when there is no signal.
ON
Vcc
Q1
OUTP
Q2
+
-
OUTN
Q4
Q3
off
off
off
off
Vcc
Q1
OUTP
6
ON
Q2
+
-
OUTN
Q4
Q3
ON
ON
OUTP
OUTN
Differential
Voltage
Across Load
Ripple
Current
BD Modulation Waveforms
• As input increases, output duty cycles are
modulated to produce a net load voltage.
OUTP
OUTN
+5V
Differential Voltage
Across Load
0V
-5V
Current
Current
Increasing
Current
Decaying
Filter-free
scheme
output
and ripple
Filterlessmodulation
modulation scheme's
output
voltagevoltage
and
current
waveforms
increased
signal level.
current
waveforms,with
example
signal
7
BD Modulation Waveforms
• As input increases, output duty cycles are
modulated to produce a net load voltage.
OUTP
OUTN
+5V
Differential Voltage
Across Load
0V
-5V
Current
Current
Increasing
Current
Decaying
Filter-free
scheme
output
and ripple
Filterlessmodulation
modulation scheme's
output
voltagevoltage
and
current
waveforms
increased
signal level.
current
waveforms,with
example
signal
8
Why Class-D? –
Class-AB Power Dissipation
• Class-AB amplifier output devices support
high voltages and currents simultaneously.
Class-AB Amplifier
PVDD
Vamp
Q1
Q4
C1
Q3
Iout
Rload
Vin
9
Q5
Q2
Class-AB Power and Efficiency
• BTL or differential output amplifier:
• For a sine output:
• Average power
Vout = Vo sin(wt)
to the load:
Iout = Vout / Rload
Pout = Vo ^ 2
2 Rload
• Average power
from the supply is
• Net power in amp:
Vps * Iout average:
Pamp = Pps – Pout
Pps = 2 Vps * Vo
• Efficiency: Pout/Pps
P Rload
• Efficiency = P Vo
4 Vps
10
Class AB Efficiency is Low
11
0%
0%
100%
20%
90%
10%
80%
40%
70%
20%
60%
60%
50%
30%
40%
80%
30%
40%
20%
100%
10%
50%
Efficiency, %
Class-AB Power and Efficiency
0%
• Dissipation is
shown relative to
maximum output
power with no
clipping.
max at output power = 40.5% of max
Pwr.Dis, % of
Max.Pout
• Ideal class-AB
amplifier power
dissipation and
efficiency.
Relative Output Power, % of Max, No Clipping
maximum 78.5% at point of clipping
Class-AB Dissipation and
Efficiency
Dissipation is ~0.56W,
and Efficiency is ~68%, at
1.2W output, near clipping
Dissipation is ~0.49W,
and Efficiency is ~29%, at
0.2W, well below clipping
Class-AB Operation (TPA6211A1)
12
Class-D Power Losses
• In Class-D amplifiers most power losses
occur in the output devices, typically FETs
with low on resistances.
VCC
VCC
ON
OFF
OFF
ON
OUTP
OUTN
OUTP
OUTN
OFF
L1
L1
15uH
15uH
ON
State 1: high-side OUTP On, low-side
OUTN On
13
ON
OFF
State 2: high-side OUTN On, low-side
OUTP On
Class-D Efficiency
• At high output levels, Class-D amplifier
efficiency depends only on a resistor ratio,
so it is essentially constant.
Efficiency =
Pout
=
Rload .
Pout + Pamp
Rload + Rfet
• Switching edges and quiescent currents
contribute only small losses, so they affect
efficiency only at the lowest power levels.
14
Typical Audio Crest Factor
& Output Power
 Ppeak 
CrestFactor  10 * log 

 PRMS 
PPK = 2.4 W
PRMS = 150 mW
PRMS = 1.2 W
• Audio signal crest factors are ~12 - 18 dB.
– At 2.4W peak, 12dB crest factor means 150mW rms!
– At 150mW, Class-AB efficiency is only ~ 24%!
– Most of the time the amplifier will run at this efficiency!
15
Class-AB vs. Class-D
Power Dissipation
• When comparing
amplifiers consider
crest factor.
• At 2.4 W peak, 12dB
crest factor, typical
RMS power is 150 mW.
• The Class-AB Amp is
24% efficient so
475mW is wasted.
• The Class-D amplifier is
greater than 80%
efficient, so less than
37.5mW is wasted.
16
RC Filter for Viewing
Class-D Output
Rflt
Cflt
Rflt
Cflt
Audio signal unclear in switching output
Class-D
RC filter reveals audio with small ripple
• Filter frequency should be 30 to 40 kHz.
• Possible RCs: 100W/47nF; 330W/18nF.
17
Class-AB and Class-D
Supply Currents
Power
Dissipation
vs.
Output
Power
18
Radiation by Cables & Traces
• Cables and traces carrying RF currents
act as antennas.
• An antenna tends to produce a quasicircular electric field around its axis.
Dipole Antenna Electric Field Strength at
Constant Radius
L = λ/2
θ
L = λ/10
<- Z AXIS ->
I
| Eθ|
<- XY PLANE ->
19
How EMI Occurs
• EMI requires a generator that couples to
another circuit where its signal is detected.
HORIZONTAL:
power/input cables
E-fields
VERTICAL:
power/input cables
output cables
output cables
E-fields
antenna
antenna
• Output cables produce EMI radiation with
horizontal orientation, and power and input
cables do so with vertical orientation.
20
Antenna Impedance
• Antenna impedance normalized to Z0,
characteristic impedance, looks like this.
• Z0 is generally several hundred ohms to
several kilohms, fairly high.
Ope n Transmission Line Z re lativ e to Zo
j Z / Zo
10
10
8
8
6
6
4
4
2
2
0
0
-2 0
1/8
-4
line -le ngth / source -wav e le ngth OR
fre que ncy / (one -wav e le ngth-fre que ncy)
-6
1/4
3/8
1/2
5/8
3/4
7/8
1.0
-4
-6
-8
-8
-10
-10
0.0
37.5
75.0
112.5
150.0
187.5
225.0
262.5
fre que ncy, M Hz, re . fo = 300 M Hz
21
-2
300.0
Test Venue and Setup
National Technical Systems’ FCC-certified 3meter chamber in Plano, Texas.
Chamber / Antenna Setup
22
Turntable / EVM Setup from Rear
EVM Layout on Turntable
• 18’ power and input lines were decoupled
with snap-on ferrites to simulate a working
situation in a product.
To 1kHz Sine Source
To 12 Vdc Power Supply
Steward 28A2024-0A0
Snap-On Beads,
2 Turns Thru Each
LinN
LinP
Inputs Fed in
Parallel
RinP
RinN
Snap-On Beads
Located 4” Along
Cables from EVM
23
RoutN
RoutP
LoutP
Output Cable Length
21”, ~0.53m
LoutN
GND
Vcc
Loads 8 ohms 50 W +
44 uH in Series, to
Simulate Loudspeakers
EMC Test
FCC Class-B Peak Scan PASS!
24
FCC Class-B
Measurement Review
Measurements we just completed
45.3 dBmV peak scan
However, quasiresult at 287MHz appears
to have only 0.7 dB margin peak result of 41.2
dBmV shows true
against limit of 46 dBmV
margin is 4.8 dB!
25
FCC Class-B Peak Scan
Comparison
What made the difference between failure
with EVM 1 and success with EVM 2?
Peak at 50 dBuV would
virtually certainly fail in
quasi-peak measurement
EVM 1, Antenna Vertical
26
Peak at 37 dBuV would
pass in quasi-peak
measurement
EVM 2, Antenna Vertical
TPA3008D2 EVM PCB
Both EVMs use the same PCB.
Top Layer
Bottom Layer
EMI Output Filters
High-Freq.
Decoupling
Power +
Output
Ground
Plane
27
Pwr
PAD
Input
Ground
Plane
The Difference: the
Ferrite Beads
• EVM 1: 2508053017Y3
(Fair-Rite), 300Ω 3A 0805.
0 Adc:
~180Ω
0.5Adc:
~17Ω
28
30MHz
• EVM 2: 2518121217Y3
(Fair-Rite), 120Ω 3A 1812.
0 Adc:
~85Ω
0.5Adc:
~41Ω
30MHz
Capacitors for EMI Filters
• Capacitor manufacturers generally provide
graphs of impedance vs. frequency.
• The graph below is by Kemet. The added
red line approximates Z of 1nF.
1nF
10nF
100nF
29
ESL (equivalent
series inductance)
is ~ 2 to 4 nH.
Capacitors to Avoid
• High-K capacitors like X5Rs are sensitivity
to temperature and even frequency.
• They also have high ranges of sensitivity
to AC and DC voltage, illustrated below.
100%
90%
80%
70%
60%
50%
40%
30%
20%
10%
0%
100%
95%
90%
85%
80%
0%
20%
40%
60%
80%
DC Bias Relative to DCV Rating
30
X5R Capacitance vs. AC Voltage
Relative Capacitance
Relative Capacitance
X5R Capacitance vs. DC Voltage
100%
0%
20%
40%
60%
80%
AC Voltage Relative to Test Voltage
100%
Inductors for EMI Filters
• Inductors saturate, and they self-resonate.
• Low SFR will make ineffective EMI filters!
31
Rules for Picking Filter
Components
• Be aware of the various shortcomings and
frequency limitations and make these part
of your design.
• Insist on full characterization before trying
a part in a filter.
• Try predicting actual attenuation from full
characterization before spending some
hours building and testing a filter.
32
Parasitics in PCB Layouts
• Parasitics that promote EMI are mostly
capacitive and inductive, coupling EMI
currents into nearby traces and degrading
decoupling and EMI filter components.
• In 1-oz copper, resistivity is ~ 0.5mΩ per
square, usually not an issue. (Resistance
of 1-oz Cu is 0.5mΩ * length / area.)
• A trace 10 mils by 100 mils has low
resistance, only about 5 milliohms.
33
PCB Trace Inductance
• Low inductance requires wide traces – so
we use planes for ground and even power!
Isolated PCB Trace
35
30
width =
width =
width =
width =
width =
width =
10 mils
10 mils
40 mils
40 mils
160 mils
160 mils
40
25
20
15
10
5
4-layer PCB width = 10 mils
4-layer PCB width = 30 mils
35
30
25
20
15
10
5
Trace Length, Inch
Inductance varies almost linearly
with length but only by a factor of
~1/2 for a 16:1 increase in width!
1.2
1.1
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
1.4
1.3
1.2
1.1
1.0
0.9
0.8
0.7
0.6
0.5
0
0.4
0
34
2-layer PCB width = 10 mils
2-layer PCB width = 30 mils
0.2
Trace Inductance, nH
40
Cu
Cu
Cu
Cu
Cu
Cu
Trace Inductance, nH
1-oz
2-oz
1-oz
2-oz
1-oz
2-oz
PCB Trace Directly Over Ground Plane
Trace Length, Inch
Inductance varies directly with length
and separation and inversely with width.
(Trace layers are next to ground plane.)
Potential Impact On
Components
• Inductance of a 0.3” long 10mil trace over
a 2-layer PCB ground plane is about 9 nH.
• This is greater than for a typical SMD cap,
2 to 4 nH, so it will dramatically degrade
performance of the capacitor!
SMD Capacitor
Equivalent Circuit
1nF SMD cap
ESL (equiv. series
inductance) = 3 nH
f.res = 92 MHz
SMD Capacitor
Equivalent Circuit
with PCB Trace
Inductance Added
1nF SMD cap
ESL (equiv. series
inductance) = 12 nH
f.res = 46 MHz
• Circuit loops also add inductance. There
is no easy measure here; smaller is better!
35
TPA3008D2 EVM PCB Review
Grounding is the single most critical factor.
Top Layer
Bottom Layer
Power +
Output
Ground
Plane
36
PwrP
AD
Input
Ground
Plane
TPA3008D2 Ferrite Bead
EMI Filters
• TPA3008D2 uses ferrite bead + capacitor
EMI output filters, which work for output
cable length up to about 21”, 0.53m.
TPA3008D2 EMI EVM
OutP
OutN
output
cables
and
loads
Ferrite beads are 2518121217Y3 from
Fair-Rite, 120 ohms at 100 MHz,
3.0A max.Idc, 1812.
Capacitors are 4.7nF 50V X7R 0603.
• In these filters a second bead on each
output damps output cable resonances.
37
Why the Second Bead?
• Without the second bead the output
impedance of the filter is essentially
inductive or capacitive.
Ferrite Bead
~ Equivalent Circuit
SMD
Capacitor
~
Equivalent
Circuit
Ferrite Bead + Shunt Cap Filter
- Filter output impedance is generally reactive
(inductive or capacitive).
- This is especially true when the ferrite bead
is in saturation, because its impedance
becomes more like that of an inductor.
• Impedance zeros of the output cable will
resonate with the shunt leg, driving its
impedance high and drastically reducing
filtering at those frequencies.
38
Inductor Filters
• Higher power and longer cables may
require inductors for compliance.
• The filter below from our TPA3100D2 EVM
(20W per channel at 10% THD) permitted
it to pass FCC-B with wide margins.
TPA3100D2
100nF
OutP
470nF
100nF
39
OutN
Inductors are A7503AY-330M
from Toko, 33uH, 1.8A,
output
11 x 13.5 mm max.
cables
and
loads
Capacitors are 50V X7R.
TPA3100D2 EVM PCB Review
Again, grounding is the most critical factor.
Top Layer
Bottom Layer
Output EMI Inductors
40
Decoupling Caps
Output EMI Caps
Additional Resources
www.ti.com/analog
www.arrow.com
41
APPENDIX 1
PCB Layout for EMC
42
PCB Layout for EMC
• Use a ground plane to minimize ground
inductances.
• Use a star ground concept centered on the
primary switching chip with the PowerPAD
as star ground center point when it is
provided.
• Place high-frequency decoupling caps as
close as feasible to the chip to minimize
trace inductance.
43
PCB Layout for EMC Cont’d.
• Place high-frequency decoupling caps and
EMI filter caps on the ground plane layer
to minimize ground return inductance.
• Confine power and output currents to a
separate section of ground plane, away
from input circuits, both digital and analog!
– EMI-inducing currents can be radiated into
either type of input.
– Switching edges can corrupt clock and data
lines, causing increases in noise and THD.
44
PCB Layout for EMC Cont’d.
• Route traces through pads of decoupling
and filter caps, not from other elements.
• Try to avoid vias in traces for high currents
and for decoupling and EMI filter caps.
Double them if they must be used in traces
for high currents. (Via impedance carries
some uncertainty.)
• Locate EMI filters at the circuits they filter,
not at a distance. The intervening traces
are antennas!
45
PCB Layout for EMC Cont’d.
• Avoid PCB trace lengths that are closely
related to wavelengths of primary power
frequencies.
• We have seen GSM interference in input
traces 1.5”, 3.8cm, long, roughly the
quarter wavelength of GSM frequency
1900MHz (wavelength ~= 0.15m or 6”!).
• In such a case, reflections reinforce each
other to create a relatively large voltage.
46
APPENDIX 2
Antennas
47
Radiation & Impedance
• Simple Antenna Radiation Pattern
• Simple Antenna Impedance Pattern
– A good engineering text on electromagnetism
is a useful reference. I used a book called
Applied Electromagnetism, by Shen & Kong,
from PWS Publishers.
48
Simple Antenna
Radiation Pattern
• A dipole antenna produces the following
cross-sectional field strength pattern at
constant radius from the antenna center.
Dipole Antenna Electric Field Strength at
Constant Radius
L = λ/2
θ
L = λ/10
<- Z AXIS ->
I
| Eθ|
<- XY PLANE ->
49
For antenna lengths less than 1/2 wavelength:
|Eo| ~= ω μ | I | L |sin θ| / ( 4π r ), where
• ω is angular frequency,
• μ is permeability of surrounding medium, in
our case generally free space,
• I is current in the antenna, defined to flow
vertically on the Z axis,
• L is antenna length,
• θ is the angle of “r” with respect to the
vertical, Z, axis, and
• r is the radius of the measurement point
from the center of the antenna.
The field vector direction is that of increasing θ.
Dipole Field Characteristics
• The pattern is like a donut with an infinitely
narrow hole.
• Field strength varies inversely with r and
directly with I, ω, L and |sin θ|.
– Antenna length less than a half wavelength
produce the field pattern shown above.
– Antenna length of a half wavelength produces
very nearly the same pattern.
– Antenna length more than a half wavelength
produces multiple lobes with nulls between.
50
Consequences of
Dipole Pattern
• Higher currents and higher frequencies
produce higher fields.
• Longer antennas produce (and collect!)
higher fields (they are more “efficient”).
• Components, wires and PCB traces lying
parallel to the axis of a current-carrying
antenna see the highest electric field
strengths and induced voltages.
51
Antennas Are Everywhere
• Input, power and output cables constitute
antennas and can radiate EMI if they are
not appropriately filtered.
• Traces on PCBs also constitute antennas
and can have the same undesirable effect.
• So, filter input, power and output lines as
required and locate filters as close as
possible to the generators they address.
52
Simple Antenna
Impedance Pattern
• Impedance of a dipole antenna over a
ground plane may be approximated using
transmission line theory.
• Characteristic impedance, which applies at
low frequency, is typically a couple of
hundred ohms to several kilohms.
• Since all current generated in an antenna
must be reflected from its ends, we can
use reflection patterns to predict relative
impedance versus frequency.
53
Open-Circuit Transmission Line
• Voltage is reflected from the open circuit in
phase with the input voltage, so
V = V0 ( e-jkz + ejkz ) = 2 V0 cos(kz).
(k is a wave number equal to 2pi / λ, wavelength.)
• Current is reflected out of phase, so
I = V0 ( e-jkz - ejkz ) / Z0 = 2j V0 sin(kz).
• Their ratio is transmission line impedance
vs. frequency, so
Z = Z0 / ( j tan(kz) ).
54
Open-Circuit Antenna
Impedance
• Impedance of an open-circuit transmission line
relative to its Z0 looks like this.
j Z / Zo
Open Transmission Line Z relative to Zo
10
10
8
8
6
6
4
4
2
2
0
0
-2 0
1/8
-4
line-length / source-wavelength OR
frequency / (one-wavelength-frequency)
-6
1/4
3/8
1/2
5/8
3/4
7/8
1.0 -2
-8
-6
-8
-10
-10
0.0
37.5
75.0
112.5 150.0 187.5 225.0 262.5 300.0
frequency, MHz, re. fo = 300 MHz
55
-4
Antenna Impedance Effects
• Remember that this is an approximation:
it’s altered by influences like end effects
and other conductors nearby (transmission
line analysis assumes infinite length and a
perfect layout!).
• Antenna impedance periodically changes
from capacitive to inductive with a zero at
each transition.
• Near its zeros antenna impedance adds
complex elements to EMI filters.
56
Antenna Impedance Effects
Cont’d.
• At the resonances that result, the antenna
will draw relatively large currents and will
radiate strongly, producing EMI peaks.
• These resonances also can interact with
EMI filters at frequencies that depend on
the phase of the filter’s output impedance.
• Damping in the environment and the EMI
filter and PCB layout, always present, will
limit magnitude of these peaks.
57
APPENDIX 3
Formulas for PCB Trace
Inductance & Via
Impedance
58
Inductance of Isolated
PCB Traces
• Inductance of an isolated PCB trace can
be computed as follows, for inches & cm.
L ~= 5l [ ln(l/(w+t))+1/2 ] nH, in
L ~= 2l [ ln(l/(w+t))+1/2 ] nH, cm
( l, w & t are trace length, width and thickness)
• Inductance is roughly linear with length.
• However, because of the logarithmic
factor, inductance is insensitive to trace
width and thickness. (Thickness
especially has little effect.)
59
PCB Trace Over a
Ground Plane
• Inductance of a PCB trace directly over a
ground plane can be computed as follows.
L ~= 5lh/w nH/in, inches
L ~= 2lh/w nH/cm, cm
( l, w & h are trace length, width and separation
from the ground plane)
• L varies directly with length and separation
from ground and inversely with width.
• It’s possible to achieve much lower
inductance in this configuration.
60
Via Impedance
• Inductance (h & d = height and diameter).
L ~= 5h [ ln(4h/d)+1 ] nH/in, inches
• Resistance (ρ is copper resistivity, 2.36e-6
ohm-inch, l is via length & A is copper
annular area)
R ~= ρl/A, inches
• 20-mil via, 1-mil plating, in 0.06” PCB:
L ~= 5*0.06*(ln(4*0.06/0.019)+1) ~= 1.1nH
R ~= ρ*0.06/(pi*(0.01^2-0.009^2)) ~= 2.4mohm
61