Download Flat Panel Display : Principle and Driving Circuit Design Outline

Outline
Flat Panel Display : Principle and
Driving Circuit Design
Ch4. Driving Circuits Design of A-Si TFT
– Gate Driving Circuit
– Source Driving Circuit
Chapter 4
– LCD-TV Driving Technology
Driving Circuit Design of A-Si TFT
– Small-Size TFT-LCD Driver IC
– Trends of Digital Interface
中興大學電機系 / 汪芳興
1
Introduction to LCD Driver IC
NCHU / EE / 汪芳興 / [email protected]
98(上)
Page 2 / 195
Driving Circuits of TFT- LCD Module
• Improved Visualization:
Y PCB
G1
– A Fundamental Market Enabler
LVDS
+5V
Connector
Connector
ASIC
LVDS
LVDS
Receiver
Receiver
Timing
Timing
Controller
Controller
Y-Driver IC
98(上)
TFT-LCD
TFT-LCD
1280*(3)*1024
1280*(3)*1024
Pixels
Pixels
G1024
D3840
D1
4 CCFL
DC/DC
DC/DC
Converter
Converter
Gamma
Correction
I/F + X PCB
X-Driver IC
DC POWER
Inverter
98(上)
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Gate Driver IC
Driving Circuits of TFT- LCD Module
• Example
• Also called scan driver or row driver
• Function
– Read in start signal
– Progressively turn on pixel TFTs on each gate line
– Turn off TFT during pixel holding period
• Design consideration
– RC delay of bus line (for large-size panel)
• Capacitive coupling driving (CC driving)
• Gate-driver in panel
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Timing of Gate Driver
Gate Driver Architecture
256 CLK
ENB
analog
VGG
VEE
CLK
SP2
S/R
S/R
S/R
S/R
S/R
level
shift
level
shift
level
shift
level
shift
level
shift
Level Shifter
buffer
buffer
buffer
buffer
buffer
Output Buffer
Out1
Out2
Out3
STP
CLK
Gate driver 1
SP1
CLK
Bi-directional Shift Register
SP1
Gate driver 2
RESET
Up/Down
VDD
digital GND
Outn
To Display Area
S/R frequency : 10k~75kHz , Output voltage range : > 12V
G1
G2
G256
G1
G2
Gate driver 1
Gate driver 2
Gate driver IC 電路方塊圖
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Key Specifications
Package of Driver IC
• Channel number
(240,256,264,270,300,308…)
• Max. operation frequency (200KHz, 500KHz)
• 2 level or 3 level driving
• Operation voltage
– digital : 5V, 3.3V
– analog : VGG>20V, VEE<-10V
• Package (TCP, COG, COF)
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Package of Driver IC
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Package of Driver IC
Page 11 / 195
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Power On/Off Sequence
Channel Number vs. Resolution
• Gate driver : No. of driver and No. of output
channel
lines
VGA
480
SVGA
600
XGA
768
SXGA
1024
UXGA
1200
98(上)
6
5
4
3
120
128
120
150
200
154
192
256
This IC is a high-voltage LCD driver,
so it may be damaged by a large
current flow if an incorrect power
sequence is used. Connecting
the drive powers, VEE & VGG,
after the logical power, VDD, is
the recommended sequence.
256
240
300
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98(上)
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CMOS Shift Register
Gate Driver Circuits
Static S/R
Example
φ
φ
IN
OUT
φ
φ
Shift register
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Level shifter Buffer
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Shift Register
Shift Register – Latch 1
/CLK
• Latch
W
X
G1
CLK
– Basic memory
element
/CLK
SP(L)
N1
– Two cross-coupled
logic inverter
– Bistable circuit
CLK
N2
ENB1
ENB2
Y
G2
Z
S1
98(上)
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Shift Register – Latch 2
98(上)
US 6,157,361
SHARP
S2
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Level Shifter – Example 1
Inverter Type
Vdd
T1
/OUT
IN
US 6,724,361 B1
Vcc
Vpp
P1
P2
Vin
Vout
T2
N1
N2
GND
GND
GND
¾ With both transistors of the inverters turned on, a current path from the supply voltage
to ground is present, resulting in undesirable power consumption.
Ref : Low Power Digital VLSI Design, A. Bellaouar, M. Elmasry
98(上)
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Level Shifter – Example 2
Vdd
Vdd
Latch Type
P3
Vcc
Output Buffer
Vdd
Vdd
M3
M6
P4
Vout
Vcc
M2
P1
Vcc
N3
P2
OUT
IN
N4
M5
Vcc
INPUT
/INPUT
Vin
OUTPUT
N1
M1
M4
GND
GND
N2
US 4,486,670
INTERSIL
GND
GND
Area ratio = e (2.7) ~ 3
¾ This circuit overcomes the problem of direct power consumption by using a latch.
98(上)
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TSMC HV Process
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Roadmap of High Voltage Technology
• Roadmap of TSMC HV technology
Source : Web of TSMC
Source : Web of TSMC
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Two-Line Scanning
• Frame inversion
• column inversion
Two-Line Scanning
• row inversion
• dot inversion
Vd,N+1
line time
(N-2)th
Vd,N-1
(N)th
Vd,N
Vd,N-2
Vg, N-2
Vd 1
Vd 2
Vd 3
Vpixel
Vg, N-1
TFT
Cst
CLC
Vg, N
~10us
~10us
1 frame ~16ms
•First pulse is for precharge
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Gate Driver Design
Scan Driver Consideration
Toshiba 15” UXGA
Dual Driving
Shut circuit was inserted
between S/R outputs and
L/S inputs to avoid over
current phenomena due to
timing difference between
right and left side scan
driver outputs.
> 40“ ?
Level shifter is divided in
two stages.
stages First stage is
made to shift high level
from 10 V to 15 V, second
stage is made to shift low
level from 0 V to –2 V..
The scan driver with both side driving effectively reduces the RC time constant of the
gate line.
Ref:SID 00’, p.1121
Ref:IDW 00’, p.167
98(上)
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Three-level Capacitive Coupling Driving
(C. C. Driving)
• Cst on gate
Three-level C. C. Driving
3-level
ΔV 1 = Vghl ×
Cgd
Clc + Cst + Cgd + Cds
ΔV 2 = Vgc ×
Cst
Clc + Cst + Cgd + Cds
ΔV 3 = Vgc ×
Cgd
Clc + Cst + Cgd + Cds
Scan line n
Clc
Cds
Vgc
Vghl
Cgd
Cst
Vcom
i.e. Vghl × Cgd = Vgc × (Cst + Cgd )
If ΔV 1 = ΔV 2 + ΔV 3
Scan line n-1
ΔV 1 = ΔV 2 + ΔV 3
98(上)
V3
V1
Pixel voltage
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V2
Vgc = Vghl ×
Cgd
Cst + Cgd
, then
• Feed-through effect is eliminated.
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98(上)
Three-level C. C. Driving
Cs on gate
GN-1(+)
Scan line n-1
Black(+)
Cds
Black(+)
Clc
Positive Driving
Data
Cente
r
White(+)
White(+) Temporal
White(-) Vcom
White(-)
Cgd
White(+) Final
White(-) Vcom
Negative Driving
Black(-)
Black(-)
Temporal Pixel
Voltage Level
Data Line
Voltage Level
98(上)
Final Pixel
Voltage Level
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Vblack(+)
Page 30 / 195
GN-1(-)
GN(+)
Ve-
VP
V6
V5
V4
Vcom
V1
Vblack(-)
V2
V3
Negative Polarity
Not valid for Vcom AC
Page 31 / 195
Ve+
Vg
GN(-)
VP
•line inversion
•low voltage source driver
•complex scan driver
Negative Driving
Black(-)
Cst
Scan line n
Negative Driving
Vg
Vcom
Positive Driving
Positive Driving
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Four-level C. C. Driving
• Cs on Gate with 3-Level Driving Scheme
Black(+)
Considerpreviouscase,
0.05× 25
ΔVgd =
= 1.47V
0.4 + 0.4 + 0.05
Cgd
0.05
Vgc = ΔVghl ×
= 25
= 2.778
Cs + Cgd
0.4 + 0.05
98(上)
Positive Polarity
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Four-level C. C. Driving
ΔV1 = (Vg + (Ve−))×
Cgd
(Cst + CLC + Cgd)
Cst
ΔV 2 = (Ve+) ×
(Cst + CLC + Cgd)
Cgd
ΔV 3 = (Ve−) ×
(Cst + CLC + Cgd)
Four-level C. C. Driving
• Cs on Gate with 4-Level Driving Scheme
ΔV 4 = (Vg − (Ve+))×
Cgd
(Cst + CLC + Cgd)
Cst
ΔV 5 = (Ve−) ×
(Cst + CLC + Cgd)
Cgd
ΔV 6 = (Ve+) ×
(Cst + CLC + Cgd)
Black(+)
Black(+)
Positive Driving
Positive Driving
Data
Center
Q ΔV1 + ΔV 2 − ΔV 3 = −ΔV 4 + ΔV 5 − ΔV 6
Cgd
∴(Ve−) − (Ve+) = 2Vg ×
Cst
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White(-)
Negative Driving
Data
Center
White(+)
White(-)
Negative Driving
Black(-)
Data Line
Voltage Level
• Pixel voltages of positive polarity and negative polarity for
LC cell are symmetry.
98(上)
White(+)
Page 33 / 195
98(上)
Black(-)
Final Pixel
Voltage Level
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Trend of Gate Driver IC
Comparison of Addressing
Frame
Row
Column
Dot
Common
voltage
AC/DC
AC/DC
DC
DC
Output
range
Low/high
voltage
Low/high
voltage
high
voltage
high
voltage
2-level
driving
V
V
V
V
3-level
driving
X/V
X/V
V
V
4-level
driving
X/V
X/V
X
X
• package ：
–
–
–
–
TCP（Tape Carrier Package）→
COG（Chip on Glass） →
COF （Chip on Film） →
GIP（Gate-Driver In Panel）。
Vcom AC/DC
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Gate-Driver In Panel
Ch4. Driving Circuits Design of A-Si TFT
• Gate Driving Circuit
• Source Driving Circuit
– Driver Architectures
– Driver Specifications
– DAC
– Output Buffer
– Low power consumption
• LCD-TV Driving Technology
• Small-Size TFT-LCD Driver IC
• Trends of Digital Interface
98(上)
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Source Driver Circuits
• Serial In, Serial Out
• Driver Architectures
• Serial In, Parallel Out
Shift Register
Shift Register
– Line-at a-time (LAAT)
Video
in
– Point-at a-time (PAAT)
Qi
Qi+1
Video in
Qi-1
Qi
Qi+1
Qi+2
Sample
capacitor
• Data Drivers Types
pixel
– Analog Data Driver
control
pixel
hold
capacitor
– Digital Data Driver
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Analog Data Driver
• Also Called Column Driver (Data Driver)
98(上)
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Shorter charging time
Page 39 / 195
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pixel
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LAAT Driver Architecture
Comparison of Analog Data Driver
For a-Si TFT-LCDs
SISO
1 phase
SIPO
1 phase
Circuits complexity
lowest
higher
Pixel charging time
TVL / Pix-H
TVL
Sample time
S/R frequency
TVL / Pix-H
1/(TVL / Pix-H )
1/(TVL / Pix-H)
No
S/R
S/R
S/R
S/R
S/R
S/R
S/R
S/R
Video in
Sample
Sample
Sample
Sample
Sample
Sample
Sample
Sample
Hold
Hold
Hold
Hold
Hold
Hold
Hold
Hold
Buffer
Buffer
Buffer
Buffer
Buffer
Buffer
Buffer
Buffer
Driver IC
TVL / Pix-H
Reformed video
needed
CLK, DIO
Array
Pixel
2
Pixel
3
Pixel
4
Pixel
5
Pixel
6
Pixel
7
Pixel
8
Pixel
Pixel
Pixel
Pixel
Pixel
Pixel
Pixel
Pixel
No
All video signals stored in A latches are written to B latches simultaneously in each horizontal
scanning period. Features: (1) has sufficient charging capability; (2) very difficult to achieve good
uniformity of output voltages of the analog buffer.
TVL :vertical line time, Pix-H :horizontal pixel number
98(上)
Pixel
1
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LAAT vs. PAAT
PAAT Driver Architecture
For some large-size LTPS TFT-LCDs
CLK, DIO
Video
in
Driver
S/R
S/R
S/R
S/R
Sample
Sample
Sample
Sample
Hold
Hold
Hold
Hold
Buffer
Buffer
Buffer
Buffer
DeMUX 1:2
DeMUX 1:2
DeMUX 1:2
DeMUX 1:2
Pixel
1
Pixel
2
Pixel
3
Pixel
4
Pixel
5
Pixel
6
Pixel
7
Pixel
8
Array Pixel
Pixel
Pixel
Pixel
Pixel
Pixel
Pixel
Pixel
A high speed and wide voltage range analog interface circuit that consumes a large amount of power
is required !! Feature : (1) less data driver ICs ; (2) shorter pixel charging time than LAAT.
98(上)
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•
•
Sanyo has been selling products using the PAAT technology. Since
small-size panels have only 234 scan line, there is sufficient time to use
PAAT technology.
SeikoSeiko-Epson has adopted the LAAT architecture because there studies
indicated that the crosstalk on a multiplexed circuit was too severe.
98(上)
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Digital Data Driver
Max. Frequency of Shift Registers
(a) Decoder type is very difficult
to achieve full gray scale
because the circuit
configuration is too complicated !
(b) DAC type is the most
promising one because it has a
less complicated configuration
while keeping full digital
interface !
Vcc=5V, Freq.<2MHz @ μn/p=70/35 cm2/Vs
(a)
Ref:SID’96 Digest. pp.673.
98(上)
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(b)
Page 45 / 195
98(上)
DIO1
complex
simple
noise immunity
high
low
gamma
correction
video signal
processing
cost
Yes
No
circuits
complexity
Compatible to
PC
ADC needed
high
low
CLK
Data
in
Data
receiver
Analog
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Architecture of Source Driver IC for
Large-Size TFT-LCDs
Comparison of Digital/Analog Data
Driver
Digital
Ref:IDW 00’ p.171
Latch Signal
S/R
S/R
S/R
Sample
Reg.
Sample
Reg.
Sample
Reg.
Sample
Reg.
sample
reg.
Hold
Reg.
Hold
Reg.
Hold
Reg.
Hold
Reg.
Hold
Reg.
Level
Shifter
Level
Shifter
Level
Shifter
Level
Shifter
Level
Shifter
Analog Part
Buffer
Output buffers
Polarity Signal
Gamma
Reference
voltages
S/R
DIO2
S/R
Digital Part
Level Shifter
Digital/Analog Converter, DAC
Buffer
Buffer
Buffer
Out1
Out2
Out3
Buffer
Outn
To LCD data line
98(上)
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Timing Diagram of Source Driver
Timing Diagram of Source Driver
128 CLKs
128 CLKs
LD(CLK1)
CLK
POL
DIO1
Positive
DIO2
Odd outputs
Data latch
P1
P2
P3
P4
P5
Driver 1
Invalid
High-Z
P127 P128 P129 P130 P131 P132 P133
1st 384 Outputs
for 1~128 Pixel
Even outputs
High-Z
Negative
High-Z
Tst
Tst
Driver 2
Output load condition :
1K
1K
1K
1K
1K
Output
2nd 384 Outputs
for 129~256 Pixel
15P
15P
15P
15P
15P
Vcom
98(上)
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Key Specifications
•
•
•
•
•
•
•
•
•
•
•
•
NCHU / EE / 汪芳興 / [email protected]
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Page 50 / 195
Channel Number vs. Resolution
Channel number (384, 402, 420, 480, 640, 720…)
Gray scale (6 bit, 8 bit, 10 bit…)
Max operation frequency (45MHz, 55MHz, 65MHz, 75MHz…)
Pixel charging time (eg. R=2k, C=20pF, 6.5us 90%, 11.5us 99.9%)
Frame/row/column/dot inversion
Output voltage deviation (±20mV, ±10mV, ±5mV, ±3mV)
Output voltage (10V, 12V, 13.5V, 15V, 18V)
Interface (TTL, RSDS, mini-LVDS)
Operation voltage (2.5V, 3.3V)
No. of Gamma reference voltage (10, 14, 18)
Package (TCP, COG, COF)
Others (data inversion, low-power mode, offset canceling, charge
sharing …etc.)
98(上)
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• Source driver : No. of driver and No. of output channel
No. of lines
VGA
SVGA
XGA
SXGA,WXGA
WXGA
SXGA+
WXGA
UXGA
WSXGA+
HDTV
QXGA
640x3
800x3
1024x3
1280x3
1366x3
1400x3
1440x3
1600x3
1680x3
1920x3
2048x3
98(上)
12
402
420
480
512
10
192
240
312
384
414
420
432
480
9
8
7
240
300
384
480
6
5
384
402 480
640
480
720
642
720
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Resolution vs. Max. Frequency
Frame rate
D/A Converter
60 Hz
60 Hz
75 Hz
75 Hz
Pixel
frequency
Horizontal
period
Pixel
frequency
Horizontal
period
VGA
25.2 MHz
31.7 μs
31.5 MHz
26.7 μs
SVGA
40 MHz
26.4 μs
49.5 MHz
21.3 μs
XGA
65 MHz
20.7 μs
78.75 MHz
16.7 μs
SXGA
108 MHz
15.6 μs
135 MHz
12.5 μs
UXGA
162 MHz
13.3 μs
202.5 MHz
10.7 μs
Higher resolution, shorter pixel charging time, higher driving frequency
98(上)
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Page 54 / 195
• Ideal N-bit DAC
‡ Digital input
Bin =
bn −1 bn − 2
b0
b1
+
+
+
+
L
21
22
2 n −1 2 n
Where bi is 1 or 0, i.e. binary, bn-1 is the MSB, and b0 is the LSB
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
NCHU / EE / 汪芳興 / [email protected]
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
Idea D/A Converter
Ideal D/A converter
Quantization error in ideal DACs
Performance limitation
Offset error
Gain error
Accuracy
98(上)
• Digital-to-Analog Converters
Page 53 / 195
Fundamentals of Data Converter
•
•
•
•
•
•
• Fundamentals of Data Converter
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
Page 55 / 195
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Idea D/A Converter (cont')
•
Idea D/A Converter (cont')
Analog output Vout is related to Bin through an
analog reference, VRef
1.
2.
3.
Vout and VRef may be voltage, current, or charge.
We assume here that they are voltage (for simplicity)
Definitions：
b
b ⎞
b
⎛b
Vout = V ref ⎜ n −1 1 + n −2 2 + L + n1−1 + 0n ⎟ = V ref × Bin ,
2
2
2 ⎠
⎝ 2
V LSB =
•
V ref
2N
, where VLSB is defined as the voltage changes
when one LSB changes.
1 LSB = 1/2N unitless definition.
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 57 / 195
Quantization Error in DACs
• An ideal 2-bit DAC example：
Input-output transfer curve. In
general, the maximum value of
Vout is not VRef but rather
-n
VRef(1-2 ) or equivalently, VRefVLSB.
• A multiplying DAC (MDAC) is realized by simply
allowing the reference signal, VREF, to be a varying
input signal along with BIN. Such an arrangement results
in Vout being proportional to the multiplication of the input
signals, BIN and VRef.
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
NCHU / EE / 汪芳興 / [email protected]
98(上)
Page 58 / 195
Performance Limitation
• Quantization error (noise) is the inherent uncertainty in
digitizing an analog value with a finite resolution converter.
It is equal to the analog output of the infinite-bit DAC
minus that of the finite-bit DAC.
• Definitions for determining the transfer
responses for DACs.
• Fig 10.1.4 & 10.1.5.
• Resolution
– The transfer response of a DAC is defined to be the
analog levels that occur for each of the digital word.
– The number of distinct analog levels corresponding
to different digital words. Thus, an N-bit resolution
implies that the converter can resolve 2N distinct
analog levels.
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
98(上)
NCHU / EE / 汪芳興 / [email protected]
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
Page 59 / 195
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 60 / 195
Offset Error
Gain Error
• Offset error is in units of LSBs.
• Offset error is the output that occurs for the input
code that should produce zero output.
Eoff ( DAC ) =
Vout
VLSB
0...0
• Gain error is the difference at the full-scale value between
the ideal and actual when the offset error has been
reduced to zero.
• Gain error is in units of LSBs.
• DACs︰
⎛ Vout
⎝ VLSB
‡ E gain ( DAC ) = ⎜
‡ 2-bit example：
1...1
−
Vout
VLSB
0...0
⎞
N
⎟ − (2 − 1)
⎠
‡ 2-bit example︰
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
NCHU / EE / 汪芳興 / [email protected]
98(上)
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
Page 61 / 195
Accuracy
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
NCHU / EE / 汪芳興 / [email protected]
98(上)
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
Page 62 / 195
Accuracy (cont’)
• Absolute accuracy
– The difference between the expected and actual
and transfer response.
– Includes 1.offset error, 2. gain error, 3. linearity
error
• Relative accuracy
– The accuracy of offset and gain errors have been
removed
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
98(上)
NCHU / EE / 汪芳興 / [email protected]
• Accuracy can be expressed as a percentage error of
full-scale value, as the effective number of bit, or as
a fraction of an LSB.
– For example, a 12-bit accuracy implies that the converter’s
error is less than the full-scale divided by 212.
– A converter may have 12-bit resolution with only 10-bit
accuracy, or 10-bit resolution with 12-bit accuracy.
– An accuracy greater than the resolution means that the
converter’s transfer response is very precisely controlled.
(better than the number of bits of resolution)
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
Page 63 / 195
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 64 / 195
D/A Converter
Digital-to-Analog Converters
• Nyquist-rate D/A converters
• Fundamentals of Data Converter
• Decoder-based DAC
• Binary-weighted converters
• Digital-to-Analog Converters
• Glitches
• Thermometer-code DACs
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
98(上)
NCHU / EE / 汪芳興 / [email protected]
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
Page 65 / 195
Nyquist-rate D/A converters
•
98(上)
NCHU / EE / 汪芳興 / [email protected]
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Decoder-Based DAC
Four main types
• Most straight forward approach
– Create 2N reference signals and pass the
appropriate signal to the output.
9 Decoder-based
9 Binary-weighted
• Three main types︰
9 Thermometer-code
– Resistor string
9 Hybrid
– Folded resistor-string
– Multiple resistor-string
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
98(上)
NCHU / EE / 汪芳興 / [email protected]
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
Page 67 / 195
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 68 / 195
Decoder-Based DAC
Resistor-string DAC
VRef
• Example 1︰a 3-bit DAC
with transmission gate,
tree-like decoder.
• Most straight forward approach
– Create 2N reference signals and pass the
appropriate signal to the output.
‡ Transmission gates might be
used rather than n-channel
switches.
• Types︰
–Resistor string
¾ Extra drain and source
capacitance (to GND) is offset
by the reduced switch resistance.
¾ Larger layout
¾ Can operate closer to positive
supply voltage.
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
NCHU / EE / 汪芳興 / [email protected]
Page 69 / 195
Resistor-string DAC (cont’)
•
•
•
b1 b1
R
R
b2
Vout
R
R
R
b2
R
Bin = b22-1+b12-2+b02-3
2N resistors
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
NCHU / EE / 汪芳興 / [email protected]
98(上)
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
Page 70 / 195
• Speed
1.
About the same speed as the transmission gate
implementation.
2.
Compact layout (no contacts are required in the tree)
– Can be estimated using open-circuit time-constant
approach (refer to microelectronics textbook written by
Monotonicity is guaranteed (if the buffer’s offset
does not depend on its input voltage)
The accuracy of this DAC depends on the type of
resistor uesd. Polysilicon (20-30 Ω/□) can have
up to 10 bits of accuracy.
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
NCHU / EE / 汪芳興 / [email protected]
R
Resistor-string DAC (cont’)
Only n-channel switches are used
98(上)
b0 b0
R
– Folded resistor-string
98(上)
R
Page 71 / 195
Sedra and Smith)
– Time-constant
≈ 3Rtr Ctr + 2 ⋅ 3Rtr Ctr + L + N ⋅ 3Rtr Ctr
= N ( N + 1) (2 ⋅ 3Rtr Ctr )
Where Rtr is on resistance of switches,
Ctr is drain or source capacitance of switches, and
N is bit number.
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
NCHU / EE / 汪芳興 / [email protected]
98(上)
Page 72 / 195
Resistor-string DAC (cont’)
Decoder-Based DAC
VRef
•
Example 2︰a 3-bit DAC with digital
decoding.
‡ Compared to example1：
1. Higher speed
2. More area for decoding circuit
‡ Speed
‹ Time-constant ≈Rtr· 2NCtr
‹ For N≦6 example 2 is faster
‹ For N>7 example 1 is faster
• Compromise between example 1 and
example 2.
¾ Folded resistor-string DAC
• Most straight forward approach
R
– Create 2N reference signals and pass the
appropriate signal to the output.
b2
R
R
b1
• Types︰
R
b0
– Resistor string
R
–Folded resistor-string
R
R
R
Vout
R
N
2 resistors (all equal-sized)
資料來源：林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 73 / 195
Folded resistor-string DAC
98(上)
NCHU / EE / 汪芳興 / [email protected]
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Folded resistor-string DAC (cont’)
•
•
•
•
Reduced the amount of digital decoding
Reduce large capacitive loading
Decoding is very similar to that for a digital memory
Example：
– 4 bit (2 bit+2 bit) DAC
– Time constant ≈ Rtr·(22Ctr)+2Rtr·(22Ctr)
• Other design examples：
– 12 bit = 6 bit + 6 bit, or 4 bit + 4 bit + 4 bit, or ……
– 10 bit = 5 bit + 5 bit, or 3 bit + 3 bit + 4 bit, or ……
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
蘇育民, “數位類比轉換器設計與測試之研究,” 暨南國際大學電機工程學系碩士論文, 2004
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
NCHU / EE / 汪芳興 / [email protected]
98(上)
蘇育民, “數位類比轉換器設計與測試之研究,” 暨南國際大學電機工程學系碩士論文, 2004
Page 75 / 195
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 76 / 195
Binary-Weighted (or Binary-Scaled)
Converter
Digital-to-Analog Converters
• An appropriate set of signals that are all related in a
binary fashion
• The binary array of signals might be voltages,
charges, or currents.
• Five main types：
• Nyquist-rate D/A converters
• Decoder-based DAC
• Binary-weighted converters
• Glitches
–
–
–
–
–
• Thermometer-code DACs
Binary-weighted resistor DAC
Reduced-resistor-ratio ladders
R-2R-based DAC
Charge-redistribution switched-capacitor DAC
Current-mode DAC
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
蘇育民, “數位類比轉換器設計與測試之研究,” 暨南國際大學電機工程學系碩士論文, 2004
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 77 / 195
Binary-Weighted Resistor DAC
98(上)
NCHU / EE / 汪芳興 / [email protected]
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Binary-weighted resistor DAC (cont’)
• 4-bit example：
•
Does not require many resistors or switches.
•
Disadvantage
1. Resistor ratio and current ratio are on the order
of 2N. If N is large, this large current ratio
requires that the switches also be scaled so that
equal voltage drops appear them.
b
b ⎞ ⎛R
⎛ b
Vout = − RFVref ⎜ − 3 − 2 − 1 ⎟ = ⎜ F Vref
⎝ 2 R 4 R 8R ⎠ ⎝ R
where : BIn = b3 2−1 + b2 2−2 + b1 2−3 +LL
2. Monotonicity is no guaranteed.
⎞
⎟ BIn
⎠
3. Prone to glitch.
資料來源： Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002, p.624
資料來源： Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002, p.624
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 79 / 195
98(上)
NCHU / EE / 汪芳興 / [email protected]
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Reduced-resistor-ratio Ladders
Reduced-resistor-ratio Ladders (cont’)
• Reduce the large resistor ratios in a binaryweighted array
• Introduce a series resistor to scale signals in
portions of the array
– Ex : VA= -1/4VRef
• An additional 4R was added such that
resistance seen to right of the 3R equals R.
• One-fourth the resistance ratio compared to
the binary-weighted case.
• Current ratio has remained unchanged
– Switches must be scaled in size.
• Repeating this procedure recursively, one can
obtain an R-2R ladder.
資料來源：林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
資料來源:林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
NCHU / EE / 汪芳興 / [email protected]
98(上)
Page 81 / 195
R-2R-based DAC
NCHU / EE / 汪芳興 / [email protected]
Page 82 / 195
R-2R-based DAC (cont’)
• Smaller size and better accuracy than a binary-sized
approach
• R-2R ladder DAC with equal currents through switches
• Slower since the internal nodes exhibit some voltage swings (as
opposed to the previous configuration where internal nodes all
remain at fixed voltage).
– Small number of components
– Resistance ratio of only 2
• 4-bit example︰ I R =
98(上)
Vref
2R
N
, and _ Vout = RF ⋅ ∑
i =1
bi ⋅ I R
= Vref
2i −1
⎛R ⎞ N b
⋅ ⎜ F ⎟∑ ii
⎝ R ⎠ i =1 2
• Current ration is
still large→ large
ratio of switch
sizes
資料來源：呂學旺, “場發射顯示器驅動電路之設計,” 國立交通大學電子工程學系碩士論文, 1994.
魯得中, “場效激發元件驅動器的設計,” 國立交通大學電子工程學系碩士論文, 2003.
資料來源： Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002, p.625
98(上)
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Page 83 / 195
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 84 / 195
Current-mode DAC
Charge-redistribution switched-capacitor DAC
• Insensitive to Op Amp input-offset voltage, 1/f noise,
and finite amplifier gain.
• An additional sign bit can be realized by
interchanging the clock phases (shown in
parentheses)
NCHU / EE / 汪芳興 / [email protected]
High-speed
Switch current to output or to ground
The output current is converted to a voltage through RF.
The upper portion of current source always remains at
ground potential.
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
98(上)
•
•
•
•
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Digital-to-Analog Converters
98(上)
•
•
• Decoder-based DAC
•
• Binary-weighted converters
A major limitation during high-speed operation
Mainly the result of different delays occurring when
switching different signals
Example︰01111……1→1000……0
1.
• Glitches
2.
• Thermometer-code DACs
NCHU / EE / 汪芳興 / [email protected]
Page 86 / 195
Glitches
• Nyquist-rate D/A converters
98(上)
NCHU / EE / 汪芳興 / [email protected]
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I1 represents the MSB current and I2 represents the sum
of (N-1) LSB currents.
The MSB current turns off slightly early, causing a glitch
of zero current.
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
NCHU / EE / 汪芳興 / [email protected]
98(上)
Page 88 / 195
Glitches (cont’)
•
Digital-to-Analog Converters
• Nyquist-rate D/A converters
Glitch disturbance can be reduced by：
• Decoder-based DAC
1. Limiting the bandwidth (placing a capacitor
across RF)
• Binary-weighted converters
2. Using a sample and hold on the output signal.
• Glitches
3. Modifying some or all of the digital word from a
• Thermometer-code DACs
binary code to a thermometer code.
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 89 / 195
Thermometer-Code DAC
•
98(上)
NCHU / EE / 汪芳興 / [email protected]
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Thermometer-Code DAC (cont’)
Digital recode the input to a thermometer-code equivalent.
• Total area required by the transistor switches is the same.
• All transistor switches are of equal sizes since they all
pass equal currents.
• Example： Thermometer-code resistor DAC
•
Advantages over its binary-weighted counterpart
1.
2.
3.
•
Low DNL errors
Guaranteed monotonicity
Reduced glitching noise
Does not increase the size of the analog circuitry compared
to a binary-weighted approach.
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 91 / 195
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
NCHU / EE / 汪芳興 / [email protected]
98(上)
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
Page 92 / 195
Thermometer-Code Current-Mode DAC
(cont’)
Thermometer-Code Current-Mode DAC
•
• Row and column decoders
In high-speed applications
1. The output current feeds directly into an off-chip
50Ω or 75Ω resistor, rather than an output opamp.
2. Cascode current sources are used to reduce currentsource variation due to voltage changes in Vout.
• Inherent monotonicity
• Good DNL errors
• INL errors depend on the placement of the
current sources
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 93 / 195
Examples of DAC 1
•
NCHU / EE / 汪芳興 / [email protected]
Page 94 / 195
Examples of DAC 2.
Chun-Yueh Huang; Tsung-Tien Hou; Hung-Yu Wang; “A 10 bit 100-MHz
current-steering DAC” ASIC, 2005. ASICON 2005. 6th international
Conference On Volume 1, 24-0 Oct. 2005 Page(s):411 - 414
98(上)
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
NCHU / EE / 汪芳興 / [email protected]
98(上)
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
Page 95 / 195
12-bit DAC Paper- A Novel Linear Digital-to-Analog Converter using
Capacitor coupled Adder for LCD Driver ICs
98(上)
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Page 96 / 195
R-String DAC
Examples of DAC 3
5
• 12 Bit Linear DAC using capacitor coupled adder
b1 b1
b2 b2
b3 b3
R/2
R
R
R
out
R
R
R
R
R/2
-19
98(上)
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Page 97 / 195
R-String DAC (cont’)
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
NCHU / EE / 汪芳興 / [email protected]
98(上)
Page 98 / 195
D/A Converter of Source Driver IC
DAC with
ROM
Decoder
資料來源：Philip E. Allen, Douglas R. Holberg, ”CMOS analog circuit design”, Oxford, 2002.
林克旯, “ADC及DAC積體電路實作,” MSD聯盟-混合訊號式積體電路設計技術推廣教育計畫.
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 99 / 195
98(上)
NCHU / EE / 汪芳興 / [email protected]
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Gamma Correction
Gamma Correction
T-V curve is non-linear.
10 Gamma Voltage
18 Gamma Voltage
Adjust the voltage of V1~V8 to obtain the desire transmittance.
資料來源： Datasheet of Novetek Corp.
98(上)
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Page 101 / 195
98(上)
z Advantages
¾Simple architecture
¾Low noise
¾Optimized gamma curve
Rail to rail OP-AMP
Gamma
Voltage
(+)
Polarity
control
signal
+
DAC P
Odd
Output
To data line
z Disadvantages
¾Large area
¾Area increase rapidly as bit number increases
¾Steady current consumption in R-string
NCHU / EE / 汪芳興 / [email protected]
Page 102 / 195
Output Buffer
R-String (ROM Decoder) DAC
98(上)
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Gamma
Voltage
(-)
Page 103 / 195
+
DAC N
98(上)
NCHU / EE / 汪芳興 / [email protected]
Even
Output
Page 104 / 195
Output Buffer
OP-AMP – An Example
P/N type OP-AMP
Gamma
Voltage
(+)
Polarity
control
signal
Odd
Output
-positive
DAC P
buffer
+
To data line
Gamma
Voltage
(-)
Even
Output
-
negative
buffer
DAC N
+
z Wide Dynamic Range (Rail to Rail )
z Low deviation
z High driving ability
Positive and negative signals are applied by DAC1 and DAC2 separately. Dynamic range of each DAC is
reduced to ½ compared with conventional ones.
NCHU / EE / 汪芳興 / [email protected]
98(上)
z Low power consumption
z Optimum OP AMP. area
Page 105 / 195
98(上)
Design of Output Amplifier
NCHU / EE / 汪芳興 / [email protected]
Page 106 / 195
Design of Output Amplifier
• OP-AMP
• CMOS OP-AMP
Vdd
Compensation
circuitry
+
+
V1
-
+
+
V2
-
V1
Vout=Av(V1-V2)
Vss
V2
Differential
Transconductance
stage
High-gain Vout Output
stage
Buffer
Vout’
Bias Circuitry
Ref: P. E. Allen, D. R. Holberg, “CMOS Analog Circuit Design,” 2nd Ed., Oxford, 2002
98(上)
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Ref: P. E. Allen, D. R. Holberg, “CMOS Analog Circuit Design,” 2nd Ed., Oxford, 2002
Page 107 / 195
98(上)
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Page 108 / 195
Model for a Nonideal Op-Amp
Classification of Op-Amp
Hierarchy
conversion
voltage to current
Current to voltage
Vout(s)=
Classic differential
amplifier
Differential-tosingle-ended load
(current mirror)
modified differential
amplifier
Source/sink
Current load
MOS diode
load
Current
stage
Av(s)[v1(s)-V2(s)]
•
•
•
•
•
•
•
Finite differential-input impedance Rid, Cid
Output resistance Rout
Common-mode input resistance Ricm
Input offset voltage Vos
Input offset current Ios=|IB1-IB2|
Common-mode rejection ratio : v1/CMRR
Noise : in2, en2
±Ac(s)[v1(s)+V2(s)]/2
/ EE / 汪芳興 / [email protected]
Ref: P. E. Allen, D. R. Holberg, “CMOS 98(上)
Analog CircuitNCHU
Design,”
2nd Ed., Oxford, 2002
voltage to current
Current to voltage
Transconductance
grounded gate
Transconductance
grounded gate
Class A (source
or sink load)
Class B (push-pull)
2nd
voltage
stage
Ref: P. E. Allen, D. R. Holberg, “CMOS Analog Circuit Design,” 2nd Ed., Oxford, 2002
Page 109 / 195
Examples of Op-Amp
• Classical two-stage
CMOS op amp
1st
voltage
stage
98(上)
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Page 110 / 195
Design of Op Amp
• Folded-cascode op
amp
• Procedures
– Choosing or creating the basic
structure
– Select dc currents and transistor
size
– Boundary conditions
1. Hand
calculation
2. Computer
simulation
• Process specification (Vt, k’, Cox…)
• Supply voltage/current and range
• Operating temperature and range
Ref: P. E. Allen, D. R. Holberg, “CMOS Analog Circuit Design,” 2nd Ed., Oxford, 2002
98(上)
NCHU / EE / 汪芳興 / [email protected]
Ref: P. E. Allen, D. R. Holberg, “CMOS Analog Circuit Design,” 2nd Ed., Oxford, 2002
Page 111 / 195
98(上)
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Page 112 / 195
Model Parameters for a Typical CMOS
Bulk Process
Design of Op Amp
– Performance requirement
•
•
•
•
•
•
•
•
•
•
•
•
Gain
Gain bandwidth
Settling time
Slew rate
Input common-mode range, ICMR
Common-mode rejection ratio, CMRR
Power-supply rejection ratio, PSRR
Output voltage swing
Output resistance
Offset
Noise
Layout area
Example
≧70 db
≧5 MHz
≦1 us
≧5 V/us
≧±1.5 V
≧60 db
≧60 db
≧±1.5 V
N/A
≦ ± 10 mV
≦100 nV/√Hz@1 kHz
≦ 5000 x (min. L)2
/ EE / 汪芳興 / [email protected]
Ref: P. E. Allen, D. R. Holberg, “CMOS 98(上)
Analog CircuitNCHU
Design,”
2nd Ed., Oxford, 2002
Page 113 / 195
Design of Op Amp
Parameter
symbol
Process description
Typical parameter value
n-channel
unit
p-channel
VT0
Threshold voltage
(VBS=0)
0.7±0.15
-0.7±0.15
V
K’
Transconductance
parameter (in sat.)
110.0±10%
50.0±10%
uA/V2
γ
Bulk threshold
parameter
0.4
0.57
V1/2
λ
Channel length
modulation parameter
0.04(L=1um) 0.05(L=1um) V-1
0.01(L=2um) 0.01(L=2um)
Surface potential at
strong inversion
0.7
2|φF|
0.8
V
Model parameters for a typical CMOS bulk process using the simple
model with values based on a 0.8um silicon-gate bulk CMOS n-well
/ EE / 汪芳興 / [email protected]
Ref: P. E. Allen, D. R. Holberg, “CMOS 98(上)
Analog CircuitNCHU
Design,”
2nd Ed., Oxford, 2002
Page 114 / 195
Frequency and Phase Response (1/3)
• For a single-loop, negative-feedback system
• Procedures
– Decide on a suitable configuration
F(s)
• Trade-off between noise, offset, power…
-
– Determine the compensation method
Vin(s)
• Especially for very large CLOAD
– Design device sizes for proper dc, ac, and
transient performance
• Hand calculation : 80% Æ important to get a feel
for sensitivity of parameter variation
• computer simulation : 20% Æ for optimization
Rule: (use of simulator) x (common sense) = constant
/ EE / 汪芳興 / [email protected]
Ref: P. E. Allen, D. R. Holberg, “CMOS 98(上)
Analog CircuitNCHU
Design,”
2nd Ed., Oxford, 2002
+
Σ
A(s)
Vout(s)
Stable Î |A(jω0°) F(jω0°)| = |L(jω0°)| < 1
where ω0° is defined as Arg[- A(jω0°) F(jω0°)] = Arg[L(jω0°)] = 0°
or Arg[- A(jω0dB) F(jω0dB)] = Arg[L(jω0dB)] > 0°
where ω0dB is defined as |A(jω0dB) F(jω0dB)| = |L(jω0dB)| = 1
Ref: P. E. Allen, D. R. Holberg, “CMOS Analog Circuit Design,” 2nd Ed., Oxford, 2002
Page 115 / 195
98(上)
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Frequency and Phase Response (2/3)
Frequency and Phase Response (3/3)
• Time response of a second-order system
• Bode Plots:
– Large P.M. results in less ‘ringing’ Æ good stability
– |A(jω) F(jω)| & Arg[- A(jω) F(jω)]
– P.M. >45° (at least), > 60° (preferable)
Phase Margin
ΦM = Arg[L(jω0dB)]
/ EE / 汪芳興 / [email protected]
Ref: P. E. Allen, D. R. Holberg, “CMOS 98(上)
Analog CircuitNCHU
Design,”
2nd Ed., Oxford, 2002
Page 117 / 195
/ EE / 汪芳興 / [email protected]
Ref: P. E. Allen, D. R. Holberg, “CMOS 98(上)
Analog CircuitNCHU
Design,”
2nd Ed., Oxford, 2002
Page 118 / 195
Miller Compensation of Op Amp
Second-order Uncompensated Op Amp
• Miller Compensation Technique
• Small-Signal Equivalent Circuit
P’1=-1/(RICI)
P’2=-1/(RIICII)
P1= -1/(gmIIRIRIICC)
If F(s)=1 (worst case)
PM << 45°
Æneed compensation
CII >> CI, CC
P2= - gmIICC /(CICII+ CCCII+ CICC) ≒ - gmII/CII
Z1 = gmII/CC
Ref: P. E. Allen, D. R. Holberg, “CMOS Analog Circuit Design,” 2nd Ed., Oxford, 2002
98(上)
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98(上)
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Page 120 / 195
Miller Compensation of Op Amp
Two Stage Op Amp with Parasitic C
• Root locus plot of the
loop gain F(s)=1
• There are more than two pole due to C1, C2, C3,......
• We will concentrate on two most dominant (small) pole
and the RHP zero.
− GI GII − ( gds2 + g ds4 )(gds6 + gds7 )
p1 ≅
• Bode plots
– P2 does not affect the
magnitude until after
|AF|<1
– Z1 increases the phase
shift
gmII CC
=
p2 ≅
− gmII − gm6
=
CII
C2
z1 ≅
gmI gm2
=
CC CC
Unit gain bandwidth：
GB ≅
/ EE / 汪芳興 / [email protected]
Ref: P. E. Allen, D. R. Holberg, “CMOS 98(上)
Analog CircuitNCHU
Design,”
2nd Ed., Oxford, 2002
Page 121 / 195
Miller Compensation Technique
g mI g m 2
=
CC
CC
/ EE / 汪芳興 / [email protected]
Ref: P. E. Allen, D. R. Holberg, “CMOS 98(上)
Analog CircuitNCHU
Design,”
2nd Ed., Oxford, 2002
Page 122 / 195
Miller Compensation Technique
• P1：miller pole and accomplish the desire
compensation
• M6 is a NMOS. CC is multiplied by the gain of the
2nd stage, gmIIRII, to give a capacitor in parallel
with RI of gmIIRIICC
• Multiplying gmIIRIICC times RI and
inverting it. Then we get
p1 ≅
gm6CC
− GI GII
=
g mII CC
• P2：output
output pole,
pole at least equal to GB and is due to the
capacitance at the output of the op amp.
• CII = CL(load capacitance).
• Since |p2| is near or greater than GB, the reactance of CC
is approximately 1/(GBCC) and is very small.
• M6 is a MOS diode and its small signal resistance is gm6-1.
• Multiplying gm6-1. by CII and inverting gives
− g mII − g m 6
p2 ≅
=
C II
C2
− ( g ds 2 + g ds 4 )( g ds 6 + g ds 7 )
g m 6 CC
/ EE / 汪芳興 / [email protected]
Ref: P. E. Allen, D. R. Holberg, “CMOS 98(上)
Analog CircuitNCHU
Design,”
2nd Ed., Oxford, 2002
Page 123 / 195
/ EE / 汪芳興 / [email protected]
Ref: P. E. Allen, D. R. Holberg, “CMOS 98(上)
Analog CircuitNCHU
Design,”
2nd Ed., Oxford, 2002
Page 124 / 195
Miller Compensation Technique
• Z1 (the RHP zero) boosts the loop gain magnitude
while causing the loop phase shift to become
more negative. Æ undesirable
• It worsens the stability of the op amp.
CC
• It comes from the two feedback path.
Output Deviation of Buffer
Vo,
absolute offset,
between chip
idea voltage
VDD
RII
Vout
+
V’’
+
-
• The signals through these two paths
may be equal and opposite and cancel,
creating the zero.
⎛ − g m 6 RII (1 / sCC ) ⎞ ⎛
RII
⎟⎟V '+⎜⎜
Vout ( s ) = ⎜⎜
⎝ RII + 1 / sCC ⎠ ⎝ RII + 1 / sCC
98(上)
V’
M6
-
Vx,
idea voltage
of gray m
Vdvo,Output deviation
Voc,
Output offset
between chip to
chip
⎞
− RII ( g m 6 / sCC − 1)
⎟⎟V ' ' =
V
RII + 1 / sCC
⎠
NCHU / EE / 汪芳興 / [email protected]
IC2
IC1
Page 125 / 195
98(上)
Phase 1
Vcaz=(Vin+Voffset)-Vin
Height
Gamma
resistor
Channel 384
Channel 383
384 channel
source driver
Channel 382
Channel 381
Voffset
Vo=Vin+Voffset
Vo=Vin+Voffset
Channel 4
Channel 3
-+
Channel 2
Channel 1
Vin
Page 126 / 195
Length
Caz +
-
Vin
NCHU / EE / 汪芳興 / [email protected]
Layout of Source Driver
Offset Cancellation
-+ +
V(m, ave),
Average output voltage
of gray m
Voffset
Channel 480
Channel 477
‹How Long the Ph1 is enough to sample the correct Voffset?
‹How Long the Ph2 is enough to charge the pixel voltage?
Gamma
resistor
Channel 479
Channel 478
Vo=Vin+Voffset-Vcaz
Vo=Vin
Channel 4
Channel 3
-+
Vin + Voffset
Channel 420
Caz
Channel 417
Channel 2
Channel 1
Gamma
resistor
Channel 419
Channel 418
Channel 1
480 channel
source driver
Channel 4
420 channel
source driver
Channel 3
Channel 2
Phase 2
Layout length/channel ＜ 2 cm / 480 = 41 um
98(上)
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Page 127 / 195
98(上)
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Page 128 / 195
TSMC HV Process
Power Saving mode (1)
Start pulse input for 1st IC
Turn on one clock
Start pulse output for 1st IC
P = fd*Cload*VDD2
Clock disable signal for 1st IC
Start pulse input for
2nd
Start pulse output for
IC
2nd
IC
I =C×
Clock disable signal for 2nd IC
dV
dt
Start pulse input for 8th IC
Start pulse output for 8th IC
Clock disable signal for 8th IC
Start pulse
1st IC
2nd IC
3rd IC
4th IC
5th IC
6th IC
7th IC
8th IC
LCD Panel
98(上)
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Page 129 / 195
98(上)
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Power Saving mode (2)
Power Saving mode (3)
Distribute clock tree
Dual edge clocking
Shift
Register
Shift
Register
Shift
Register
Shift
Register
Shift
Register
Shift
Register
Shift
Register
Shift
Register
Shift
Register
Shift
Register
Shift
Register
Shift
Register
clock
clock
Clock 1
Clock 2
Page 130 / 195
CK1
XCK1
CK2
XCK2
1
2 3 4
Clock 1
Clock 2
98(上)
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Page 131 / 195
98(上)
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Page 132 / 195
Charge Sharing
Charge Sharing
• Because only half of the charges with positive potential are
recycled, power saving efficiency of the previous charge
sharing is theoretically limited to 50%.
Vgmama+
-
DAC P
O.P.
+
Data
Charge Sharing
Control signal
Vgmama-
-
DAC N
Odd
Output
To data
line
Even
Output
O.P.
+
Data
98(上)
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Page 133 / 195
Waveform of LCD Driver
98(上)
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Page 134 / 195
Power Consumption of LCD Source Driver
Power Consumption
/ IC
Large size panel
4~8mA
CNS for Car
2~3mA
DSC, Game, PDA
0.5 ~ 2 mA
Mobile-phone
< 0.3mA,
(power saving )
98(上)
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Page 135 / 195
98(上)
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Outline
Trend of LCD Source Driver
Ch4. Driving Circuits Design of A-Si TFT
z 6-bit / 8-bit to 10 bit resolution
z 384 / 480 to >500 output channels for SXGA+ /
WSXGA+ / UXGA / WUXGA panel
– Gate Driving Circuit
– Source Driving Circuit
z TTL to RSDS/mini-LVDS data interface for low power
and EMI issue
– LCD-TV Driving Technology
z 10V to 18V for wide-view-angle panel
– Small-Size TFT-LCD Driver IC
z Reduced chip size for cost down
– Trends of Digital Interface
z Low power consumption
z TV application
98(上)
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Page 137 / 195
98(上)
LCD TV vs. LCD Monitor
NCHU / EE / 汪芳興 / [email protected]
LCD TV Electronics
LCD Control Board
LCD Module
ASIC
Antenna
RF in
Composite
video
YPbPr
Analog
RGB
Digital
DVI
OSD
Tuner
Y/C
YCbCr
Sync
Sep.
Video
Decoder
ADC
Antenna
Cable
NCHU / EE / 汪芳興 / [email protected]
Page 139 / 195
(T-CON,
RSDS/TTL
transmitter)
De-interlace
cable
Scaler
TTL/
LVDS/
TMDS
MiniMini-LVDS/
RSDS/
Gamma Volt. Gen. Data driver
TTL/
γ
PCB with data bus
Display Area
HDTV : 1920x1080; 1280x720
WXGA : 1366x768/1280x768
SDTV : 720x480
TMDS
Rx
MCU
98(上)
Page 138 / 195
Set-Top Box
DC/DC
power
Inverter
Back light unit
Scan driver
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 140 / 195
Digital TV System
LCD-TV Control IC
• Trumpion-Zipro Chip
Scaler IC
VSB/COFDM
Demodulator
Digital Tuner
MPEG2
Decoder
POD/CI
32MB DDR
SDRAM
98(上)
NCHU / EE / 汪芳興 / [email protected]
LCD TV Block Diagram
NTSC/PAL
SVC208
SVC230
TUNER
SAW
LA7585
LA7587/88
LA7565
VIF/SIF
SWITCH
LC875448A
LC87F54C8A
MICON
Complex Devices
(PNP+NPN)
Power MOSFET
CPH5506
Diodes
FSS140/132/134
CPH3304/3414
SBS004/005/006
SBE001/SBE002
98(上)
LA4266/4267/4268
LA4276/4277/4278
LC41XX
LC758XX
LCD
DRIVER
PANEL 1/F
γ CORRECT
スキャンコンバータ
Audio/video Switch
RESET
AF/Output
NCHU / EE / 汪芳興 / [email protected]
Page 142 / 195
De-interlace
POWERAMP
EXTERNAL
SOUND
LA7221
2SC3950
ビデオ出力ドライバ
98(上)
• example
ELECTRONIC
VOL.
RGB DECODER
LA76818
LA863228
EXTERNAL
VIDEO
LA4260/61
LA4265/70
LA4280/82
STK401STK401-XXX
Page 141 / 195
OSD
CONTROLLER
DCDC-DC
CONVERTER
LCD
PANEL
LC74723M
LC74725M
LC74781M
LC74782M
LC74883M
DCDC-AC
INVERTER
BACK LIGHT
LA5661/LA5662
FTS2005/FW332/ECH8605~6
2SD1804T/2SD1802T
2SC57062SC5706-PM/2SJ503/2SJ485
CPH3109/3116/3216/3205/3212/5504
NCHU / EE / 汪芳興 / [email protected]
Page 143 / 195
2SA2039/2SC5566/2SC5707
CPH3115/3109
CPH5702/CPH5706
98(上)
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Page 144 / 195
Black Insertion
Luminance of LCD vs. CRT
• Emulate impulse-type display
– Turn off backlight
– Insert black data or clear the data on pixel
• Double frame rate (60 -> 120 -> 240 Hz)
Display Quality for Moving Picture ： Which is Better ？
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 145 / 195
Adjustable Gamma Curve
Vref [1:18]
switch
γ−2.2
R2 string
NCHU / EE / 汪芳興 / [email protected]
Page 146 / 195
Adjustable Gamma Curve
example
Gamma control code
(from T-CON)
γ−1.8
R1 string
98(上)
8-bit Data
V0~V255
DAC
Vout
γ−2.6
R3 string
Use external Gamma control code to control final Gamma resistance rings
γ−1.8
98(上)
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Page 147 / 195
γ−2.2
98(上)
γ−2.6
NCHU / EE / 汪芳興 / [email protected]
Page 148 / 195
Overdrive for LCD TV
Over 8-bit Color Depth
Conventional
Over-8 bit data
Overdrive
Gray Level
Gray Level
Gn'
new DAC
with
γ-R string
Vref [1:18]
Gn
Gn
Vout (over-8 bit resolution)
Ideal Response /
Voltage by Driver
Gn-1
Voltage by Driver
Gn-1
Overdrive Response
Actual Response
8 bit Source driver IC Æ 10 bit ! Æ 12 bit ?
98(上)
NCHU / EE / 汪芳興 / [email protected]
n-2
n-1
n
Page 149 / 195
n+1 n+2 n+3
n-2 n-1
n
n+1 n+2 n+3
Frame
NCHU / EE / 汪芳興 / [email protected]
98(上)
Frame
Page 150 / 195
Overdrive Circuits
Effect on LC Overdrive
FFD ( Feedforward Driving )Method by Mitsubishi
Dn+1
Data
Lookup Table
Data’
Dn
Memory
Read/Write
Ctrl
address
Control ckts
Tr+ Tf ≈ 25ms (ON/OFF)
Tr, Tf < 20ms (Gray Level)
98(上)
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Page 151 / 195
98(上)
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Page 152 / 195
Other Technologies
Dynamic Contrast Enhancement
¾
Over-8 bit color depth
¾
Dynamic gamma correction
¾
Dynamic backlight control
• 10 bit color depth (source driver IC)
– 1.07 billion colors
• LED backlight
– R, G, B mixed LED BLU
• NTSC ratio > 100%
• R/G/B color sequential method (CF-free)
Normal image
bright image
98(上)
• Thin module thickness
brightness
brightness
brightness
• Power consumption
• Digital Image Processing
– Sony:WEGA engine…
dark image
NCHU / EE / 汪芳興 / [email protected]
Page 153 / 195
LCD TV Trends
Size
15-20”
15-20”
Brightness/C.R.
250/400
400~550/500~700
Response Time
25~16ms
Viewing angle
Resolution
25~16ms
20-30”
16~12ms
Page 154 / 195
Ch4. Driving Circuits Design of A-Si TFT
challenge
LCD TV
NCHU / EE / 汪芳興 / [email protected]
Outline
Technology trends and challenges
LCD Monitor
98(上)
32-65”
– Source Driving Circuit
>550/>2k
< 8 ms
140/160
160/160
176/176
XGA~UXGA
SVGA~WXGA
WUXGA~HDTV
– Gate Driving Circuit
– LCD-TV Driving Technology
COST
– Small-Size TFT-LCD Driver IC
– Trends of Digital Interface
2002~2003
98(上)
2004~2008
NCHU / EE / 汪芳興 / [email protected]
Page 155 / 195
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 156 / 195
Small-Size TFT-LCD Driver IC
Applications of Small Size Display
Controller
0.25 μm
DC/DC
converter
0.6 μm HV
TFT
TFT
Source
driver
0.5 μm
Controller +
Source driver
Gate
driver
0.6 μm HV
DC/DC converter
+ gate driver
HV
4 Chip
TCP/COF Type
Single chip
HV
2 Chip
COG/COF Type
Hitachi
Samsung
奇景
聯詠
智寶科技
1 Chip
COG Type
小尺寸TFT-LCD驅動晶片解決方案演進圖
NCHU / EE / 汪芳興 / [email protected]
Page 157 / 195
Architecture of One Chip TFT-LCD
Driver IC
98(上)
G177
G176
G175
:::
G2
G1
G0
– 2.5V Block：OSC, SRAM, APR, Some Logic
– 5V Block：I/O, Source, Regulator, Charge Pump (PWR)
VGH
VGL
Vgoff
GVDD
VGS
AVDD
– 32/40V Block：Gate, Regulator, Charge Pump (PWR)
(176+2)CH
Gate Driver
Gate
control
Power
Supply
Circuit
Source Driver
SRAM
Gamma
adjusting
and
graylevel
generator
64
SRAM
Built-in GRAM
132x18x176 bit
Address
counter
Read / Write
Data latch
18
Timing
Generator
PWR
APR
Index
register
OSC
Control
register
18
SPI
98(上)
Control
Data
Page 159 / 195
R/W
NCHU / EE / 汪芳興 / [email protected]
18
System Interface
18/16/9/8-bit parallel, 3-pin SPI
Vcom
VDD
VDD3
98(上)
396 Channel
Source Driver
Latch circuit
Gate Driver
GAMA64
PWR
Page 158 / 195
Single Chip TFT-LCD Driver IC for
Mobile Phone Application
• Simple IC Sketch for TFT Mobile Phone
Gate Driver
NCHU / EE / 汪芳興 / [email protected]
S396
S395
S394
:
:
:
S3
S2
S1
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 160 / 195
Voltage Setting
Voltage Waveforms
VGH(+9 ~ +16.5V)
BT2-0
VCI(2.5V ~ 3.3V)
VCI1
VDD(2.0V ~ 3.3V)
VSS(0V)
VREG1OUT
GVDD(+3.0~ +5.0V)
VCOM4-0
VRH3-0
-1 times
VC3=0
VRL3-0
VGH(+9 ~ +16.5V)
AVDD(+3.5~ +5.5V)
BT2-0
VC2-0
AVDD(+3.5~ +5.5V)
GVDD(+3.0~ +5.0V)
VDD(2.0V ~ 3.3V)
VcomH(+3.0~ VREG1OUT)
sn(source output)
VcomH(+3.0~ VREG1OUT)
VDV4-0
-1 times
VCOM
VcomL(VCL+0.5 to 1.0V)
VSS(0V)
VcomL(VCL+0.5 to 1.0V)
VCL
VC3=1
VRL3-0
VgoffH (~to -5.0V)
VgoffH (to -5.0V)
Gn (Gate output)
VgoffL(VGL+0.5 to -5V)
VREG2OUT
VgoffL(VGL+0.5 to -5V)
VGL (-16.5 to -9V)
電壓波形設定示意圖
Note :
adjust the conditions of AVDD-GVDD > 0.5V, VcomL-VCL > 0.5V, and VgoffL-VGL > 0.5V with loads because
They differ depending on the display load to be driven. In addition, Vci can be directly input to Vci1.
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 161 / 195
98(上)
Power Pins Description (1/2)
Symbol
I/O
VDD
-
NCHU / EE / 汪芳興 / [email protected]
Page 162 / 195
Power Pins Description (2/2)
Description
System power supply.
As NT3911 has internal regulator, VDD range varies with each mode.
Non-regulated mode (PregB = 1) : +2.0 ~ +2.5 V
Regulated mode (PregB = 0) : +2.0V
Symbol
I/O
Description
VcomR
I
A reference voltage of VcomH.
When VcomH is externally adjusted, halt the internal adjuster of VcomH by setting the register and insert a variable resistor
between VREG1OUT and VSS. When this pin is not externally adjusted, leave it open and adjust VcomH by setting the
internal register.
VcomH
O
This pin indicates a high level of Vcom generated in driving the Vcom alternation.
Connect this pin to the capacitor for stabilization.
VcomL
O
When the Vcom alternation is driven, this pin indicates a low level of Vcom. An internal register can be used to adjust the
voltage. Connect this pin to a capacitor for stabilization.
When the VCOMG bit is low, the VcomL output stops and a capacitor for stabilization is not needed.
VGH
O
A positive power output pin for gate driver, internal charge-pump circuits, bias circuits, and operational amplifiers. Connect a
capacitor for stabilization.
Connect this pin to VCI3 pin. When not using a charge-pump circuit 2, leave it open.
VGL
O
A Negative power output pin for gate driver, bias circuits, and operational amplifiers.
Connect a capacitor for stabilization. When internal VGL generator is not used, connect an external-voltage power supply
higher than -15.0 V.
Vgoff
I
Power supply pin for off level for gate of TFT.
Connect this pin to VgoffOUT. When VgoffOUT is not used, connect an external-voltage power supply higher than -TBD V.
VgoffOUT
O
An power output pin for gate driver.
This pin is a negative voltage for the gate off level. Alternation can be synchronized by M pin. Set the internal register
according to the structure of the TFT-display retention volume.
For the amplitude at the alternation driving, this pin outputs a voltage between VcomH and VcomL with the VgoffL reference
voltage..
VgoffH
O
When the Vgoff alternation is driven, this pin indicates a high level of Vgoff. Connect a capacitor for stabilization. When the
CAD bit is low, the VgoffH output stops and a capacitor for stabilization is not needed.
VgoffL
O
When the Vgoff alternation is driven, this pin indicates a low level of Vgoff. Connect a capacitor for stabilization. An internal
register can be used to adjust the voltage.
System power supply for regulator as external power.
(VDD3: +2.5 ~ +3.3 V)
VDD3
-
AVDD
I/O
A power output pin for source driver block that is generated from power block.
Connect a capacitor for stabilization. (AVDD: +3.5 ~ +5.5 V)
Connect this pin to VCI2 pin. When not using a charge-pump circuit 1, leave it open.
GVDD
I/O
Standard level for grayscale voltage generator.
Connect a capacitor for stabilization.
VCI
I/O
An internal reference power supply for VREG1OUT/VREG2OUT.
Connect VDD when VDD = 2.5 to 3.3 V.
Connect a 2.5 to 3.3 V external-voltage power supply when VDD = 2.0 to 2.5 V.
VSS
-
System ground (0V)
AVSS
-
System ground level for analog circuit block.
VCL
I/O
REGP
I/O
Input pins for reference voltages of VREG1OUT when the internal reference-voltage generation circuit is not used. Leave
these pins open when the internal reference-voltage generation circuit is used.
VREG1OUT
O
This pin outputs a reference voltage for VREG1 between AVDD and VSS. When the internal reference voltage is not used,
the reference voltage can be generated from the voltage of REGP. Connect this pin to a capacitor for stabilization.
When this pin is not used, leave it open.
VREG2OUT
O
This pin outputs a reference voltage for VREG2 between VSS and VGL When the internal reference voltage is not used, the
reference voltage can be generated from the voltage of REGN. Connect this pin to a capacitor for stabilization. When
this pin is not used, leave it open.
VcomOUT
O
A power supply for the TFT-display counter electrode.
The alternating cycle can be set by the M pin. Connect this pin to the TFT-display counter electrode.
This pin is also used as equalizing function: When EQ = “High” period, all source driver’s outputs (S1 to S396) are short to
Vcom level (Hi-z).98(上)
In case of VcomLNCHU
< 0V, equalizing
function/ [email protected]
must not be used. (Set EQ bit (R07h) to bePage
“00” 163
for / 195
/ EE / 汪芳興
preventing the abnormal function.)
A power supply pin for generating VcomL. When VcomL is higher than VSS, outputs
VSS level.
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 164 / 195
Power Definition (1/2)
Power Definition(2/2)
(For the analog circuit)
(For the regulator circuit)
Parameter
Symbol
LCD Supply Voltage
Min.
Typ.
Max.
Unit
AVDD
3.5
-
5.5
V
VGH
9
-
16.5
V
VGL
-16.5
-
-9
V
Conditions
Symbol
Parameter
For the analog circuit power
Min.
Typ.
Max.
Unit
Reference voltage for internal
digital power
RVDD
1.95
2.0
2.1
V
RVDD driving current
IRVDD
-
200
500
uA
Conditions
VDD3=3.3V
VGOFF
-16
-
-5
V
GVDD
3
-
5
V
Reference voltage of
VREG1OUT
REGP
2.46
2.5
2.56
V
Internal reference power supply
voltage
VCI
2.5
-
3.3
V
Reference voltage for
grayscale voltage generator
VREG1
OUT
5.71
5.83
5.95
V
Output Voltage deviation
Vod
-
±20
-
mV
Source Driver
VREG1OUT driving current
IVREG1
1
-
2.5
mA
Output Offset between Chips
Voc
-
±20
-
mV
Source Driver
-6.73
V
0.1
-
GVDD-0.1
V
S1 ~ S396
VREG2
OUT
-6.6
Vdr
Reference voltage output for
gate driver
-6.47
Dynamic Range of Output
Source Driver Driving Current of
Outputs
IVREG2
-500
-
-100
uA
50
-
-
uA
S1 ~ S396; Vo=4.5V v.s 3.5V
AVDD=5V, Gradation output
VREG2OUT driving current
ISOH
GVDD driving current
IGVDD
100
-
150
uA
Gate Driver Sinking Current of
Outputs
IGOL
| -250 |
-
-
uA
G0 ~ G177; Vo=-12V v.s -11.5V
VGH-VGOFF=30V
High level reference voltage of
Vgoff
VgoffH
-0.85
-0.83
-0.81
V
VCI=3.3V, VC2-0=”100”, VRH3-0=”1001”,
VDV4-0=”01101”
Gate Driver Driving Current of
Outputs
IGOH
250
-
-
uA
G0 ~ G177; Vo=18V v.s 17.5V
VGH-VGOFF=30V
Low level reference voltage of
Vgoff
VgoffL
-6.47
-6.6
-6.73
V
VCI=3.3V, VRL3-0=”0001”
Power consumption for Standby mode
Isc
-
-
5
uA
No load, VDD3=3V, VDD=2V, VCI=2.7V,
VBS=VSS and all operating is stopped
High level reference voltage of
Vcom
VCOMH
4.57
4.66
4.75
V
VCI=3.3V, VC2-0=”100”, VRH3-0=”1001”,
VCM4-0=”10101”
IVDD
-
200
500
uA
Low level reference voltage of
Vcom
VCOML
-1.13
-1.11
-1.09
V
VCI=3.3V, VC2-0=”100”, VRH3-0=”1001”,
VDV4-0=”01101”
IVCI
-
1.5
2.0
mA
Operating Current
(VDD =2.0V, VDD3=3V, VSS =0V,
NCHU / EE / 汪芳興 / [email protected]
98(上)
Page 165 / 195
•
VREG1
OUT
Amplification
Circuit 1
(GVDD
adjustment)
G VD D
output
amplifier
REGN
VcomH
adjustment
circuit
REGP
C1 5
GVDD
C9
GVDD
G R AM
Vcom
Amplitude
adjustment
circuit
MS B
R5
Regulator
VcomH
output
amplifier
Voltage
adjustment
circuit
VCOMH
C14
C17
C18
C16
Chargepump
circuit 1
C4
C3
C2
C21M
C21P
C22M
C22P
C23M
C23P
VGH
Chargepump
circuit 2
VCOML
VgoffH
amplitude
adjustment
circuit
VCI2
AVDD
P KP 02
PKP 01
PK P00
P KP 12
PKP 11
PK P10
P KP 22
PKP 21
PK P20
P KP 32
PKP 31
PK P30
P KP 42
PKP 41
PK P40
P KP 52
PKP 51
PK P50
P RP 02
PRP 01
PR P00
P RP 12
PRP 11
PR P10
V RP0 3
V RP 02
VRP 01
VR P00
V RP1 3
V RP 12
VRP 11
VR P10
P KN02
PK N01
P KN0 0
P KN12
PK N11
P KN1 0
P KN22
PK N21
P KN2 0
P KN32
PK N31
P KN3 0
P KN42
PK N41
P KN4 0
P KN52
PK N51
P KN5 0
P RN02
PR N01
P RN0 0
P RN12
PR N11
P RN1 0
VR N03
V RN02
VR N01
V RN0 0
VR N13
V RN12
VR N11
V RN1 0
VCOMOUT
VcomL
output
amplifier
C12C12+
AVDD
VgoffL
output
amplifier
VgoffH
output
amplifier
P o sitive
p o larity
re g ister
C10
VGOFFH
C19
V RP1 4
VGOFFOUT
C1
VGH
C6
∫
VGL
VCI3
C31M
C31P
VGL
C7
VC1
C5
Vci4
C41C41+
VCL
L SB
R4
R3
R2
R1
R0
G5
G4
G3
G2
G1
G0
B5
B4
B3
B2
B1
B0
C12
VCOMR
Vci1
C11C11+
The NT3911 provides the gamma adjustment function to display 262,144 colors
simultaneously. The gamma adjustment executed by the gradient adjustment
register and the micro-adjustment register that determines 8 grayscale levels.
VCI
VCI
VCIOUT
A dj u s t V C O M H
v olt ag e ( w he n
using an extemal
var ia ble resistor)
Page 166 / 195
GAMMA ADJUSTMENT FUNCTION
C1
1
Amplification
Circuit2
(Vgoff
adjustment)
VCI=3.3V, VRL3-0=”0001”
NCHU / EE / 汪芳興 / [email protected]
98(上)
Power Supply Circuits
VREG2
OUT
VCI=3.3V, VC2-0=”100”, VRH3-0=”1001”
(Reference for system design)
TA=25 oC)
C2
1
VCI=3.3V, VC2-0=”100”
VGOFFL
Chargep ump
circuit 3
C20
N eg ative
p o larity
re g ister
VDD
VSS
VCI
VSS
Chargepump
circuit 4
C3
98(上)
NCHU / EE / 汪芳興 / [email protected]
Configuration of the Internal Power-supply Circuit
Page 167 / 195
VRN 14
98(上)
V0
6
6
6
V1
8
G raysc ale
Am p lifie r
64
V 63
64 g ra ysc ale
c o ntro l
<R>
6 4 g ra ysca le
co ntro l
<G >
64 g rays cale
co n tro l
<B >
L CD d river
LC D drive r
LCD driver
R
G
B
LC D
NCHU / EE / 汪芳興 / [email protected]
Page 168 / 195
Scan Mode
G1
A
ODD
G1
G2
EVEN
TFT
Panel
G175
G176
G175
G176
G2
NT3911
G1
B
ODD
G1
G2
EVEN
TFT
Panel
G175
G176
G175
G176
G2
NT3911
Structure of Grayscale Amplifier
• Gate scan mode of NT3911 is set by SM and GS bit. GS
bit determines the scan direction whether the gate driver
scans forward or reverse direction. SM bit determines the
method of display division (Even/Odd or Upper/Lower
division drive). Using this function, various connections
between NT3911 and the liquid crystal panels can be
accomplished
SM
• The structure of the grayscale amplifier is shown as below. Determine
8-level (VIN0-VIN7) by the gradient adjuster and the micro adjustment
register. Each level is split by the internal ladder resistance and level
between V0 to V63 is generated.
G r a d ie n t a d ju s t m e n t
r e g is te r
P R P /N 0
P R P /N 1
G VDD
3
3
M ic r o a d ju s t m e n t r e g is t e r ( 6 X 3 b i t s )
P K P /N 0
P K P /N 1
P K P /N 2
P K P /N 3
P K P /N 4
P K P /N 4
3
3
3
3
3
3
O s c il la t i o n a d ju s t m e n t
r e g is t e r
V R P /N 0
V R P /N 1
4
C
0
TFT
Panel
G175
0
V IN P /N 1
A
G176
V IN P /N 2
8 to 1
s e le c t o r
0
G176
G175
1
B
G2
G176ÆG175ÆG174ÆG173Æ…ÆG4ÆG3ÆG2ÆG1
D
TFT
Panel
G175
V IN P /N 4
8 to 1
s e le c to r
1
0
C
G1ÆG3ÆG5Æ…ÆG173ÆG175
ÆG2ÆG4ÆG6Æ…ÆG174ÆG176
1
1
D
G176ÆG174ÆG172Æ…ÆG4ÆG2
ÆG175ÆG173ÆG171Æ…ÆG3ÆG1
V8
V9
V IN P /N 3
8 to 1
s e le c t o r
Ladder
r e s is t a n c e
NT3911
G1
V1
V2
V3
G1ÆG2ÆG3ÆG4Æ…ÆG173ÆG174ÆG175ÆG176
G2
G1
V0
Scan Mode
GS
8 to 1
s e le c t o r
G1
5
V IN P /N 0
V20
V21
G ra y s c a le
a m p l if i e r
V43
V44
V IN P /N 5
8 to 1
s e le c t o r
V55
V56
V57
G2
G176
G175
G176
G2
NT3911
V IN P /N 6
8 to 1
s e le c to r
V62
V IN P /N 7
V63
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 169 / 195
Gamma Adjustment Register
Grayscale Number
b) Amplitude adjustment
98(上)
Grayscale Voltage
Grayscale Voltage
Grayscale Voltage
Grayscale Number
a) Gradient adjustment
Grayscale Number
c) Reference adjustment
NCHU / EE / 汪芳興 / [email protected]
98(上)
VG S
NCHU / EE / 汪芳興 / [email protected]
Page 170 / 195
Chip Size & Pad Dimensions
• This block has the register to set up the grayscale voltage adjusting
to the gamma specification of the LCD panel. These registers can
independently set up to positive/negative polarities and there are 4type of register groups to adjust gradient and amplitude on number
of the grayscale, characteristics of the grayscale voltage. (average
<R><G><B> are common.) The following figure indicates the
operation of each adjusting register.
Grayscale Voltage
G1
Items
Pad name.
Chip size
Pad size
Size
X
Y
-
20720
2500
Input Pad
Output Pad
Dummy Pad
1,195,239,742
54
36
100
70
80
80
Unit
um
Grayscale Number
d) Micro-adjustment
Page 171 / 195
• NOTES:
• Scribe line included in this chip size (Scribe line: 120um)
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 172 / 195
Product Applications
Process Feature for One Chip Driver IC
• TFT Driver IC for Mobile Phone
Technology
0.25um
1P3M (MIM Process)
LV (Dual Gate) / HV
Oxide Thickness
40V HVMOS Structure
LV / HV Well Structure
Isolation
Gate Material
S / D Area
Capacitor
High Rs Poly
2.5V / 5V / 40V (+/-20V)
48A / 110A / 1000A
LDMOS
Retrograde / Drive-In Well + NBL
STI / P-EPI
Poly (S/D Implant Doped) + Salicide
Salicide
MIM
400~2000 Ohm
• STN Driver IC for Mobile Phone
– NT7523為Hi-Fas CSTN One Chip Driver IC，使用0.25μm
2.5/5/32V Process
Source : Novatek training material
98(上)
Source : Novatek training material
NCHU / EE / 汪芳興 / [email protected]
Page 173 / 195
NCHU / EE / 汪芳興 / [email protected]
98(上)
LV and HV Device Type
Page 174 / 195
LTPS TFT-LCD Driver ICs
Q LV and HV Device Type and Related Data (1)
Ò 2.5 / 5V Devices Characterization
Salicide Structure
Ò2.5 / 5V Devices Cross Section Sketch
Source Driver IC
Simplified
Source Driver IC
T-Con
Gate Driver IC
Triple Well Structure
Gate Driver IC
T-Con
STI Field Isolation
A-Si TFT-LCD
Conventional
a-Si TFT-LCD
De-MUX
LTPS
TFT-LCD
LTPS TFT-LCD
Example 1
T-Con
Gate Driver IC
5 LV Devices ( 2.5V NMOS and PMOS, 5V NMOS and PMOS, 2.5V Native NMOS )
Source Driver IC
LTPS
TFT-LCD
LTPS TFT-LCD
Example 2
非晶矽與多晶矽 TFT-LCD 驅動方式比較圖
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 175 / 195
Source : Novatek training material
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 176 / 195
Outline
Interfaces in TFT-LCD Monitor
Ch4. Driving Circuits Design of A-Si TFT
– Gate Driving Circuit
– Source Driving Circuit
– LCD-TV Driving Technology
– Small-Size TFT-LCD Driver IC
– Trends of Digital Interface
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 177 / 195
Digital vs. Analog
98(上)
NCHU / EE / 汪芳興 / [email protected]
98(上)
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Page 178 / 195
Panel Input Interface
Page 179 / 195
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 180 / 195
Comparison of TTL，RSDS，and MiniLVDS
TTL vs. LVDS
Bus lines
8-bit
6-bit
Voltage Swing
Frequency
XGA
SXGA
UXGA
LCD
Application
PCB Area
T-CON Pins
Driver IC
Input Pins
6-bit
6-bit
TTL
24X2
18X2
3.3V
RSDS
24
18
200mV
Mini-LVDS
12
10
200mV
32.5M
(2 ports)
54M
(2 ports)
~SXGA
67M
( 1 port)
54M
(2 ports)
81M
~ UXGA
67M
(2 ports)
108M
(2 ports)
162M
~ QXGA
1
~100
~80
0.7~0.8
~64
~60
0.5~0.7
~100
~50
Remark
Lower amplitude for
reducing EMI
Remark: RSDS and Mini-LVDS use twin-pair lines
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 181 / 195
98(上)
NCHU / EE / 汪芳興 / [email protected]
Page 182 / 195
8 Bit RGB RSDS Data
RSDS Definition
• Reduced Swing Differential Signal
–
–
–
–
–
±200 mV swing (typical)
2:1 mux - 2 data per clock cycle
100 ohm differential terminals
Voffset = 1.2 V
RGB data and clock only
Start pulse
R0 R1
Invalid data
R2 R3
R4 R5
R6 R7
G0 G1
• Apply to bus between T-Con and
source drivers
Invalid data
G2 G3
G4 G5
G6 G7
– Reduce EMI
– Reduce power consumption
– Reduce source driver bus width
B0 B1
Invalid data
B2 B3
B4 B5
B6 B7
98(上)
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System Diagram ：
RSDS Configuration
RSDS Features and Benefits
Single-end
• Reduced pin T-Con counts
– Enable smaller area PCBs
• Reduced number of components
– Small area PCBs
– Lower cost
Front/Back
– Number of components：TTL:RSDS = 190:101 (in
14.1” XGA) Æ 46.8％ reduction
• Reduced number of PCB layers
– Number of layers：TTL :RSDS = 6:4
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Mini-LVDS
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Mini-LVDS
6 bit data,
5 pairs
Data Bus Structure
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Mini-LVDS
RSDS vs. Mini-LVDS
RSDS
8 bit data,
PCB Size
6 pairs
TCON
Larger
XGA: TQFP64/80
SXGA/UXGA:TQFP128/144
Same
Driver Size
DIO(Start Pulse)
Frequency Limitation
Resolution Limitation <= UXGA/Dual Buses
Possible Driver Vendors More
Mini-LVDS
Smaller
TQFP100
Same
No
QXGA or larger/Dual Buses
Limited
RSDS is the major interface
Mini-LVDS become important for high resolution panel
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Point to Point Differential Signaling
(PPDS)
• PPDS is based largely on the
RSDS™
• Advantages:
– The total number of input signals
for each column driver is greatly
reduced.
• 8-bit RSDS system: 12 data pairs
and the clock pair by each column
driver.
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PPDSTM
• Advantages: (cont.)
– improved signal integrity.
Parameter
RSDS
PPDS
Differential Signal
Level
±200mV
±200mV
Common Mode
Voltage
1.3V
0.8V
Typical Output
Current
2mA
2mA
Transmission Lines
50Ω
50Ω
• typical RSDS bus architecture
• Î vias and stubs on every signal
line Î creates a large number of
impedance discontinuities.
• point to point systemÎ no vias and
stubsÎ data signal maintain higher
levelÎ Higher color depth
– Major improvement in EMI
Table 1. Comparison of PPDS and RSDS levels
• In a PPDS system: signal data pair
and a clock pair
• Due to the incoming LVDS clock
and PPDS clock operating at
different frequency.
• 26 in RSDS Î 4 in PPDS
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Summaries
PPDS Protocol
Technology Trends of Large-Size TFT-LCD Source Driver IC
• The total reduction in
data signals from an
RSDS based system
to the PPDS
10-bit
(1024灰階)
architecture.
• The protocol is split
into 5 required interval
and 1 optional interval.
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Memo
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