Download Charge pump: adaptive hysteretic control with modular switches

Charge pump: adaptive hysteretic control with
modular switches
Ricardo Madureira
Valter Sádio
Marcelino Santos
IST-UTL
Lisbon, Portugal
[email protected]
IST-UTL / INESC-ID
Lisbon, Portugal
[email protected]
IST-UTL / INESC-ID, Silicongate
Lisbon, Portugal
[email protected]
Abstract— A charge pump boost converter is proposed. To
achieve an acceptable output voltage ripple without
jeopardizing transient response and start-up time, a method of
parallelizing switches is used, controlled by a Mealy machine.
The result is an adaptive hysteretic control that uses modular
switches. With light loads, the number of modules is reduced to
one per switch. During high current load transients, all modules
are used, thus tripling the output current capability. The
proposed charge pump supplies a regulated voltage of 3.3V,
from an input voltage of 2.5V. The IP includes soft-start (inrush current limitation), soft-recovery and short-circuit
protection on all output pins. It was designed with TSMC
130nm LowPower technology.
Keywords — charge pump; switched capacitors; hysteretic
control; digital control; inrush current; soft-recovery; controlled
start-up
I. INTRODUCTION
To implement a charge pump able to convert 2.5V into
3.3V, a voltage doubler structure was used, showed in Fig.1.
In a voltage doubler charge pump, transient response is
limited by the output equivalent resistance [1], given by (1).
Vin
Rsw2
Rsw1
Cfly
Rsw4
Rsw3
Vout
switcher’s resistance or the flying capacitor (CFLY in Fig.1
and in equation (1)).
The maximum efficiency of the charge pump doubler is
presented in (2) and the voltage ripple [2] depends on the
frequency of operation, current load and output capacitance
(3).
(2)
(3)
The current consumption of the driving circuit of the
charge pump is, typically, the most significant and must be
minimized to achieve higher efficiency. This consumption is
proportional to the capacitance of the driver’s stage, to the
frequency of operation and to the square of the driving
voltage. This last parameter must be equal to the highest
voltage available in the circuit, to properly turn ON and OFF
the PMOS switches.
Lowering the frequency of operation or the driving stage
capacitance (including the main switch) is possible. But,
lowering the frequency of operation will increase voltage
ripple, and less driving capacitance will limit the transient
response of the charge pump, by limiting the output
resistance to a higher value. The solution proposed in this
work is to divide the switches, enabling low ripple and fast
transient response.
Cout
II. CONTROL METHOD
Fig. 1. Voltage doubler topology.
(1)
This resistance is modulated to achieve output voltage
regulation by varying the frequency of operation, the
Several methods are used nowadays to regulate output
voltage in charge pumps, each
with
advantages
and
disadvantages. Constant frequency control [1] with two
sequenced phases of operation is used with a
transconductance amplifier that compares the output voltage
with a voltage reference and controls the amount of charge
injected in the flying capacitors. The output capacitor is
charged when the output voltage is too low. This control
scheme has a well defined noise frequency spectrum that can
be filtered, but has poor efficiency with light loads. Current
mode control [3, 4] can be implemented with a current
sensing circuit, usually a series sensing resistor, with large
power loss, or with a current sensing transistor in parallel
with the power transistor that demands additional circuitry.
This kind of control is used with duty cycle close to
conversion ratio to achieve low output voltage ripple in high
load conditions.
Pulse Skipping [5,6] is used with constant switching
frequency and constant duty cycle with charge and discharge
modes of operation. The charge pump operates on charge
mode when the output voltage drops below a predetermined
threshold voltage, otherwise operates with minimum supply
current in discharge mode, where the output capacitor
supplies charge to the load. This control method is very good
to improve efficiency with light loads and has fast response
to load variations but produces high output voltage ripple.
Burst Mode [7] is a control methodology that, when the
load current load is less than a prescribed threshold, the clock
is turned on and off for a number of clock cycles that depend
on the current absorbed by the load. This method allows high
efficiency with light loads but produces high output voltage
ripple and high tonal spectrum.
The solution proposed in this work is to divide the
switches, so that for lower output currents only 1/3 of the
switching strength is used, and for higher currents or during
transients, full switching strength is used. This way, driver
capacitance is reduced for most of the output current range of
the charge pump. To further increase efficiency, a break
before make circuit is used to drive the main switches.
A PMOS switch module, in Fig.2, and a NMOS switch
module, in Fig.3, were designed, that include the driving
circuit, driven by low voltage digital signals. These modules
can be parallelized in different number. In the proposed
charge pump, they were combined in triplets, for each charge
pump switch. During normal operation only one module per
switch is activated. Whenever the current demand is higher,
all three are used.
The control was implemented with a Mealy machine. If
the output is low during two or more pumping periods, all
switches are used. If the output voltage stays higher than the
desired value for less than two pumping cycles, only one
switch is used. This method allows reducing the output
resistance of the charge pump to one third for load transients,
thus improving dynamic response.
In the diagram presented in the Fig. 4 the box Switch
Module represents either a PMOS switch or an NMOS
switch, depending on the relevant phase. Two slightly
different blocks were design for Sw2 position. The difference
is that the PMOS module switches in this position must allow
being current limited, so that they can behave like current
sources during start-up, short-circuit and soft-recovering
events.
Low voltage domain
(DVDD)
Highest voltage domain
(Vin if Vout < Vin
or
Vout if Vout > Vin)
Mp1
Feedback to
digital ctrl block
Mn1
Non-overlap circuit
avoids shoot-through
current in the driver
Fig. 2. Simplified PMOS module circuit diagram.
Highest voltage domain
(Vin if Vout < Vin
or
Vout if Vout > Vin)
Low voltage domain
(DVDD)
Mp1
Crtl input
Feedback to
digital ctrl
block
Mn1
Non-overlap circuit
avoids shoot-through
current in the driver
Fig. 3. Simplified NMOS module circuit diagram.
Vout
Cout
Phase
Generator
Feedback
comparator
switch
module
+
switch
module
Z
Vref
-1
switch
module
Fig. 4. Simplified block diagram of the main control circuit.
III. ANCILLARIES
A. Highest voltage selection
The PMOS switching devices must be biased and driven by
the highest voltage available in the circuit. During start-up
the highest voltage available is VIN, after that, VOUT is the
highest one. Therefore, a voltage selection circuit must be
used. For bulk biasing, a simple OR diode circuit is enough,
but to properly supply the driving circuitry, a low impedance
selection circuit is needed. Fig. 5 presents the block diagram
of the voltage selection circuit. A passive OR circuit is used
in parallel with an active OR circuit. The impedance of the
selection devices is such that allows the driven circuit to
operate without significant voltage drops.
B. Soft-start, short-circuit protection and soft-recovery
When the charge pump is first turned on, the flying and
output capacitors can be fully discharged. This means that
when the switches are closed in order to charge the flying
capacitor and when they change to the second phase, to
transfer the charge do the output capacitor, the voltage
difference will be high. This high voltage difference will
originate current peaks in the order of VIN/RSW, which can
reach several amps. Also, when supplying digital cores, it is
important to control the ramp-up output voltage, so that the
digital circuits start properly. To limit the in-rush current and
to ramp the output voltage the following method is used.
available in the circuit, in this situation, V IN. Also that is the
voltage applied to the Sw2 source terminal, therefore, it
remains saturated.
The same circuit also gives short-circuit protection for all
output pins of the charge pump. When the output current
increases beyond regulation limits, the output voltage begins
to fall, due to the output resistance of the charge pump. When
the output voltage falls below VIN less a threshold voltage,
the current limitation circuit is engaged and switches are
forced to the start-up position. This mechanism protects the
circuit from short-circuits in the load and in the capacitors.
Mp1
A comparator will sense the output and input voltages. If
the output voltage is lower than the input voltage, the
comparator output signals the control block to enter in startup mode.
‘1’
Sw2
Mn1
Passive OR
‘0’
Vhigh
Vin
Vin
Vout
Active OR
Vout
Vin
Comparator
Levelshifter
with shcmitt-trigger
Fig. 7. Switch 2 current limitation.
Fig. 5. Highest voltage selection circuit.
Vin
Rsw1
Rsw2
Rsw3
Rsw4
To control
block
Start-up
comparator
output
+
Vout
Vreg
Start-up
comparator
Vbulk
Vin
Fig. 6. Start-up comparator.
When signaled to enter start-up mode, the control block
will close switches 3 and 4, putting CFLY and COUT in parallel,
and limit the current capability of switch 2. The control
voltage applied to the gate of this switch is limited by the
circuit presented in Fig. 7. The current limit is configurable,
in order to control the start-up time with different external
capacitances.
The driving transistors Mp1 and Mn1, in Fig.7, are turned
off and the Sw2 is forced into saturation [8]. The current
reference circuit is supplied by V IN, so the output voltage is
compared with VIN less a threshold voltage, in Fig. 6, to
ensure that the internal reference currents do not lose
compliance. The driving voltage is the highest voltage
Vreg
Vout
Vin-Vth
Continuous charge
t=(Vin - Vth).(Cout + Cfly) / Ilimit
Discrete charge
τ=(Rout)Cout
Fig. 8. Start-up sequence.
To lower the current consumption during normal
operation, the start-up comparator is shut-down by the
highest voltage selection circuit. Whenever the output voltage
becomes higher than VIN, start-up circuit is disabled.
C. Feedback voltage circuit
The feedback voltage comparator used in the proposed
charge pump is a standard clocked comparator. The
comparator uses low voltage transistors, thus the reference
voltage is not the usual bandgap voltage of approximately
1.2V but only 600mV. Also due to the usage of low voltage
transistors, the inputs of the comparator are clamped by diode
connected high voltage NMOS, to protect them from higher
voltages.
latch
latch
other input, coming from a voltage reference circuit, has also
low impedance at moderate frequencies. The capacitors are
formed by fingered metal layers placed on top of the poly
resistors from the resistive voltage divider, so no extra silicon
area is needed.
(3)
The feedback circuit is trimmed, and provides two extra
output voltages: Vsnshigh and Vsnslow. These voltages are
used by an external comparator, sharing the same 600mV
reference voltage, to assert a Vok signal.
outp
outn
reset
vp
vn
Vsns
latch
resetz
latch
C1
R1
C2
R2
Vsnshigh
Vfb
HV
HV
Clamping
Output voltage
comparator
Vsnslow
Cin
Fig. 9. Output voltage feedback comparator.
3:8
decoder Trim[0:2]
Fig. 11. Output voltage feedback.
clk
reset
latch
(
resetz
(
)
)
(
)
(4)
Fig. 10. Reset and latch generation.
To reduce kick-back to the inputs, a complementary
switch is used to perform the reset of the internal nodes of the
comparator. By using complementary switches, the charge
injection of each one, when they are turned ON or OFF, is
canceled by each other.
Due to process variations, reset and latch phases can
become off 180º phase. To guarantee that the latch phase
always start after the completeness of the reset phase a
starved inverter is used to generate it , presented in Fig. 10.
The output of the voltage comparator is held by a
standard latch circuit, during the reset phase.
The output voltage is sensed by a compensated voltage
feedback resistive divider, as presented in Fig.11. Two
capacitors are placed in parallel with the two branches of the
resistive divider with two proposes. First, they allow
compensating the capacitance effect of the input node of the
comparator that would otherwise form a pole in the sensing
circuit at a frequency given by (3). Secondly, these capacitors
will raise the capacitance seen by the input of the comparator,
therefore reducing kickback effects. It is assumed that the
(5)
To compensate the response of the resistive divider, the
capacitance in the lower branch, formed by the added
capacitance plus the input capacitance of the comparator (5),
must be equal to C1*(R1/R2) (4).
E. Vok and UVLO
Two important signals are commonly used with switched
regulators: Vok and UVLO.
Vok is asserted when the output voltage is within certain
limits of the desired voltage. This allows to start-up the
supplied circuit only when the supplying circuit is fully
operational. In the proposed charge pump this signal is
generated with a comparator with hysteresis, which shares the
same voltage divider of the charge pump, taking advantage of
the two additional voltages Vsnshigh and Vsnslow. Fig. 12
presents the block diagram of the comparator with hysteresis.
Vsnslow
Vsnshigh
-
Vok
+
Vref
Averaged input current.
Fig. 12. Vok generator circuit.
Vin
+
UVLOz
-
en
Vsnshigh
Vref
UVLO
Vsnslow
Fig. 15. Start-up simulation.
dvdd
Vbgok
Fig. 13. UVLO generator circuit.
UVLO signal is used to disable the charge pump
operation when the input voltage V IN is not high enough to
sustain the proper operation of the circuit. The objective is to
avoid erratic behavior of the switching circuit. This signal is
of utmost importance when no soft-start circuit is used. When
VIN is ramping up, without a soft-start circuit the switching
regulator will soon start to demand high peak currents from
the supply voltage. These peak currents can interfere with the
ramp up of the input voltage and cause oscillations during
start-up. Despite the inclusion of a start-up circuit, it was
decided to implement also an UVLO circuit.
IV. Results
The circuit was simulated using HSPICE. In the
simulation results, presented below, ideal 1uF and 100nF
flying and output capacitors were used. Input voltage is 3.3V
(TYP), 2.25V (MIN) and 2.94V (MAX). The external
frequency supplied to the charge pump is 1 MHz.
Fig. 16. Short-circuit protection and soft-recovery.
Fig. 15 presents the average input current during start-up
(middle plot). Fig. 16 presents an output short-circuit,
followed by soft-recovery. The load of 330Ω is shortcircuited during 100µs. The upper plot shows the output
voltage, the middle plot shows the current in the load and the
third plot shows the input current.
Fig. 17 presents the transient response from 1mA to
10mA. Table I resumes the obtained results.
Fig. 17. Load transient from 1mA to 10mA.
Table I. Simulation results
Parameter
ACKNOWLEDGMENT
MIN TYP MAX
Unit
Output Voltage
3.15
3.3
3.45
V
Output Voltage Accuracy
-5
-
+5
%
Output voltage ripple (1 mA)
-
-
20
mVpp
Output voltage ripple (10 mA)
-
-
60
mVpp
In rush and short-circuit current
(programmable)
25
-
150
mA
Start-up time
-
100
200
s
Load Transient (1-10mA)
-
2
-
mV/mA
Load Regulation (0-50mA)
-
0.8
-
mV/mA
Quiescent current
-
5
10
A
(no load)
Efficiency
65
%
Core area
(Excluding Vok/UVLO circuits and
PADS)
-
0.13
-
mm2
V. Conclusions
An adaptive hysteretic control that uses modular switches
was proposed. With light loads, the number of modules is
reduced to one per switch. During high current load
transients, all modules are used, thus tripling the output
current capability. The proposed charge pump supplies a
regulated voltage of 3.3V, from an input voltage of 2.5V. It
was designed to provide a maximum permanent output
current of 10mA. When operating at full strength, the output
current capability extends to more than 30mA. Output
voltage ripple with low ESR capacitors (20mΩ) of 1μF and
100nF (COUT and CFLY) is 20mV with 1mA load current and
60mV with 10mA load. The typical efficiency with
maximum output current is 65%, 98% of the maximum
theoretical value. The quiescent current without load is less
than 10μA. The typical power dissipation at full load is 17.8
mW. The IP includes soft-start (in-rush current limitation),
soft-recovery and short-circuit protection on all output pins. It
was designed with TSMC 130nm LowPower technology.
This circuit can be used in standard applications like USBOTG 3.3V.
This work was partially funded by the Portuguese
National Foundation of Science and Technology through the
project PTDC/EEA-ELC/113902/2009 and through the
PIDDAC Program funds (INESC-ID multiannual funding).
REFERENCES
[1] B. Robert Gregoire, “A Compact Switched-Capacitor
Regulated Charge Pump Power Supply”, IEEE Journal of
Solid-state Circuits, vol. 41, pp. 1944-1953, August 2006.
[2] Jae-Youl Lee et al, “A Regulated Charge Pump With
Small Ripple Voltage and Fast Start-Up”, IEEE Journal of
Solid-state Circuits, vol. 41, pp. 425-432, February 2006.
[3] Shao Bin; Yang Yujia; Wang Ying; Hong Zhiliang, “High
Efficiency, inductorless step-down DC-DC converter”, ASIC,
2005, ASICON 2005. 6th International conference, vol. 1, no.,
pp. 395-398, 24-0 Oct. 2005.
[4] Cheung Fai Lee; Mok, P.K.T., "A monolithic currentmode CMOS DC-DC converter with on-chip current-sensing
technique," Solid-State Circuits, IEEE Journal of , vol.39,
no.1, pp. 3-14, Jan. 2004.
[5] Saiz-Vela, A.; Miribel-Catala, P.; Puig-Vidal, M.;
Samitier, J., "An electron mobility independent pulse
skipping regulator for a programmable CMOS charge pump,"
Circuits and Systems, 2005. ISCAS 2005. IEEE International
Symposium on , vol., no., pp. 117-120 Vol. 1, 23-26 May
2005.
[6] Valter Sádio; Abílio Parreira; Marcelino Santos, "A
100mA fractional step-down charge pump with digital
control," Proc. SPIE, vol. 7363, May 2009.
[7] Budaes, M.; Goras, L., "Burst Mode Switching
Mechanism for an Inductorless DC-DC Converter,"
Semiconductor Conference, 2007. CAS 2007. International,
vol.2, no., pp.463-466, Oct. 15 2007-Sept. 17 2007.
[8] Erich Bayer, Hans Schmeller, “Charge Pump with Active
Cycle Regulation – Closing the Gap between Linear and Skip
Modes”, Power Electronics Specialists Conference, vol. 3,
pp. 1497-1502, June 2000.